JP2019146419A - Inverter control device and motor driving system - Google Patents

Abstract

To provide an inverter control device capable of achieving a fast speed response, and a motor driving system.SOLUTION: An inverter control device comprises: an inverter 1; a current detector 3; a speed control circuit 28 including a first PI controller which calculates a torque command value so that a rotational speed of a motor 2 equals a rotational speed command value; a current command value generation circuit 27 which calculates a current command value equivalent to a current response value on the basis of the torque command value; a current control circuit 25 which calculates an output voltage output from the inverter 1 so that the current command value equals the current response value; a second PI controller which outputs a torque command estimation value with the difference between the rotational speed command value and a rotational speed approximate value as an input; and a motor machine model circuit which calculates the rotational speed approximate value using the torque command estimation value. A gain used by the first PI controller comprises an approximate speed estimation circuit 29 equivalent to the gain used in the first PI controller, and a speed estimation circuit which calculates a speed estimation value of the motor 2 using the rotational speed approximate value.SELECTED DRAWING: Figure 1

Description

  Embodiments described herein relate generally to an inverter control device and an electric motor drive system.

  In a rotation angle sensorless control device for a permanent magnet synchronous motor (PMSM) or a synchronous reluctance motor (SynRM), a method for estimating a rotation angle is used. In speed estimation, a system called PLL (Phase Locked Loop) is known that estimates speed by performing PI control so that an evaluation index corresponding to an axis error becomes zero.

  In addition, a primary frequency estimator has been proposed that calculates an estimated primary frequency from a q-axis secondary magnetic flux induced voltage and feeds it to a PI controller in a feedforward manner in a speed sensorless control device for an induction motor (IM).

Japanese Patent No. 3692085

“Sensorless vector control of AC drive system”, edited by the IEEJ / Survey of Technical Committee on Organizing Sensorless Vector Control, Ohm, September 2016

  When a response of high-speed speed estimation is required, there is a possibility of stepping out unless the gain of the PI controller is increased in the PLL control. On the other hand, when the gain of the PI controller is increased, overcontrol occurs, and the control becomes oscillating.

  On the other hand, in the permanent magnet synchronous motor (PMSM) and the synchronous reluctance motor (SynRM) as well as the induction motor (IM), the estimated speed calculated from the q-axis induced voltage is given by feedforward, thereby the PI controller. It is possible to realize a high speed response without increasing the gain. However, in permanent magnet synchronous motors (PMSM) and synchronous reluctance motors (SynRM) that mainly use reluctance torque, it is difficult to calculate the estimated speed from the q-axis induced voltage because the q-axis induced voltage changes according to the load. Met.

  Embodiments of the present invention have been made in view of the above circumstances, and an object thereof is to provide an inverter control device and an electric motor drive system that realize a high-speed speed response.

  In the inverter control device according to the embodiment, the rotation speed and the rotation speed command value of the electric motor connected to the inverter, the current detector that detects the current response value output from the inverter, and the inverter are equal to each other. A speed control circuit including a first PI controller that calculates a torque command value of the electric motor; a current command value generation circuit that calculates a current command value corresponding to the current response value based on the torque command value; and the current command A current control circuit for calculating an output voltage output from the inverter so that the current response value is equal to the current response value, and a difference between the rotational speed command value and the rotational speed approximate value is input and a torque command estimated value is output. A second PI controller, and a motor machine model circuit that calculates the approximate rotational speed value using the estimated torque command value, wherein the second PI controller Gains need includes a schematic speed estimation circuit is equivalent to the gain used by the first 1PI controller, using the rotational speed approximate value, and a speed estimation circuit for calculating a velocity estimation value of the motor, the.

FIG. 1 is a block diagram schematically illustrating a configuration example of an inverter control device and an electric motor drive system according to the first embodiment. FIG. 2 is a block diagram schematically showing a configuration example of the speed control circuit of the inverter control device shown in FIG. FIG. 3 is a block diagram schematically showing a configuration example of the schematic speed estimation circuit shown in FIG. FIG. 4 is a block diagram schematically showing a configuration example of the speed / rotation phase angle estimation circuit shown in FIG. FIG. 5 is a block diagram schematically showing a configuration example of the rotational phase angle error calculation circuit shown in FIG. FIG. 6 is a block diagram schematically showing another configuration example of the schematic speed estimation circuit shown in FIG. FIG. 7 is a block diagram schematically showing another configuration example of the motor machine model shown in FIG. FIG. 8 is a block diagram schematically illustrating a configuration example of the inverter control device and the motor drive system according to the second embodiment. FIG. 9 is a block diagram schematically showing a configuration example of the speed estimation circuit shown in FIG.

Hereinafter, an inverter control device and an electric motor drive system according to a plurality of embodiments will be described with reference to the drawings.
FIG. 1 is a block diagram schematically illustrating a configuration example of an inverter control device and an electric motor drive system according to the first embodiment.

The electric motor drive system of the present embodiment includes an inverter control device, an inverter 1 and a synchronous machine 2.
The synchronous machine 2 is, for example, a synchronous reluctance motor (SynRM) including a stator and a rotor. The synchronous machine 2 is an AC motor that is driven by a three-phase AC current supplied from the inverter 1, generates a magnetic field by a three-phase AC current flowing in each excitation phase, and generates a torque by a magnetic interaction with the rotor. is there.

  The inverter 1 includes, for example, an inverter main circuit (none of which is shown) including a DC power source (DC load) and two power semiconductor elements of each phase of U phase, V phase, and W phase. . The two power semiconductor elements in each phase are connected in series between a DC line (DC link) connected to the positive electrode of the DC power supply and a DC line (DC link) connected to the negative electrode of the DC power supply. The operation of the power semiconductor element of the inverter 1 is controlled by a gate command received from an inverter control device described later. The inverter 1 is a three-phase AC inverter that outputs a U-phase current iu, a V-phase current iv, and a W-phase current iw to a synchronous machine 2 that is an AC load. Further, the inverter 1 can also charge the power generated by the synchronous machine 2 to a DC power source.

  The inverter control device of the present embodiment may be configured by hardware, may be configured by software, or may be configured by combining hardware and software. The inverter control device may include, for example, at least one processor and a memory in which a program executed by the processor is recorded.

  The inverter control device of this embodiment includes a current detector 3, a PWM modulation circuit 22, a coordinate conversion (UVW / dcqc) circuit 23, a coordinate conversion (dcqc / UVW) circuit 24, a current control circuit 25, a speed A rotational phase angle estimation circuit 26, a current command generation circuit 27, a speed control circuit 28, and an approximate speed estimation circuit 29 are provided.

  The current detector 3 includes at least two-phase current response values (for example, U-phase) among the three-phase AC currents (U-phase current value iu, V-phase current value iv, and W-phase current value iw) output from the inverter 1. The current value iu and the W-phase current value iw) are detected. The current response value detected by the current detector 3 is input to the coordinate conversion (UVW / dcqc) circuit 23.

  The coordinate conversion (UVW / dcqc) circuit 23 is a vector conversion circuit that performs coordinate conversion from the three-phase fixed coordinate system to the dcqc axis rotational coordinate system using the rotational phase angle estimated value θest. The coordinate conversion (UVW / dcqc) circuit 23 converts the U-phase current value iu and W-phase current value iw of the three-phase fixed coordinate system into the dc-axis current value idc and the qc-axis current value iqc of the dcqc-axis rotation coordinate system. Output.

  The dcqc axis rotation coordinate system is an estimated rotation coordinate system. The d-axis is a vector axis having the smallest static inductance in the rotor of the synchronous machine 2, and the q-axis is a vector axis that is an electrical angle and orthogonal to the d-axis. On the other hand, the estimated rotational coordinate system corresponds to the d axis and the q axis at the estimated position of the rotor. That is, the vector axis rotated by the rotation angle error Δθ from the d axis is the dc axis, and the vector axis rotated by the rotation angle error Δθ from the q axis is the qc axis.

The speed control circuit 28 includes, for example, a PI controller, and a torque command T so that the speed command ω ^* supplied from the outside is equal to the speed estimated value ωest supplied from the speed / rotation phase angle estimation circuit 26. ^* Calculate and output.

FIG. 2 is a block diagram schematically showing a configuration example of the speed control circuit of the inverter control device shown in FIG.
The speed control circuit 28 includes a subtractor 31, a PI controller (first PI controller) 32, and a torque limiting circuit 33.

The subtracter 31 subtracts the estimated speed value ωest from the speed command ω ^*, and outputs a speed error Δω (= ω ^* −ωest) as a calculation result.
The PI controller 32 receives the speed error Δω from the subtractor 31 and calculates the torque command calculation value T ^* cal so that the speed error Δω becomes zero.

The torque limit circuit 33 limits the torque command calculation value T ^* cal with the torque limit value Tlim, and outputs a torque command value T ^* that is a value after the limit. The torque limit value Tlim may be a preset value or a value input from the outside.
The current command generation circuit 27 generates and outputs dq axis current commands idc ^* and iqc ^* using the torque command T ^* and the estimated speed ωest.

The current control circuit 25 includes current response values idc and iqc obtained by vector conversion of the current response values iu and iw detected by the current detector 3, and current command values idc ^* and iqc ^* generated by the current command generation circuit 27. And fundamental wave voltage command values (modulation rate command values) vdc ^* and vqc ^* are determined so as to realize current command values idc ^* and iqc ^* .

The coordinate conversion (dcqc / UVW) circuit 24 is a vector conversion circuit that performs coordinate conversion from the dcqc axis rotation coordinate system to the three-phase fixed coordinate system using the rotation phase angle estimated value θest. The coordinate conversion (dcqc / UVW) circuit 24 converts the voltage command values vdc ^* and vqc ^* of the dcqc axis rotation coordinate system into voltage command values (modulation rate command values) vu ^* , vv ^* and vw ^* of the three-phase fixed coordinate system. Convert and output.

The PWM modulation circuit 22 modulates voltage command values vu ^* , vv ^* , and vw ^* for driving the synchronous machine 2 with a triangular wave PWM, and turns on (conducts) / off (non-conducts) each phase power semiconductor element of the inverter 1. Outputs a gate signal that is a continuity command.
The approximate speed estimation circuit 29 calculates a speed estimation FF (feed forward) value ωestFF using the speed command ω ^* .

FIG. 3 is a block diagram schematically showing a configuration example of the schematic speed estimation circuit shown in FIG.
The approximate speed estimation circuit 29 includes a subtractor 41, a PI controller (second PI controller) 42, a torque limiting circuit 43, and a motor machine model circuit 44.

The subtractor 41 subtracts the speed estimation FF value ω ^* estFF from the speed command ω ^* to calculate a speed error Δω ^* .
The PI controller 42 calculates the torque command calculation estimated value T ^* estcal so that the speed error Δω ^* becomes zero. Here, the proportional gain and integral gain used in the PI controller 42 are the same as the proportional gain and integral gain used in the PI controller 32 of the speed control circuit 28.

The torque limit circuit 43 limits the torque command calculation estimated value T ^* estcal with the torque limit value Tlim and outputs the torque command estimated value T ^* est. Here, the torque limit value Tlim is the same value as the torque limit value Tlim used in the torque limit circuit 33 used in the speed control circuit 28.

The motor machine model circuit 44 includes a divider 48 and an integrator 49.
The divider 48 divides the torque command estimated value T ^* est by the inertia estimated value Jest and outputs the result.
The integrator 49 integrates the value T ^* est / Jest output from the divider 48, calculates the speed estimation FF value ω ^* estFF, and outputs it. As the inertia estimated value Jest, a fixed value adjusted by speed control can be used. That is, the speed estimation FF value ω ^* estFF can be expressed as the following equation.

The speed estimation FF value ω ^* estFF is an approximate rotational speed value of the synchronous machine 2 calculated by the motor machine model circuit 44.
The speed / rotation phase angle estimation circuit 26 calculates the speed estimation value ωest and the rotation phase angle estimation value θest using the speed estimation FF value ω ^* estFF, voltage commands vdc ^* , vqc ^*, and current response values idc, iqc.

FIG. 4 is a block diagram schematically showing a configuration example of the speed / rotation phase angle estimation circuit shown in FIG.
FIG. 5 is a block diagram schematically showing a configuration example of the rotational phase angle error calculation circuit shown in FIG.
The speed / rotation phase angle estimation circuit 26 includes a rotation phase angle error calculation circuit 51, a PI controller 52, an adder 53, and an integrator 54.

The rotational phase angle error calculation circuit 51 includes a phase difference δ setting circuit 60, a γ voltage calculation circuit 61, a γ voltage estimation circuit 62, and a subtractor 63.
The phase difference δ setting circuit 60 receives the torque command value T ^* as an input, and sets a phase angle δ in the vicinity where the voltage change is greatest when an error occurs between the rotor phase estimation value and the rotor phase. . The phase difference δ setting circuit 60 may include a set value table for the phase angle δ that stores a set value for the phase angle δ that is analytically or experimentally obtained in advance according to the torque command value T ^*. Good.

The γ voltage calculation circuit 61 receives the voltage commands vdc ^* and vqc ^* and the phase difference δ, and calculates the γ voltage Vγ by the following formula.
Vγ = vdc ^* × cos (δ) + vqc ^* × sin (δ)
The γ voltage estimation circuit 62 calculates the γ voltage estimated value VγH by using the current response values idc and iqc of the dcqc axis rotation coordinate system, the speed estimated value ωest, and the phase difference δ as inputs.

The γ voltage estimation circuit 62 calculates the dcqc axis voltage estimated values VdcH and VqcH by the following formula.

  In the above formula, R is a winding resistance, Ldc is a dc axis inductance, Lqc is a qc axis inductance, φPM is a permanent magnet magnetic flux, and s is a Laplace operator.

The γ voltage estimation circuit 62 calculates the γ voltage estimated value VγH by the following equation using the dcqd axis voltage estimated values VdcH and VqcH obtained by the above equation.
VγH = VdcH × cos (δ) + VqcH × sin (δ)
The subtracter 63 subtracts the estimated γ voltage value VγH from the γ voltage Vγ, and outputs a voltage error ΔVγ. The voltage error ΔVγ is a voltage error in the γ-axis direction shifted from the dc axis by the phase difference δ. When the rotational phase angle error Δθ is zero, the rotational phase angle estimated value θest follows the actual rotational phase angle θ. That is, the rotational phase angle error Δθ is an error included in the estimated rotational phase angle value θest, and the rotational phase angle error Δθ is zero when the voltage error ΔVγ is zero.

  The rotational phase angle error calculation circuit 51 has, for example, mathematical formulas and tables representing the relationship between the voltage error ΔVγ and the rotational phase angle error Δθ in advance, and calculates and outputs the rotational phase angle error Δθ based on the voltage error ΔVγ. be able to.

  The relationship between the rotational phase angle error Δθ and the voltage error ΔVγ differs depending on the inductance characteristics of the synchronous machine 2. For example, when the synchronous machine 2 is a synchronous reluctance motor (SynRM), the relationship between the rotational phase angle error Δθ and the voltage error ΔVγ is such that when the rotational phase angle error Δθ is zero, the voltage error ΔVγ is zero and the rotational phase angle error Δθ is zero. This is a sine wave having a maximum voltage error ΔVγ when the angular error Δθ is ± 90 °. For example, when the synchronous machine 2 is a permanent magnet synchronous motor (PMSM), the relationship between the rotational phase angle error Δθ and the voltage error ΔVγ is such that the voltage error ΔVγ is zero when the rotational phase angle error Δθ is zero, This is a sine wave having a maximum voltage error ΔVγ when the rotational phase angle error Δθ is ± 180 °. The rotation phase angle error calculation circuit 51 is not limited to the above method, and may calculate a value corresponding to the rotation phase angle error Δθ.

The PI controller 52 receives the rotational phase angle error Δθ, performs the PI control so that the rotational phase angle error Δθ becomes zero, and the speed estimation FB (feedback) value ω ^* estFB. Is calculated. The speed estimated FB value ω ^* estFB corresponds to an error of the speed estimated FF value ω ^* estFF with respect to the speed estimated value ωest.

The adder 53 adds the speed estimated FB value ω ^* estFB and the speed estimated FF value ω ^* estFF to calculate the speed estimated value ωest. The integrator 54 integrates the speed estimation value ωest to calculate the rotation angle estimation value θest.

In the inverter control device and the motor drive system, when the speed command ω ^* changes, the approximate speed estimation circuit 29 calculates and outputs a speed estimation FF value ω ^* estFF corresponding thereto. At this time, the proportional gain and the integral gain used in the PI controller 42 are the same as the proportional gain and the integral gain used in the PI controller 32 of the speed control circuit 28, so that the motor machine model circuit 44 is connected to the actual synchronous machine 2. If it matches the response of the mechanical system, the speed estimation FF value ω ^* estFF matches the actual speed. The speed estimation FF value ω ^* estFF is used by the adder 53 of the speed / rotation phase angle estimation circuit 26 to calculate the speed estimation value ωest.

When using the value obtained by feedback when calculating the estimated speed value ωest, the rotational angle error Δθ, which is the input of the PI controller 52, increases when the speed command ω ^* changes. On the other hand, in the present embodiment, when the speed estimation value ωest is calculated, the speed estimation FF value ω ^* estFF works as feedforward, so that the rotational phase angle error Δθ increases even when the speed command ω ^* changes. There is nothing.
Therefore, in this embodiment, even if the gain of the PI controller 52 that receives the rotational phase angle error Δθ is not increased, step-out does not occur and a high-speed speed response can be realized.

  It is difficult for the motor machine model circuit 44 to completely match the actual response of the mechanical system of the synchronous machine 2. However, even if the motor machine model circuit 44 and the response of the mechanical system of the actual synchronous machine 2 do not completely match, it is possible to reduce the error between the speed estimated value ωest and the actual speed ω. In particular, when a high speed response is required, step-out can be prevented.

As described above, according to this embodiment, by using the same gain and the same torque limit as the speed control circuit 28 in the approximate speed estimation circuit 29, the primary delay of the speed command ω ^* and the speed command ω ^* is simply fed. It is possible to estimate with less actual speed ω and error than when adding in the forward direction.

That is, according to the present embodiment, it is possible to provide an inverter control device and an electric motor drive system that realize a high speed response.
In addition, although the example which employ | adopted the synchronous reluctance motor as the synchronous machine 2 was shown in the said embodiment, the same effect is acquired also with the permanent magnet synchronous motor using a reluctance torque.

  Next, modifications of the inverter control device and the electric motor drive system according to the first embodiment will be described. The modification described below is different in the configuration of the approximate speed estimation circuit 29 from the inverter control device of the first embodiment described above.

FIG. 6 is a block diagram schematically showing another configuration example of the schematic speed estimation circuit shown in FIG.
In this example, the approximate speed estimation circuit 29 includes a PI controller 45 with a torque limit instead of the PI controller 42 and the torque limit circuit 43 shown in FIG. Except this point, the configuration is the same as the configuration example shown in FIG.

The PI controller 45 calculates the torque command estimated value T ^* est by PI control so that the speed estimation error Δω ^* est becomes zero.
The PI controller 45 includes a proportional gain multiplier 451, an integral gain multiplier 452, an integrator 453, a limiter 454, an adder 455, and subtractors 456 and 457.

The proportional gain multiplier 451 multiplies the speed estimation error Δω ^* est by the proportional gain P and outputs the result.
The integral gain multiplier 452 multiplies the speed estimation error Δω ^* est by the integral gain I and outputs the result.
The subtractor 457 (first subtractor) calculates and outputs the difference (ΔT ^* est) between the input value and the output value of the limiter 454.
The subtracter (second subtractor) 456 outputs a value obtained by subtracting the calculation result (ΔT ^* est) of the subtractor 457 from the value (I × Δω ^* est) output from the integral gain multiplier 452.

The integrator 453 integrates and outputs the value (I × Δω ^* est−ΔT ^* est) output from the subtractor 456.
The adder 455 adds the output value (P × Δω ^* est) from the proportional gain multiplier 451 and the output value of the integrator 453 (∫ (I × Δω ^* est−ΔT ^* est) dt) and outputs the result. .
The limiter 454 receives the value output from the adder 455 and outputs a torque command estimated value T ^* est that is limited so as not to exceed the output limit of the inverter 1.

  For example, when there is no subtractor 456 in the preceding stage of the integrator 453, the control error caused by limiting the torque command estimated value cannot be made zero, and the integrator 453 continues to add errors. Therefore, as shown in FIG. 6, the difference between the input value and the output value of the limiter 454 is previously subtracted from the input of the integrator 453, thereby preventing the integrator 453 from continuing to add errors. it can.

For example, the motor machine model circuit 44 does not coincide with the actual mechanical system of the synchronous machine 2, and particularly when the actual synchronous machine 2 has a load torque, the estimated speed FF value ω ^* estFF is greater than the actual speed ω. Will become bigger. In the example shown in FIG. 5, when there is a load torque and the torque is limited by the torque limiting circuit 33 of the speed control circuit 28, the error between the actual speed ω and the speed estimated FF value ω ^* estFF may increase. Disappear. As a result, when the limit state changes to the non-limit state (non-limit state), it is possible to avoid destabilizing (breaking down) the control according to the amount of continued error addition.

  Next, another modification of the inverter control device and the motor drive system of the first embodiment will be described. The modification described below is different in the configuration of the motor machine model circuit 44 of the inverter control device of the first embodiment described above.

FIG. 7 is a block diagram schematically showing another configuration example of the motor machine model shown in FIG.
In this example, the motor machine model circuit 44 calculates a speed estimation FF value ω ^* estFF in consideration of friction such as a bearing, and more accurately models the actual mechanical system of the synchronous machine 2.
The motor machine model circuit 44 includes a divider 48, an integrator 49, a subtracter 441, and a multiplier 442.

The divider 48 divides the torque command estimated value T ^* est by the inertia estimated value Jest and outputs the result.
The multiplier 442 outputs the product of the speed estimation FF value ω ^* estFF output from the integrator 49 and the friction coefficient Dest.
The subtractor 441 outputs a difference obtained by subtracting the output value (Dest × ω ^* estFF) of the multiplier 442 from the output value (T ^* est / Jest) of the divider 48.

The integrator 49 integrates the value output from the divider 48 (T ^* est / Jest−Dest × ω ^* estFF), and calculates and outputs the speed estimation FF value ω ^* estFF. As the inertia estimated value Jest, a fixed value adjusted by speed control can be used. That is, the speed estimation FF value ω ^* estFF can be expressed as the following equation.

As described above, by calculating the speed estimation FF value ω ^* estFF in consideration of the friction in the synchronous machine 2, an error between the speed estimation FF value ω ^* estFF and the actual speed value can be reduced.
That is, according to the above-described modification, it is possible to provide an inverter control device and an electric motor drive system that can achieve the same effects as those of the first embodiment and perform more reliable sensorless control. It is. Furthermore, if the characteristics of the load coupled to the motor are known, the characteristics can be further reduced from the value output from the divider 48 (T ^* est / Jest−Dest × ω ^* estFF), so that a more accurate speed FF Value can be calculated and speed response can be improved. For example, when a fan pump is joined to the motor, it can be regarded as a load that changes with the third power (ω ³ ) of the rotational speed ω of the motor, and a value considering this characteristic is subtracted from the output value of the divider 48. By setting the value, a more accurate speed FF value can be calculated and the speed response can be improved.

  Next, an inverter control device and an electric motor drive system according to the second embodiment will be described in detail with reference to the drawings. In the following description, the same components as those in the first embodiment described above are denoted by the same reference numerals and description thereof is omitted.

FIG. 8 is a block diagram schematically illustrating a configuration example of the inverter control device and the motor drive system according to the second embodiment.
The electric motor drive system of this embodiment includes an inverter control device, an inverter 1 and an induction motor 4.

The induction motor 4 is an AC motor that includes, for example, a stator and a rotor and is driven by a three-phase AC current supplied from the inverter 1.
The inverter control device of this embodiment includes a current detector 3, a PWM modulation circuit 22, a coordinate conversion (UVW / dcqc) circuit 23, a coordinate conversion (dcqc / UVW) circuit 24, a current control circuit 25, a speed An estimation circuit 102, a current command generation circuit 27, a speed control circuit 28, an approximate speed estimation circuit 29, a slip frequency calculation circuit 101, a speed estimation circuit 102, an adder 103, and an integrator 104 are provided. ing.

The slip frequency calculation circuit 101 calculates a slip frequency command ωs ^* from the current commands idc ^* and iqc ^* . For example, it calculates with the following formula.
Here, R ₂ represents a secondary resistance, and L ₂ represents a secondary self-inductance.

Speed estimation circuit 102 receives current command iqc ^* , current iq, and speed estimation FF value ω ^* estFF, and outputs speed estimation value ωest.

FIG. 9 is a block diagram schematically showing a configuration example of the speed estimation circuit shown in FIG.
The speed estimation circuit 102 includes a subtracter 71, a PI controller 72, and an adder 73.
The subtractor 71 subtracts the current iq from the current command iq ^* to calculate and output a q-axis current deviation Δiq.
The PI controller 72 calculates and outputs the speed estimated FB value ω ^* estFB so that the q-axis current deviation Δiq becomes zero.

The adder 73 calculates the speed estimated value ωest by adding the speed estimated FF value ω ^* estFF and the speed estimated FB value ω ^* estFB, and outputs the result.
The adder 103 adds the estimated speed value ωest the slip frequency command .omega.s ^* calculating and outputting a primary frequency instruction .omega.1 ^*.
The integrator 104 integrates the primary frequency command ω1 ^* to calculate and output the d-axis phase θ. The d-axis phase θ output from the integrator 104 is input to the coordinate conversion (UVW / dcqc) circuit 23 and the coordinate conversion (dcqc / UVW) circuit 24, and is used for vector conversion.

Also in the present embodiment, as in the first embodiment described above, in the approximate speed estimation circuit 29, when the speed command ω ^* changes, the speed estimation FF value ω ^* estFF is calculated accordingly. At this time, since the same gain and torque limit as the speed control circuit 28 are used, if the motor machine model circuit 44 matches the actual response of the mechanical system of the induction motor 4, the speed estimation FF value ω ^* estFF is It matches the actual speed. The speed estimation FF value ω ^* estFF is used for calculation of the speed estimation value ωest in the adder 73 of the speed estimation circuit 102.

When the speed command omega ^* is changed, usually the q-axis current deviation Δiq is the input of the PI controller 72 is increased, the speed estimated FF value ω ^* estFF is that serve as a feedforward velocity command omega ^* is The q-axis current deviation Δiq does not change in the same way as when it has not changed. For this reason, the step-out does not occur even if the gain of the PI controller 72 is not increased, and a high speed response can be realized.

For example, in the conventionally proposed method, a value corresponding to the speed estimation FF value ω ^* estFF using the q-axis secondary magnetic flux induced voltage is calculated. However, in this method, the estimation is correctly performed when the speed ω is small. Difficult to do. On the other hand, in the inverter control device and the motor drive system of the present embodiment, feedforward can be added regardless of the speed, and a high speed response can be realized.
That is, according to this embodiment, the same effect as that of the first embodiment described above can be obtained.

Note that the approximate speed estimation circuit 29 described in the first embodiment may be applied to the inverter control device and the motor drive system of the present embodiment.
In the present embodiment, an example of an inverter control device and an electric motor drive system including the induction motor 4 is shown. However, in the sensorless control of PMSM that does not use reluctance torque, a high speed response in a low speed range is realized in the same manner. can do.

  Although several embodiments of the present invention have been described, these embodiments are presented by way of example and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.

  DESCRIPTION OF SYMBOLS 1 ... Inverter, 2 ... Synchronous machine, 3 ... Current detector, 4 ... Induction motor, 22 ... PWM modulation circuit, 23 ... Coordinate conversion circuit, 24 ... Coordinate conversion circuit, 25 ... Current control circuit, 26 ... Speed and rotation phase Angle estimation circuit, 27 ... Current command generation circuit, 28 ... Speed control circuit, 29 ... General speed estimation circuit, 31 ... Subtractor, 32 ... PI controller (first PI controller), 33 ... Torque limiting circuit (first torque) Limit circuit), 41 ... subtractor, 42 ... PI controller (second PI controller), 43 ... torque limit circuit (second torque limit circuit), 44 ... motor machine model circuit, 45 ... PI controller with torque limit, 48 ... divider, 49 ... integrator, 51 ... rotational phase angle error calculation circuit, 52 ... PI controller, 53 ... adder, 54 ... integrator, 60 ... phase difference δ setting circuit, 61 ... γ voltage calculation circuit, 62 ... γ voltage estimation circuit, 63 ... Arithmetic unit 71 ... Subtractor 72 ... PI controller 73 ... Adder 101 ... Frequency arithmetic circuit 102 ... Speed estimation circuit 103 ... Adder 104 ... Integrator 441 ... Subtractor 442 ... Multiplier 451 ... Proportional gain multiplier, 452 ... Integral gain multiplier, 453 ... Integrator, 454 ... Limiter, 455 ... Adder, 456 ... Subtractor, 457 ... Subtractor.

Claims (5)

An inverter;
A current detector for detecting a current response value output from the inverter;
A speed control circuit including a first PI controller that calculates a torque command value of the electric motor so that the rotational speed of the electric motor connected to the inverter is equal to the rotational speed command value;
A current command value generation circuit for calculating a current command value corresponding to the current response value using the torque command value;
A current control circuit for calculating an output voltage output from the inverter so that the current command value and the current response value are equal;
A second PI controller that inputs a difference between the rotational speed command value and the approximate rotational speed value and outputs a torque command estimated value; a motor machine model circuit that calculates the approximate rotational speed value using the estimated torque command value; A gain used in the second PI controller is equivalent to a gain used in the first PI controller;
An inverter control device comprising: a speed estimation circuit that calculates an estimated speed value of the electric motor using the approximate rotational speed value. The speed control circuit includes a first torque limiting circuit that limits the torque command value;
2. The inverter control device according to claim 1, wherein the approximate speed estimation circuit includes a second torque limiting circuit that limits the torque command estimated value with the same value as the first torque limiting circuit. The speed control circuit includes a torque limiting circuit that limits the torque command value;
The second PI controller outputs a product obtained by multiplying the difference between the rotational speed command value and the rotational speed approximate value by a proportional gain, and a limiter that limits the torque command estimated value by the same value as the torque limiting circuit. A proportional gain multiplier; an integral gain multiplier that outputs a product obtained by multiplying a difference between the rotational speed command value and the approximate rotational speed value by an integral gain; and the torque command estimated value before being limited by the limiter; A first subtractor that outputs a difference from the torque command estimated value after being limited by the limiter; and a first subtracter that outputs a difference obtained by subtracting the output value of the first subtractor from the output value of the integral gain multiplier. 2 subtractors, an integrator that integrates and outputs the output value of the second subtractor, and an addition that supplies the sum obtained by adding the output value of the proportional gain multiplier and the output value of the integrator to the limiter With a vessel Inverter control device according to claim 1, wherein. An inverter control device according to any one of claims 1 to 3,
An electric motor drive system comprising the electric motor.   5. The electric motor drive system according to claim 4, wherein the electric motor is a synchronous reluctance motor.