JP4380437B2 - Control device and module for permanent magnet synchronous motor - Google Patents

Description

The present invention relates to a control device and a module for a permanent magnet synchronous motor , and more particularly to a control device and a module for a permanent magnet synchronous motor suitable for vector control of a field weakening region .

  As a conventional technique of the field control method of the field weakening region, there is a method of converting the d-axis current command value described in JP-A-8-182398 into a table and making the d-axis and q-axis current control a proportional calculation method, As described in JP-A-2002-95300, a terminal voltage of an electric motor is obtained from a d-axis and q-axis current control unit, and a deviation between the command value of the terminal voltage and the terminal voltage is calculated by proportional / integral calculation. There is a method for calculating the current command value.

JP-A-8-182398 JP 2002-95300 A

However, in the method described in JP-A-8-182398, since current control is a proportional calculation method, current according to the current command value is not generated, and torque accuracy is deteriorated.
In the method described in Japanese Patent No. 95300, the torque response tends to deteriorate because the d-axis current command is generated slowly.

The present invention has been made in view of the above points, and an object, even in the field weakening磁域, to provide a control device and a module for a permanent magnet synchronous motor that can achieve high precision and high response is there.

In order to achieve the above object, the controller for a permanent magnet synchronous motor according to the present invention provides a second d-axis and q-axis current calculated from the first d-axis and q-axis current command values and the detected current value. A control device for a permanent magnet synchronous motor that controls an output voltage value of a power converter that drives a permanent magnet synchronous motor according to a command value and a frequency command value, wherein the output voltage command value in a field weakening region and the output voltage A field weakening command calculation unit that uses the integral calculation value of the deviation from the value as the first d-axis current command value, and integral calculation of the deviation between the output voltage command value and the output voltage value in the field weakening region Is characterized in that the integral gain is corrected according to the frequency command value.

In order to achieve the above object, the module of the present invention includes the control device for the permanent magnet synchronous motor and a power converter that converts direct current into alternating current .

ADVANTAGE OF THE INVENTION According to this invention, the control apparatus and module of a permanent-magnet synchronous motor which can implement | achieve a highly accurate and highly responsive motor torque can be obtained even in a field- weakening region .

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

FIG. 1 shows an example of the configuration of a field-weakening vector control device for a permanent-magnet electric motor that is an embodiment of the present invention. 1 is a permanent magnet synchronous motor, 2 is a power converter that outputs a voltage proportional to the voltage command values Vu ^* , Vv ^* , and Vw ^* of a three-phase AC, 21 is a DC power source, 3 is a three-phase AC current Iu, Iv, A current detector capable of detecting Iw, 4 a magnetic pole position detector capable of detecting a position detection value θi for each electric angle of 60 ° of the motor, and 5 a frequency calculation unit for calculating a frequency command value ω ₁ ^* from the position detection value θi, 6 is a phase calculation unit for calculating the rotational phase command θc ^* of the motor from the position detection value θi and the frequency command value ω ₁ ^* , and 7 is the rotation of the detected values Iuc, Ivc, Iwc of the three-phase alternating currents Iu, Iv, Iw. A coordinate conversion unit for outputting the detected current values Idc and Iqc of the d-axis and the q-axis from the phase command θc ^* , and 8 is the _first difference from the deviation between the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ ^* field weakening command operation for calculating a d-axis current command value Id ^* , 9 outputs a second d-axis current command value Id ^** according to a difference of the first d-axis current command value Id ^* and the d-axis current detection value Idc, which is the output of the field weakening command calculating section d An axis current command calculation unit 10 outputs a second q axis current command value Iq ^** in accordance with the deviation between the first q axis current command value Iq ^* and the q axis current detection value Iqc. , 11 is a voltage vector calculation unit for calculating the voltage command values Vd ^* , Vq ^* based on the electric constant of the electric motor 1, the second current command values Id ^** , Iq ^**, and the frequency command value ω ₁ ^* , An output voltage calculation unit for calculating the output voltage value V ₁ ^* of the power converter from the voltage command values Vd ^* and Vq ^* , and 13 is a three-phase AC voltage command from the voltage command values Vd ^* and Vq ^* and the rotation phase command θc ^*. This is a coordinate conversion unit that outputs values Vu ^* , Vv ^* , and Vw ^* .

  First, the basic operation of the voltage control and phase control of the vector control method when the field-weakening command calculation, which is a feature of the present invention, is used will be described.

In the voltage control, the output voltage calculation unit 12 in FIG. 1 calculates the output voltage value V ₁ ^* using the d-axis and q-axis voltage command values Vd ^* and Vq ^* as shown in Equation ₁ ^. .

In field weakening command calculation unit 8, the output voltage value V ₁ ^* is, so to match the output voltage command value V ₁ ^* _ref at field weakening磁域, calculating a first d-axis current command value Id ^* To do.

In addition, the voltage vector calculation unit 11 calculates the d-axis and q-axis voltage command values Vd ^* and Vq ^* using the second d-axis and q-axis current command values and motor constants shown in advance in Expression 2 ^. Control the converter output voltage.

Here, R ₁ ^* is a resistance setting value, Ld ^* is a d-axis inductance setting value, Lq ^* is a q-axis inductance setting value, and Ke ^* is an induced voltage constant setting value.

  On the other hand, in the phase control, the magnetic pole position detector 4 can grasp the magnetic pole position at every electrical angle of 60 degrees. In this embodiment, the position detection value θi at this time is

  Here, i = 0, 1, 2, 3, 4, and 5.

The frequency calculation unit 5 calculates an average rotation frequency ω ₁ ^* (hereinafter referred to as a frequency command value) in the shortest 60 ° section from the position detection value θi.

  Here, [Delta] [theta] = [theta] i- [theta] (i-1), and [Delta] t is the time until the position detection signal in the 60-degree section is detected.

Further, the phase calculation unit 6 controls the reference phase of the electric motor 1 by calculating the rotational phase command θc ^* as shown in Equation 5 using the position detection value θi and the frequency command ω ₁ ^* .

  The above is the basic operation of voltage control and phase control.

  Next, the field-weakening command calculation unit 8 based on the feedback control method, which is a feature of the present invention, will be described with reference to FIG.

In the field weakening command calculation unit 8, the deviation between the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ ^{* in} the field weakening field is input to the integration calculation unit 81 having a constant integral gain K, and the integral calculation is performed. Done. The calculated value is input to a limiter calculating unit 82 that limits the positive side to “zero”, and the output value becomes the first d-axis current command Id ^* .

  Next, the effect which this invention brings about is demonstrated by a present Example.

Consider a case in which the first d-axis current command value Id ^* is controlled to “zero” in the control device of FIG. 1 (when field-weakening command calculation is not performed).

V ₁ ^* output from the voltage vector calculation unit 11 is obtained by substituting Equation 2 into Equation 1.

Also, when the saturation value of V ₁ ^* and V ₁ ^* _max, the relation of the number 7 in the voltage saturation range.

Here, by rearranging Equation 7, a quadratic equation for the frequency command ω ₁ ^* can be obtained,

数８より、Ｖ₁ ^*が飽和する際のω₁ ^*を求めることができる。

From Equation 8, ω ₁ ^* when V ₁ ^* is saturated can be obtained.

Here, FIG. 3 shows the relationship between the motor torque τ and the frequency command ω ₁ ^* when Id ^** = Id ^* = 0 and Iq ^** = τ / KT.

  Here, τ is a motor torque, and KT is a torque coefficient.

The solid line shown in FIG. 3 is a boundary line where V ₁ ^* is saturated. The upper side of the boundary line is a saturated region, and the lower side is a non-saturated region, which is a range where actual operation is possible.

For this reason, in the vector control in which the d-axis current command value Id ^* is set to “zero”, there is a problem that the operation range in the high speed range is limited to be low.

In this embodiment, the output voltage value V ₁ ^* is, so to match the output voltage command value V ₁ ^* _ref at weakening磁域calculates a first d-axis current command value Id ^*, the Id ^{* Is used} to create the second d-axis current command value Id ^** and to calculate the voltage vector.

Here, the output voltage command value V ₁ ^* _{ref in the} field weakening region is set as shown in Equation 10.

As a result, the voltage command values Vd ^* and Vq ^* are calculated by the voltage vector calculation unit 11 so that the output voltage value V ₁ ^* is not saturated (is a value smaller than V ₁ ^* _max ). The operating range can be expanded.

  When the present invention is applied, current can be generated according to the current command value, so that highly accurate torque control can be realized, and the operating range can be expanded as shown in FIG.

  When high torque is required during torque control operation, it is necessary to flow a large current commensurate with the torque. When high torque is required for a continuous time, the winding resistance value R inside the motor increases with time due to heat generated by the motor current. Then, since the resistance set value calculated by the voltage vector calculation unit and the actual resistance value do not match, the voltage required for the motor cannot be supplied. As a result, the current necessary for generating torque does not flow and the torque is insufficient. There is concern about falling into

  Therefore, as shown in FIG. 1 of the present embodiment, by having the current command calculation unit upstream of the vector calculation unit, the output voltage is controlled so that the motor current matches the current command value, the fluctuation of the motor constant, It is possible to provide a control apparatus for an AC motor that does not cause torque shortage from a low speed range without being affected by mounting errors such as Hall elements.

  FIG. 5 shows another embodiment of the present invention.

The present embodiment is a control device for a permanent magnet same-machine electric motor of a system in which the integral gain of the field weakening command calculation unit by the feedback control system is changed by the frequency command ω ₁ ^* .

In FIG. 5, 1-7, 9-13, and 21 are the same as FIG. Reference numeral 8a denotes a field weakening command calculation unit that automatically corrects an integral gain when the difference between V ₁ ^* _ref and V ₁ ^* is integrated according to the frequency command ω ₁ ^* .

  Next, the field weakening command calculation unit 8a, which is a feature of the present invention, will be described with reference to FIG.

In field weakening command calculation unit 8a, the deviation of the output voltage command value V ₁ ^* _ref and output voltage V ₁ in the field weakening磁域is input to the integration unit 8a1 with constant integral gain K, the integral calculation is Done. At that time, the integral gain K is automatically corrected by the frequency ω ₁ . The output value of the integral calculation unit 8a1 is input to the limiter calculation unit 8a2 that limits the positive side to “zero”, and the output value becomes the first d-axis current command Id ^* .

Using this current command value Id ^* , a second current command value Id ^** is created, and voltage command values Vd ^* and Vq ^* are calculated to control the converter output voltage.

  Here, the effect which this invention brings about is demonstrated.

When the integral gain K used in the field weakening command calculation is constant, the closed loop transfer function G _φ (s) from V ₁ ^* _ref to Id ^* at no load (Iq ^* = 0) is

Here, s is a Laplace operator. From Equation 11, it can be seen that Id ^* occurs with a first-order lag, its response time constant _Tφ is _Equation 12, and _Tφ changes with the frequency command ω ₁ ^* .

  Therefore, the integral gain K of 8a1 is calculated as shown in Equation 13.

Here, ω _c is the control response angular frequency (rad / s) of the field weakening command calculation. Then, the new transfer function G _φ ′ (s) is

となる。ここで、新しい応答時定数Ｔ_φ′は、

It becomes. Here, the new response time constant T _φ ′ is

である。これより、Ｔ_φ′は周波数指令ω₁ ^*と無関係に設定することができ、より高応答の効果を得ることができる。

It is. Thus, T _φ ′ can be set regardless of the frequency command ω ₁ ^*, and a higher response effect can be obtained.

In addition, the control device for the permanent magnet same-machine electric motor of the method of changing the integral gain of the field weakening command calculation unit by the feedback command method of the present embodiment with the frequency command ω ₁ ^* is shown in FIG. The present invention can also be applied to a control system other than a control system having a current command calculation unit in the upstream portion.

  FIG. 7 shows another embodiment of the present invention. The present embodiment is a field-weakening vector control device for a permanent-magnet same-machine electric motor when a feedforward method is used for a field-weakening command calculation unit.

  In FIG. 7, components 1 to 7, 9 to 13, and 21 are the same as those in FIG.

  With reference to FIG. 8, the field weakening command calculation unit 8b based on the feedforward control method, which is a feature of the present invention, will be described.

  This actual exception is obtained in advance by a calculation of a d-axis current command generated when there is no load.

In the field weakening command calculation unit 8b, the calculation unit 8b1 subtracts the induced voltage command value (= ω ₁ ^* Ke ^* ) from the output voltage command value V ₁ ^* _ref in the field weakening region, and the subtraction value is represented by ω _1. Division operation is performed by the multiplication value of ^* and Ld ^* . The output value of the arithmetic unit 8b1 is input to the first-order lag filter 8b2. Further, the output value of 8b2 is input to a limiter calculation unit 8b2 that limits the positive side to “zero”, and the output value becomes the first d-axis current command Id ^* .

Using this current command value Id ^* , a second current command value Id ^** is created, and voltage command values Vd ^* and Vq ^* are calculated to control the converter output voltage.

In the high speed region, even if the torque is “zero”, V ₁ ^* is saturated only by the induced voltage command value (= ω ₁ ^* Ke ^* ) of Vq ^* .

If the d-axis current command value necessary to get out of the voltage saturation region is Id ^* _ff0 ,

これにより、８ｂ２の一次遅れフィルタ時定数Ｔを、数（１７）のように設定することで、フィードフォワード制御方式でも、実施例２と同様な効果を得ることができる。

Thus, by setting the first-order lag filter time constant T of 8b2 as shown in the equation (17), the same effect as that of the second embodiment can be obtained even in the feedforward control method.

  Note that the field weakening vector control device for the permanent magnet same-machine electric motor when the feed-forward method is used for the field weakening command calculation unit of the present embodiment, the current command to the upstream of the voltage vector calculation unit as shown in FIG. The present invention can also be applied to a control system other than a control system having an arithmetic unit.

  FIG. 9 shows another embodiment of the present invention. The present embodiment is a control device for a permanent magnet same-machine electric motor when a feedforward control method and a feedback control method are used for a field weakening command calculation unit.

In FIG. 9, components 1 to 7, 9 to 13, and 21 are the same as those in FIG. The field weakening command calculation unit 8c using the feedforward control method and the feedback control method, which are features of the present invention, will be described with reference to FIG.

In the field weakening command calculation unit 8c, the calculation unit 8c1 subtracts the induced voltage command value (= ω ₁ ^* Ke ^* ) from the output voltage command value V ₁ ^* _ref in the field weakening region, and the subtraction value is represented by ω _1. ^A division operation is performed with the product of ^* and Ld ^* .

The output value of the calculation unit 8c1 is input to the first-order lag filter 8c2. Further, the output value of 8c2 is input to a limiter calculation unit 8c3 that limits the positive side to “zero”, and the output value becomes Id ^* _ff .

At the same time, the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ are input to the integration calculation unit 8c4 having a constant integral gain K, and the integration calculation is performed. At that time, the integral gain K is automatically corrected by the frequency ω ₁ .

The output value of the integral calculation unit 8c4 is input to the limiter calculation unit 8c5 that limits the positive side to “zero”, and the output value becomes Id ^* _fb .

Therefore, as shown in equation (18), the first d-axis current command Id ^* is calculated from the added value of the feedforward control output value Id ^* _ff and the feedback control output value Id ^* _fb .

  This method also operates in the same manner as in the above-described embodiment and can obtain a higher response effect.

  Similarly, the control device for the permanent magnet same-machine motor when the feedforward control method and the feedback control method are used for the field weakening command calculation unit of the present embodiment is an upstream portion of the voltage vector calculation unit as shown in FIG. The present invention is also applicable to a control system other than a control system having a current command calculation unit.

  In the first to fourth embodiments, the three-phase AC currents Iu to Iw detected by the expensive current detector 3 are detected. However, the present invention can also be applied to a control device that performs inexpensive current detection. it can.

  FIG. 11 shows this embodiment. In FIG. 11, constituent elements 1, 2, 4 to 7, 8a, 9 to 13, and 21 are the same as those in FIG.

  Reference numeral 14 denotes a current estimation unit that estimates the three-phase AC currents Iu, Iv, and Iw that flow through the electric motor 1 from the DC current IDC that flows through the input bus of the power converter.

  Using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate conversion unit 7 calculates current detection values Idc, Iqc for the d-axis and the q-axis.

Even in such a current sensorless control method, since Id ^* and Idc, and Iq ^* and Iqc coincide with each other, it is apparent that the operation is the same as in the above-described embodiment and the same effect is obtained.

  Further, in this embodiment, the method of FIG. 6 is used for the field weakening command calculation unit, but the same effect can be obtained even if the methods of FIGS. 2, 8, and 10 are used.

  FIG. 12 shows another embodiment of the present invention.

  This embodiment is applied to a control device that performs inexpensive current detection and omits the magnetic pole position detector.

  In FIG. 12, constituent elements 1, 2, 7, 8a, 9 to 13, 21 are the same as those in FIG.

Reference numeral 6 'denotes a phase calculation unit that integrates the frequency command ω ₁ ^* to calculate the rotation phase command θc ^* .

  Reference numeral 14 denotes a current estimation unit that estimates three-phase AC currents Iu, Iv, and Iw that flow through the synchronous motor from a DC current IDC that flows through the input bus of the power converter.

  Using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate conversion unit 7 calculates current detection values Idc, Iqc for the d-axis and the q-axis.

Reference numeral 15 denotes a phase error Δθc (= θc ^* −θ) which is a deviation between the rotational phase command θc ^* and the rotational phase θ of the electric motor 1 based on the voltage command values Vd ^* and Vq ^* and the current detection values Idc and Iqc. It is a phase error calculation part to estimate.

Reference numeral 16 denotes a frequency estimation unit that calculates ω ₁ ^** so that the phase error Δθc is set to “zero”. It is clear that even such a position and current sensorless control system operates in the same manner as in the above-described embodiment, and the same effect can be obtained.

  Further, in this embodiment, the method of FIG. 6 is used for the field weakening command calculation unit, but the same effect can be obtained even if the methods of FIGS. 2, 8, and 10 are used.

  An example in which the present invention is applied to a module will be described with reference to FIG. The present example shows an embodiment of the first example. Here, the frequency calculation unit 5, the phase calculation unit 6, the coordinate conversion unit 7, the field weakening command calculation unit 8, the d-axis current command calculation unit 9, the q-axis current command calculation unit 10, the voltage vector calculation unit 11, the output voltage The calculation unit 12 and the coordinate conversion unit 13 are configured using a one-chip microcomputer. Further, the one-chip microcomputer and the power converter are housed in one module configured on the same substrate. The module here means “standardized structural unit” and is composed of separable hardware / software components. In addition, although it is preferable to comprise on the same board | substrate on manufacture, it is not limited to the same board | substrate. Thus, the circuit board may be configured on a plurality of circuit boards built in the same housing. In other embodiments, the same configuration can be adopted.

  As described above, according to the present invention, a highly accurate and highly responsive motor torque can be realized even in a weak field region, and an inexpensive current detection system and a magnetic pole position detector are omitted. Even in such a system, it is possible to provide a field-weakening vector control device for a permanent magnet synchronous motor that can be applied in common.

The block diagram of the field-weakening vector control apparatus of the permanent-magnet synchronous motor which shows one Example of this invention. Explanatory drawing of the field weakening command calculating part 8 in the control apparatus of FIG. An example of a voltage saturation characteristic diagram when the field weakening command calculation unit 8 is not provided. An example of a voltage saturation characteristic diagram when the field weakening command calculation unit 8 is inserted. The block diagram of the field-weakening vector control apparatus of the permanent-magnet synchronous motor which shows the other Example of this invention. FIG. 6 is an example of an explanatory diagram of a field weakening command calculation unit 8a in the control device of FIG. 5. The block diagram of the field-weakening vector control apparatus of the permanent-magnet synchronous motor which shows the other Example of this invention. FIG. 8 is an example of an explanatory diagram of a field weakening command calculation unit 8b in the control device of FIG. The block diagram of the field-weakening vector control apparatus of the permanent-magnet synchronous motor which shows the other Example of this invention. FIG. 10 is an example of an explanatory diagram of a field weakening command calculation unit 8c in the control device of FIG. 9; The block diagram of the field-weakening vector control apparatus of the permanent-magnet synchronous motor which shows the other Example of this invention. The block diagram of the field-weakening vector control apparatus of the permanent-magnet synchronous motor which shows the other Example of this invention. The block diagram at the time of applying the Example of this invention to a module.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Permanent magnet synchronous motor, 2 ... Power converter, 3 ... Current detector, 4 ... Magnetic pole position detector, 5 ... Frequency calculating part, 6 ... Phase calculating part, 7, 13 ... Coordinate converting part, 8, 8a, 8b, 8c ... field weakening command calculation unit, 9 ... d-axis current command calculation unit, 10 ... q-axis current command calculation unit, 11 ... voltage vector calculation unit, 12 ... output voltage calculation unit, 14 ... current estimation unit, 15 ... Phase error calculation unit, 21 ... DC power supply, IDC ... Input DC bus current detection value, Id ^* ... First d-axis current command value, Id ^** ... Second d-axis current command value, Iq ^* ... First the q-axis current command value, Iq ^** ... second q-axis current command value, V ₁ ^* _ref ... field weakening output voltage command value at磁域, V ₁ ^* ... output voltage value, .theta.c ^* ... rotational phase command, ω ₁ ^* … Frequency command.

Claims (7)

In accordance with the second d-axis and q-axis current command values calculated from the first d-axis and q-axis current command values and the detected current value, and the frequency command value, the power converter for driving the permanent magnet synchronous motor A control device for a permanent magnet synchronous motor for controlling an output voltage value,
Weakening the output voltage command value磁域 and the integral calculation value of the difference between the output voltage value, it comprises a field weakening command calculation unit to the first d-axis current command value,
The control device for a permanent magnet synchronous motor , wherein the integral calculation of the deviation between the output voltage command value and the output voltage value in the field weakening region corrects the integral gain according to the frequency command value . A control device for a permanent magnet synchronous motor that controls an output voltage value of a power converter that drives a permanent magnet synchronous motor according to a current command value and a frequency command value of a d-axis and a q-axis,
A field weakening command calculation unit having an integral calculation value of a deviation between the output voltage command value in the field weakening field and the output voltage value as a d-axis current command value;
The control device for a permanent magnet synchronous motor , wherein the integral calculation of the deviation between the output voltage command value and the output voltage value in the field weakening region corrects the integral gain according to the frequency command value . A control device for a permanent magnet synchronous motor that controls an output voltage value of a power converter that drives a permanent magnet synchronous motor according to a current command value and a frequency command value of a d-axis and a q-axis,
The d-axis current command value is calculated from the output voltage command value in the field weakening region, the frequency command value, and the motor constant,
Addition of the integral calculation value of the deviation between the output voltage command value and the output voltage value of the field weakening region and the value calculated by the output voltage command value of the field weakening region, the frequency command value and the motor constant A control device for a permanent magnet synchronous motor , wherein the value is a d-axis current command value . In claim 3,
The control device for a permanent magnet synchronous motor , wherein the integral calculation of the deviation between the output voltage command value and the output voltage value in the field weakening region corrects the integral gain according to the frequency command value . In claim 3 ,
In the calculation based on the output voltage command value and frequency command value of the field weakening region and the motor constant, the induced voltage command value of the motor is subtracted from the output voltage command value of the field weakening region, and the subtraction value is used as the frequency command value. A control device for a permanent magnet synchronous motor , wherein a division operation is performed by a multiplication value of d-axis inductance . In claim 1 ,
The current detection value reproduces the motor current from the input DC bus current detection value of the power converter.
A control device for a permanent magnet synchronous motor, wherein A module comprising: the permanent magnet synchronous motor control device according to any one of claims 1 to 6; and a power converter that converts direct current into alternating current .

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