EP2345104B1 - Differenz-dipolantennensystem mit einer koplanaren strahlungsstruktur und sender-/empfängereinrichtung - Google Patents

Differenz-dipolantennensystem mit einer koplanaren strahlungsstruktur und sender-/empfängereinrichtung Download PDF

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EP2345104B1
EP2345104B1 EP08875626A EP08875626A EP2345104B1 EP 2345104 B1 EP2345104 B1 EP 2345104B1 EP 08875626 A EP08875626 A EP 08875626A EP 08875626 A EP08875626 A EP 08875626A EP 2345104 B1 EP2345104 B1 EP 2345104B1
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Prior art keywords
differential
dipole
antenna
strip
antenna system
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French (fr)
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EP2345104A1 (de
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Raffi Bourtoutian
Christophe Delaveaud
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Commissariat a lEnergie Atomique et aux Energies Alternatives CEA
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Commissariat a lEnergie Atomique CEA
Commissariat a lEnergie Atomique et aux Energies Alternatives CEA
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/203—Strip line filters
    • H01P1/2039—Galvanic coupling between Input/Output
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/16—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole
    • H01Q9/28—Conical, cylindrical, cage, strip, gauze, or like elements having an extended radiating surface; Elements comprising two conical surfaces having collinear axes and adjacent apices and fed by two-conductor transmission lines
    • H01Q9/285—Planar dipole

Definitions

  • the present invention relates to a differential dipole antenna system adapted for high bandwidth differential signal transmission / reception applications. It also relates to a corresponding transmission and / or reception device.
  • Radio frequency transmit / receive systems powered by differential electrical signals are very attractive for current and future wireless communications systems, especially for autonomous communicating object concepts.
  • a differential supply is a supply of two signals of equal amplitude in phase opposition. It helps reduce, or even eliminate, unwanted "common mode” noise in transmit and receive systems.
  • a non-differential power supply causes the radiation of an undesired cross component due to the common mode flowing on the non-symmetrical power cables.
  • the use of a differential power supply eliminates the cross-radiation of the measurement cables and thus makes it possible to obtain reproducible measurements independent of the measurement context as well as perfectly symmetrical radiation diagrams.
  • the "push-pull" power amplifiers whose structure is differential have several advantages, such as the doubling of the output power and the elimination of higher order harmonics.
  • the low noise differential amplifiers offer several perspectives in terms of reduction of the noise factor.
  • the use of a differential structure prevents unwanted triggering of the oscillators by common mode noise.
  • US 2005/162240 A1 discloses a differential feed antenna including a bandpass filter.
  • the electric dipole antenna is the most naturally conceivable differential antenna. It is an antenna constituted by two identical and symmetrical arms, powered by two signals of equal amplitude and in phase opposition. Recently, the thick dipoles known for their large bandwidths have been fully utilized for high-speed communications, in accordance with the various UWB (Ultra Wide Band) communication standards aimed at broad bandwidth communications. When used in non-symmetrical devices these antennas show problems of common-mode noise whose differential supply makes it possible to overcome.
  • UWB Ultra Wide Band
  • these antennas are also advantageously made in coplanar technology, especially in differential CPS (CoPlanar Stripline) technology.
  • differential CPS CoPlanar Stripline
  • the differential CPS technology makes it possible to take advantage of the differential structures while allowing a simple coplanar integration with discrete elements: it is not necessary to create via-type connections to connect the elements to each other.
  • the absence of a ground plane also makes it possible to envisage a simple and less disturbing connection with other coplanar differential elements. As a result, more and more differential devices are designed using this technology.
  • the invention therefore relates more precisely to an antenna comprising, on one and the same face of a dielectric substrate, a first half of a thick radiating dipole, a first conductive strip of a bi-band supplying a differential signal, this first band conductor being connected to the first thick radiating dipole half, a second thick radiating dipole half and a second conductive strip of the bi-ribbon supply line, this second conductive strip being connected to the second thick radiating dipole half.
  • Such a differential dipole antenna is for example described in the document "Differential and single ended elliptical antennas for 3.1-10.6 GHz ultra wideband communication", by Powell et al, IEEE Antennas and Propagation Society International Symposium Proceedings, vol. 3, pp. 2935-2938 (2004) ).
  • the thick dipole comprises two radiating halves of elliptical shape fed by a differential bi-ribbon line. It provides operation in a frequency range from 3.1 to 10.6 GHz for UWB type applications.
  • the standard WiMedia UWB allocates a bandwidth of between 4.2 and 4.8 GHz in Europe, to ensure compatibility with standards Americans.
  • the thick dipole comprises two half-disc shaped radiating halves fed by two conductive strips of a differential bi-band line.
  • the term "thick dipole” any dipole whose radiating halves occupy a compact geometric surface, such as a polygon (in particular a triangle), an ellipse, a disk, a half ellipse or a half disk.
  • a dipole antenna is thick and slow transition field lines between his arms, the more it has a significant bandwidth.
  • Several geometric shapes make it possible to reach more or less important bandwidths.
  • a "butterfly" type antenna whose arms are triangular in shape, has a relative bandwidth, defined by the relation ⁇ f / f 0 where ⁇ f is the bandwidth and f 0 is the central operating frequency of the bandwidth. the antenna, of the order of 20%.
  • An elliptical antenna may, in some cases, have a relative bandwidth exceeding 100%.
  • the aforementioned antennas are fairly compact and wide bandwidth but they generally have the dimension of a half wave at the low frequency of operation, or 30 to 40 mm at 4 GHz. In many applications where a very strong miniaturization is required, however, they remain too large. In particular, commonly used applications are those using USB wireless communication protocols, on very small USB cards for which the aforementioned dimensions are not suitable.
  • an antenna must generally be connected to a bandpass filtering device.
  • an antenna is a device that emits and receives electromagnetic power.
  • a bandpass filter is then used to limit the frequency band in which the antenna will emit or receive electromagnetic signals. This makes it possible to reduce the noise picked up out of band and to prevent the interference of the signals transmitted or received by the antenna with the signals emitted by other communication systems operating on other sometimes adjacent frequency bands.
  • the European patent application published under the number EP 1 548 872 plans to realize a filter antenna in multilayer technology.
  • the radiating element of the antenna is placed on an upper layer and a coupled resonator filter is made on a multiplicity of lower layers of the structure between the radiating structure and a ground plane.
  • this filter antenna has a narrow bandwidth due to the use of a patch antenna.
  • its implementation requires the control of a multilayer technology quite expensive and difficult to implement.
  • Such a wideband differential filter antenna is nevertheless described in the document " Co-designed UWB CPS filter-antenna system "by Yang et al, IEEE Antenna Propagation International Symposium Proceedings, June 2007, pages 1433-1436 .
  • This filter antenna is made using differential CPS technology.
  • the filtering device of this antenna provides the impedance matching of the high impedance loop antenna used.
  • This differential filter antenna thus has several advantages such as the elimination of impedance matching circuits and the suppression of baluns.
  • the filtering device of this antenna provides the impedance matching and the symmetrization of the loop antenna, there is really no joint design of these two elements since neither the antenna which is an ordinary loop antenna, nor the filter which is made by rectilinear impedance jump bands, are optimized in size. Indeed, the filter antenna assembly made in this document occupies a large size, of the order of a guided wavelength, which makes it difficult to integrate into current portable telecommunications systems.
  • the invention aims to overcome at least some of the aforementioned problems and constraints by providing a differential antenna system of optimized size in coplanar technology.
  • the subject of the invention is therefore a differential dipole antenna system comprising, on one and the same face of a dielectric substrate, a first half of a thick radiating dipole, a first conductive strip of a bi-ribbon supply line.
  • differential signal this first conductive strip being connected to the first thick radiating dipole half, a second thick radiating dipole half and a second conductive strip of the bi-ribbon supply line, this second conductive strip being connected to the second half a thick radiating dipole
  • the antenna system further comprising on said same face an additional conducting strip forming a short circuit connecting the first and second half of the thick dipole, and a resonant differential filtering device whose bandwidth is designed to combine with the resonance generated by the short circuit so as to produce an impedance matching of the antenna.
  • the short circuit behaves like an impedance matching network and provides resonance at a lower frequency than the natural resonant frequency of the antenna.
  • the operating wavelengths increase. In other words, for a given high operating wavelength, the size of the antenna system is significantly reduced to dimensions less than half the apparent wavelength.
  • this joint design of a shorted antenna and a resonant filtering device cleverly allows the filtering device to broaden the bandwidth of the antenna, and the antenna to improve the rejection properties off. band of the filtering device.
  • the additional conductive strip is rectilinear and arranged in a direction orthogonal to the main direction of the feed line.
  • the additional conductive strip is disposed at a predetermined distance from a feed point of the two halves of the radiating dipole by the bi-ribbon supply line, this distance being chosen sufficiently small to shift towards the low frequencies. a resonance generated by the short circuit on the radiating dipole.
  • the first and second thick radiating dipole halves are of semi elliptical, elliptical or triangular shape.
  • the resonant filtering device comprises a pair of coupled resonators disposed on the same face, each resonator comprising two conductive strips positioned symmetrically with respect to an axis of said same face, these two conductive strips being respectively connected to two conductors of a bi-ribbon port connecting to a bi-ribbon line for transmitting a differential signal.
  • each conductive band of each resonator is folded back on itself so as to form a capacitive coupling between its two ends.
  • each conductive strip makes it possible to envisage a smaller filter size, in particular a filter length less than half the apparent wavelength, for geometric reasons. Furthermore, the fact that this refolding is designed to form a capacitive coupling between the two ends of each conductive strip creates at least one additional frequency transmission zero ensuring high bandwidth and out-of-band rejection performance. filtering device. Finally, the capacitive coupling by folding also generating a magnetic coupling, the size of each conductive strip can be further reduced while ensuring the same filtering function of the assembly.
  • a differential dipole antenna system may further comprise a quarter-wave line with two coplanar conductive strips arranged to connect, in impedance matching, the bi-ribbon line. supplying the antenna to the filtering device, this quarter-wave line being shaped in the form of a printed circuit for presenting structural discontinuities generating at least one impedance jump and at least one capacitive coupling between its two conductive strips so as to reproduce a quarter-wave phase shift.
  • the invention also relates to a device for transmitting and / or receiving a broad bandwidth signal, comprising an antenna system as defined above.
  • Broad bandwidth means a signal transmitted or received for high speed communication in accordance with one of the various UWB communication standards for broad bandwidth communications.
  • the subject of the invention is also a differential dipole antenna comprising, on one and the same face of a dielectric substrate, a first half of a thick radiating dipole, a first conducting strip of a bi-ribbon signal supply line. differential, this first conductive strip being connected to the first thick radiating dipole half, a second thick radiating dipole half and a second conductive strip of the bi-ribbon supply line, this second conductive strip being connected to the second half of radiating dipole, the antenna further comprising on said same face a additional conductive strip forming a short circuit connecting the first half and the second half of the thick dipole, and being adapted to be connected to a resonant differential filtering device to form an antenna system as defined above.
  • the differential dipole antenna 10 illustrated on the figure 1 comprises, on the same face 12 of a dielectric substrate, a first antenna arm 14 and a second antenna arm 16, arranged symmetrically with respect to an axis D.
  • the first antenna arm 14 includes a first half 18 of thick radiating dipole and a first conductive strip 20 of a bi-band differential signal supply line.
  • the first half 18 of thick radiating dipole is more precisely, in the example illustrated in this figure, a half ellipse whose major axis is parallel to the axis D and constituting one of the lateral edges of the face 12 of the dielectric substrate on which is printed the antenna 10: in the repository of the figure 1 it is more precisely the left lateral edge.
  • the first conductive strip 20 is of rectilinear shape and extends parallel to and near the axis D, on the side of the first half 18 of thick radiating dipole.
  • One of its ends forms a first conductor of a bi-ribbon port 24 for connection to an external differential device (not shown).
  • the other 26 of its ends has a cubit to the left to connect the first conductive strip 20 to the convex portion of the first half 18 of thick radiating dipole, at the minor axis of the half ellipse.
  • the second antenna arm 16 has a second half 28 of thick radiating dipole and a second conductive strip 30 of the bi-band differential signal supply line.
  • the second half 28 of thick radiating dipole is more precisely, in the example illustrated in this figure, a half ellipse whose major axis is parallel to the axis D and constituting the right lateral edge of the face 12 of the dielectric substrate on which is printed the antenna 10.
  • the second conductive strip 30 is of rectilinear shape and extends parallel to and near the axis D, on the side of the second half 28 of thick radiating dipole.
  • One of its ends forms the second conductor of the bi-ribbon port 24 for connection to an external differential device.
  • the other 34 of its ends comprises a cubit to the right to connect the second conductive strip 30 to the convex portion of the second half 28 of thick radiating dipole, at the minor axis of the half ellipse.
  • a feed point P of the differential dipole antenna 10 is defined as being the intersection between the axis D and the axis of the upper edges of the two cubits 26 and 34 whose direction is orthogonal to the axis D.
  • the differential dipole antenna 10 is generally square in shape. If it consisted simply of the two arms previously described, each side of this square shape would be of the order of half an apparent wavelength.
  • the dipole antenna 10 further comprises, on the same face 12 of the dielectric substrate, an additional conductive strip 36 connecting the first half 18 and the second half 28 of the thick dipole.
  • the additional conductive strip 36 forms a short circuit between the first 18 and second 28 halves of the thick dipole. It is of thickness w of rectilinear shape and of principal direction orthogonal to the axis D, that is to say orthogonal to the main direction of the two conductive strips of the bi-band differential supply line, or parallel to the direction of the upper edges of the two cubits 26 and 34. It is located at a distance d from the feeding point P.
  • This short circuit makes it possible to obtain a significant reduction in the total area of the antenna. Indeed, it behaves like an impedance matching network and provides a resonance at a frequency lower than the natural resonance frequency of the antenna 10 if it was simply constituted by the two antenna arms 14 and 16.
  • the operating wavelengths increase. In other words, for a given high operating wavelength, the size of the antenna is significantly reduced. More precisely, it is thus possible to gain 60% in each dimension, that is to say to design a square-shaped antenna of which each side is of the order of one-fifth of an apparent wavelength. .
  • the graphic shown on the figure 2 represents the characteristic of a frequency response in reflection of the differential dipole antenna 10 previously described for operating frequencies close to 5 GHz.
  • the reflection coefficient S 11 of the frequency response has a resonance at 5.6 GHz.
  • the reflection coefficient S 11 of the frequency response has a greater resonance at 5.2 GHz.
  • the reflection coefficient S 11 of the frequency response has an even greater resonance at 4.6 GHz.
  • the distance d between the additional conductive short-circuiting strip 36 and the supply point P must be chosen sufficiently small to shift the resonance generated by the short-circuit on the radiating dipole towards the low frequencies and to achieve miniaturization. desired, but large enough to maintain an acceptable bandwidth depending on the desired use of the antenna 10.
  • the conductive strips of the feed line are selected 1.5 mm wide and spaced from each other by 0.25 mm.
  • the half ellipses of the two dipole halves have a major axis of 8.5 mm and a small axis of 7 mm.
  • the width w of the short-circuit 36 is chosen at 0.5 mm and the distance d is adjustable to vary the resonance generated by the short-circuit according to the desired application or reduction.
  • a differential dipole antenna having a surface of 17 x 17.85 mm is thus obtained.
  • This size makes it possible to consider integrating the antenna into small communicating devices too.
  • the antenna has an impedance matching between
  • the graphic shown on the figure 3 represents the characteristic of a transmission frequency response of the differential dipole antenna 10 previously described for operating frequencies close to 5 GHz.
  • the transmission coefficient S 21 of this frequency response has a significant rejection slope in the low band, notably much larger than in the high band.
  • the differential dipole antenna 10 can then be likened to a high-pass filter of the first order.
  • this filter frequency response antenna is suitable for integration with a bandpass filter, since the frequency response of the antenna can contribute to improve the low band rejection of such a filter.
  • this filter must also be chosen so that it can adapt the impedance of the antenna which is reduced by the addition of the strongly resonant short-circuit.
  • the short-circuited antenna can be modeled by an equivalent electrical circuit 40 illustrated in FIG. figure 4 .
  • the addition of the short circuit 36 to the antenna initially not short-circuited creates in fact an L-type resonator C added in parallel to the input impedance Z of the antenna initially not short-circuited.
  • This electrical circuit 40 modeling the short-circuited antenna therefore comprises two son conductors 42 and 44 between which is disposed a parallel circuit LC 46 modeling the resonator type L, C. These two son are connected to one of their ends to the impedance load Z of the antenna 10 considered without its short circuit. The other two free ends are intended to be connected to an external dipole not shown. Lead wire 44 is conventionally shown to be further connected to ground.
  • a differential dipole antenna such as that which has been described above thus advantageously comprises a resonant differential filtering device whose bandwidth is designed to combine with the resonance generated by the short circuit. to produce an impedance matching of the antenna.
  • a filter differential dipole antenna system takes advantage, on the one hand, of the strong resonance introduced by the short-circuit of the antenna to reinforce the low-band filtering of the device.
  • differential bandpass filtering directly connected to the antenna and, on the other hand, the bandwidth of the filtering device to better adapt the antenna and expand its bandwidth.
  • the filter device in the differential dipole antenna described above, it is advantageously designed in coplanar technology.
  • it may comprise a pair of coupled resonators disposed on the same face of a dielectric substrate, each resonator comprising two conductive strips positioned symmetrically with respect to a plane perpendicular to said same face, these two conductive strips being respectively connected to two conductors of a bi-ribbon port connecting to a bi-ribbon line for transmitting a differential signal.
  • This filtering device can for example be designed according to the example illustrated by the figure 12 of the document "Broadband and compact coupled coplanar stripline filters with impedance steps", by Ning Yang et al, IEEE Transactions on Microwave Theory and Techniques, vol. 55, No. 12, December 2007 .
  • the filtering device is thus improved in compactness by folding each conductive strip of each resonator of the filtering device on itself so as to form a capacitive coupling between its two ends. This makes it possible in the end to obtain an ultra miniature filtering antenna that can be powered by differential broadband signals.
  • the differential filter device 50 with coupled resonators shown in FIG. figure 5 comprises at least one pair of resonators 52 and 54, coupled to one another by capacitive coupling and arranged on the same plane face 56 of a dielectric substrate.
  • the first resonator 52 consisting of a bi-ribbon line portion, is connected to two conductors E1 and E2 of a bi-ribbon connection port to a transmission line of a differential signal.
  • These two conductors E1 and E2 of the bi-ribbon port are symmetrical about an axis D 'through which passes a plane perpendicular to the plane face 56 and forming a virtual electric ground plane. They are of a width w 'and distant from each other by a distance s, these two parameters s and w' defining the impedance of the bi-ribbon port.
  • the second resonator 54 also consisting of a bi-ribbon line portion, is connected to two conductors S1 and S2 of a bi-ribbon connection port to a transmission line of a differential signal.
  • These two conductors S1 and S2 of the bi-ribbon port are also symmetrical with respect to the axis D '.
  • the two resonators 52 and 54 are themselves symmetrical with respect to an axis perpendicular to the axis D '. Therefore, the filter device 50 is symmetrical between its differential input and output so that these can be quite inverted.
  • the two conductors E1 and E2 will be chosen by convention as being the dual-band input port of the filtering device 50, for receiving an unfiltered differential signal.
  • the two conductors S1 and S2 will be conventionally selected as the dual-band output port of the filter device 50, for providing the filtered differential signal.
  • the first resonator 52 comprises two conductive strips identified by their references LE1 and LE2. These two conductive strips LE1 and LE2 are positioned symmetrically with respect to the axis D '. They are respectively connected to the two conductors E1 and E2 of the input port.
  • the second resonator 54 comprises two conductive strips identified by their references LS1 and LS2. These two conductive strips LS1 and LS2 are also positioned symmetrically with respect to the axis D '. They are respectively connected to the two conductors S1 and S2 of the output port.
  • the capacitive coupling of the two resonators 52 and 54 is ensured by the arrangement in opposite but non-contact of their respective pairs of conductive strips.
  • the conductive strips LE1 and LS1 located on the same side with respect to the axis D ', are arranged vis-à-vis at a distance e from one another.
  • This distance e between the two resonators 52 and 54 mainly influences the bandwidth of the filtering device 50 and has a side effect on its characteristic impedance.
  • the distance e must be small enough to increase the bandwidth but also large enough not to generate unwanted reflection within the bandwidth.
  • each conductive strip must be of length ⁇ / 4, where ⁇ is the apparent wavelength, for a substrate considered, corresponding to the frequency high operating filter device.
  • the conductive strips LE1, LE2, LS1 and LS2 are advantageously folded back on themselves so as to locally form additional capacitive and magnetic couplings between their two ends.
  • the size of the filtering device 50 is thus reduced for at least two reasons: the collapses geometrically generate a size reduction of the assembly, but moreover, thanks to the capacitive and magnetic couplings, the size of each conductive strip can be further reduced. while ensuring a good functioning of the resonators.
  • This capacitive and magnetic coupling further generates a feedback between the input and the output of each conductive strip, so as to create one or more additional transmission zeros at frequencies higher than the upper limit of the bandwidth of the filter device 50 The high band rejection is thus improved.
  • the four conductive strips are generally annular in shape, their ends being folded back to the interior of this annular general shape over a portion of predetermined length thereof.
  • the folding of the ends of each conductive strip is located on a portion of this conductive strip disposed vis-à-vis the other conductive strip of the same resonator.
  • the folds of ends of the conductive strips LE1 and LE2 are arranged vis-a-vis on both sides of the axis D 'and in the vicinity thereof.
  • the conductive strip LE1 is generally rectangular in shape and consists of rectilinear conductive segments.
  • a first segment LE1 having a first free end of the conductive strip LE1 extends inwardly of the rectangle formed by the conductive strip over a length L in a direction orthogonal to the axis D '.
  • a second segment LE1 2 connected to this first segment at right angles, is part of the side of the rectangle parallel to the axis D 'and close to it.
  • a third segment LE1 3 connected to this second segment at right angles, constitutes the side of the rectangle orthogonal to the axis D 'and connected to the conductor E1 of the input port.
  • a fourth segment LE1 4 connected to this third segment at right angles, constitutes the side of the rectangle parallel to the axis D 'and close to an outer edge of the substrate.
  • a fifth segment LE1 5 connected to this fourth segment at right angles, constitutes the side of the rectangle orthogonal to the axis D 'and opposite to the side LE1 3 .
  • a sixth segment LE1 6 connected to this fifth segment at right angles, constitutes as the second segment LE1 2 a portion of the side of the rectangle parallel to the axis D 'and close thereto.
  • the segments LE1 1 and LE1 7 are spaced a constant distance e s along their entire length which ensures their capacitive coupling.
  • the conductive strip LE1 may also be seen as consisting of a folded main conductive strip connected at one of its ends to the conductor E1, this main conductive strip comprising the segments LE1 1 , LE1 2 and the portion of the segment LE1 3 located between the segment LE1 2 and the conductor E1, and a stub type branch folded on the main conductive strip, this stub type branch comprising the other part of the segment LE1 3 , and the segments LE1 4 to LE1 7 .
  • the stub type derivation is then considered as placed at the junction between the main conductive strip and the conductor E1. It should theoretically have a total length of ⁇ / 4, but the capacitive and magnetic couplings generated by the folding of the conductive strip LE1 on itself can reduce this length, especially 10 to 20% on the derivation in "stub" .
  • segment LE1 4 makes it possible to bring together the segments LE1 3 and LE1 5 , but also the segments LE1 3 and LE1 1 , or the segments LE1 5 and LE1 7 , so as to multiply the number of capacitive and magnetic couplings generated by the folding of the conductive strip LE1 on itself. These multiple couplings improve the operation of the filtering device 50.
  • the coupling length L between the two folded ends ie the two segments LE1 1 and LE1 7 , mainly influences the bandwidth of the filtering device 50, but also has a side effect on the high band rejection. The more it increases, the lower the bandwidth but the higher the band rejection is improved.
  • the distance e s between the two folded ends mainly influences the high band rejection of the filtering device 50: the smaller it is, the higher the high band rejection is improved. It should be noted, however, that this distance can not be less than a limit imposed by the precision of the etching of the conductive strip LE1 on the substrate.
  • the conductive strip LE2 consists, like the conductive strip LE1, of seven conductive segments LE2 1 to LE2 7 disposed on the plane face 56 of the substrate symmetrically to the seven segments LE1 1 to LE1 7 with respect to the axis D '.
  • the two conductive strips LE1 and LE2 are spaced a constant distance e 1 , corresponding to the distance separating the segments LE1 2 and LE1 6 , on the one hand, the segments LE2 2 and LE2 6 , on the other hand.
  • This distance e 1 mainly influences the impedance of the first resonator 52, that is to say the input impedance of the filtering device 50, but also has a side effect on the bandwidth of the filtering device 50. More it increases, the more the impedance increases and less markedly, the more the bandwidth is reduced.
  • the conductive strips LS1 and LS2 each consist, like the conductive strips LE1 and LE2, of seven segments. conductors LS1 1 to LS1 7 and LS2 1 to LS2 7 respectively, printed on the flat face 56 of the substrate symmetrically to the segments of the conductive strips LE1 and LE2 relative to this axis.
  • the two conductive strips LS1 and LS2 are spaced a constant distance e 2 equal to e 1 , corresponding to the distance separating the segments LS1 2 and LS1 6 , on the one hand, of the segments LS2 2 and LS2 6 , on the other hand.
  • This distance e 2 also mainly influences the impedance of the second resonator 54, that is to say the output impedance of the filtering device 50, but also has a side effect on the bandwidth of the filtering device 50. More it increases, the more the impedance increases and less markedly, the more the bandwidth is reduced.
  • the distance e separating the two resonators 52 and 54 corresponds to the distance separating the segments LE1 5 and LE2 5 , on the one hand, from the segments LS1 5 and LS2 5 , on the other hand.
  • the capacitive coupling between the two resonators 52 and 54 is thus established over the entire length of the segments LE1 5 and LE2 5 , on the one hand, and the segments LS1 5 and LS2 5 , on the other hand.
  • the graphic shown on the figure 6 represents the characteristic of a frequency response in transmission and reflection of the filtering device described above.
  • the reflection coefficient S 11 of this frequency response shows a bandwidth of -10 dB (generally accepted definition of the bandwidth in reflection) of between about 3.2 and 4.4 GHz.
  • -10 dB generally accepted definition of the bandwidth in reflection
  • the bandwidth is widened by the presence of two distinct reflection zeros within this bandwidth, these two zeros being due to the presence of the two coupled resonators remote from e in the filtering device 50.
  • the portion of curve S 11 situated between these two reflection zeros can go back up to -10 dB, which generates a separation of the enlarged bandwidth into two separate bandwidths. Therefore, the distance e should not be too small not to cause reflection greater than -10 dB in the extended bandwidth.
  • the transmission coefficient S 21 of the frequency response shows a bandwidth of -3 dB (generally accepted definition of the bandwidth in transmission), between about 2.7 and 4.5 GHz, as well as two transmission zeros at about 5.1 and 6.9 GHz.
  • the figure 7 schematically presents an equivalent electrical circuit of a differential dipole filter antenna according to the second aspect of the invention.
  • a first inverter 60 represents an impedance jump from Z 0 to Z 1 at the input of the filter device 50.
  • the impedance Z 0 is determined by the parameters s and w 'of the conductors E 1 and E 2 of the input port of the filter device 50, while the impedance Z 1 is determined in particular by the distance e 1 between the conductive strips LE 1 and LE 2.
  • a second inverter 62 represents the corresponding impedance jump, from Z 1 to Z 0 , at the output of the filtering device 50.
  • the first and second coupled resonators 52 and 54 are each represented by an LC circuit with capacitance C and inductance L in parallel. These two LC circuits are connected, on the one hand, respectively to the first and second inverters 60 and 62 and, on the other hand, to ground.
  • a feedback circuit LC 64 with capacitance C1 and inductance L1. parallel, connected, on the one hand, to the junction 66 between the first resonator 52 and the first inverter 60 and, on the other hand, to the junction 68 between the second resonator 54 and the second inverter 62.
  • This LC feedback circuit 64 improves the high band rejection of the filtering device 50 by adding one or more transmission zeros in the high frequencies.
  • the junction of the radiating antenna 10 and the filtering device 50 is modeled in this circuit by the connection of the inverter 62 to the free ends of the two conductor wires 42 and 44 of the electrical circuit 40, via the ground with respect to the lead wire 44
  • the addition of the short circuit in the structure of the antenna creates a resonator resonant low frequency: the LC parallel circuit 46.
  • the addition of this resonator filtering device 50 increases its order and improves its performance. Indeed, it creates in the bandwidth of the filtering device an additional reflection zero which contributes to the broadening of the bandwidth of the set and to an improvement of the impedance matching in the bandwidth.
  • the resonance of the short circuit being low frequency it helps to improve the rejection of the filter device which has a moderate rejection in its lower band.
  • an improved compactness filtering differential dipole antenna may further include a quarter-wave line for improving impedance matching between the filter device and the radiating part of the filter element. the antenna.
  • this quarter wave line is itself improved compactness. It is arranged between the filtering device and the radiating part of the antenna so as to connect, in impedance matching, the bi-ribbon supply line of the antenna to one of the dual-ribbon ports of the device. filtering.
  • Such a quarter-wave line with improved compactness and capable of transmitting a differential signal is represented on the figure 8 . It is shaped as a printed circuit to have structural discontinuities generating at least one impedance jump and at least one capacitive coupling between its two conducting strips thus performing the same functions as a conventional quarter wave line.
  • a quarter-wave bi-ribbon line 70 comprises two conductive strips 72 and 74 disposed on the same plane face 76 of a dielectric substrate.
  • the conductive strip 72 comprises a first end E'1 and a second end S'1.
  • the second conductive strip 74 comprises a first end E'2 and a second end S'2.
  • the first two ends E'1 and E'2 of the two conductive strips 72 and 74 respectively form two conductors of a first bi-ribbon port 78 for connection to a first external differential device (not shown in this figure) and the two seconds ends S'1 and S'2 of the two conductive strips respectively form two conductors of a second bi-ribbon port 80 for connection to a second external differential device (not shown in this figure).
  • the ends E'1 and E'2, on the one hand, and S'1 and S'2, on the other hand, are symmetrical with respect to an axis D "of the plane face 76.
  • the capacitive coupling and the impedance jumps of the bi-ribbon line 70, conferring on it a quarter-wave line phase shift, are directly generated by structural discontinuities themselves generating an inductance and a capacitance. More specifically, these structural discontinuities comprise, on the one hand, linearity breaks of the conductive strips 72 and 74 and, on the other hand, additional conductive branch formations extending from the conductive strips 72 and 74.
  • the breaks in linearity make it possible to vary the distance between the two conductive strips for achieving at least one impedance jump.
  • the first conductive strip 72 has several linearity breaks allowing a portion 72A of this conductive strip 72 to be further from the axis D "than the portions E '1 and S'1 forming the ends of this conductive strip. 72, while maintaining the portions E'1, S'1 and 72A parallel to the axis D ".
  • These linearity breaks are made by a portion 72B of the conductive strip 72, extending laterally and orthogonally to the axis D "of an end of the portion E'1 towards one end of the portion 72A, and by a portion 72C of the conductive strip 72, extending laterally and orthogonally to the axis D "of the other end of the portion 72A towards one end of the portion S'1.
  • the second conductive strip 74 has several linearity breaks allowing a portion 74A of this conductive strip 74 to be further from the axis D "than the portions E'2 and S'2 forming the ends of this strip. while maintaining the portions E'2, S'2 and 74A parallel to the axis D.
  • These linearity failures are formed by a portion 74B of the conductive strip 74, extending laterally and orthogonally to the D "axis of one end of the portion E'2 to one end of the portion 74A, and a portion 74C of the conductive strip 74, extending laterally and orthogonal to the axis D" of the other end of the portion 74A towards one end of the portion S'2.
  • the bi-ribbon line 70 has a first structure discontinuity, increasing the distance between its two conductive strips 72 and 74, made by the portions 72B and 74B, for performing a first impedance jump. by increasing this impedance. Indeed, the impedance increases with the distance between the two conductive strips.
  • It also has a second structure discontinuity, reducing the distance between its two conductive strips 72 and 74, made by the portions 72C and 74C, for performing a second impedance jump by reducing this impedance.
  • additional conductive branches extending from the conductive strips 72 and 74 make it possible to create at least one interdigitated capacitance for carrying out the capacitive coupling between the two conductive strips 72 and 74.
  • an interdigitated capacitance is formed by two conductive fingers 72D and 74D extending parallel to each other and orthogonal to the axis D ", facing each other over at least a part of their length.
  • the conductive finger 72D consists of a portion of rectilinear conductive strip, one end of which is secured to the portion 72A of the first conductive strip 72 and the other end remains free, while the conductive finger 74D consists of a portion of rectilinear conductive strip whose one end is integral with the portion 74A of the second conductive strip 74 and the other end remains free.
  • the pair of conductive fingers therefore extends laterally inwardly of the rectangular zone defined above from the portions 72A and 74A of the two conductive strips 72 and 74, which takes advantage of the area of the substrate in which the bi-ribbon line 70 has a greater spacing between its conductive strips 72 and 74 to form the interdigitated capacitance.
  • the length l of the bi-ribbon line 70 thus produced is substantially less than the length of a bi-ribbon quarter-wave line of the state of the art which would consist of two rectilinear and parallel conductive strips, thanks to the discontinuities of structure.
  • the bi-ribbon line 70 has a better compactness while maintaining the same characteristics as a bi-ribbon quarter-wave line of the state of the art.
  • One of the two bi-ribbon ports of the filtering device 50 is connected to one of the two bi-ribbon ports of the quarter-wave line 70 which performs an impedance inverter function.
  • the other of the two bi-ribbon ports of the quarter-wave line 70 is connected to the bi-ribbon port 24 of the dipole antenna 10.
  • the example shown in this figure is designed to operate in the 4.2-5 GHz frequency band allocated to UWB broadband communications in Europe.
  • the overall size of the filter antenna 82 square thus produced is about one-fifth of an apparent wavelength for each side.
  • these dimensions are practically those of the short-circuited antenna alone illustrated on the figure 1 , the filtering device 50 contributing to the miniaturization of the antenna ensuring its impedance matching low frequency.
  • the graphic shown on the figure 10 represents the comparative characteristics of a frequency response in reflection of the radiating antenna 10, the filtering device 50 and the filtering antenna 82.
  • the reflection coefficient S 11 of the frequency response of the filtering antenna 82 has a bandwidth at -10 dB which is considerably larger than that of the filtering device 50 alone or of the radiating antenna 10 alone.
  • the reflection coefficient S 11 of the frequency response of the radiating antenna 10 alone is not adapted to the desired UWB application, but to a narrower band between 4.45 and 5.05 GHz.
  • the filtering device alone is adapted between 4.25 and 4.9 GHz.
  • the combination of the radiating antenna and the filtering device, by an impedance matching effect of the radiating antenna is adapted between 4.15 and 5 GHz, the desired frequency band.
  • the graphic shown on the figure 11 represents the comparative characteristics of a frequency response in transmission of the radiating antenna 10, the filtering device 50 and the filtering antenna 82.
  • the transmission coefficient S 21 of the frequency response of the filtering antenna 82 has a bandwidth at -3 dB which is much more selective than that of the filtering device 50 alone.
  • the low and high band rejections are also improved and rebalanced by the combination of the first-order high-pass filter effect of the short-circuited antenna and the initial asymmetric filtering of the filtering device 50.
  • the short circuit has a first effect on the radiating antenna itself while allowing its miniaturization, but also a second effect on the filtering antenna by acting on the bandwidth of the filtering to improve the band rejections. low and high and allow transmission / reception of broadband differential signals.
  • the aforementioned double effect of the short circuit on the filter antenna described above is not limited to this form of dipole antenna.
  • Other forms of thick radiating dipoles are also suitable, whether they are low, medium or wide bandwidth.
  • the figure 12 represents a differential filtering dipole antenna 82 'resulting from a joint embodiment of a butterfly-type shorted radiating antenna 10', of the filtering device 50 shown in FIG. figure 5 and the quarter-wave line 70 shown on the figure 8 .
  • Its two dipole halves are of triangular shape and connected to the bi-ribbon feed line of the antenna by one of their vertices, for a relatively low bandwidth.
  • the figure 13 represents a differential filtering dipole antenna 82 "resulting from a joint realization of an elliptic-type short-circuited radiating antenna 10", of the filtering device 50 represented on FIG. figure 5 and the quarter-wave line 70 shown on the figure 8 . Its two halves of dipole are elliptical in shape and connected to the bi-ribbon feed line of the antenna at one end of their minor axis, for a high bandwidth.
  • the filtering device 50 described above is a good solution to be integrated in these different types of antennas, thanks to its asymmetrical frequency response particularly suitable for a design with shorted antennas, but also because it allows to reach a wide range of relative bandwidths ranging from 15% to 70%. That said, other filters having a similar asymmetric frequency response are also suitable.
  • differential dipole antenna such as one of those described above can achieve a much better compactness and a much smaller size than the known differential dipole antennas realized in differential CPS technology, while retaining the possibility of being able to transmit and receive broadband differential signals in accordance with the requirements of the UWB communication applications.
  • the coplanar structure of this differential dipole antenna also facilitates its realization in hybrid technology and its integration in monolithic technology with structures comprising discrete elements mounted on the surface.
  • it is simple to design it in integration with a bandpass filtering device produced in coplanar technology, as has been illustrated by several examples, by chemical or mechanical etching on substrates with low or high permittivity according to the applications and performances. required.
  • This antenna could in particular be manufactured on a low cost substrate, but in this case the losses generated could reduce its performance. However, this solution may remain valid for certain applications intended for the general public.
  • This antenna can also find applications in the millimeter frequency band where its small size and its high performance allow it to be integrated at low cost in monolithic technology with active transmit or receive circuits.
  • the filter antenna thus produced then has optimum characteristics in terms of size, bandwidth, radiation, consumption and rejection of noises and interfering signals.

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Claims (10)

  1. Differential-Dipolantennensystem (10; 82; 82'; 82"), umfassend auf der gleichen Seite (12) eines dielektrischen Substrats eine erste Hälfte (18) eines dicken strahlenden Dipols, ein erstes leitendes Band (20) einer Zweiband-Leitung zur Versorgung mit einem Differentialsignal, wobei dieses erste leitende Band (20) mit der ersten Hälfte (18) des dicken strahlenden Dipols verbunden ist, eine zweite Hälfte (28) des dicken strahlenden Dipols und ein zweites leitendes Band (30) der Zweiband-Versorgungsleitung, wobei dieses zweite leitende Band (30) mit der zweiten Hälfte (28) des dicken strahlenden Dipols verbunden ist, dadurch gekennzeichnet, dass es Folgendes umfasst :
    - auf der gleichen Seite (12) ein zusätzliches leitendes Band (36), das einen Kurzschluss bildet, der die erste Hälfte (18) und die zweite Hälfte (28) des dicken Dipols miteinander verbindet, und
    - eine Resonanz-Differentialfiltereinrichtung (50), deren Bandbreite dafür ausgelegt ist, mit der Resonanz kombiniert zu werden, die vom Kurzschluss erzeugt wird, um eine Antennenimpedanzanpassung zu bewirken.
  2. Differential-Dipolantennensystem (10; 82; 82'; 82") nach Anspruch 1, wobei das zusätzliche leitende Band (36) geradlinig ist und in einer Richtung angeordnet ist, die rechtwinklig zur Hauptrichtung der Versorgungsleitung (20, 30) ist.
  3. Differential-Dipolantennensystem (10; 82; 82'; 82") nach Anspruch 1 oder 2, wobei das zusätzliche leitende Band (36) in einem vorbestimmten Abstand (d) von einem Versorgungspunkt (P) der zwei Hälften (18, 28) des strahlenden Dipols von der Zweiband-Versorgungsleitung (20, 30) angeordnet ist, wobei dieser Abstand (d) ausreichend gering gewählt wird, um eine vom Kurzschluss erzeugte Resonanz am strahlenden Dipol zu den niedrigen Frequenzen hin zu verschieben.
  4. Differential-Dipolantennensystem (10; 82; 82'; 82") nach einem der Ansprüche 1 bis 3, wobei die erste (18) und die zweite Hälfte (28) des dicken strahlenden Dipols eine elliptische oder halbelliptische Form aufweisen.
  5. Differential-Dipolantennensystem (10; 82; 82'; 82") nach einem der Ansprüche 1 bis 3, wobei die erste (18) und die zweite Hälfte (28) des dicken strahlenden Dipols eine dreieckige Form aufweisen.
  6. Differential-Dipolantennensystem (10; 82; 82'; 82") nach einem der Ansprüche 1 bis 5, wobei die Resonanz-Differentialfiltereinrichtung (50) ein Paar gekoppelte Resonatoren (52, 54) aufweist, die auf der gleichen Seite (56) angeordnet sind, wobei jeder Resonator (52, 54) zwei leitende Bänder (LE1, LE2, LS1, LS2) umfasst, die auf symmetrische Weise in Bezug auf eine Achse der gleichen Seite (56) angeordnet sind, wobei diese zwei leitenden Bänder (LE1, LE2, LS1, LS2) mit zwei jeweiligen Leitern (E1, E2, S1, S2) eines Zweiband-Anschlusses zur Verbindung mit einer Zweiband-Leitung zur Übertragung eines Differentialsignals verbunden sind.
  7. Differential-Dipolantennensystem (10; 82; 82'; 82") nach Anspruch 6, wobei jedes leitende Band (LE1, LE2, LS1, LS2) jedes Resonators (52, 54) auf sich selbst zurückgeklappt ist, um eine kapazitive Kopplung zwischen seinen zwei Enden zu bilden.
  8. Differential-Dipolantennensystem (10; 82; 82'; 82") nach einem der Ansprüche 1 bis 7, umfassend eine Viertelwellenleitung (70) mit zwei koplanaren leitenden Bändern (72, 74), die derart angeordnet ist, dass die Zweiband-Versorgungsleitung (20, 30) der Antenne unter Impedanzanpassung mit der Filtereinrichtung (50) verbunden wird, wobei diese Viertelwellenleitung in Form einer gedruckten Schaltung derart ausgebildet ist, dass Strukturunterbrechungen (72B, 72C, 72D, 74B, 74C, 74D) vorhanden sind, die mindestens einen Impedanzsprung und mindestens eine kapazitive Kopplung zwischen ihren zwei leitenden Bändern (72, 74) erzeugen, um eine Viertelwellen-Phasenverschiebung zu reproduzieren.
  9. Vorrichtung zum Senden und/oder Empfangen eines Signals mit großer Bandbreite, umfassend ein Antennensystem nach einem der Ansprüche 1 bis 8.
  10. Differential-Dipolantenne (10), umfassend auf der gleichen Seite (12) eines dielektrischen Substrats eine erste Hälfte (18) eines dicken strahlenden Dipols, ein erstes leitendes Band (20) einer Zweiband-Leitung zur Versorgung mit einem Differentialsignal, wobei dieses erste leitende Band (20) mit der ersten Hälfte (18) des dicken strahlenden Dipols verbunden ist, eine zweite Hälfte (28) des dicken strahlenden Dipols und ein zweites leitendes Band (30) der Zweiband-Versorgungsleitung, wobei dieses zweite leitende Band (30) mit der zweiten Hälfte (28) des dicken strahlenden Dipols verbunden ist, dadurch gekennzeichnet, dass sie auf der gleichen Seite (12) ein zusätzliches leitendes Band (36) umfasst, das einen Kurzschluss bildet, der die erste Hälfte (18) und die zweite Hälfte (28) des dicken Dipols miteinander verbindet, und dass sie geeignet ist, mit einer Resonanz-Differentialfiltereinrichtung (50) verbunden zu werden, um ein Antennensystem nach einem der Ansprüche 1 bis 8 zu bilden.
EP08875626A 2008-11-07 2008-11-07 Differenz-dipolantennensystem mit einer koplanaren strahlungsstruktur und sender-/empfängereinrichtung Active EP2345104B1 (de)

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PCT/FR2008/001573 WO2010052377A1 (fr) 2008-11-07 2008-11-07 Systeme d'antenne dipole differentielle a structure rayonnante coplanaire et dispositif d'emission/reception

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JP2015008354A (ja) * 2013-06-24 2015-01-15 富士通株式会社 伝送装置および高周波フィルタ
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CN110459867A (zh) * 2019-08-13 2019-11-15 昆山恩电开通信设备有限公司 一种具有抑制高频寄生辐射功能的超宽带低频辐射单元
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CN105449379B (zh) * 2015-11-30 2018-04-13 华南理工大学 一种能抑制高频谐波的滤波天线

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EP2345104A1 (de) 2011-07-20
US8704723B2 (en) 2014-04-22
WO2010052377A1 (fr) 2010-05-14
ES2396006T3 (es) 2013-02-18

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