EP2184801A1 - Vorrichtung zur Abdichtung eines Öffnungsfeldes eines Kraftfahrzeugs mit Hilfe eines Schiebepaneels, und entsprechendes Kraftfahrzeug - Google Patents

Vorrichtung zur Abdichtung eines Öffnungsfeldes eines Kraftfahrzeugs mit Hilfe eines Schiebepaneels, und entsprechendes Kraftfahrzeug Download PDF

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Publication number
EP2184801A1
EP2184801A1 EP09175192A EP09175192A EP2184801A1 EP 2184801 A1 EP2184801 A1 EP 2184801A1 EP 09175192 A EP09175192 A EP 09175192A EP 09175192 A EP09175192 A EP 09175192A EP 2184801 A1 EP2184801 A1 EP 2184801A1
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EP
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Prior art keywords
filtering device
differential
resonator
resonators
conductive strip
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EP09175192A
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English (en)
French (fr)
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EP2184801B1 (de
Inventor
Raffi Bourtoutian
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Commissariat a lEnergie Atomique et aux Energies Alternatives CEA
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Commissariat a lEnergie Atomique CEA
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/203—Strip line filters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/307—Individual or coupled radiating elements, each element being fed in an unspecified way
    • H01Q5/314—Individual or coupled radiating elements, each element being fed in an unspecified way using frequency dependent circuits or components, e.g. trap circuits or capacitors
    • H01Q5/335—Individual or coupled radiating elements, each element being fed in an unspecified way using frequency dependent circuits or components, e.g. trap circuits or capacitors at the feed, e.g. for impedance matching
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/16—Resonant antennas with feed intermediate between the extremities of the antenna, e.g. centre-fed dipole
    • H01Q9/28—Conical, cylindrical, cage, strip, gauze, or like elements having an extended radiating surface; Elements comprising two conical surfaces having collinear axes and adjacent apices and fed by two-conductor transmission lines
    • H01Q9/285—Planar dipole

Definitions

  • the present invention relates to a differential filtering device with coupled resonators. It also relates to a filter antenna comprising at least one filtering device of this type.
  • Radio frequency transmit / receive systems powered by differential electrical signals are very attractive for current and future wireless communications systems, especially for autonomous communicating object concepts.
  • a differential supply is a supply of two signals of equal amplitude in phase opposition. It helps reduce, or even eliminate, unwanted "common mode” noise in transmit and receive systems.
  • a non-differential power supply causes the radiation of an undesired cross component due to the common mode flowing on the non-symmetrical power cables.
  • the use of a differential power supply eliminates the cross-radiation of the measurement cables and thus makes it possible to obtain reproducible measurements independent of the measurement context as well as perfectly symmetrical radiation diagrams.
  • the "push-pull" power amplifiers whose structure is differential have several advantages, such as the doubling of the output power and the elimination of higher order harmonics.
  • the low noise differential amplifiers offer several perspectives in terms of reduction of the noise factor. Also, the use of a differential structure prevents unwanted triggering of the oscillators by common mode noise.
  • baluns involves several disadvantages: increasing congestion, cost and adding additional losses thus reducing the overall performance of the system.
  • Another problem lies in the difficulty of achieving broad bandwidth baluns, that is to say capable of ensuring a perfect transformation of a non-differential signal into a differential signal over the entire bandwidth. They can cause the creation of common mode signals and degrade the overall operation of the system. This results in a great need to make filters directly in differential technology to overcome all the disadvantages caused by the use of baluns.
  • EP 0 542 917 B1 presents a differential ring filter coupled in micro ribbon technology. This filter comprises two coupled micro ribbons that can transmit a differential signal.
  • this coupled ring filter made in micro-band technology has a narrow bandwidth and is therefore not suitable for broadband telecommunications requiring very large bandwidths.
  • the invention thus relates more precisely to a differential filtering device comprising a pair of coupled resonators disposed on the same face of a dielectric substrate, each resonator comprising two conductive strips positioned symmetrically with respect to a plane perpendicular to the face on which the resonator is arranged, these two conductive strips being respectively connected to two conductors of a bi-ribbon connection port to a transmission line of a differential signal.
  • This filter comprises two coplanar resonators, each having a bi-ribbon line portion consisting of two rectilinear conductive strips parallel and symmetrical with respect to a plane perpendicular to the plane of the resonators. This plane of symmetry represents a virtual mass plane for the filter because of its differential character.
  • Each conductive strip has a length which corresponds to a quarter of the apparent wavelength in the filter substrate at the high operating frequency of the filter.
  • the two conductive strips of the same resonator are connected, at one of their two ends, respectively to two conductors of a bi-ribbon port for connection to a transmission line of a differential signal. They therefore each keep a free end. Capacitive coupling of the two resonators is then achieved by the arrangement vis-à-vis the free ends of their respective conductive strips.
  • the bandpass filtering is performed, on the one hand, by the impedance jumps between each pair of conductive strips and the port to which it is connected and, on the other hand, by the capacitive coupling of the two resonators.
  • Such a topology makes it possible to achieve high bandwidths with high out-of-band rejection for filters of order 2, 3 or 4.
  • the arrangement opposite the two pairs of rectilinear and parallel conductive strips implies a dimension of the filter close to half apparent wavelength at the high operating frequency, which is relatively compact. This compactness can even be optimized by choosing a substrate whose dielectric properties can reduce the apparent wavelength.
  • some applications, especially small autonomous communicating objects, require even more compact filters.
  • the invention therefore relates to a differential filtering device with coupled resonators, comprising a pair of coupled resonators arranged on the same face of a dielectric substrate, each resonator comprising two conductive strips positioned symmetrically with respect to a plane perpendicular to the face on which the resonator is arranged, these two conductive strips being respectively connected to two conductors of a bi-ribbon connection port to a transmission line of a differential signal, characterized in that each conducting band of each resonator is folded on itself so as to form a capacitive coupling between its two ends.
  • each conductive strip makes it possible to envisage a smaller filter size, in particular a filter length less than half the apparent wavelength, for geometric reasons. Furthermore, the fact that this refolding is designed to form a capacitive coupling between the two ends of each conductive strip creates at least one additional frequency transmission zero ensuring high bandwidth and out-of-band rejection performance. filtering device. Finally, the capacitive coupling by folding also generating a magnetic coupling, the size each conductive strip can be further reduced while ensuring the same filtering function of the assembly.
  • the two resonators of the pair are coupled by the arrangement with respect to their respective conductive strips disposed on the same side with respect to said plane of symmetry, over respective length portions of these folded conductive strips.
  • the capacitive coupling of the two resonators is thus improved, by not being limited to the coupling of the ends of the conductive strips.
  • each conductive strip of each resonator is generally annular in shape, its ends being folded inside the generally annular shape over a portion of predetermined length thereof, the folding of the ends being located on a portion of the conductive strip disposed opposite the other conducting band of the resonator.
  • the portion of length on which the folding is performed may be chosen to set a certain desired bandwidth of the filtering device.
  • each conductive strip of each resonator is of generally rectangular shape.
  • each conductive strip of each resonator is generally square.
  • At least a portion of the conductive strip portions forming the sides of the generally rectangular or square shape of each conductive strip has additional folds.
  • the additional folds are directed inwardly of the generally rectangular or square shape.
  • the two conductive strips of one of the two resonators are spaced a first distance between them and the two conductive strips of the other of the two resonators are separated by a second distance between them, this second distance being different from the first distance so that the filtering device performs an additional impedance matching function by presenting an output impedance different from its input impedance.
  • the filtering device can be used to directly connect two different impedance circuits, such as an antenna and an active circuit.
  • the invention also relates to a differential dipole filter antenna comprising at least one filtering device as defined above.
  • a differential dipole filter antenna according to the invention may comprise a radiating structure shaped to integrate in its external dimensions said filtering device.
  • the coupled resonator differential filtering device 10 shown in FIG. figure 1 comprises at least one pair of resonators 12 and 14, coupled together by capacitive coupling and disposed on the same plane face 16 of a dielectric substrate.
  • the first resonator 12 consisting of a bi-ribbon line portion, is connected to two conductors E1 and E2 of a bi-ribbon connection port to a transmission line of a differential signal.
  • These two conductors E1 and E2 of the bi-ribbon port are symmetrical with respect to a plane P perpendicular to the plane face 16 and forming a virtual electric ground plane. They are of a width w and distant from each other by a distance s, these two parameters s and w defining the impedance of the bi-ribbon port.
  • the second resonator 14 also consisting of a bi-ribbon line portion, is connected to two conductors S1 and S2 of a bi-ribbon connection port to a transmission line of a differential signal.
  • These two conductors S1 and S2 of the bi-ribbon port are also symmetrical with respect to the virtual electrical ground plane P.
  • the filtering device 10 is symmetrical between its differential input and its output so that these can be totally reversed.
  • the two conductors E1 and E2 will be chosen by convention as being the bi-band input port of the filtering device 10, for receiving an unfiltered differential signal.
  • the two conductors S1 and S2 will be conventionally chosen as the bi-band output port of the filter device 10, for the supply of the filtered differential signal.
  • the first resonator 12 comprises two conductive strips identified by their references LE1 and LE2. These two conductive strips LE1 and LE2 are positioned symmetrically with respect to the virtual electrical ground plane P. They are respectively connected to the two conductors E1 and E2 of the input port.
  • the second resonator 14 comprises two conductive strips identified by their references LS1 and LS2. These two conductive strips LS1 and LS2 are also positioned symmetrically with respect to the virtual electrical ground plane P. They are respectively connected to the two conductors S1 and S2 of the output port.
  • Capacitive coupling of the two resonators 12 and 14 is ensured by the arrangement vis-à-vis but without contact of their respective pairs of conductive strips.
  • the conductive strips LE1 and LS1 located on the same side with respect to the virtual electrical ground plane P, are arranged vis-a-vis at a distance e from one another.
  • the conductive strips LE2 and LS2, situated on the other side with respect to the virtual electrical ground plane P, are arranged facing each other at the same distance e from each other.
  • This distance e between the two resonators 12 and 14 mainly influences the bandwidth of the filtering device 10 and has a side effect on its characteristic impedance.
  • the bandwidth is enlarged by the appearance of two distinct reflection zeros within this bandwidth, corresponding to two distinct resonant frequencies, when e is small enough to achieve the capacitive coupling between the two resonators.
  • the lower the distance e the more the two reflection zeros created move away from each other, thus widening the bandwidth.
  • the distance e must be small enough to increase the bandwidth but also large enough not to generate unwanted reflection within the bandwidth.
  • each conductive strip must be of length ⁇ / 4, where ⁇ is the apparent wavelength, for a substrate considered, corresponding to the frequency high operating filter device.
  • the conductive strips LE1, LE2, LS1 and LS2 are advantageously folded back on themselves so as to locally form additional capacitive and magnetic couplings between their two ends.
  • the size of the filtering device 10 is thus reduced for at least two reasons: the collapses geometrically generate a size reduction of the assembly, but moreover, thanks to the capacitive and magnetic couplings, the size of each conductive strip can be further reduced. while ensuring a good functioning of the resonators.
  • This capacitive and magnetic coupling further generates a feedback between the input and the output of each conductive strip, so as to create one or more additional transmission zeros at frequencies higher than the upper limit of the bandwidth of the filter device 10. The high band rejection is thus improved.
  • the four conductive strips are of generally annular shape, their ends being folded inside this annular general shape over a portion of predetermined length thereof.
  • the folding of the ends of each conductive strip is located on a portion of this conductive strip disposed vis-à-vis the other conductive strip of the same resonator.
  • the folds of ends of the conductive strips LE1 and LE2 are arranged vis-à-vis on both sides of the plane of symmetry P and in the vicinity thereof.
  • the conductive strip LE1 is generally rectangular in shape and consists of rectilinear conductive segments.
  • a first segment LE1 1 having a first free end of the conductive strip LE1 extends inwardly of the rectangle formed by the conductive strip over a length L in a direction orthogonal to the virtual ground plane P.
  • a second segment LE1 2 connected to this first segment at right angles, is part of the side of the rectangle parallel to the virtual ground plane P and close to it.
  • a third segment LE1 3 connected to this second segment at right angles, constitutes the side of the rectangle orthogonal to the virtual ground plane P and connected to the conductor E1 of the input port.
  • a fourth segment LE1 4 connected to this third segment at right angles, constitutes the side of the rectangle parallel to the virtual ground plane P and close to an outer edge of the substrate.
  • a fifth segment LE1 5 connected to this fourth segment at right angles, constitutes the side of the rectangle orthogonal to the virtual ground plane P and opposite the side LE1 3 .
  • a sixth segment LE1 6 connected to this fifth segment at right angles, constitutes as the second segment LE1 2 a portion of the side of the rectangle parallel to the virtual ground plane P and close to it.
  • the segments LE1 1 and LE1 7 are spaced a constant distance e S over their entire length which ensures their capacitive coupling.
  • the conductive strip LE1 may also be seen as consisting of a folded main conductive strip connected at one of its ends to the conductor E1, this main conductive strip comprising the segments LE1 1 , LE1 2 and the portion of the segment LE1 3 located between the segment LE1 2 and the conductor E1, and a stub-type branch folded on the main conductive strip, this stub-type branch comprising the other part of the segment LE1 3 , and the segments LE1 4 to LE1 7 .
  • the "stub" type branch is then considered to be placed at the junction between the main conducting strip and the conductor E1. It should theoretically have a total length of ⁇ / 4, but the capacitive and magnetic couplings generated by the folding of the conductive strip LE1 on itself can reduce this length, especially 10 to 20% on the derivation in "stub" .
  • segment LE1 4 makes it possible to bring together the segments LE1 3 and LE1 5 , but also the segments LE1 3 and LE1 1 , or the segments LE1 5 and LE1 7 , so as to multiply the number of capacitive and magnetic couplings generated by the folding of the conductive strip LE1 on itself. These multiple couplings improve the operation of the filtering device 10.
  • the coupling length L between the two folded ends ie the two segments LE1 1 and LE1 7 , mainly influences the bandwidth of the filter device 10, but also has a side effect on the high band rejection. The more it increases, the lower the bandwidth but the higher the band rejection is improved.
  • the distance e S between the two folded ends mainly influences the high-band rejection of the filtering device 10: the smaller it is, the higher the rejection at high band is improved. It should be noted, however, that this distance can not be less than a limit imposed by the precision of the etching of the conductive strip LE1 on the substrate.
  • the conductive strip LE2 consists, like the conductive strip LE1, of seven conductive segments LE2 1 to LE2 7 disposed on the plane face 16 of the substrate symmetrically to the seven segments LE1 1 to LE1 7 with respect to the virtual ground plane P.
  • the two conductive strips LE1 and LE2 are spaced a constant distance e 1 , corresponding to the distance separating the segments LE1 2 and LE1 6 , on the one hand, the segments LE2 2 and LE2 6 , on the other hand.
  • This distance e 1 mainly influences the impedance of the first resonator 12, that is to say the input impedance of the filtering device 10, but also has a side effect on the bandwidth of the filtering device 10. More it increases, the more the impedance increases and less markedly, the more the bandwidth is reduced.
  • the two resonators 12 and 14 being symmetrical with respect to an axis normal to the virtual ground plane P located on the plane face 16, the conductive strips LS1 and LS2 each consist, as the conductive strips LE1 and LE2, of seven conductive segments LS1 1 to LS1 7 and LS2 1 to LS2 7 respectively, printed on the flat face 16 of the substrate symmetrically to the segments of the conductive strips LE1 and LE2 by report to this axis.
  • the two conductive strips LS1 and LS2 are spaced a constant distance e 2 equal to e 1 , corresponding to the distance separating the segments LS1 2 and LS1 6 , on the one hand, of the segments LS2 2 and LS2 6 , on the other hand.
  • This distance e 2 also mainly influences the impedance of the second resonator 14, that is to say the output impedance of the filtering device 10, but also has a side effect on the bandwidth of the filtering device 10. More it increases, the more the impedance increases and less markedly, the more the bandwidth is reduced.
  • the distance e separating the two resonators 12 and 14 corresponds to the distance separating the segments LE1 5 and LE2 5 , on the one hand, from the segments LS1 5 and LS2 5 , on the other hand.
  • the capacitive coupling between the two resonators 12 and 14 is thus established over the entire length of the segments LE1 5 and LE2 5 , on the one hand, and the segments LS1 5 and LS2 5 , on the other hand.
  • the figure 2 schematically presents an equivalent electric circuit of the filtering device 10 previously described.
  • a first inverter 20 represents an impedance jump, from Z 0 to Z 1 , at the input of the filtering device 10.
  • the impedance Z 0 is determined by the parameters s and w of the conductors E1 and E2 of the port input, while the impedance Z 1 is determined in particular by the distance e 1 between the conductive strips LE 1 and LE 2.
  • a second inverter 22 represents the corresponding impedance jump, from Z 1 to Z 0 , at the output of the filtering device 10.
  • the first and second coupled resonators 12 and 14 are each represented by an LC circuit with capacitance C and inductance L in parallel. These two LC circuits are connected, respectively, respectively to the first and second inverters 20 and 22 and, on the other hand, to ground.
  • the folding of the conductive strips LE1, LE2, LS1 and LS2 creates additional couplings, inside each resonator but also between the resonators, which can be represented by a feedback LC circuit 24, with capacitance C1 and inductance L1 in parallel, connected, on the one hand, to the junction 26 between the first resonator 12 and the first inverter 20 and, on the other hand, to the junction 28 between the second resonator 14 and the second inverter 22.
  • This LC feedback circuit 24 improves the high band rejection of the filter device 10 by adding one or more transmission zeros in the high frequencies.
  • the graphic shown on the figure 3 represents the characteristic of a frequency response in transmission and reflection of the filtering device described above.
  • the reflection coefficient S 11 of this frequency response shows a bandwidth of -10 dB (generally accepted definition of the bandwidth in reflection) of between about 3.2 and 4.4 GHz.
  • the bandwidth is widened by the presence of two distinct reflection zeros within this bandwidth, these two zeros being due to the presence of the two coupled resonators remote from e in the filtering device 10.
  • the transmission coefficient S 21 of the frequency response shows a bandwidth of -3 dB (generally accepted definition of the bandwidth in transmission), between about 2.7 and 4.5 GHz, as well as two transmission zeros at about 5.1 and 6.9 GHz.
  • One of these two out-of-band transmission zeros is due to the coupling between the two resonators of the filter device 10 over the entire length of their portions LE1 5 , LE2 5 on the one hand and LS1 5 , LS2 5 on the other hand .
  • the other of these two transmission zeros is due to the additional intra-resonator couplings created by the folding of the conductive strips on themselves.
  • These two transmission zeros cause a high rejection of the high band filter and an asymmetry of the frequency response due to the low band mean rejection. But this asymmetry may be advantageous, especially for a direct integration application of the filtering device 10 in a differential antenna. Indeed, such antennas generally have high resonances low frequency and therefore equivalent to high-pass filters, which compensates for the asymmetry of the filter device 10 by improving its low band rejection.
  • a second embodiment of a differential filtering device according to the invention is shown schematically on the figure 4 .
  • This device 10 ' comprises a pair of resonators 12' and 14 ', coupled together by capacitive coupling and disposed on the same plane face 16 of a dielectric substrate. These two resonators are similar to those, 12 and 14, of the device of the figure 1 .
  • the two resonators 12 'and 14' are not symmetrical with respect to an axis normal to the plane P situated on the flat face 16.
  • the distance e 1 separating the two conducting strips LE1 and LE2 of the first resonator 12 ' is distinct from the distance e 2 between the two conductive strips LS1 and LS2 of the second resonator 12'.
  • the distance e 2 is greater than the distance e 1 .
  • the capacitive coupling between the two resonators 12 'and 14' is not broken so far. Indeed, due to the folding of the conductive strips on themselves, they remain in vis-à-vis at least a portion of their length, more specifically at least a portion of the lengths LE1 5 and LS1 5, d the one hand, and lengths LS2 and LE2 5 5, on the other hand. In comparison with the existing one, it would not be possible, for example, to conceive of such a difference between the distances e 1 and e 2 in the filtering device described with reference to FIG.
  • these distances e 1 and e 2 make it possible respectively to adjust the input and output impedances of the filtering device 10 ', it is possible to design a bandpass filtering device which also fulfills an adaptation function of impedances between the circuits to which it is intended to be connected.
  • the distance e 1 thus generates an input impedance Z 1 smaller than the output impedance Z 2 generated by the distance e 2 .
  • This second embodiment allows the direct integration of a filtering device according to the invention with differential antennas and active circuits. different impedance differentials. Note, however, that such a direct integration with a single filter device works all the better that the difference between the impedances Z 1 and Z 2 is small.
  • a set of several filtering devices according to the second embodiment of the invention added in series can be used to facilitate impedance matching between very different impedance circuits.
  • Such a set with two filtering devices is for example represented diagrammatically on the figure 5 .
  • an amplifier 30 is connected to the input of a first filtering device 32, via the input port 34 of this first filtering device. Since the impedance of the amplifier 30 has a value Z 1 , the first filtering device 32 is designed, by adjusting the distance between the folded conductive strips of its first resonator, to present a conjugate value input impedance Z 1 * thus ensuring a maximum power transfer between the first filtering device 32 and the amplifier 30.
  • An antenna 36 is connected to the output of a second filtering device 38, via the output port 40 of this second filtering device. Since the impedance of the antenna 36 has a value Z 2 , the second filtering device 38 is designed, by adjusting the distance between the folded conductive strips of its second resonator, to present a conjugate value output impedance Z 2 * thus ensuring maximum power transfer between the second filtering device 38 and the antenna 36.
  • the two filtering devices 32 and 38 are connected together, either directly or indirectly via a quarter-wave line 42 fulfilling an inverter function, the output of the first filtering device 32 and the input of the second device filtering device 38 being designed, by adjusting the distance between the folded conductive strips of the second resonator of the first filtering device 32 and the distance between the folded conductive strips of the first resonator of the second filtering device 38, to present the same impedance Z 0 .
  • This same impedance Z 0 ensures the adaptation of impedances and can be chosen so as to ensure the best possible rejection.
  • the adaptation of the impedances Z 1 and Z 2 which can be very different is through an intermediate impedance Z 0 through the set comprising the two asymmetric filtering devices 32 and 38.
  • a third embodiment of a differential filtering device according to the invention is shown schematically on the figure 6 .
  • This filter device 10 "comprises a pair of resonators 12" and 14 ", coupled together by capacitive coupling and disposed on the same plane face 16 of a dielectric substrate.
  • the two resonators 12 "and 14" are symmetrical with respect to an axis normal to the plane P situated on the plane face 16. Consequently, the distance e 1 between the two conductive strips LE1 and LE2 of the first resonator 12 "is equal to the distance e 2 between the two conductive strips LS1 and LS2 of the second resonator 14".
  • these two distances could be different, as in the second embodiment, for the filtering device to further fulfill an impedance matching function.
  • this third embodiment is distinguished from the first and second embodiments by the general shape of the folded conductive strips.
  • the four conductive strips are of generally annular shape, their ends being folded inside this annular general shape over a portion of predetermined length thereof, but they are more precisely of shape. general square.
  • each of them has additional folding on at least a portion of the sides of the square general shape.
  • the conductive strip LE1 comprises three additional folds LE1 8 , LE1 9 and LE1 10 in the three sides of the square general shape not having the folding of its two ends.
  • the three additional folds are directed towards the inside of the square general shape. They are for example in the form of niche.
  • the conductive strips LE2, LS1 and LS2 have the same additional folds, LE2 referenced 8, 9 and LE2 LE2 10 to the conductive strip LE2; LS1 8 , LS1 9 and LS1 10 for the conductive strip LS1; LS2 8 , LS2 9 and LS2 10 for the conductive strip LS2.
  • each conductive strip LE1, LE2, LS1 and LS2 implies a generally square shape of the filtering device 10 ", so the compactness of the latter is optimal.
  • the additional folds create additional capacitive and magnetic couplings that can further improve the performance of the filter device 10 ".
  • the length L of the folding of the two ends of each conductive strip within its overall square shape can be adjusted to adjust the bandwidth of the filter device 10 ".
  • a filtering device according to the invention is not limited to the embodiments described above. Other geometrical shapes are possible for a filtering device according to the invention, from the moment they provide for a folding of each conductive strip of each resonator on itself so as to form a capacitive coupling between its two ends.
  • FIGS. 7 to 9 illustrate schematically three examples of differential filter dipole antennas each advantageously integrating at least one filtering device such as those described above.
  • the filtering dipole antenna 50 represented on the figure 7 comprises on the one hand a radiating electric dipole 52 and on the other hand a filtering device 54 such as that described with reference to FIG. figure 1 .
  • the electric dipole 52 is more precisely a coplanar thick dipole etched on a substrate and whose radiating structure is of elliptical shape. This type of dipole is very wide bandwidth.
  • the relative bandwidth defined by the relation ⁇ f / f 0 where ⁇ f is the width of the bandwidth and f 0 the central operating frequency of the antenna, may exceed 100%.
  • the two arms of the dipole 52 are directly connected to the two conductors of the output port of the filtering device 54.
  • the dipole 52 and the filtering device 54 could be connected via a quarter-wave line: this would provide a filter antenna with improved performance.
  • the two conductors of the input port of the filter device 54 are for their part to be supplied with a differential signal.
  • the filtering dipole antenna 60 shown in FIG. figure 8 comprises on the one hand a radiating electric dipole 62 and on the other hand a filtering assembly comprising two filtering devices 64 and 66 such as that described with reference to the figure 6 .
  • the electric dipole 62 is more precisely a coplanar thick dipole etched on a substrate and whose radiating structure is of "butterfly" shape. More specifically, the radiating structure of the dipole has a thin portion, in a central zone of the antenna comprising the connection to the filtering devices 64 and 66, which widens outwardly of the antenna on both sides of the dipole.
  • This type of radiating dipole is medium bandwidth. Its relative bandwidth ⁇ f / f 0 is of the order of 20%.
  • the two arms of the dipole 62 are directly connected to the two conductors of the output port of the first filtering device 64.
  • the dipole 62 and the first filtering device 64 could be connected via a quarter-turn line. wave.
  • the two conductors of the input port of the first filtering device 64 are directly connected to the two conductors of the output port of the second filtering device 66.
  • the first filtering device 64 and the second filtering device 66 could be connected via a quarter-wave line to obtain a higher order, higher performance filter.
  • the two conductors of the input port of the second filter device 66 are for their part to be supplied with a differential signal.
  • the filter dipole antenna 70 represented on the figure 9 comprises on the one hand a radiating electric dipole 72 and on the other hand a filtering assembly comprising two filtering devices 74 and 76 identical to the two devices 64 and 66.
  • the electric dipole 72 is more precisely a coplanar thick dipole etched on a substrate and whose radiating structure is of "butterfly" shape.
  • it differs from the electric dipole 62 especially in that the two broad ends of its radiating structure, oriented towards the outside of the antenna, are shaped to integrate in their external dimensions (ie greater length and greater width) the two filter devices 74 and 76. This results in a further gain in compactness of the filter antenna 70 relative to the filter antenna 60.
  • a differential dipole filter antenna according to the invention is smaller than a conventional corresponding antenna, thanks to the better compactness of the filtering devices used.
  • a differential dipole filter antenna according to the invention is more efficient because it may comprise a larger number of filtering devices to achieve an even higher order filtering, thus more efficient in terms of bandwidth.
  • the coplanar structure of this filtering device further facilitates its realization in hybrid technology and its integration in monolithic technology with structures comprising discrete elements mounted on the surface.
  • it is simple to design it in integration with a differential dipole antenna with broadband coplanar radiating structure, as has been illustrated by several examples, by chemical or mechanical etching on substrates with low or high permittivity according to the desired applications and performance. .
  • This filtering device can also find applications in the millimetric frequency band where its small size and its strong performances allow it to be integrated in monolithic technology with antennas and active circuits.

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EP09175192.5A 2008-11-07 2009-11-06 Differentialfiltervorrichtung mit koplanar gekoppelten Resonatoren und Filterantenne mit einer entsprechenden Vorrichtung Active EP2184801B1 (de)

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CN105680127A (zh) * 2016-04-27 2016-06-15 上海海事大学 基于信号干扰理论的差分带通滤波器
CN106654551A (zh) * 2016-11-18 2017-05-10 深圳市共进电子股份有限公司 无线电子设备及其pcb板
CN108717994A (zh) * 2018-04-18 2018-10-30 西安电子科技大学 一种应用于wlan频段的新型平面双频带通滤波器天线
CN110444840A (zh) * 2019-08-06 2019-11-12 西安电子科技大学 基于枝节负载谐振器的双频差分带通滤波器
CN111864321A (zh) * 2020-08-14 2020-10-30 中国电子科技集团公司第五十四研究所 一种基于枝节加载槽线谐振器的平衡式双通带滤波器
CN112186345A (zh) * 2020-09-17 2021-01-05 华南理工大学 一种基于谐振器型偶极子的三阶滤波基站天线
CN113889754A (zh) * 2021-09-29 2022-01-04 重庆大学 一种紧凑的单层差分馈电滤波透明天线
CN114597622A (zh) * 2022-02-25 2022-06-07 南京恒电电子有限公司 双通带平衡滤波耦合器
CN114824715A (zh) * 2022-03-29 2022-07-29 中国人民解放军国防科技大学 基于矩形微同轴结构的w波段滤波功分器
CN115051154A (zh) * 2022-07-27 2022-09-13 重庆邮电大学 一种基于开口阶梯槽的差分宽带端射滤波天线

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WO2010052377A1 (fr) * 2008-11-07 2010-05-14 Commissariat A L'energie Atomique Et Aux Energies Alternatives Systeme d'antenne dipole differentielle a structure rayonnante coplanaire et dispositif d'emission/reception
JP5377778B2 (ja) * 2011-05-11 2013-12-25 パナソニック株式会社 電磁共鳴結合器
FR3033103A1 (fr) * 2015-02-24 2016-08-26 Univ Paris Diderot Paris 7 Dispositif resonateur electrique tridimensionnel de type inductance-capacite
CN107331958B (zh) * 2017-08-11 2023-06-27 深圳山脊技术有限公司 一种圆极化滤波天线
TWI648950B (zh) * 2018-02-27 2019-01-21 台郡科技股份有限公司 Differential filter microstrip line structure capable of suppressing common mode signals
CN112909460B (zh) * 2021-01-18 2022-04-19 电子科技大学 同时具有共模和差模信号无反射特性的平衡式微带滤波器
US11817630B2 (en) 2021-09-17 2023-11-14 City University Of Hong Kong Substrate integrated waveguide-fed Fabry-Perot cavity filtering wideband millimeter wave antenna
CN116598738B (zh) * 2023-07-17 2023-10-13 成都华兴汇明科技有限公司 一种四端口选频网络及其构造的微波振荡器

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CN105680127A (zh) * 2016-04-27 2016-06-15 上海海事大学 基于信号干扰理论的差分带通滤波器
CN105680127B (zh) * 2016-04-27 2018-06-19 上海海事大学 基于信号干扰理论的差分带通滤波器
CN106654551A (zh) * 2016-11-18 2017-05-10 深圳市共进电子股份有限公司 无线电子设备及其pcb板
CN108717994A (zh) * 2018-04-18 2018-10-30 西安电子科技大学 一种应用于wlan频段的新型平面双频带通滤波器天线
CN110444840A (zh) * 2019-08-06 2019-11-12 西安电子科技大学 基于枝节负载谐振器的双频差分带通滤波器
CN111864321A (zh) * 2020-08-14 2020-10-30 中国电子科技集团公司第五十四研究所 一种基于枝节加载槽线谐振器的平衡式双通带滤波器
CN112186345B (zh) * 2020-09-17 2022-02-15 华南理工大学 一种基于谐振器型偶极子的三阶滤波基站天线
CN112186345A (zh) * 2020-09-17 2021-01-05 华南理工大学 一种基于谐振器型偶极子的三阶滤波基站天线
CN113889754A (zh) * 2021-09-29 2022-01-04 重庆大学 一种紧凑的单层差分馈电滤波透明天线
CN113889754B (zh) * 2021-09-29 2023-12-19 重庆大学 一种紧凑的单层差分馈电滤波透明天线
CN114597622A (zh) * 2022-02-25 2022-06-07 南京恒电电子有限公司 双通带平衡滤波耦合器
CN114597622B (zh) * 2022-02-25 2024-04-09 南京恒电电子有限公司 双通带平衡滤波耦合器
CN114824715A (zh) * 2022-03-29 2022-07-29 中国人民解放军国防科技大学 基于矩形微同轴结构的w波段滤波功分器
CN114824715B (zh) * 2022-03-29 2023-09-29 中国人民解放军国防科技大学 基于矩形微同轴结构的w波段滤波功分器
CN115051154A (zh) * 2022-07-27 2022-09-13 重庆邮电大学 一种基于开口阶梯槽的差分宽带端射滤波天线
CN115051154B (zh) * 2022-07-27 2023-07-18 重庆邮电大学 一种基于开口阶梯槽的差分宽带端射滤波天线

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US8284001B2 (en) 2012-10-09
FR2938379A1 (fr) 2010-05-14
EP2184801B1 (de) 2013-07-24

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