EP0673565B1 - ERZEUGUNG EINES KOHERENTSIGNALS UNTER VERWENDUNG EINES ADAPTIVEN FILTERS FüR AUSLOSUNG UND SYNCHRONE DETEKTION IN EINEM DIGITALEN FUNKEMPFäNGER - Google Patents

ERZEUGUNG EINES KOHERENTSIGNALS UNTER VERWENDUNG EINES ADAPTIVEN FILTERS FüR AUSLOSUNG UND SYNCHRONE DETEKTION IN EINEM DIGITALEN FUNKEMPFäNGER Download PDF

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Publication number
EP0673565B1
EP0673565B1 EP93924765A EP93924765A EP0673565B1 EP 0673565 B1 EP0673565 B1 EP 0673565B1 EP 93924765 A EP93924765 A EP 93924765A EP 93924765 A EP93924765 A EP 93924765A EP 0673565 B1 EP0673565 B1 EP 0673565B1
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Prior art keywords
signal
input
coherent
receiver
coupled
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Expired - Lifetime
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EP93924765A
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English (en)
French (fr)
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EP0673565A1 (de
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John Elliott Whitecar
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Ford Motor Co Ltd
Ford Motor Co
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Ford Motor Co Ltd
Ford Motor Co
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04H—BROADCAST COMMUNICATION
    • H04H20/00—Arrangements for broadcast or for distribution combined with broadcast
    • H04H20/44—Arrangements characterised by circuits or components specially adapted for broadcast
    • H04H20/46—Arrangements characterised by circuits or components specially adapted for broadcast specially adapted for broadcast systems covered by groups H04H20/53-H04H20/95
    • H04H20/47—Arrangements characterised by circuits or components specially adapted for broadcast specially adapted for broadcast systems covered by groups H04H20/53-H04H20/95 specially adapted for stereophonic broadcast systems
    • H04H20/48—Arrangements characterised by circuits or components specially adapted for broadcast specially adapted for broadcast systems covered by groups H04H20/53-H04H20/95 specially adapted for stereophonic broadcast systems for FM stereophonic broadcast systems
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D1/00—Demodulation of amplitude-modulated oscillations
    • H03D1/22—Homodyne or synchrodyne circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H04B1/1646—Circuits adapted for the reception of stereophonic signals
    • H04B1/1653—Detection of the presence of stereo signals and pilot signal regeneration

Definitions

  • the present invention relates in general to digital radio receivers employing coherent (i.e., synchronous) detection, and more specifically to the generation of a coherent signal using adaptive notch filter techniques, without use of a phase-locked loop.
  • Coherent detection i.e., synchronous detection
  • This type of demodulation requires that the receiver have a signal available with the same phase and frequency as the original modulating signal used prior to signal transmission.
  • Coherent detection is used to demodulate mono and stereo commercial AM broadcasts and the stereo difference channel in commercial FM stereo broadcasts.
  • the coherent signal used in coherent detection is typically generated using a phase-locked loop (PLL).
  • FM stereo broadcasts include the transmission of a pilot signal which is isolated by the receiver and input to the phase-locked loop.
  • the PLL locks to the carrier signal which is transmitted as the center frequency of the AM broadcast.
  • DSP digital signal processing
  • an analog tuner to generate a signal at a lower frequency (e.g., a demodulated signal or an intermediate frequency signal) which can be represented using a lower sample rate.
  • a standard analog tuner is employed to generate an FM intermediate frequency (IF) of 10.7 MHz which is FM detected to form an analog FM baseband signal.
  • the baseband signal has a frequency range of from 50 Hz to 53 kHz and is digitized in an analog-to-digital converter (ADC).
  • a pilot signal is isolated from the FM baseband signal using a digital bandpass filter. The pilot signal is then input into a digital phase-locked loop for generating a coherent digital signal.
  • Digital phase-locked loops suffer several disadvantages. As in any phase-locked loop, the requirement for a loop filter tends to slow down the locking process of the phase-locked loop and also limits the maximum capture range. Furthermore, a digital phase-locked loop employs excessive amounts of processing time and software code.
  • the present invention has the advantages of faster locking to a desired signal, a wider capture range, more efficient operation, and reduced software code requirements.
  • the invention generates a coherent signal using adaptive notch filter techniques.
  • a radio receiver generating a coherent signal in response to a stereo input signal including a sum channel, a difference channel, and a pilot signal comprises summing means having first and second inputs and an output for generating a notch-filtered signal according to the difference between the signals applied to the first and second inputs.
  • the first input is coupled to receive said stereo input signal and the second input is coupled to receive the coherent signal.
  • a signal source provides a reference signal having a frequency about equal to the input pilot frequency.
  • Adaptive means receives the reference signal and the notch-filtered signal for generating the coherent signal such that the notch-filtered signal is minimized.
  • Synchronous detector means coupled to the summing means and the adaptive means demodulate at least a portion of the notch-filtered signal in response to the coherent signal.
  • the reference signal may be comprised of a pilot signal or an intermediate frequency carrier signal. Minimization of the notched-filtered signal is achieved using a least mean squared (LMS) method.
  • LMS least mean squared
  • Figure 1 is a block diagram showing a prior art radio receiver employing a phase-locked loop.
  • Figure 2 shows the transmission spectrum of an FM stereo multiplex broadcast.
  • Figure 3 is a block diagram showing an adaptive-notched filter for generating a coherent signal for FM stereo reception.
  • Figure 4 is a block diagram showing an adaptive-notched filter generating a coherent signal for use in reception of AM stereo broadcasts.
  • FIG. 5 is a block diagram showing the adaptive line enhancement (ALE) filter of Figure 4.
  • ALE adaptive line enhancement
  • FIG. 1 shows a prior art digital receiver employing a digital phase-locked loop (PLL).
  • An antenna 10 receives broadcast radio signals. These radio frequency (RF) signals are coupled to an analog tuner 11 which generates an intermediate frequency (IF) signal.
  • IF intermediate frequency
  • the IF signal is applied to an FM detector 12 where it is demodulated to generate an analog FM multiplex (MPX) signal at baseband.
  • MPX analog FM multiplex
  • the analog MPX signal is passed through a low-pass filter 13 to an analog-to-digital converter (ADC) 14.
  • ADC analog-to-digital converter
  • Low-pass filter 13 prevents signal aliasing in the digital conversion.
  • a digital FM MPX signal is provided from ADC 14 to a digital signal processing (DSP) block 15.
  • DSP digital signal processing
  • the spectrum of the demodulated FM MPX signal includes a stereo sum channel 30, a stereo difference channel including sidebands 31 and 32, and a stereo pilot signal 33 at a pilot frequency f p of 19 kHz.
  • the stereo difference channel is amplitude modulated to form a double-sideband suppressed-carrier signal.
  • the suppressed carrier is at frequency 2f p (i.e., 38 kHz) which is recovered from the stereo pilot signal by doubling it.
  • a low-pass filter 16 recovers the stereo sum channel from the FM MPX signal and provides an L+R signal to a stereo decoder matrix 17.
  • a pilot bandpass filter 18 isolates the pilot signal from the FM MPX signal and provides the pilot signal to a phase-locked loop (PLL) 20.
  • the filtered pilot signal is coupled to one input of a digital phase comparator 21.
  • the output of phase comparator 21 is coupled through a digital loop filter 22 to a digital voltage-controlled oscillator (VCO) 23.
  • VCO digital voltage-controlled oscillator
  • the output of VCO 23 is coupled to the input of a doubler 24 and to a second input of phase comparator 21.
  • the output from VCO 23 has a phase and frequency equal to the pilot signal but with a constant amplitude and low noise content.
  • the frequency of the VCO signal is doubled in doubler 24.
  • the reconstructed carrier signal is provided to the input of a synchronous detector/mixer 25 for demodulating the stereo difference channel of the FM MPX signal.
  • the demodulated difference channel is passed through a low-pass filter 26 to produce an L-R signal coupled to a second input of matrix 17.
  • the L+R and L-R signals are added and subtracted in matrix 17 to produce right and left stereo signals as is known in the art.
  • DSP block 15 The functions shown in DSP block 15 are implemented using software instructions in a DSP microprocessor.
  • a relatively large amount of software instructions are required in implementing the digital phase comparator, digital loop filter, and digital VCO.
  • memory space for storing the software instructions and data is relatively large and the execution time required in the DSP microprocessor for implementing the phase-locked loop is relatively long.
  • the digital loop filter is required in order to provide stability for the phase-locked loop, its presence lengthens the time required for phase-locking and thereby limits the maximum capture range of the phase-locked loop.
  • the present invention as shown in Figure 3 avoids these difficulties associated with phase-locked loops.
  • the digital FM MPX signal is input to an adaptive-notch filter 40 which is implemented with DSP sottware instructions.
  • a summer 41 receives the digital MPX signal at one input.
  • Adaptive-notch filter 40 attempts to minimize the magnitude of a notched-filter signal output from summer 41 as follows.
  • a reference signal source 42 provides a reference signal approximately equal to the frequency of the signal desired to be locked onto, i.e., the 19 kHz pilot signal.
  • the frequency of the reference signal need only be approximately equal to the pilot frequency since any difference in frequency is compensated for by the adaption of the filter.
  • capture time is improved if an accurate reterence signal is used.
  • Reference source 42 can comprise an oscillator or clock for generating a 19 kHz signal or may be the pilot signal itself derived from a bandpass filter.
  • the reference signal, designated x(i) is input to a multiplier 43 and to a 90 phase shifter 44 to produce a phase-shifted signal designated y(i).
  • Signals x(i) and y(i) can thus be represented as cosine and sine signals, respectively.
  • Signal y(i) is input to a multiplier 45.
  • a set of weights w 1 (i) and w 2 (i) are coupled to the second inputs of multipliers 43 and 45, respectively.
  • the outputs of multipliers 43 and 45 are summed in a summer 47, the output of which provides the coherent cancelling signal which is connected to a subtracting input on summer 41.
  • the notched-filter output from summer 41 provides an error signal e(i) to adaption block 46.
  • Adaption block 46 also receives inputs of x(i) and y(i) for calculating the weight values of weights w 1 (i) and w 2 (i). Weights w 1 and w 2 are adjusted to change the resultant phase of the coherent cancelling signal to match the pilot signal in the FM MPX signal.
  • the coherent signal from summer 47 is coupled to an automatic gain control (AGC) block 50 for adjusting the coherent signal to a peak magnitude equal to one (e.g., by multiplying each coherent signal sample by the reciprocal of an average value of the coherent signal averaged over several cycles).
  • AGC automatic gain control
  • the AGC-adjusted coherent signal is coupled to a frequency doubler 51.
  • the coherent signal is coupled to one input of a multiplier 52 and to the input of a 90 phase shifter 53.
  • the phase-shifted signal is coupled to a second input of multiplier 52. Since the coherent signal is a sinusoidal signal, multiplier 52 performs the multiplication equivalent to a sine times a cosine resulting in an output signal equal to the cosine at twice the frequency of the coherent signal.
  • the frequency-doubled signal is provided to one input of synchronous detector (i.e., mixer) 25.
  • the notch-filtered signal from summer 41 is coupled to the input of low-pass filter 16 and to the first input of synchronous detector 25.
  • the notch-filtered signal has the FM stereo pilot signal stripped away, allowing low-pass filter 15 to be simplified and/or to provide improved performance.
  • FIG. 4 An alternative embodiment is shown in Figure 4 for receiving an AM signal, specifically an AM stereo signal using quadrature encoding.
  • the adaptive notch filter is supplemented with an adaptive line enhancer to improve capture time and capture range.
  • an adaptive line enhancer is employed to provide the reference signal for the adaptive notch filter.
  • the adaptive line enhancer functions as a passband filter with a variable center frequency for enhancing the carrier frequency in the AM IF signal.
  • the AM IF signal from an ADC (not shown) is coupled to the input of summer 41 and to the input of an adaptive line enhancer (ALE) 55.
  • the enhanced carrier frequency signal provides the input signal x(i) to the adaptive notch filter.
  • the AM IF signal is coupled to the input of an in-phase synchronous detector (I-detector) 56.
  • the gain-adjusted coherent signal from AGC block 50 is coupled to a second input of I-detector 56. Synchronous detection generates an in-phase demodulated output I. In monophonic broadcasts, nothing further is required.
  • a quadrature-phase detector (Q-detector) 57 also receives the IF signal.
  • the gain-adjusted coherent signal from AGC block 50 is phase shifted by 90 in a phase shifter 58.
  • the shifted signal is provided to a second input of Q-detector 57 which produces a quadrature output signal Q.
  • output signal Q is the stereo difference signal.
  • the I and Q output signals can then be decoded into stereo signals in a stereo decoder matrix.
  • ALE 55 is shown in greater detail in Figure 5.
  • ALE 55 takes the form of a recursive filter to provide a high Q factor to greatly attenuate the sideband signals in the AM signal.
  • the input AM IF signal a(i) is coupled to one input of a summer 60 and to the input of a unit delay 61.
  • the output of unit delay 61 is coupled to one input of a multiplier 62, the input of a unit delay 70, and one input of an RLS adaption block 63.
  • Adaption block 63 provides a weight w 3 (i) multiplied by a constant "k” to a second input of multiplier 62 which multiplies the product k " w 3 (i) by the unit delayed signals from unit delay 61 and provides the result to one input of a summer 64.
  • the output of unit delay 70 is coupled to one input of a multiplier 65 having its second input receiving a constant weight value w 4 and providing the product to a second input of summer 64.
  • the output of summer 64 is coupled to a subtracting input on summer 60. Further, the output of summer 64 provides the output signal x(i) of ALE 55 to the adaptive notch filter. The output of summer 60 is an error signal which is further coupled to adaption block 63.
  • a recursive portion of ALE 55 includes a unit delay 66 receiving output signal x(i).
  • the unit delayed output of unit delay 66 is coupled to one input of a multiplier 67, the input of a unit delay 68, and to an input of adaption block 63.
  • Adaption block 63 provides weight w 3 (i) to a second input of multiplier 67 which has its output coupled to summer 64.
  • Unit delay 68 has its output coupled to an input of a multiplier 69.
  • a fixed weight w 5 is provided to the second input of multiplier 69 and the product of multiplier 69 is coupled to an input of summer 64.
  • Adaption block 63 performs a recursive least mean squares (RLS) method to minimize the error signal error(i).
  • RLS least mean squares
  • Constant weights w 4 and w 5 determine the Q factor which is preferably about 175, while weight w 3 deter- mines the center frequency.

Claims (7)

  1. Ein digitaler Stereo-Multiplex-Rundfunkempfänger zur Erzeugung eines kohärenten Signals in Reaktion auf ein Stereoeingabesignal, das einen Summenkanal, einen Differenzkanal und ein Pilotsignal einschließt, der Empfänger umfassend:
    eine Summiervorrichtung (41) mit einem ersten und einem zweiten Eingang sowie einem Ausgang zur Erzeugung eines durch einen Schmalband-Sperrfilter geleiteten Signals entsprechend der Differenz zwischen den Signalen, die in diesen ersten und zweiten Eingang eingespeist wurden, wobei dieser erste Eingang gekoppelt ist, um dieses Stereoeingabesignal zu empfangen, und dieser zweite Eingang gekoppelt ist, um das kohärente Signal zu empfangen;
    eine Signalquelle (42), die ein Referenzsignal mit einer Frequenz liefert, die ungefähr gleich dem eingespeisten Pilotsignal ist;
    eine adaptive Vorrichtung (46), die dieses Referenzsignal und dieses durch den Schmalband-Sperrfilter geleitete Signal empfängt, um ein kohärentes Signal so zu erzeugen, daß dieses durch den Schmalband-Sperrfilter geleitete Signal minimiert wird; und
    eine synchrone Detektorvorrichtung (25), die mit dieser Summiervorrichtung (41) und dieser adaptiven Vorrichtung (46) gekoppelt ist, um zumindest einen Teil dieses durch den Schmalband-Sperrfilter geleiteten Signals in Reaktion auf dieses kohärente Signal zu demodulieren.
  2. Ein Empfänger nach Anspruch 1, worin diese adaptive Vorrichtung (46) eine Vielzahl von Gewichtsfaktoren einschließt, die durch Verwendung eines LMS-Verfahrens minimiert werden.
  3. Ein Empfänger nach Anspruch 1 oder 2, worin diese Signalquelle (42) aus einem festen Oszillator besteht.
  4. Ein Empfänger nach Anspruch 1 oder 2, worin diese Signalquelle aus einem variablen Bandpaß (55) besteht, der zum Empfangen dieses Eingabesignals gekoppelt ist.
  5. Ein Empfänger nach Anspruch 4, worin dieser variable Bandpaß (55) aus einem adaptiven Leitungsverstärkungsfilter besteht.
  6. Ein Empfänger nach irgendeinem der vorhergehenden Ansprüche, worin die Signalquelle (42) erste und zweite Signale liefert, deren Frequenz jeweils im wesentlichen gleich dem Pilotsignal ist, wobei diese ersten und zweiten Signale im wesentlichen zueinander in Phasenquadratur stehen;
    wobei die adaptive Vorrichtung (46) so angeordnet ist, daß sie ausgewählte Gewichtsfaktoren auf diese ersten und zweiten Signale anwendet; und
    ferner eine Summiervorrichtung (47) zum Summieren der gewichteten ersten und zweiten Signale bereitgestellt ist, um dieses kohärente Signal zu erzeugen.
  7. Ein Empfänger nach irgendeinem der vorhergehenden Ansprüche, worin dieser Demodulierer einen Frequenzverdoppler (51) zum Verdoppeln der Frequenz dieses kohärenten Signals umfaßt sowie einen Mixer (25), um dieses verdoppelte Signal und dieses durch den Schmalband-Sperrfilter geleitete Signal zu mischen.
EP93924765A 1992-12-14 1993-11-15 ERZEUGUNG EINES KOHERENTSIGNALS UNTER VERWENDUNG EINES ADAPTIVEN FILTERS FüR AUSLOSUNG UND SYNCHRONE DETEKTION IN EINEM DIGITALEN FUNKEMPFäNGER Expired - Lifetime EP0673565B1 (de)

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
US07/990,143 US5357574A (en) 1992-12-14 1992-12-14 Coherent signal generation in digital radio receiver
PCT/GB1993/002342 WO1994014246A1 (en) 1992-12-14 1993-11-15 Generation of a coherent signal using an adaptive filter for cancelling and synchronous detection in a digital radio receiver
US990143 1997-12-12

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EP0673565A1 EP0673565A1 (de) 1995-09-27
EP0673565B1 true EP0673565B1 (de) 1996-09-18

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US (2) US5357574A (de)
EP (1) EP0673565B1 (de)
JP (1) JP3397792B2 (de)
AU (1) AU677256B2 (de)
CA (1) CA2151715A1 (de)
DE (1) DE69304933T2 (de)
WO (1) WO1994014246A1 (de)

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AU677256B2 (en) 1997-04-17
EP0673565A1 (de) 1995-09-27
DE69304933T2 (de) 1997-02-06
US5467399A (en) 1995-11-14
JPH08504549A (ja) 1996-05-14
CA2151715A1 (en) 1994-06-23
AU5431094A (en) 1994-07-04
JP3397792B2 (ja) 2003-04-21
WO1994014246A1 (en) 1994-06-23
DE69304933D1 (de) 1996-10-24
US5357574A (en) 1994-10-18

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