CN1520669A - 在ofdm接收机中补偿载波频率偏移的方法和系统 - Google Patents
在ofdm接收机中补偿载波频率偏移的方法和系统 Download PDFInfo
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Abstract
一种正交频分多址(OFDM)接收机,使用N个二阶锁相环(62-66,74-78),共享一个公用的积分器(72),(其中,N是系统中导频的数目)。N个二阶锁相环(62-66,74-78)跟踪出独立相位误差,以便所述导频相位信息的建设性平均。同时,通过共享公用的积分器(72),OFDM接收机利用了在多个导频之间对噪音进行平均的好处,以便获得比较干净的频率偏移估计值。OFDM接收机还可以通过计算(84)选定一对导频的相位差,并跟踪算出的相位差随着时间的变化速率来补偿FFT窗口漂移。算出的相位差用来在超过预定相位差阈值之后控制上游FFT窗口的位置。跟踪的变化速率用来连续调整下游均衡器引出线的相位。
Description
本发明涉及正交频分多址(OFDM)信号的处理。
无线局域网(WLAN)是一种作为在建筑物或校园内有线局域网的延伸或替代而实现的灵活的数据通信系统。利用电磁波,WLAN在空中发送和接收数据,减少了导线连接的必要性。因而,WLAN把数据连接与用户的移动性结合起来,通过简化配置,使移动局域网成为可能。已经利用便携式终端(例如笔记本电脑)来传送和接收实时数据而从提高生产效率中得到好处的某些行业是数字家用网络、医疗护理、零售业、制造业和仓储业。
WLAN制造厂家在设计WLAN时有若干种传输技术可供选择。某些示例性技术是多载波系统、扩频系统、窄带系统和红外系统。尽管每种系统都各有利弊,但是一种特定类型的多载波系统,正交频分多址(OFDM),已经证明是对无线局域网异常有用的。
对于通过一个信道有效地传输数据而言,OFDM是一种健壮的技术。该技术在传输数据的信道频带内采用多个子载波频率(子载波)。为了分开和隔离子载波频谱,从而避免载波间干扰(ICI),传统的频分多址(FDM)可能浪费一部分信道频带,与此相比,OFDM的这些子载波是为了最优频带效率而安排的。相反,尽管OFDM子载波的频谱在OFDM信道频带内明显重叠,但是OFDM仍旧允许调制在每一个子载波上的信息被分辨和恢复。
与传统的传输技术相比,借助于OFDM信号通过信道传输数据还提供其他几个优点。这些优点中的一些是允许多路径延迟扩频和频率选择性衰落、有效利用频谱、简化子信道均衡和良好的干扰特性。
尽管OFDM表现出这些优点,但是OFDM传统的实现也表现出几个困难和实际限制。一个困难是确定和校正载波频率偏移的问题,OFDM同步的一个主要方面。理想的是,接收载波频率fcr,应该准确匹配发射载波频率fct。但若不满足这个条件,则不匹配就会在接收的OFDM信号中造成非零的载波频率偏移Δfc。OFDM信号对这样的载波频率偏移非常敏感,这会造成OFDM子载波正交性的损失,并导致载波间干扰(ICI)和在接收机上恢复数据位误码率(BER)的严重增大。
许多OFDM标准要求在用户数据中嵌入导频(已知值)的发送。在传统的OFDM系统中,一般对导频相位信息进行平均,以便在噪音环境中改善闭环载波频率偏移跟踪。例如,导频相位平均可以用来产生载波频率偏移估计,而后者本身又可用以调整均衡器引出线的相位旋转,以便减小以致消除载波频率偏移的影响。这种技术的一个缺点是,在时变信道存在的情况下,导频的相位可以独立变化。更具体地说,所有导频相位共享一个,如上所述,由发射机载波频率和接收机载波频率之间不匹配造成的公共的代表载波频率偏移的相位旋转。但是,时变信道存在时,每一个导频相位也可以包含一个由随时间而变化的发射信道引起的独立的相位旋转。这个独立的导频相位旋转潜在地造成导频相位破坏性平均,而后者本身又会破坏载波频率偏移估计的推算。被破坏的载波频率偏移估计可能降低使用该估计值补偿实际载波频率偏移的任何处理单元(例如,均衡器)的性能。本发明旨在校正这个问题。
接收机采样时钟的频率还可能略微不同于发射机采样时钟的频率。若有频率差异,则FFT窗口相对于接收信号的定位可能随着时间逐渐漂移。这个时域漂移会造成所接收的OFDM子载波在频域上的相位旋转。这种相位旋转可以在由OFDM接收机恢复的用户数据中产生差错。本发明也旨在校正这个问题。
正交频分多址(OFDM)接收机使用共享一个公用积分器的N个二阶锁相环(其中N是系统中导频的数目)。N个二阶锁相环跟踪出独立的导频相位旋转,以有助于导频相位信息的建设性平均。同时,通过共享一个公用的积分器,OFDM接收机利用了在多个导频间进行噪音平均的好处,以便获得比较干净的频率偏移的估计值。OFDM接收机还可以通过计算选中的一对导频之间的相位差并跟踪计算相位差随着时间的变化速率来补偿FFT窗口的漂移。算出的相位差用来控制超过预定相位差阈值后上游FFT窗口的位置。所跟踪的改变速率用来连续调整下游均衡器引出线的相位。
图1是示例性OFDM接收机的方框图;
图2是举例说明按照本发明的OFDM符号帧内训练序列、用户数据、和导频信号的放置示意图;
图3是举例说明按照本发明的OFDM接收机的载波频率偏移补偿系统的方框图;
图4是举例说明本发明与图1的示例性OFDM接收机的集成的方框图;
图5和6是公共的、独立的和调整后的相位误差的示意图。
从以下以举例方式给出的描述中,本发明的特征和优点将变得明显。
参见图1,典型OFDM接收机10的第一个元件是RF(射频)接收机12。RF接收机12存在许多变化,而且在先有技术中是众所周知的,但一般RF接收机12包括天线14、低噪音放大器(LNA)16、RF带通滤波器18、自动增益控制(AGC)电路20、RF混频器22、RF载波频率本机振荡器24和IF(中频)带通滤波器26。
RF接收机12通过天线14在RF OFDM调制的载波通过信道后与其连接。然后,将其与由RF本机振荡器24产生的频率fcr的接收机载波混频,RF接收机12把RF OFDM调制的载波降频变换,获得接收的IF OFDM信号。接收机载波和发射机载波之间的频率差造成载波频率偏移Δfc。
接收的IF OFDM信号连接到混频器28和混频器30,分别与同相IF信号及90°相移(正交)IF信号混频,以产生同相和正交OFDM信号。馈入混频器28的同相IF信号是由IF本机振荡器32产生的。馈入混频器30的90°相移(正交)IF信号是由IF本机振荡器32的同相IF信号在提供给混频器30之前通过90°相移器34而产生的。
然后同相和正交OFDM信号分别进入模数转换器(ADC)36和38,在这里以时钟电路40决定的采样速率fck_r数字化。ADC 36和38产生分别构成同相和正交离散时间OFDM信号的数字化样本。接收机的采样速率和发射机的采样速率之间的差值是采样速率偏移ΔfCK=fck_r-fck_t。
然后令来自ADC36和38的未经滤波的同相和正交离散时间OFDM信号分别通过数字低通滤波器42和44。低通滤波器42和44的输出分别为接收的OFDM信号滤波后的同相和正交样本。这样,所接收的OFDM信号转换为同相(qi)和正交(pi)样本,分别代表复数OFDM信号的实数值和虚数值分量,ri=qi+jpi。这些接收的OFDM信号的同相和正交(实数值和虚数值)样本的送到FFT 46。注意,在接收机10的某些传统的实施例中,模数转换在IF混频过程之前进行。在这样的实现中,混频过程涉及数字混频器和数字频率合成器的使用。还应注意,在接收机10的许多传统的实现中,在滤波之后进行数模转换。
为了恢复在每个OFDM符号间隔过程中用来调制子载波的频域子符号的序列,FFT 46完成接收OFDM信号的快速富里叶转换(FFT)。然后,FFT 46把这些子符号的序列送到译码器48。
译码器48从FFT 46传送给它的频域子符号序列恢复所发送的数据位。这个恢复过程是通过频域子符号的译码进行的,以便获得理想地匹配馈送给OFDM发射机的数据位流。这个译码过程可以包括软维特比码和/或里德-索罗蒙码,例如,以便从块和/或卷积编码的子符号恢复数据。
回到图2,其中示出本发明的示例性OFDM符号帧50。符号帧50包括训练序列或符号52,其中包含对OFDM载波中的每个子载波的已知发射值和预定数目的循环前缀54和用户数据56对。例如,所提出的ETSI-BRAN HIPERLAN/2(欧洲)和IEEE 802.11a(美国)无线电标准,结合在此作参考,指定64个已知值或子符号(亦即52个非零值和12个零值)给训练序列(亦即,建议的IEEE标准的建议ETSI标准的“训练符号C”和“长OFDM训练符号”)的选定训练符号。用户数据56具有预定数目的导频58,还包含已知的发送值,嵌入在预定的子载波中。例如,建议的ETSI和IEEE标准具有4个导频,位于仓(bin)或子载波±7和±21中。尽管本发明是针对符合建议的ETSI-BRAN HIPERLAN/2(欧洲)和IEEE802.11a(美国)无线局域网标准的接收机的操作描述的,但在其他OFDM系统中实现本发明传授技术是在本专业的技术人员能力范围之内。
现参见图3,其中示出本发明的示例性实施例。尽管本发明举例说明为不同于图1的OFDM接收机的的元件,但如图4所示和下面将要讨论的,本专业的技术人员不难想出本发明可与OFDM接收机的元件集成。但是,为清晰和便于引用起见,并为了便于理解本发明,本发明举例说明为独特的载波频率偏移补偿系统。
现参见图3,其中示出载波频率偏移补偿系统60。应该注意系统60可用软件、硬件以及它们的组合实现。系统60包括共享一个积分器的多个二阶锁相环。正如下面将要更详细讨论的,多个锁相环使消除因时变信道引起的独立的导频相位误差(亦即,相位旋转)成为可能,从而便于导频相位信息的建设性平均,以产生载波频率偏移估计。还应该注意到,共享一个公共的积分器,利用了多个导频的噪音平均的优点来产生比较干净的载波频率偏移估计。
更具体地说,有N个二阶锁相环(PLL),其中N代表由系统60处理的导频数目。每个二阶PLL都包括一个反旋转器或复数乘法器62、相位误差检测器64、比例增益级66、求和器74、数字控制的振荡器(NCO)76和Sin/Cos查找表78。各二阶PLL还共享一个平均单元68、积分增益级70和连接在每一个PLL的相位检测器64和求和器74之间的积分器72。Sin/Cos查找表80可以连接到积分器72的输出和均衡器82的输入。此外,相位差计算器84、比较器86和FFT窗口偏差校正器88电路可以连接到二阶PLL的NCO的输出,正如下面将要更详细讨论的。
在操作中,用户数据段56的每一个导频58由单独的PLL处理,并与用户数据段56的其他导频58平均。更具体地说,每一个反旋转器62都把所接收的导频乘以复数(代表独立相位误差校正),以驱使独立的相位误差趋向于零。
每个反旋转器62都把处理后的导频送到相位误差检测器64。每个相位误差检测器64都产生一个导频相位误差。产生相位误差的一个示例性途径是计算已知导频理想相位和所接收的导频实测相位之间的差值。其他相位误差信号产生技术的使用,正如本专业的技术人员已知的,是在本发明的范围内的。在每个PLL中,把相位误差送到相关的比例增益级66,以及共享的平均单元68。每个比例增益级66都对所接收的相位误差(代表导频独立的相位旋转)进行缩放至预定的每个PLL的相关NCO 76可用的增量。平均单元68对给定用户数据段中所有导频接收的相位误差值进行平均,并将该平均误差(代表给定用户段所有导频的平均相位旋转)送到积分增益级70。正如下面将要更详细讨论的,积分增益级70对平均相位误差缩放到预定的每一个NCO 76以及Sin/Cos查找表80可用的增量。积分器72对从积分增益级70接收的缩放后的平均相位误差进行积分,并输出代表多个用户数据段中所有导频的公共相位误差的积分缩放后的相位误差。应该注意,正如下面将要讨论的,在达到某个相位锁定之前,积分缩放后的相位误差的一部分是由于导频独立的相位旋转引起的。
每个PLL的求和器74求出从相关比例增益级66接收的独立相位误差和从积分器72接收的公共相位误差之和。所得数值若通过由时变信道引起的独立导频相位旋转进行调整,则代表所有导频的公共相位误差。现参见图5和6,针对两个PLL示出了公共相位误差、独立相位误差和调整后的相位误差。
由每个求和器74输出的调整后的相位误差被送到相关的NCO 76,它对所接收的相位误差随着时间进行累计。给定PLL的每个查找表78都把相关NCO 76的输出转换为相位复矢量(phasor)。相位复矢量送回相关的反旋转器62,后者把下一个接收的导频乘以相位复矢量以旋转导频,使得独立相位误差趋向于零。
除了二阶锁相环之外,Sin/Cos查找表也连接到积分器72,以便在预定的锁定状态之后把积分器72的输出转换为载波频率偏移估计值,后者由均衡器82用来调整均衡器引出线的相位旋转。应该注意,载波频率偏移估计值可以送到其他处理单元(未示出),用以校正载波频率偏移。一个示例性的锁定条件是积分器72的输出在预定的时间周期内落在预定范围内。这种条件的出现指示PLL已经达到稳态,而独立相位误差已经消除。应该注意,最好用单个分时PLL代替图3所示的多个PLL处理导频,并从导频消除独立相位误差。
如上所述,接收机采样时钟的频率可能略微不同于发射机采样时钟的频率。若有频率差,则FFT窗口相对于接收的信号的定位可能随着时间逐渐漂移。FFT窗口漂移将造成所接收的OFDM子载波的相位旋转。相位旋转可能在由OFDM接收机恢复的用户数据中产生误差。相位差计算器84、相位差估计器86的速率和FFT窗口同步单元88电路旨在补偿和校正FFT窗口偏移。
更具体地说,相位差计算器84计算来自给定一对NCO 76输出的值之间的差值。这个差值等于用户数据段中给定一对导频子载波之间的相位差。应该注意,NCO值在达到±pi之后将会回卷(ROLL OVER)。因此,相位差计算器84跟踪每个NCO 76数值超过±pi的次数,以便精确计算给定一对NCO 76输出数值之间的相位差。算出的相位差送到相位差比率估计器86和FFT窗口同步单元88。FFT窗口同步单元88比较算出的相位差和阈值相位差(例如,代表一个样本FFT窗口偏移的相位差),并若算出的相位差超过阈值相位差,则控制上游FFT窗口的位置(例如,窗口偏移一个样本)。这样,当算出的相位差超过预定的阈值相位差时,上游FFT的FFT窗口偏移量可周期地进行修正。估计器86跟踪多个用户数据段之间算出的相位误差变化速率。估计器86产生均衡器调整值,它可以(例如,通过求和器74)与积分器72的输出结合,以便针对给定一对NCO 76输出的值之间相位差的连续变化,补偿从Sin/Cos查找表输出的载波频率偏移。这样,下游均衡器的均衡器引出线可以连续旋转,以便补偿逐渐漂移的FFT窗口。
现参见图4,如图所示,本发明与图1的示例性OFDM接收机集成。更具体地说,系统60连接到FFT 46的输出和补偿载波频率误差(例如,图3的均衡器82和/或前端频率偏移校正装置(未示出))的处理装置的输入。另外,系统60的输出反馈回FFT 46。采用这样的电路,系统60从由FFT 46接收的OFDM样本提取导频,并产生没有因时变信道引起的独立相位误差的频率偏移量估计值。系统60还处理所提取的导频,以便补偿(例如,在均衡器82中)和校正(例如,在FFT 46中)FFT窗口的漂移。
Claims (20)
1.一种处理正交频分多址(OFDM)接收机中OFDM信号用的方法,所述方法包括以下步骤:
接收OFDM信号;
从所接收的OFDM信号中提取多个导频;
处理所提取的多个导频,以便减少独立相位误差,所述独立相位误差代表时变信道引起的独立导频相位旋转(62-78);以及
在所述独立相位误差减小到预定的数值后,从所述多个处理后的导频产生载波频率偏移估计值(80)。
2.权利要求1的方法,其特征在于,它还包括以下步骤:
利用载波频率偏移估计值校正载波频率偏移(82)。
3.权利要求1的方法,其特征在于,所述处理步骤包括让多个导频中的每一个通过独特的锁相环(62-66,74-78)。
4.权利要求1的方法,其特征在于,所述处理步骤包括让多个导频中的每一个通过时分锁相环。
5.权利要求1的方法,其特征在于,所述产生载波频率偏移估计值的步骤包括以下步骤:
计算多个导频的公共相位误差,所述公共相位误差代表由载波频率偏移引起的平均导频相位旋转(68,70);
对公共相位误差进行积分,以便减小噪音的影响(72);以及
从积分的公共相位误差产生载波频率偏移估计值(80)。
6.权利要求1的方法,其特征在于,所述预定值基本上为零。
7.权利要求1的方法,其特征在于,所述预定值选择得使所产生的载波频率偏移估计值基本上恒定。
8.权利要求1的方法,其特征在于,还包括以下步骤:
计算预定一对所提取的导频之间的相位差(84);
把所算出的相位差与阈值比较,所述阈值代表预定FFT窗口的偏移量;以及
若算出的相位差超过阈值,则调整上游FFT模块的FFT窗口(88)。
9.权利要求8的方法,其特征在于,所述产生载波频率偏移估计值的步骤还包括以下步骤:
跟踪算出的相位差的变化速率(86);以及
调整所产生的载波频率偏移估计值,来补偿跟踪的变化速率,所跟踪的变化速率代表因FFT窗口偏移造成的相位旋转(74)。
10.权利要求1的方法,其特征在于,所述OFDM接收机在无线局域网适配器、家用网络终端、便携式终端和台式终端中的一种中实现。
11.一种处理正交频分多址(OFDM)信号的系统,所述系统包括:
处理模块(62-78),用以从OFDM信号接收多个导频,并减少每个导频的独立相位误差,所述独立相位误差代表时变信道引起的独立导频相位旋转;和
载波频率偏移估计器模块(80),连接到所述处理模块,所述载波频率偏移估计器模块在每个导频的独立相位误差减小到预定值后从所述多个导频产生载波频率偏移估计值。
12.权利要求11的系统,其特征在于,所述OFDM接收机在无线局域网适配器、家用网络终端、便携式终端和台式终端中的一种中实现。
13.权利要求11的系统,其特征在于,所述预定值基本上为零。
14.权利要求11的系统,其特征在于,所述预定值选择得使所产生的载波频率偏移估计值基本上恒定。
15.权利要求11的系统,其特征在于,所述处理模块包括多个锁相环(62-66,74-78),每个锁相环专门用于用户数据段内的预定的导频。
16.权利要求11的系统,其特征在于,所述处理模块包括时分锁相环,为处理用户数据段内所有导频而锁定。
17.权利要求11的系统,其特征在于,所述载波频率偏移估计器模块包括:
平均单元(68),用以计算多个导频的公共相位误差,所述公共的相位误差代表由载波频率偏移引起的平均导频相位旋转;
积分单元(72),对所述公共相位误差进行积分,以便减小噪音的影响。
18.权利要求11的系统,其特征在于,所述处理模块还包括:
相位差计算器(84),计算用户数据段内两个预定的导频之间的相位差;和
FFT窗口同步单元(88),若算出的相位差超过预定值,则用以调整上游FFT模块的定位。
19.权利要求18的系统,其特征在于,所述载波频率偏移模块还包括:
相位差的变化速率估计器(86),连接到相位差计算器(84),用以跟踪算出的相位差在多个用户数据段之间的变化;所述估计器(86)针对因漂移的FFT窗口造成的相位旋转,补偿所产生的载波频率偏移。
20.一种正交频分多址(OFDM)信号的系统,所述系统包括:
从OFDM信号提取多个导频用的装置;
从所提取的导频消除(62-78)独立相位误差用的装置,所述独立相位误差代表因时变信道造成的独立相位旋转;
处理所提取的导频以便补偿FFT窗口漂移用的装置(84,88);
在把每个导频的独立相位误差减小到预定水平和补偿FFT窗口漂移之后,从多个导频产生载波频率偏移估计。
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CN105516043A (zh) * | 2015-11-26 | 2016-04-20 | 中国电子科技集团公司第三十研究所 | 一种基于ofdm的多载波通信系统频偏估计方法及装置 |
CN105516043B (zh) * | 2015-11-26 | 2018-11-16 | 中国电子科技集团公司第三十研究所 | 一种基于ofdm的多载波通信系统频偏估计方法及装置 |
CN108289071A (zh) * | 2018-01-03 | 2018-07-17 | 深圳市极致汇仪科技有限公司 | 一种相位跟踪方法及相位跟踪系统 |
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CN113645169A (zh) * | 2020-05-11 | 2021-11-12 | 大唐移动通信设备有限公司 | 正交频分复用多载波系统载波相位跟踪方法及装置 |
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US20040156349A1 (en) | 2004-08-12 |
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BRPI0117039B1 (pt) | 2015-12-08 |
WO2003001760A1 (en) | 2003-01-03 |
BR0117039A (pt) | 2004-07-27 |
JP2004531168A (ja) | 2004-10-07 |
EP1400083B1 (en) | 2010-05-05 |
US8184523B2 (en) | 2012-05-22 |
KR100802973B1 (ko) | 2008-02-14 |
EP1400083A1 (en) | 2004-03-24 |
KR20040014571A (ko) | 2004-02-14 |
JP4864286B2 (ja) | 2012-02-01 |
DE60142082D1 (de) | 2010-06-17 |
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