WO2022183537A1 - 五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法 - Google Patents

五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法 Download PDF

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WO2022183537A1
WO2022183537A1 PCT/CN2021/082038 CN2021082038W WO2022183537A1 WO 2022183537 A1 WO2022183537 A1 WO 2022183537A1 CN 2021082038 W CN2021082038 W CN 2021082038W WO 2022183537 A1 WO2022183537 A1 WO 2022183537A1
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fault
phase
tolerant
permanent magnet
motor
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PCT/CN2021/082038
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English (en)
French (fr)
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张丽
朱孝勇
徐磊
张超
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江苏大学
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Priority to US18/013,568 priority Critical patent/US11888419B2/en
Priority to GB2200428.7A priority patent/GB2606611B/en
Publication of WO2022183537A1 publication Critical patent/WO2022183537A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/06—Rotor flux based control involving the use of rotor position or rotor speed sensors
    • H02P21/10—Direct field-oriented control; Rotor flux feed-back control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H02P21/0017—Model reference adaptation, e.g. MRAS or MRAC, useful for control or parameter estimation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H02P25/22—Multiple windings; Windings for more than three phases
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/028—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the motor continuing operation despite the fault condition, e.g. eliminating, compensating for or remedying the fault
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

Definitions

  • the invention belongs to the technical field of multi-phase motor fault-tolerant control, and in particular relates to an open-circuit unified fault-tolerant control method for a five-phase permanent magnet fault-tolerant motor vector and direct torque control drive system.
  • Multi-phase permanent magnet motors have the advantages of high efficiency, high power density, wide speed regulation range, low torque ripple and strong fault tolerance, and have been widely concerned and applied in electric vehicles, hybrid vehicles and other electric drive fields.
  • Vector control (VC) and direct torque control (DTC) have attracted much attention due to their high driving performance.
  • VC Vector control
  • DTC direct torque control
  • the failure of the motor drive system will affect the normal operation of the entire application system, and even cause a safety accident. Therefore, it is of great practical significance to study the fault-tolerant control of the multi-phase permanent magnet motor drive system and improve the reliability of the motor drive system.
  • the document "Fault-tolerant direct torque control of five-phase FTFSCW-IPM motor based on analogous three-phase SVPWM for electric vehicle applications” (IEEE Transaction on Vehicular Technology, 2017) aims at five One-phase open-circuit fault of phase permanent magnet motor, the corresponding SVPWM fault-tolerant control method is proposed, which overcomes the shortcomings of the fault-tolerant control system based on hysteresis control.
  • this type of fault-tolerant control strategy is based on the voltage vector reconstruction after the fault, and the algorithm is relatively complex , is not conducive to engineering.
  • the present invention proposes an open-circuit unified fault-tolerant control method based on a CPWM-based five-phase permanent magnet fault-tolerant motor vector and direct torque control drive system, starting from essentially revealing the fault-tolerant mechanism. Improve the stable operation ability of the motor drive system, so that the control system not only has good open-circuit fault fault-tolerant operation performance, but also has good dynamic and static performance, anti-interference ability and robustness, and can be applied to various new control algorithms. Minimize reconfiguration of the control system and save controller CPU memory resources under different faults.
  • the technical solution adopted in the present invention is: an open-circuit unified fault-tolerant control method for a five-phase permanent magnet fault-tolerant motor vector and direct torque control drive system, comprising the following steps:
  • Step 1) establishing the current mathematical model of the five-phase permanent magnet fault-tolerant motor during normal operation
  • the five-phase permanent magnet fault-tolerant motor runs under normal operating conditions, and the expressions of its currents i A , i B , i C , i D and i E are as follows:
  • ⁇ 72°
  • ⁇ is the rotor position electrical angle
  • i d1 * , i q1 * are the direct-axis and quadrature-axis components of the fundamental current given value
  • id3 * , i q3 * are the third harmonic current given values Direct axis and quadrature axis components
  • Step 2) Obtain the fault-tolerant current of one-phase open circuit, non-adjacent two-phase open circuit and adjacent two-phase open circuit, and analyze the fault-tolerant mechanism;
  • Step 3 Use the difference between the given speed and the detected actual speed to construct a high-quality torque controller to obtain a high-quality torque given T e * , so as to suppress the motor torque ripple after the fault, and at the same time, the load disturbance, the system Parameter changes and electromagnetic torque ripple factors caused by faults are taken into account;
  • Step 4) Detect the five-phase currents i A , i B , i C , i D and i E of the five-phase permanent magnet fault-tolerant motor, and obtain the current components i d1 , i q1 , i q1 , i d3 , i q3 ;
  • the current component in the two-phase rotating coordinate system of the five-phase permanent magnet fault-tolerant motor is expressed as:
  • Step 5 Utilize the current components i d1 , i q1 and the high-quality torque given T e * under the two-phase rotating coordinate system to calculate the quadrature-axis fundamental wave voltage of the vector control VC drive system and the direct torque control DTC drive system ;
  • Step 6 Based on the third harmonic current zero control strategy, use the current components i d3 and i q3 under the two-phase rotating coordinate system to obtain the third harmonic voltage of the AC-direction axis;
  • Step 7) Calculate the phase voltage of the winding under the fault mode based on the fault-tolerant mechanism and the AC-DC axis voltage;
  • Step 8) According to the back EMF of the motor and the winding phase voltage in the fault mode, obtain the given phase voltage of the five-phase winding in the fault-tolerant operation mode;
  • Step 9) The five-phase winding is given a five-phase voltage command through the voltage source inverter, and the pulse width modulation CPWM mode is adopted to realize the disturbance-free operation of the five-phase permanent magnet fault-tolerant motor VC and the DTC drive system under any open-circuit fault condition.
  • the present invention proposes a unified fault-tolerant control strategy for open-circuit faults suitable for VC and DTC drive systems for the first time, which essentially reveals the fault-tolerant mechanism.
  • the fault-tolerant operation of the system in different open-circuit fault states is realized, thereby solving the problem that the fault-tolerant control schemes corresponding to various new basic control algorithms will also change and become complicated.
  • the invention is based on the design of high-quality torque controller, so that the control system still has high-quality output torque even under open-circuit fault, and at the same time has good dynamic and static performance, anti-interference ability and robustness under normal and fault operation.
  • the performance of the motor drive system is comprehensively improved.
  • the present invention realizes the disturbance-free operation in the case of open-circuit fault based on the pulse width modulation CPWM method, which can effectively solve the problems of high current harmonic content, large torque ripple and low switching frequency caused by the traditional fault-tolerant control method based on hysteresis comparator.
  • the traditional fault-tolerant control is generally based on the technical restraint of ensuring that the magnetomotive force before and after the fault is equal to the fundamental wave component of the AC-DC axis current, and the action mechanism of the third-harmonic AC-DC axis current before and after the fault is considered to ensure that the stator flux linkage track is a circle At the same time, the current quality under fault-tolerant operation is further improved.
  • the current component is zero, which solves the problem of large loss of the motor under sudden load or fault operating conditions, and effectively improves the efficiency of the motor drive system.
  • the proposed method has a small amount of calculation, is simple and easy to implement, and is beneficial to the engineering and practical application of the new theory.
  • FIG. 1 is a schematic structural diagram of a five-phase permanent magnet fault-tolerant motor of the present invention
  • FIG. 2 is a schematic block diagram of a unified fault-tolerant control strategy for open-circuit faults according to the present invention
  • Figure 3 shows the current vector diagrams in normal and fault-tolerant operation;
  • (a) is the current vector diagram in normal operation;
  • (b) is the current vector diagram in the case of an open-circuit fault of phase A;
  • (c) is the non-adjacent phase A , the current vector diagram of C two-phase open-circuit fault;
  • (d) is the current vector diagram of the adjacent phase A, B two-phase open-circuit fault;
  • Fig. 4 is the structural block diagram of the fault-tolerant control drive system of five-phase permanent magnet fault-tolerant motor based on CPWM of the present invention; (a) fault-tolerant VC drive system; (b) fault-tolerant DTC drive system;
  • Fig. 5 is the simulation waveform of the DTC drive system when the load and system parameters are changed under normal working conditions of the present invention; (a) current waveform; (b) torque waveform; (c) rotational speed waveform; (d) stator flux linkage amplitude value waveform; (e) AC and direct axis current waveform;
  • Fig. 6 is the simulation waveform of the VC drive system of the present invention under normal operation, A, C two-phase open-circuit fault without fault-tolerant operation and fault-tolerant operation; (a) current waveform; (b) torque waveform; (c) rotational speed waveform ;
  • Fig. 7 is the simulation waveform of the DTC drive system of the present invention under normal operation, A and C two-phase open-circuit fault without fault-tolerant operation and fault-tolerant operation; (a) current waveform; (b) torque waveform; (c) rotational speed waveform ; (d) Amplitude waveform of stator flux linkage;
  • FIG. 8 is a simulation waveform of the DTC drive system of the present invention under normal operation, non-fault-tolerant operation under A-phase open-circuit fault, and fault-tolerant operation.
  • Fig. 9 is the simulation waveform of the DTC drive system of the present invention under normal operation, A and B two-phase open-circuit fault without fault-tolerant operation and fault-tolerant operation; (a) current waveform; (b) torque waveform; (c) rotational speed waveform .
  • the control object of the present invention is a schematic structural diagram of a five-phase permanent magnet fault-tolerant motor, including a stator, a rotor, a permanent magnet, an armature tooth, a fault-tolerant tooth, and an armature winding;
  • the circumferential intervals of the rings are evenly distributed, and the tooth width of the armature teeth and the tooth width of the fault-tolerant teeth are not equal;
  • the armature teeth are wound with armature winding coils, which are single-layer concentrated windings, and between two adjacent single-layer concentrated windings It is isolated by fault-tolerant teeth;
  • the permanent magnets are embedded in the rotor, which is distributed in a "V" shape; the total number of teeth of the armature teeth and fault-tolerant teeth is 20, and the number of poles of the permanent magnet is 18; because the stator adopts a single-layer concentrated winding, it is greatly reduced.
  • the magnetic resistance of the motor's direct-axis magnetic circuit reduces the salient pole rate of the motor, making the inductance of the motor's direct-axis nearly equal; the stator part is provided with fault-tolerant teeth, which can effectively realize the electrical, magnetic and thermal isolation between the windings of each phase, and has a strong Fault tolerance performance.
  • the principle block diagram of the unified fault-tolerant control strategy for open-circuit faults of the present invention includes obtaining high-quality torque reference, obtaining AC and DC axis reference voltage, analyzing the fault-tolerant mechanism to obtain fault-tolerant current, and obtaining fault winding voltage based on the fault-tolerant mechanism. Based on the back EMF to obtain the fault tolerance voltage.
  • the specific implementation steps of the open-circuit fault unified fault-tolerant control strategy principle of the present invention include:
  • Step 1) Establish a current mathematical model of the five-phase permanent magnet fault-tolerant motor during normal operation.
  • the five-phase permanent magnet fault-tolerant motor runs under normal operating conditions, and its current expression is as follows:
  • ⁇ 72°
  • ⁇ is the rotor position electrical angle
  • i d1 * , i q1 * are the direct-axis and quadrature-axis components of the fundamental current given value
  • id3 * , i q3 * are the third harmonic current given values Direct and quadrature axis components.
  • Step 2 Obtain fault-tolerant currents in different fault modes (one-phase open circuit, non-adjacent two-phase open circuit, and adjacent two-phase open circuits), and analyze the fault-tolerant mechanism.
  • the magnetomotive force expression of the five-phase permanent magnet fault-tolerant motor under normal operating conditions is:
  • MMF 1 Ni A + ⁇ Ni B + ⁇ 2 Ni C + ⁇ 3 Ni D + ⁇ 4 Ni E (2)
  • ⁇ cos ⁇ +jsin ⁇ ; N is the number of winding turns; i A , i B , i C , i D , i E are the A, B, C, D, and E phase currents of the motor during normal operation.
  • MMF 2 ⁇ Ni B1 + ⁇ 2 Ni C1 + ⁇ 3 Ni D1 + ⁇ 4 Ni E1 (3)
  • i B1 , i C1 , i D1 , i E1 are the B, C, D, and E phase currents when the A phase fails.
  • the fault-tolerant current under the open-circuit fault of the adjacent two-phase and non-adjacent two-phase can be calculated separately. It is worth noting that when a two-phase open-circuit fault occurs, there will be no third harmonic current in the five-phase permanent magnet fault-tolerant motor, so it can be ignored.
  • the fault-tolerant current after non-adjacent (A, C) two-phase open-circuit fault is:
  • i B2 , i D2 , i E2 are the B, D, and E phase currents when the A and C phases are faulty.
  • the fault-tolerant current after the adjacent (A, B) two-phase open-circuit fault is:
  • i C3 , i D3 , i E3 are the C, D, and E phase currents when the A and B phases are faulty.
  • the current vector diagrams in normal and fault-tolerant operation are obtained accordingly, as shown in Figure 3, it can be found that in the fault-tolerant mode, the fault current vector is zero, the phase of the non-fault phase current vector is shifted and the magnitude increases responsively. Therefore, when the five-phase permanent magnet fault-tolerant motor fails, the corresponding fault-tolerant control strategy is used to ensure that the winding current satisfies the relationship in Figure 3, and the motor can run without disturbance under different faults, that is, the five-phase permanent magnet fault-tolerant motor drive system fault. fault tolerance mechanism.
  • Step 3 Use the difference between the given speed and the detected actual speed to construct a high-quality torque controller to obtain a high-quality torque given T e * , so as to suppress the motor torque ripple after the fault, and at the same time, the load disturbance, the system Factors such as parameter changes and electromagnetic torque ripple caused by faults are taken into account.
  • ⁇ is the mechanical angular velocity
  • B is the friction coefficient
  • J is the moment of inertia
  • T L is the load torque
  • T b represents the non-pulsating component of the electromagnetic torque
  • ⁇ T e represents the pulsating component of the electromagnetic torque caused by the fault. Therefore, T b is designed to guarantee the system performance of the five-phase motor, and ⁇ T e is considered as the uncertainty factor of the system.
  • the high-quality torque control strategy can eliminate the disturbance ⁇ T e , so the torque ripple of the five-phase permanent magnet fault-tolerant motor can be suppressed after the fault.
  • equation (7) can be rewritten as:
  • the high-quality torque controller is designed as:
  • T m is the maximum value of TL and is greater than zero; ⁇ is a constant greater than zero.
  • the constructed high-quality torque controller comprehensively considers factors such as load disturbance (T L ), system parameter changes (J and B), and electromagnetic torque ripple ( ⁇ T e ) caused by faults.
  • the torque controller can not only suppress the electromagnetic torque ripple caused by faults, but also has good anti-interference performance against uncertain factors such as load disturbance and system parameter changes.
  • Step 4) Detect the five-phase currents i A , i B , i C , i D and i E of the five-phase permanent magnet fault-tolerant motor, and obtain the current components i d1 , i q1 , i q1 , i d3 , i q3 .
  • the current component in the two-phase rotating coordinate system of the five-phase permanent magnet fault-tolerant motor is expressed as:
  • Step 5 Using the current components i d1 , i q1 and the high-quality torque given T e * in the two-phase rotating coordinate system to calculate the quadrature-axis fundamental wave voltage of the VC drive system and the DTC drive system.
  • the given voltage of quadrature fundamental wave of VC drive system is acquired.
  • Figure 4(a) shows the fault-tolerant VC drive system.
  • the specific implementation steps for obtaining the quadrature fundamental wave given voltage of the VC drive system include:
  • the given direct-axis voltage u d1 * is obtained after the difference between the given direct-axis current zero and the direct-axis current id1 through the PI regulator;
  • the quadrature axis current i q1 * can be obtained by the following formula:
  • pr is the number of pole pairs of the motor
  • ⁇ f is the amplitude of the permanent magnet flux linkage
  • the given voltage of quadrature fundamental wave of DTC drive system is acquired.
  • Figure 4(b) shows the fault-tolerant DTC drive system.
  • the specific implementation steps of obtaining the quadrature fundamental wave given voltage of the DTC drive system include:
  • L s is the stator inductance
  • the magnitude and phase of the stator flux linkage can be obtained as:
  • the essence of the electromagnetic torque of the five-phase permanent magnet motor is the result of the interaction between the rotor magnetic field and the stator magnetic field, namely:
  • ⁇ is the angle between the stator flux linkage and the rotor flux linkage, that is, the phase angle of the stator flux linkage.
  • the present invention compares the direct-axis current i d1 with zero, and then uses the PI regulator as the stator flux linkage. Therefore, the given stator flux linkage can be adaptively adjusted according to the load conditions to ensure The direct-axis current component of the motor running under different working conditions is zero.
  • Step 6 Based on the third harmonic current zero control strategy, the current components i d3 and i q3 in the two-phase rotating coordinate system are used to obtain the third harmonic voltage of the AC-direction axis.
  • the current components i d3 and i q3 in the two-phase rotating coordinate system are compared with zero, respectively, and then the corresponding third harmonic voltage of the AC and DC axes is obtained after passing through the PI regulator.
  • Step 7) Calculate the winding phase voltage in the fault mode based on the fault tolerance mechanism and the AC-DC axis voltage.
  • phase voltage expression of the five-phase permanent magnet fault-tolerant motor is:
  • i x is the phase current
  • e x is the opposite potential.
  • the winding phase voltage expression under A-phase winding fault is:
  • u Ae3 , u Be3 , u Ce3 , u De3 , and u Ee3 are the phase voltages of the five-phase windings without considering the back EMF when the A and B two-phase windings fail.
  • Step 8) According to the back EMF of the motor and the winding phase voltage in the fault mode, obtain the given phase voltage of the five-phase winding in the fault-tolerant operation mode.
  • the back EMF under the open-circuit fault of the five-phase permanent magnet fault-tolerant motor is consistent with the back EMF during normal operation. Due to the small change in the amplitude of the permanent magnet flux linkage of the five-phase permanent magnet fault-tolerant motor and the small harmonic content of the back EMF, its reverse The electric potential can be expressed as:
  • ⁇ f is the amplitude of the permanent magnetic flux linkage
  • ⁇ e is the electrical angular velocity
  • the given fault-tolerant phase voltage expression under the two-phase winding fault of A and C is:
  • the given fault-tolerant phase voltage expression under the two-phase winding fault of A and B is:
  • Step 9) The five-phase winding is given a five-phase voltage command through the voltage source inverter, and the pulse width modulation CPWM method is used to realize the disturbance-free operation of the five-phase permanent magnet fault-tolerant motor VC and the DTC drive system under any open-circuit fault condition.
  • Figure 5 shows the simulation results of the operation when the load and system parameters are changed under normal conditions.
  • the operating conditions of the motor are as follows: the speed is 800r/min, the initial load is 2N ⁇ m, the sudden change to 8N ⁇ m in 0.3s, and the moment of inertia becomes twice the original. It can be found from the simulation results that the current of the drive system of the present invention is relatively sinusoidal; the sudden load and moment of inertia have little effect on the operation of the system, indicating that the present invention has better resistance to load disturbance and system parameter changes; in addition, during the entire operation process, The stator flux linkage can be adaptively changed to ensure that the direct axis current is zero and improve the motor efficiency.
  • Figures 6 and 7 show the simulation results of the VC and DTC drive systems in the case of non-adjacent two-phase (A, C) open-circuit faults, respectively.
  • the operating conditions of the motor are as follows: the speed is 500 r/min, the load is 5 N m, the motor failure time is 0.2 s, and the fault-tolerant control strategy is adopted at 0.3 s. It can be seen that after the fault occurs in 0.2s, the fault phase current becomes zero, the electromagnetic torque pulsation and flux linkage pulsation increase significantly, and the rotational speed begins to oscillate.
  • FIGS. 8 to 9 show the simulation results of the DTC drive system in the case of one-phase (A) open-circuit fault and two adjacent two-phase (A, B) open-circuit faults, respectively. It can be seen that the effectiveness of the unified fault-tolerant control strategy is adopted after the fault.
  • the present invention provides an open-circuit unified fault-tolerant control method for a five-phase permanent magnet fault-tolerant motor vector and direct torque control drive system.
  • the invention essentially reveals the fault-tolerant mechanism, and based on the pulse width modulation CPWM mode, proposes a unified fault-tolerant control strategy for open-circuit faults suitable for vector control and direct torque control drive systems, and effectively solves the fault-tolerant control corresponding to various new basic control algorithms.
  • the solution will also change and complicate the problem; at the same time, a high-quality torque controller is designed, so that the control system still has high-quality output torque even under open-circuit faults, and has good dynamic performance under normal and faulty operation.
  • the static performance, anti-interference ability and robustness comprehensively improve the operation performance of the motor drive system; the present invention does not need to change the coordinate transformation and additional compensation voltage, that is, does not need to change the structure of the control system, only needs to change the control strategy of one of the modules , which can realize fault-tolerant operation under different faults, simplify the controller algorithm, minimize the reconstruction of the control system and save the controller CPU memory resources under different faults in the true sense; breaking through the traditional fault-tolerant control is generally based on the AC-DC-axis current base

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Abstract

本发明公开了一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法。所述控制方法的核心部分为开路故障统一控制策略，所述开路故障统一控制策略原理结构包括获取高品质转矩给定、获取交直轴给定电压、分析容错机理以获取容错电流、基于容错机理以获取故障绕组电压、根据反电势以获取容错电压。本发明从本质上揭示容错机理，基于脉宽调制CPWM方式，提出一种适用于矢量控制和直接转矩控制驱动系统的开路故障统一容错控制策略，有效解决各种新型基本控制算法对应的容错控制方案也将随之多变与复杂化的问题。

Description

五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法 技术领域
本发明属于多相电机容错控制技术领域，尤其涉及一种用于五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法。
背景技术
随着全球环境问题日益突出，电动汽车受到越来越多的关注。电动汽车作为载人设备，其安全性和可靠性至关重要。多相永磁电机具有高效率、高功率密度、宽调速范围、低转矩脉动和强容错能力等优点，在电动汽车、混合动力汽车等电力驱动领域中得到广泛关注和应用。矢量控制(VC)和直接转矩控制(DTC)因其高驱动性能倍受人们的关注。然而，电机驱动系统发生故障将影响到整个应用系统的正常工作，甚至发生安全事故。因此，对多相永磁电机驱动系统进行容错控制研究，提高电机驱动系统的可靠性具有重要的现实意义。
目前，国内外学者对多相永磁电机容错控制策略的研究主要集中于开路故障。文献“五相永磁同步电机容错控制策略”(电机与控制学报，2014)提出了一种基于铜耗最小原则和铜耗相等原则的五相永磁电机容错电流控制策略；中国发明专利《一种基于新型容错开关表的五相永磁同步电机容错直接转矩控制方法》(专利号CN201910669271.9)和文献“Openphasefault-tolerant direct torque control technique for five-phase induction motor drives”(IEEETransactiononIndustrialElectronics，2017)提出了一种基于虚拟空间电压矢量的容错开关表的容错直接转矩控制方法。但由于上述方法均是基于传统滞环比较控制提出的，存在着开关频率不固定、转矩和磁链脉动大等问题。为克服滞环比较控制带来的问题，文献 “Fault-tolerant direct torque control of five-phase FTFSCW-IPM motor based on analogous three-phase SVPWM for electric vehicle applications”(IEEE Transaction on Vehicular Technology，2017)针对五相永磁电机一相开路故障，提出了相应的SVPWM容错控制方法，克服了基于滞环控制的容错控制系统的缺点，但是，该类容错控制策略是基于故障后电压矢量重构，算法相对复杂，不利于工程化。为此，文献“Remedial field-oriented control of five-phase fault-tolerant permanent-magnet motor by using reduced-order transformation matrices”(IEEETransactiononIndustrialElectronics，2017)和中国发明专利《一种五相永磁直线电机一相开路容错直接推力控制方法》(专利号CN201810025607.3)分别基于VC和DTC提出了一种CPWM容错方法，无需重构故障后的电压矢量，但是由于采用两套坐标变换矩阵，实际上电机正常运行与故障容错运行时采用两套独立控制算法，仍未从本质上实现简化控制器算法和故障时最小化重构控制系统。总体来说，现有的CPWM容错控制方法没有从本质上揭示容错机理，随着近年来各种新型算法的涌现，其相应的容错控制方案也将随之多变与复杂化。
发明内容
发明目的：针对现有技术中存在的问题，本发明从本质上揭示容错机理出发，提出了一种基于CPWM的五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，提高电机驱动系统的稳健运行能力，使得控制系统不但具有较好的开路故障容错运行性能，还具备良好动静态性能、抗干扰能力和鲁棒性，同时可以适用于各种新型的控制算法，实现不同故障下最小化重构控制系统与节省控制器CPU内存资源。
技术方案：为实现上述发明目的，本发明所采用的技术方案是：一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，包括如下步骤：
步骤1)建立五相永磁容错电机正常运行时的电流数学模型；
五相永磁容错电机运行于正常运行工况下，其电流i A、i B、i C、i D和i E表达式如下式所示：
Figure PCTCN2021082038-appb-000001
其中，γ＝72°，θ为转子位置电角度，i d1 *、i q1 *为基波电流给定值直轴和交轴分量，i d3 *、i q3 *为三次谐波电流给定值直轴和交轴分量；
步骤2)获取一相开路、非相邻两相开路和相邻两相开路的容错电流，分析容错机理；
步骤3)利用给定转速与检测到的实际转速之差，构建高品质转矩控制器获取高品质转矩给定T e *，以抑制故障后的电机转矩脉动，同时将负载扰动、系统参数变化以及由故障引起的电磁转矩脉动因素均考虑进去；
步骤4)检测五相永磁容错电机的五相电流i A、i B、i C、i D和i E，经Clark和Park变换得到两相旋转坐标系下的电流分量i d1、i q1、i d3、i q3；
五相永磁容错电机的两相旋转坐标系下的电流分量表示为：
Figure PCTCN2021082038-appb-000002
步骤5)利用两相旋转坐标系下的电流分量i d1、i q1以及高品质转矩给定T e *，以计算矢量控制VC驱动系统和直接转矩控制DTC驱动系统的交直轴基波电压；
步骤6)基于三次谐波电流为零控制策略，利用两相旋转坐标系下的电流分量i d3、i q3以获取交直轴三次谐波电压；
步骤7)基于容错机理以及交直轴电压，计算故障模式下的绕组相电压；
步骤8)根据电机反电势和故障模式下的绕组相电压，获取容错运行模式下的五相绕组给定相电压；
步骤9)将五相绕组给定五相电压指令经电压源逆变器，采用脉宽调制CPWM方式实 现五相永磁容错电机VC与DTC驱动系统任意开路故障情况下的无扰运行。
本发明具有以下有益效果：
1)本发明首次提出一种适用于VC和DTC驱动系统的开路故障统一容错控制策略，从本质上揭示容错机理，只需根据五相永磁容错电机驱动系统的交直轴电压给定，即可实现该系统在不同开路故障状态的容错运行，从而解决了各种新型基本控制算法对应的容错控制方案也将随之多变与复杂化的问题。
2)本发明基于高品质转矩控制器设计，使得控制系统即使在开路故障下仍拥有高品质的输出转矩，同时在正常和故障运行下均具有良好动静态性能、抗干扰能力和鲁棒性，综合提高了电机驱动系统的运行性能。
3)本发明基于脉宽调制CPWM方式实现开路故障情况下的无扰运行，能够有效解决传统基于滞环比较器的容错控制方法带来的电流谐波含量高、转矩脉动大、开关频率不固定等问题；并且相对于基于SVPWM的容错控制方法，无需判别扇区和计算故障下的空间电压矢量重构，大大简化了控制算法；此外，相对于现有的基于CPWM的容错控制方法，无需变更坐标变换和额外补偿电压，即不需要改变控制系统的结构，只需改变其中一个模块的控制策略，即可实现不同故障下的容错运行，简化了控制器算法，做到真正意义上的不同故障下最小化重构控制系统与节省控制器CPU内存资源。
4)本发明中突破传统容错控制一般基于交直轴电流基波分量保证故障前后磁动势相等的技术束缚，考虑三次谐波交直轴电流在故障前后的作用机理，以确保定子磁链轨迹为圆形的同时，进一步提高故障容错运行下的电流质量。
5)本发明基于“i d＝0”控制和磁链自适应给定控制策略，实现两相旋转坐标系下的容错控制，确保电机运行于不同工况下包括故障运行工况下的直轴电流分量为零，解决了电机在突变负载或故障运行工况的损耗大的问题，有效提高了电机驱动系统的 效率。
6)所提出方法计算量小，简单易于实现，有利于新理论的工程化和实用化。
附图说明
图1为本发明五相永磁容错电机的结构示意图；
图2为本发明开路故障统一容错控制策略原理框图；
图3给出了正常和容错运行时的电流矢量图；(a)为正常运行时的电流矢量图；(b)为A相开路故障时的电流矢量图；(c)为非相邻相A、C两相开路故障时的电流矢量图；(d)为相邻相A、B两相开路故障时的电流矢量图；
图4为本发明基于CPWM的五相永磁容错电机容错控制驱动系统的结构框图；(a)容错VC驱动系统；(b)容错DTC驱动系统；
图5为本发明正常工况下变负载和变系统参数时的DTC驱动系统运行仿真波形；(a)电流波形；(b)转矩波形；(c)转速波形；(d)定子磁链幅值波形；(e)交直轴电流波形；
图6为本发明VC驱动系统在正常运行、A、C两相开路故障下无容错运行和容错运行情况下的仿真波形；(a)电流波形；(b)转矩波形；(c)转速波形；
图7为本发明DTC驱动系统在正常运行、A、C两相开路故障下无容错运行和容错运行情况下的仿真波形；(a)电流波形；(b)转矩波形；(c)转速波形；(d)定子磁链幅值波形；
图8为本发明DTC驱动系统在正常运行、A相开路故障下无容错运行和容错运行情况下的仿真波形。(a)电流波形；(b)转矩波形；(c)转速波形；
图9为本发明DTC驱动系统在正常运行、A、B两相开路故障下无容错运行和容错运行情况下的仿真波形；(a)电流波形；(b)转矩波形；(c)转速波形。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅用于解释本发明，并不用于限定本发明。
如图1所示，本发明的控制对象为五相永磁容错电机的结构示意图，包括定子、转子、永磁体、电枢齿、容错齿、电枢绕组；电枢齿和容错齿沿定子内圈周向间隔均匀分布，并且电枢齿的齿宽和容错齿的齿宽不相等；电枢齿上绕有电枢绕组线圈，为单层集中绕组，两相邻的单层集中绕组之间由容错齿进行隔离；转子内嵌入永磁体，呈“V”型分布；电枢齿和容错齿的总齿数为20，永磁体的极数为18；由于定子采用单层集中绕组，大大降低了电机直轴磁路的磁阻，从而降低了电机的凸极率，使得交直轴电感接近相等；定子部分设置容错齿，能够有效实现各相绕组间的电、磁、热隔离，具有较强的容错性能。
如图2所示，本发明开路故障统一容错控制策略原理框图，包括获取高品质转矩给定、获取交直轴给定电压、分析容错机理以获取容错电流、基于容错机理以获取故障绕组电压、根据反电势以获取容错电压。本发明开路故障统一容错控制策略原理的具体实施步骤包括：
步骤1)建立五相永磁容错电机正常运行时的电流数学模型。
五相永磁容错电机运行于正常运行工况下，其电流表达式如下式所示：
Figure PCTCN2021082038-appb-000003
其中，γ＝72°，θ为转子位置电角度，i d1 *、i q1 *为基波电流给定值直轴和交轴分量，i d3 *、i q3 *为三次谐波电流给定值直轴和交轴分量。
步骤2)获取不同故障模式下(一相开路、非相邻两相开路和相邻两相开路)的容错电流，分析容错机理。
五相永磁容错电机正常运行工况下磁动势表达式为：
MMF 1＝Ni A+ηNi B+η 2Ni C+η 3Ni D+η 4Ni E  (2)
其中，η＝cosγ+jsinγ；N为绕组匝数；i A、i B、i C、i D、i E为电机正常运行时的A、B、C、D、E相电流。
当五相永磁容错电机发生单相(A相)开路故障时，故障相A相的电流为零，此时磁动势表达式为：
MMF 2＝ηNi B1+η 2Ni C1+η 3Ni D1+η 4Ni E1  (3)
其中，i B1、i C1、i D1、i E1为A相故障时的B、C、D、E相电流。
根据故障前后磁动势相等和故障容错电流幅值相等原则，可得A相发生开路后其余非故障相的电流分配情况，如下式所示：
Figure PCTCN2021082038-appb-000004
其中，
Figure PCTCN2021082038-appb-000005
同理，利用故障前后磁动势相等原则，可分别计算得到相邻两相和非相邻两相开路故障下的容错电流。值得注意的是，发生两相开路故障时，五相永磁容错电机将不存在三次谐波电流，故可将其忽略。
非相邻(A、C)两相开路故障后的容错电流为：
Figure PCTCN2021082038-appb-000006
其中，i B2、i D2、i E2为A、C两相故障时的B、D、E相电流。
相邻(A、B)两相开路故障后的容错电流为：
Figure PCTCN2021082038-appb-000007
其中，i C3、i D3、i E3为A、B两相故障时的C、D、E相电流。
根据式(1)和(4)～(6)的电流表达式，相应地得到正常和容错运行时的电流矢量图，如图3所示，可以发现，在故障容错模式下，故障电流矢量为零，而非故障相电流矢量的相位发生了平移，并且幅值响应地增大了。因此，当五相永磁容错电机发生故障时，通过相应的容错控制策略保证绕组电流满足图3的关系，可保证电机在不同故障下无扰运行，即为五相永磁容错电机驱动系统故障时的容错机理。
步骤3)利用给定转速与检测到的实际转速之差，构建高品质转矩控制器获取高品质转矩给定T e *，以抑制故障后的电机转矩脉动，同时将负载扰动、系统参数变化以及由故障引起的电磁转矩脉动等因素均考虑进去。
五相永磁容错电机转矩与转速之间关系为：
Figure PCTCN2021082038-appb-000008
其中，ω为机械角速度，B为摩擦系数，J为转动惯量，T L为负载转矩。
对于五相永磁容错电机系统处于故障模式下，其电磁转矩的表达式为
T e＝T b+ΔT e  (8)
其中，T b代表电磁转矩无脉动分量，ΔT e代表电磁转矩由故障引起的脉动分量。所以，设计T b以保证五相电机的系统性能，而将ΔT e被认为是系统的不确定性因素。采用高品质转矩控制策略可消除扰动ΔT e，故而可抑制五相永磁容错电机在故障后转矩脉动。
假设ΔT e＝α 1T b，其中，α 1未知，但是有界限，其最大值为α 1m，故α 1满足|α 1|≤α 1m<1。因此，式(7)可改写为：
Figure PCTCN2021082038-appb-000009
其中，B m和J m分别为B和J的最大值，并且大于零，可以根据电机系统极端环境得到相应的数值；α 2＝1-(J m/J)，α 2的取值范围为：0≤α 2<1。令e＝ω-ω *，ω *为转子给定角速度，则：
ω＝e+ω *  (10)
即而可得：
Figure PCTCN2021082038-appb-000010
根据强鲁棒控制规律，设计高品质转矩控制器为：
Figure PCTCN2021082038-appb-000011
其中，
Figure PCTCN2021082038-appb-000012
T m为T L的最大值，并且大于零；ε为大于零的常数。
可见，所构建的高品质转矩控制器综合考虑负载扰动(T L)、系统参数变化(J和B)以及由故障引起的电磁转矩脉动(ΔT e)等因素，因此，该高品质转矩控制器不仅可以抑制故障引起的电磁转矩脉动，同时对负载扰动、系统参数变化等不确定因素具有较好的抗干扰性能。
步骤4)检测五相永磁容错电机的五相电流i A、i B、i C、i D和i E，经Clark和Park变换得到两相旋转坐标系下的电流分量i d1、i q1、i d3、i q3。
五相永磁容错电机的两相旋转坐标系下的电流分量表示为：
Figure PCTCN2021082038-appb-000013
步骤5)利用两相旋转坐标系下的电流分量i d1、i q1以及高品质转矩给定T e *，以计算 VC驱动系统和DTC驱动系统的交直轴基波电压。
第一部分，VC驱动系统的交轴基波给定电压获取。
图4(a)给出了容错VC驱动系统。VC驱动系统的交轴基波给定电压获取具体实现步骤包括：
(1)采用i d＝0控制，给定直轴电流零与直轴电流i d1之差经PI调节器后得到直轴电压给定u d1 *；
(2)获取交轴电流i q1 *，将i q1 *与直轴电流i q1之差经PI调节器后得到直轴电压给定u q1 *。
交轴电流i q1 *可由下式得到：
Figure PCTCN2021082038-appb-000014
其中，p r为电机的极对数，ψ f为永磁磁链幅值。
第二部分，DTC驱动系统的交轴基波给定电压获取。
图4(b)给出了容错DTC驱动系统。DTC驱动系统的交轴基波给定电压获取具体实现步骤包括：
(1)利用上述两相旋转坐标系下的电流分量i d1、i q1来计算定子磁链大小、相位以及估算转矩；
定子磁链的交直轴分量的表达式为：
Figure PCTCN2021082038-appb-000015
其中，L s为定子电感。
由上式可得定子磁链幅值大小和相位，为：
Figure PCTCN2021082038-appb-000016
由于五相永磁容错电机的交直轴电感接近相等，其估算转矩可由下式得到：
Figure PCTCN2021082038-appb-000017
(2)将给定转矩T e *与计算转矩之差经转速PI调节器后得到转矩角增量Δδ，并通过磁链自适应给定控制策略得到定子磁链的给定值ψ s *；
五相永磁电机的电磁转矩本质是转子磁场与定子磁场相互作用的结果，即有：
Figure PCTCN2021082038-appb-000018
其中，
Figure PCTCN2021082038-appb-000019
为转子磁场矢量；
Figure PCTCN2021082038-appb-000020
为定子磁场矢量；δ为定子磁链与转子磁链的夹角，即定子磁链相位角。
对上式两边求导，得：
Figure PCTCN2021082038-appb-000021
可见，转矩偏差ΔT e与转矩角增量Δδ之间具有非线性关系，因此，转矩角Δδ可由ΔT e通过PI调节器后得到。
另外，定子磁链给定若为定值，当电机空载或突加重载运行时，需要额外的直轴电流分量来维持定子磁链不变。额外的直轴电流分量会增加电机损耗，降低系统效率。为解决上述问题，本发明将直轴电流i d1与零作差比较后经PI调节器作为定子磁链给定，因此，给定的定子磁链可根据负载情况进行自适应地调节，以确保电机运行于不同工况下的直轴电流分量为零。
(3)将定子磁链大小和相位、转矩角增量Δδ以及定子磁链的给定值ψ s *通过预期电压计算，得到两相旋转坐标系上的交直轴基波电压参考值u d1 *和u q1 *。
根据五相永磁容错电机交直轴电压方程，得到交直轴基波电压参考值的表达式为
Figure PCTCN2021082038-appb-000022
步骤6)基于三次谐波电流为零控制策略，利用两相旋转坐标系下的电流分量i d3、i q3 以获取交直轴三次谐波电压。
具体的，将两相旋转坐标系下的电流分量i d3、i q3分别与零作差比较后经PI调节器后得到相应的交直轴三次谐波电压。
步骤7)基于容错机理以及交直轴电压，计算故障模式下的绕组相电压。
五相永磁容错电机的相电压表达式为：
Figure PCTCN2021082038-appb-000023
其中，x＝A,B,C,D,E，R s为定子电阻，i x为相电流，e x为相反电势。
则上式可改写为：
Figure PCTCN2021082038-appb-000024
当五相容错电机发生故障，上式可用交直轴电压的形式表示，则可得不同故障下的绕组相电压表达式。A相绕组故障下的绕组相电压表达式为：
Figure PCTCN2021082038-appb-000025
并且，u A1＝0，其中，u Ae1、u Be1、u Ce1、u De1、u Ee1为A相绕组故障时未考虑反电势情况下的五相绕组相电压；u ed1 *＝u d1 *-e d1 *，u eq1 *＝u q1 *-e q1 *；u ed3 *＝u d3 *-e d3 *，u eq1 *＝u q3 *-e q3 *；e d1 *、e q1 *、e d3 *、e q3 *分别为五相反电势交直轴分量，可通过相反电势经过五相静止坐标系到两相旋转坐标系的坐标变换矩阵得到。
A、C两相绕组故障下的绕组相电压表达式为：
Figure PCTCN2021082038-appb-000026
A、B两相绕组故障下的绕组相电压表达式为：
Figure PCTCN2021082038-appb-000027
其中，u Ae3、u Be3、u Ce3、u De3、u Ee3为A、B两相绕组故障时未考虑反电势情况下的五相绕组相电压。
步骤8)根据电机反电势和故障模式下的绕组相电压，获取容错运行模式下的五相绕组给定相电压。
五相永磁容错电机开路故障下的反电势与正常运行时的反电势一致，由于五相永磁容错电机的永磁磁链幅值变化很小以及反电势的谐波含量很小，其反电势可表示为：
Figure PCTCN2021082038-appb-000028
其中，ψ f为永磁磁链幅值，ω e为电角速度。
将反电势代入式(21)，可得到不同故障模式下相应的容错电压给定值。A相绕组故障下的给定容错相电压表达式为：
Figure PCTCN2021082038-appb-000029
并且，u A *＝e A。
A、C两相绕组故障下的给定容错相电压表达式为：
Figure PCTCN2021082038-appb-000030
A、B两相绕组故障下的给定容错相电压表达式为：
Figure PCTCN2021082038-appb-000031
因此，只需已知五相永磁容错电机驱动系统的直轴电压u d *和交轴电压u q *，基于给定容错相电压表达式(26)～(28)，即可实现该系统在电机绕组开路故障状态的容错运行。
步骤9)将五相绕组给定五相电压指令经电压源逆变器，采用脉宽调制CPWM方式实现五相永磁容错电机VC与DTC驱动系统任意开路故障情况下的无扰运行。
值得注意的是，无论采用何种基本控制算法，只需将相应的给定交直轴电压经过开路故障统一容错控制策略，即可实现五相永磁容错电机开路故障情况下的无扰运行，避免了不同新型的基本控制算法导致对应的容错控制策略复杂的问题。
图5给出了正常工况下变负载和变系统参数时的运行仿真结果。电机运行工况为：转速为800r/min，初始负载为2N·m，0.3s时突变为8N·m，并且转动惯量变为原来的两倍。由仿真结果可以发现，本发明驱动系统电流较为正弦；突变负载和转动惯量对系统的运行几乎没有影响，说明本发明具有较好抗负载扰动和系统参数变化能力；此外，在整个运行过程中，定子磁链能够自适应地变化，以保证直轴电流为零，提高电机效率。
图6和图7分别给出了VC和DTC驱动系统在非相邻两相(A、C)开路故障情况下的仿真结果。电机运行工况为：转速500r/min，负载为5N·m，电机故障发生时间为0.2s，0.3s时采用容错控制策略。可以看出在0.2s发生故障后，故障相电流变为零，电磁转矩脉动和磁链脉动明显增大，并且转速开始震荡。当在0.3s采用容错控制策略时，转速迅速恢复至参考转速，电流更加正弦，并且与公式(5)中的理论值一致，转矩也迅速稳定，相对于故障下的转矩脉动大大降低了，并且，磁链的脉动也明显下降。此外，图8～图9分别给出了DTC驱动系统在一相(A)开路故障和相邻两相(A、B)开路故障情况下的仿真结果。可见故障后采用统一容错控制策略的有效性。
综上，本发明的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法。本发明从本质上揭示容错机理，基于脉宽调制CPWM方式，提出一种适用于矢量控制和直接转矩控制驱动系统的开路故障统一容错控制策略，有效解决各种新型基本控制算法对应的容错控制方案也将随之多变与复杂化的问题；同时设计高品质转矩控制器，使得控制系统即使在开路故障下仍拥有高品质的输出转矩，同时在正常和故障运行下均具有良好动静态性能、抗干扰能力和鲁棒性，综合提高了电机驱动系统的运行性能；本发明无需变更坐标变换和额外补偿电压，即不需要改变控制系统的结构，只需改变其中一个模块的控制策略，即可实现不同故障下的容错运行，简化了控制器算法，做到真正意义上的不同故障下最小化重构控制系统与节省控制器CPU内存资源；突破传统容错控制一般基于交直轴电流基波分量保证故障前后磁动势相等的技术束缚，考虑三次谐波交直轴电流在故障前后的作用机理，以确保定子磁链轨迹为圆形的同时，进一步提高故障容错运行下的电流质量；基于“i d＝0”控制和磁链自适应给定控制策略，确保电机运行于不同工况下包括故障运行工况下的直轴电流分量为零，解决了电机在突变负载或故障运行工况的损耗大的问题，有效提高了电机驱动系统的效率。
以上实施例仅用于说明本发明的设计思想和特点，其目的在于使本领域内的技术人员能够了解本发明的内容并据以实施，本发明的保护范围不限于上述实施例。所以，凡依据本发明所揭示的原理、设计思路所作的等同变化或修饰，均在本发明的保护范围之内。

Claims (9)

  1. 一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，包括如下步骤：
    步骤1)建立五相永磁容错电机正常运行时的电流数学模型；
    五相永磁容错电机运行于正常运行工况下，其电流i A、i B、i C、i D和i E表达式如下式所示：
    Figure PCTCN2021082038-appb-100001
    其中，γ＝72°，θ为转子位置电角度，i d1 *、i q1 *为基波电流给定值直轴和交轴分量，i d3 *、i q3 *为三次谐波电流给定值直轴和交轴分量；
    步骤2)获取一相开路、非相邻两相开路和相邻两相开路的容错电流，分析容错机理；
    步骤3)利用给定转速与检测到的实际转速之差，构建高品质转矩控制器获取高品质转矩给定T e *，以抑制故障后的电机转矩脉动，同时将负载扰动、系统参数变化以及由故障引起的电磁转矩脉动因素均考虑进去；
    步骤4)检测五相永磁容错电机的五相电流i A、i B、i C、i D和i E，经Clark和Park变换得到两相旋转坐标系下的电流分量i d1、i q1、i d3、i q3；
    五相永磁容错电机的两相旋转坐标系下的电流分量表示为：
    Figure PCTCN2021082038-appb-100002
    步骤5)利用两相旋转坐标系下的电流分量i d1、i q1以及高品质转矩给定T e *，以计算矢量控制VC驱动系统和直接转矩控制DTC驱动系统的交直轴基波电压；
    步骤6)基于三次谐波电流为零控制策略，利用两相旋转坐标系下的电流分量i d3、i q3以获取交直轴三次谐波电压；
    步骤7)基于容错机理以及交直轴电压，计算故障模式下的绕组相电压；
    步骤8)根据电机反电势和故障模式下的绕组相电压，获取容错运行模式下的五相绕组给定相电压；
    步骤9)将五相绕组给定五相电压指令经电压源逆变器，采用脉宽调制CPWM方式实现五相永磁容错电机VC与DTC驱动系统任意开路故障情况下的无扰运行。
  2. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述步骤2)具体实现包括：
    五相永磁容错电机正常运行工况下磁动势表达式为：
    MMF 1＝Ni A+ηNi B+η 2Ni C+η 3Ni D+η 4Ni E
    其中，η＝cosγ+jsinγ；N为绕组匝数；i A、i B、i C、i D、i E为电机正常运行时的A、B、C、D、E相电流；
    当五相永磁容错电机发生单相开路故障时，假如为A相，故障相A相的电流为零，此时磁动势表达式为：
    MMF 2＝ηNi B1+η 2Ni C1+η 3Ni D1+η 4Ni E1
    其中，i B1、i C1、i D1、i E1为A相故障时的B、C、D、E相电流；
    根据故障前后磁动势相等和故障容错电流幅值相等原则，同时考虑到三次谐波电流，可得A相发生开路后其余非故障相的电流分配情况，如下式所示：
    Figure PCTCN2021082038-appb-100003
    其中，
    Figure PCTCN2021082038-appb-100004
    同理，利用故障前后磁动势相等原则，可分别计算得到相邻两相和非相邻两相开路故障下的容错电流，发生两相开路故障时，五相永磁容错电机将不存在三次谐波电流，故可将其忽略；非相邻(A、C)两相开路故障后的容错电流为：
    Figure PCTCN2021082038-appb-100005
    其中，i B2、i D2、i E2为A、C两相故障时的B、D、E相电流；
    相邻(A、B)两相开路故障后的容错电流为：
    Figure PCTCN2021082038-appb-100006
    其中，i C3、i D3、i E3为A、B两相故障时的C、D、E相电流；
    当五相永磁容错电机发生故障时，通过相应的容错控制策略保证绕组电流满足上述容错电流，可保证电机在不同故障下无扰运行，即为五相永磁容错电机驱动系统故障时的容错机理。
  3. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述步骤3)利用给定转速与检测到的实际转速之差，构建高品质转矩控制器获取高品质转矩给定T e *的具体过程包括：
    五相永磁容错电机转矩与转速之间关系为：
    Figure PCTCN2021082038-appb-100007
    其中，ω为机械角速度，B为摩擦系数，J为转动惯量，T L为负载转矩；
    对于五相永磁容错电机系统处于故障模式下，其电磁转矩的表达式为
    T e＝T b+ΔT e
    其中，T b代表电磁转矩无脉动分量，ΔT e代表电磁转矩由故障引起的脉动分量，所以，设计T b以保证五相电机的系统性能，而将ΔT e被认为是系统的不确定性因素，采用高品质转矩控制策略可消除扰动ΔT e，故而可抑制五相永磁容错电机在故障后转矩脉动；
    假设ΔT e＝α 1T b，其中，α 1未知，但是有界限，其最大值为α 1m，故α 1满足|α 1|≤α 1m<1，五相永磁容错电机转矩与转速之间的关系式可改写为：
    Figure PCTCN2021082038-appb-100008
    其中，B m和J m分别为B的最大值和J的最小值，并且大于零，可以根据电机系统极端环境得到相应的数值；α 2＝1-(J m/J)，α 2的取值范围为：0≤α 2<1，令e＝ω-ω *，ω *为转子给定角速度，则：
    ω＝e+ω *
    即而可得：
    Figure PCTCN2021082038-appb-100009
    根据强鲁棒控制规律，设计高品质转矩控制器为：
    Figure PCTCN2021082038-appb-100010
    其中，
    Figure PCTCN2021082038-appb-100011
    T m为T L的最大值，并且大于零；|e|为机械角速度与转子给定角速度之差的绝对值；|λ|为λ的绝对值；ε为大于零的常数。
  4. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述步骤5)利用两相旋转坐标系下的电流分量i d1、i q1以及高品质转矩给定T e *，以计算VC驱动系统和DTC驱动系统的交直轴基波电压的具体实现包括：
    第一部分，VC驱动系统的交轴基波给定电压获取；
    (1.1)采用i d＝0控制，给定直轴电流零与直轴电流i d1之差经PI调节器后得到直轴电压给定u d1 *；
    (1.2)获取交轴电流i q1 *，将i q1 *与直轴电流i q1之差经PI调节器后得到直轴电压给定u q1 *，
    交轴电流i q1 *可由下式得到：
    Figure PCTCN2021082038-appb-100012
    其中，p r为电机的极对数，ψ f为永磁磁链幅值；
    第二部分，DTC驱动系统的交轴基波给定电压获取；
    (2.1)利用上述两相旋转坐标系下的电流分量i d1、i q1来计算定子磁链大小、相位以及估算转矩；
    定子磁链的交直轴分量的表达式为：
    Figure PCTCN2021082038-appb-100013
    其中，L s为定子电感；
    由上式可得定子磁链幅值大小和相位，为：
    Figure PCTCN2021082038-appb-100014
    由于五相永磁容错电机的交直轴电感接近相等，其估算转矩可由下式得到：
    Figure PCTCN2021082038-appb-100015
    (2.2)将给定转矩T e *与计算转矩之差经转速PI调节器后得到转矩角增量Δδ，并通过磁链自适应给定控制策略得到定子磁链的给定值ψ s *；
    五相永磁电机的电磁转矩本质是转子磁场与定子磁场相互作用的结果，即有：
    Figure PCTCN2021082038-appb-100016
    其中，
    Figure PCTCN2021082038-appb-100017
    为转子磁场矢量；
    Figure PCTCN2021082038-appb-100018
    为定子磁场矢量；δ为定子磁链与转子磁链的夹角，即定子磁链相位角；
    对上式两边求导，得：
    Figure PCTCN2021082038-appb-100019
    故转矩偏差ΔT e与转矩角增量Δδ之间具有非线性关系，因此，转矩角Δδ可由ΔT e通过PI调节器后得到；
    另外，定子磁链给定若为定值，当电机空载或突加重载运行时，需要额外的直轴电流分量来维持定子磁链不变；发明将直轴电流i d1与零作差比较后经PI调节器作为定子磁链给定，因此，给定的定子磁链可根据负载情况进行自适应地调节，以确保电机运行于不同工况下的直轴电流分量为零；
    (2.3)将定子磁链大小和相位、转矩角增量Δδ以及定子磁链的给定值ψ s *通过预期电压计算，得到两相旋转坐标系上的交直轴基波电压参考值u d1 *和u q1 *；
    根据五相永磁容错电机交直轴电压方程，R s为定子电阻，得到交直轴基波电压参考值的表达式为
    Figure PCTCN2021082038-appb-100020
  5. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述步骤6)的具体实现方式是将两相旋转坐标系下的电流分量i d3、i q3分别与零作差比较后经PI调节器后得到相应的交直轴三次谐波电压。
  6. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述步骤7)的具体实现包括：
    五相永磁容错电机的相电压表达式为：
    Figure PCTCN2021082038-appb-100021
    其中，x＝A,B,C,D,E，R s为定子电阻，i x为相电流，e x为相反电势；
    则上式可改写为：
    Figure PCTCN2021082038-appb-100022
    当五相容错电机发生故障，上式可用交直轴电压的形式表示，则可得不同故障下的绕组相电压表达式，A相绕组故障下的绕组相电压表达式为：
    Figure PCTCN2021082038-appb-100023
    并且，u A1＝0，其中，u Ae1、u Be1、u Ce1、u De1、u Ee1为A相绕组故障时未考虑反电势情况下的五相绕组相电压；u ed1 *＝u d1 *-e d1 *，u eq1 *＝u q1 *-e q1 *；u ed3 *＝u d3 *-e d3 *，u eq1 *＝u q3 *-e q3 *；e d1 *、e q1 *、e d3 *、e q3 *分别为五相反电势交直轴分量，可通过相反电势经过五相静止坐标系到两相旋转坐标系的坐标变换矩阵得到；
    A、C两相绕组故障下的绕组相电压表达式为：
    Figure PCTCN2021082038-appb-100024
    其中，u Ae2、u Be2、u Ce2、u De2、u Ee2为A、C两相绕组故障时未考虑反电势情况下的五相绕组相电压；
    A、B两相绕组故障下的绕组相电压表达式为：
    Figure PCTCN2021082038-appb-100025
    其中，u Ae3、u Be3、u Ce3、u De3、u Ee3为A、B两相绕组故障时未考虑反电势情况下的五相绕组相电压。
  7. 根据权利要求6所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述步骤8)的具体实现包括：
    五相永磁容错电机开路故障下的反电势与正常运行时的反电势一致，由于五相永磁容错电机的永磁磁链幅值变化很小以及反电势的谐波含量很小，其五相反电势可表示为：
    Figure PCTCN2021082038-appb-100026
    其中，ψ f为永磁磁链幅值，ω为电角速度；
    将反电势代入五相永磁容错电机的相电压表达式，可得到不同故障模式下相应的容错电压给定值，A相绕组故障下的给定容错相电压表达式为：
    Figure PCTCN2021082038-appb-100027
    并且，u A *＝e A；
    A、C两相绕组故障下的给定容错相电压表达式为：
    Figure PCTCN2021082038-appb-100028
    A、B两相绕组故障下的给定容错相电压表达式为：
    Figure PCTCN2021082038-appb-100029
    因此，只需已知五相永磁容错电机驱动系统的直轴电压u d *和交轴电压u q *，基于给定容错相电压表达式，即可实现该系统在电机绕组开路故障状态的容错运行。
  8. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，所述开路故障统一容错控制策略还适用于其他基于采用脉宽调制CPWM方式的驱动控制系统的开路故障容错运行。
  9. 根据权利要求1所述的一种五相永磁容错电机矢量与直接转矩控制驱动系统的开路统一容错控制方法，其特征在于，矢量与直接转矩控制驱动系统的开路统一容错控制方法还适用于五相永磁直线电机控制系统。
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