WO2018018672A1 - 五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法 - Google Patents

五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法 Download PDF

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WO2018018672A1
WO2018018672A1 PCT/CN2016/095627 CN2016095627W WO2018018672A1 WO 2018018672 A1 WO2018018672 A1 WO 2018018672A1 CN 2016095627 W CN2016095627 W CN 2016095627W WO 2018018672 A1 WO2018018672 A1 WO 2018018672A1
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phase
fault
circuit
short
current
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PCT/CN2016/095627
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English (en)
French (fr)
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周华伟
刘国海
吉敬华
赵文祥
陈前
陈龙
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江苏大学
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Priority to US15/748,926 priority Critical patent/US10574164B2/en
Publication of WO2018018672A1 publication Critical patent/WO2018018672A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60K—ARRANGEMENT OR MOUNTING OF PROPULSION UNITS OR OF TRANSMISSIONS IN VEHICLES; ARRANGEMENT OR MOUNTING OF PLURAL DIVERSE PRIME-MOVERS IN VEHICLES; AUXILIARY DRIVES FOR VEHICLES; INSTRUMENTATION OR DASHBOARDS FOR VEHICLES; ARRANGEMENTS IN CONNECTION WITH COOLING, AIR INTAKE, GAS EXHAUST OR FUEL SUPPLY OF PROPULSION UNITS IN VEHICLES
    • B60K1/00—Arrangement or mounting of electrical propulsion units
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/50—Vector control arrangements or methods not otherwise provided for in H02P21/00- H02P21/36
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/06—Linear motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/06—Linear motors
    • H02P25/064—Linear motors of the synchronous type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H02P25/22—Multiple windings; Windings for more than three phases
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/0243—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the fault being a broken phase
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/027—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the fault being an over-current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/032—Preventing damage to the motor, e.g. setting individual current limits for different drive conditions
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60Y—INDEXING SCHEME RELATING TO ASPECTS CROSS-CUTTING VEHICLE TECHNOLOGY
    • B60Y2400/00—Special features of vehicle units
    • B60Y2400/60—Electric Machines, e.g. motors or generators

Definitions

  • the invention relates to a fault-tolerant control method for a non-adjacent two-phase fault of a five-phase permanent magnet motor, in particular to a fault-tolerant vector control method for a non-adjacent two-phase short-circuit of a five-compatible permanent magnet linear motor. It is suitable for aerospace, electric vehicles, deep sea, medical equipment, etc. where high reliability and dynamic performance of the motor are required.
  • fault-tolerant control is to optimize the fault-tolerant current for different applications, so that the output thrust or torque of the motor in the fault state is as smooth as possible, and the performance of the motor reaches or approaches the performance before the fault.
  • the Chinese invention patent application number 201510059387.2 patent "a short-circuit fault-tolerant control method for five-compatible short-circuit permanent magnet motor” is directed to a five-compatible misaligned surface-mounted permanent magnet rotating motor, which decomposes the effect of short-circuit fault on motor torque into two parts: One part is the effect of open circuit fault on torque; the other part is the influence of short circuit current on torque.
  • the principle of the magnetomotive force before and after the fault and the principle of the same magnitude of the current after the fault are used to optimize the phase current of the remaining non-faulty phase; for the torque ripple caused by the short-circuit current, the magnetomotive force after the fault is zero and The principle of minimum copper loss is used to find the non-fault phase compensation current; the last two parts of the current are added to obtain the current command of the remaining non-fault phase. According to the obtained residual non-fault phase current, the current hysteresis control strategy is adopted to control the five-compatible mis-surface-mounted permanent magnet rotating motor.
  • the method is used to suppress the magnitude of the residual non-fault phase compensation current caused by the short-circuit phase current to cause torque fluctuation, which is constant regardless of the motor speed, and the sum of the currents of the remaining non-fault phases is not zero.
  • this method is not suitable for fault-tolerant operation in the case of two-phase faults (open circuit, short circuit or one phase open one phase short circuit).
  • the commonly used fault-tolerant control method is to calculate the fault-tolerant current and then use the current hysteresis strategy to control.
  • this method has problems such as disordered switching frequency, large noise, and poor dynamic performance of the motor, and is not suitable for occasions where the power is large and the dynamic performance of the motor is high.
  • 201510661212.9 In-line hybrid magnetic material fault-tolerant cylindrical linear motor short-circuit fault-tolerant vector control
  • the same method is used to optimize the residual non-fault current, and then the vector control strategy is used to realize the vector operation of the one-phase short-circuit fault condition of the motor.
  • this method achieves high fault tolerance, high dynamic performance, and good current followability of this type of motor system under short-circuit fault conditions, this method cannot achieve two-phase fault (open circuit, short circuit or one phase open one phase short circuit) fault. In case the fault-tolerant vector runs.
  • the purpose of the invention is to overcome the fault of the adjacent two phases of the motor (open circuit, short circuit or one phase open circuit one phase short circuit), the current fault tolerance strategy uses the current hysteresis control to cause the inverter switching frequency to be disordered, the motor response speed is decreased, the dynamic performance is poor, The current cannot be accurately followed, the noise is severe, the traditional current PI control is difficult to adjust the parameters due to the contradiction between response fastness and overshoot, and the fault-tolerant operation of the two-phase fault condition cannot be realized by the existing fault-tolerant vector control strategy.
  • the non-adjacent two-phase short-circuit fault-tolerant vector control method for the five-phase permanent magnet embedded fault-tolerant linear motor used in the invention realizes accurate estimation of back EMF, reduces controller parameter adjustment difficulty, and realizes that the motor system is in phase High fault-tolerant performance, high dynamic performance, and good current in the case of adjacent two-phase faults (open, short, or one-phase open-phase short-circuit)
  • follow-up reduce CPU overhead, achieve constant inverter switching frequency, reduce noise, and improve the non-adjacent two-phase fault of the five-phase permanent magnet embedded fault-tolerant linear motor of the present invention (open circuit, short circuit or one phase open phase) Dynamic performance and reliability in the short circuit condition. It can solve any faults in the adjacent two phases of the motor (short circuit, open circuit, one phase short circuit and one phase open circuit), which has good versatility.
  • a fault-tolerant vector control method for non-adjacent two-phase short-circuit of a five-phase permanent magnet embedded fault-tolerant linear motor includes the following steps:
  • Step 1 Establish a five-phase permanent magnet embedded fault-tolerant linear motor model
  • Step 2 The permanent magnet embedded error-tolerant linear motor is divided into five phases: A, B, C, D, and E.
  • A, B, C, D, and E When the motor has a B-phase and E-phase short-circuit fault, it is assumed that only the open fault of the B phase and the E phase of the motor occurs.
  • the traveling wave magnetomodynamic force is invariant before and after the motor fault and the residual non-fault phase current is zero, then the adjacent two-phase C-phase and D-phase current amplitudes are equal as constraints, and the B-phase sum is obtained.
  • Step 3 According to the non-fault phase current, obtain the generalized Clark transform matrix T post and the inverse transform matrix of the three non-fault phase natural coordinate systems to the two-phase stationary coordinate system transformation Transposed matrix
  • Step 4 using the non-fault phase current to suppress the thrust fluctuation caused by the fault phase short-circuit current, and obtaining the short-circuit compensation current for suppressing the thrust phase fluctuation caused by the fault phase short-circuit current, and transforming the current by using the generalized Clark transform matrix T post Short-circuit compensation current to the two-phase stationary coordinate system;
  • Step 5 using a generalized Clark transform matrix T post to transform the remaining three-phase non-faulty phase current sampled in the natural coordinate system to the current on the two-phase stationary coordinate system, and subtracting the current from the current obtained in step 4 Obtaining (i ⁇ , i ⁇ ), transforming (i ⁇ , i ⁇ ) to the current on the synchronous rotating coordinate system using the Parker transformation matrix C 2s/2r ;
  • step 5 subtracting the residual three-phase non-fault phase current sampled in the natural coordinate system with the short-circuit compensation current to obtain (i' A , i' C , i' D ), using the generalized Clark transform matrix T post and The Parker transformation matrix C 2s/2r transforms (i' A , i' C , i' D ) into a feedback current on a synchronous rotating coordinate system;
  • Step 6 Establish a mathematical model of the five-phase permanent magnet embedded fault-tolerant linear motor in the synchronous rotating coordinate system without the adjacent two-phase short-circuit fault state;
  • Step 7 Design a first-order inertia feedforward voltage compensator to obtain a feedforward compensation voltage, and the difference between the current command and the feedback current is added by the current internal mode controller to control the voltage and the feedforward compensation voltage to obtain a synchronous rotating coordinate system.
  • Voltage command using the Parker inverse transformation matrix C 2r/2s to convert the voltage command to a voltage on a two-phase stationary coordinate system
  • Step 8 adopt And the C 2r/2s and the mover permanent magnet flux design back potential observer observe the non-fault opposite potential, and find the fault opposite potential according to the non-fault opposite potential;
  • Step 9 To ensure that the motor output is used to suppress the short-circuit compensation current of the non-fault phase caused by the short-circuit current, according to the relationship between the B-phase short-circuit current and the B-opposing potential, and the relationship between the E-phase short-circuit current and the E-side potential and the short-circuit compensation current.
  • Mathematical expression defining the short-circuit compensation voltage of the remaining three-phase non-faulty phase, and transforming the compensation voltage to the short-circuit compensation voltage on the two-phase stationary coordinate system by using the generalized Clark transform matrix T post ;
  • Step 10 adding a voltage command on the two-phase stationary coordinate system and the short-circuit compensation voltage to obtain a voltage command
  • Generalized Clark inverse transformation matrix Voltage command Transform the voltage command to the natural coordinate system
  • step 10 using the generalized Clark inverse transformation matrix Voltage command in two-phase stationary coordinate system Transform the voltage command to the natural coordinate system Then, the short-circuit compensation voltage of the remaining three-phase non-fault phase is added, and finally, the opposite potentials of the remaining non-fault phases are respectively added to obtain a desired phase voltage command.
  • Step 11 the desired phase voltage command obtained in step 10
  • the CPWM modulation method is used to realize the fault-tolerant vector undisturbed operation of the five-phase permanent magnet embedded fault-tolerant linear motor without adjacent two-phase short-circuit fault.
  • the short circuit compensation current in step 4 is only set to zero, and the short circuit compensation voltage in step 9 is set to zero, and the fault tolerant vector control method can make the five phase permanent
  • the magnet embedded fault-tolerant linear motor operates fault-tolerant without adjacent two-phase open circuit faults.
  • the control method can make the five-phase permanent magnet embedded error-tolerant linear motor operate fault-tolerant in the case of B-phase open circuit and E-phase short-circuit fault.
  • the control method can make the five-phase permanent magnet embedded fault-tolerant linear motor operate fault-tolerant in the case of B-phase short circuit and E-phase open circuit fault.
  • the invention can not only ensure fault-tolerant operation of the motor in the case of non-adjacent two-phase short-circuit fault, but also can generate any fault (open circuit, short circuit or one-phase open circuit one-phase short circuit) in the non-adjacent two phases.
  • the dynamic performance, current following performance and the performance under normal conditions are consistent under the fault-tolerant operation of the motor.
  • the invention can not only effectively suppress the motor thrust fluctuation, but also more importantly, under the premise of ensuring that the motor output thrust is equal before and after the motor is not adjacent to two phases (open circuit, short circuit or one phase open circuit one phase short circuit).
  • the dynamic performance, current following performance and normal performance of the motor are consistent under fault-tolerant operation, and no complicated calculation is required.
  • the voltage source inverter has a constant switching frequency, low noise, and low CPU overhead. Can solve this type of motor is not adjacent to two Any fault that occurs in the phase (short circuit, open circuit, one phase short circuit, one phase open circuit) has good versatility.
  • the generalized Clark transform matrix and the Parker transform matrix derived from the residual non-faulty phase current vector in the present invention can leave the remaining non-faulty phase in the state of non-adjacent two-phase faults (open circuit, short circuit or one phase open one phase short circuit).
  • the steady-state current is subtracted from the short-circuit compensation current and then converted to the synchronous rotating coordinate system according to the equal amplitude, and the current is not pulsating.
  • the traditional Clark transform matrix and the Parker transform matrix can only transform the current of the remaining non-faulty phase to the pulsating current on the synchronous rotating coordinate system in the non-adjacent two-phase fault state.
  • the combination of the Clarke transformation matrix and the Parker transformation matrix realizes the transformation from the natural coordinate system of the non-faulty phase to the synchronous rotating coordinate system in the state of non-adjacent two-phase faults (open circuit, short circuit or one-phase open one-phase short circuit). It creates a precondition for fault-tolerant vector control in the case where the motor is not adjacent to two phases.
  • the combination of the transposed matrix and the Parker inverse transformation matrix of the Clark transform matrix and the permanent magnet flux linkage of the mover realizes the non-adjacent two-phase fault of the motor (open circuit, short circuit or one-phase open-phase one-phase short circuit).
  • the back EMF is accurately estimated to achieve fault-tolerant vector operation in the case of non-adjacent two-phase faults (open, short or one-phase open-phase short-circuit).
  • the current internal model controller is combined with the generalized Clark inverse transformation matrix, the Parker inverse transformation matrix, the back EMF observer and the first-order inertia feedforward voltage compensator.
  • the nonlinear strong coupling system under two-phase fault (open circuit, short circuit or one-phase open-phase one-phase short circuit) is transformed into a first-order inertial system, which reduces the difficulty of setting the controller parameters and ensures that the motor system is not adjacent to two phases.
  • the current following performance, steady state performance, and dynamic performance in the fault state make the motor dynamic performance, steady state performance and performance before the motor fault consistent, and can achieve no overshoot and fast response.
  • Non-adjacent two-phase fault tolerant vector control strategy back EMF estimation strategy, current internal mode control strategy, first-order inertia feedforward voltage compensation strategy, CPWM modulation technology combined with five-phase permanent magnet embedded fault-tolerant linear motor,
  • the fault-tolerant performance, dynamic performance and steady-state performance of the motor in the state of non-adjacent two-phase failure are greatly improved, and the CPU overhead is saved. Reduces noise and reduces compared to current hysteresis control The difficulty of electromagnetic compatibility design.
  • the motor has high control precision, good current following performance, high motor efficiency, fast output response speed and thrust pulsation before the fault in the state of non-adjacent two-phase failure (open circuit, short circuit or one-phase open-phase one-phase short circuit).
  • FIG. 1 is a schematic structural view of a five-phase permanent magnet embedded fault-tolerant linear motor according to an embodiment of the present invention
  • FIG. 2 is a schematic diagram of a vector control strategy of a five-phase permanent magnet embedded fault-tolerant linear motor according to an embodiment of the present invention
  • FIG. 3 is a first schematic diagram of the control of the B-phase and E-phase short-circuit fault-tolerant vector of the five-phase permanent magnet embedded fault-tolerant linear motor according to an embodiment of the present invention
  • FIG. 4 is a schematic diagram of the control principle of the B-phase and E-phase short-circuit fault-tolerant vector control of the five-phase permanent magnet embedded fault-tolerant linear motor according to the embodiment of the present invention
  • FIG. 5 is a phase current waveform of a fault-free and fault-tolerant vector operation in the case of a B-phase and an E-phase short-circuit fault according to an embodiment of the present invention
  • FIG. 6 is a thrust waveform of a fault-free and fault-tolerant vector operation in the case of a B-phase and an E-phase short-circuit fault according to an embodiment of the present invention
  • FIG. 7 is a current waveform on a synchronous rotating coordinate system in a step of a thrust command during a faultless operation according to an embodiment of the present invention
  • FIG. 8 is a waveform of a motor output thrust when a thrust command step is during a faultless operation according to an embodiment of the present invention
  • FIG. 10 is a waveform diagram of the output thrust of the motor when the thrust command step is in the process of fault-tolerant operation of phase B and phase E of the embodiment of the present invention
  • Step 1 Establish a five-phase permanent magnet embedded fault-tolerant linear motor model.
  • a schematic diagram of a five-phase permanent magnet embedded fault-tolerant linear motor includes a primary 1 and a secondary 2 .
  • the primary 1 includes a pole piece 4, an armature tooth 6, a fault-tolerant tooth 5, and a concentrated winding coil 9, and 10 of the armature teeth 6 and the fault-tolerant teeth 5 are embedded with a rare earth permanent magnet 8 on the secondary 2, the primary 1
  • the two end teeth 7 of the primary 1 are asymmetrical and wider than the fault-tolerant teeth and the armature teeth.
  • the zero sequence of c 0 -(max(u i )+min(u i ))/2 is injected into the five-phase sinusoidal modulated wave.
  • the CPWM method of voltage harmonics (u i is a five-phase sinusoidal modulation wave per phase function) and the five-phase SVPWM method can obtain the same flux linkage control effect. Therefore, the present invention performs pulse width modulation using a CPWM method based on injection of zero-sequence voltage harmonics.
  • the five-phase permanent magnet embedded fault-tolerant linear motor is powered by a voltage source inverter.
  • the motor is divided into five phases A, B, C, D, and E.
  • the vector control strategy of CPWM technology based on zero-sequence voltage harmonic injection is adopted.
  • the zero sequence current control is zero, and the control block diagram is shown in Figure 2.
  • the phase winding current can be expressed as
  • the traveling wave magnetomotive force (MMF) generated by the motor can be expressed as
  • N is the effective number of turns of the stator windings of each phase.
  • Step 2 When the motor has a B-phase and E-phase short-circuit fault, it is assumed that only the open-circuit fault occurs in the B-phase and the E-phase of the motor. According to the principle that the traveling wave magnetomotive force is unchanged before and after the motor fault and the sum of the remaining non-faulted phase currents is zero. Constraint condition, then the adjacent two-phase C-phase and D-phase current amplitudes are equal as the constraint conditions, and the non-faulty phase current of the fault-tolerant operation of the motor after the B-phase and E-phase open-circuit faults are obtained.
  • the B phase and the E phase have a short circuit fault.
  • the remaining non-faulty phase current of the motor compensates for the short-circuit fault phase, resulting in the loss of normal thrust of the two phases.
  • the phase current is zero, and the traveling wave magnetomotive force inside the motor is generated by the remaining three-phase non-faulty phase winding, which can be expressed as
  • the motor windings are connected in a star shape and their center point is not connected to the center point of the DC bus voltage. Therefore, the sum of the phase currents of the windings is zero. Optimize the non-faulty phase current based on the principle that the amplitudes of adjacent two-phase currents are equal, assuming
  • I C and I D are the C-phase and D-phase current amplitudes, respectively.
  • the non-fault phase current is optimized by the above constraints, and the phase current command of the fault-tolerant operation of the motor is
  • Equation (5) can be expressed as a matrix
  • Step 3 According to the non-fault phase current, obtain the generalized Clark transform matrix T post and the inverse transform matrix of three rows and two columns of three non-fault phase natural coordinate systems to two rows and three columns of two-phase stationary coordinate system transformation. Transposed matrix
  • the transformation matrix of the two-phase stationary coordinate system to the remaining non-faulty phase natural coordinate system is defined as
  • Step 4 Using the non-fault phase current to suppress the thrust fluctuation caused by the fault phase short-circuit current, and obtain the short-circuit compensation current (i′′ A , i′′ C , i′′ D of the non-fault phase for suppressing the thrust phase fluctuation caused by the fault phase short-circuit current .
  • the short-circuit compensation current (i′′ A , i′′ C , i′′ D ) is transformed into a short-circuit compensation current (i′′ ⁇ , i′′ ⁇ ) on the two-phase stationary coordinate system by using the generalized Clark transform matrix T post .
  • the transformation matrix C 2s/2r and the inverse transformation matrix C 2r/2s defining the two-phase stationary coordinate system to the synchronous rotating coordinate system are respectively
  • the non-fault phase current is used to suppress the thrust fluctuation caused by the short-circuit phase current.
  • ⁇ fB is the angle between the opposite potential of B and the short-circuit current of the phase
  • ⁇ fE is the angle between the E opposite potential and the short-circuit current of the phase
  • ⁇ ⁇ v / ⁇ , v linear motor motor motion electric velocity
  • ⁇ is the pole distance.
  • x A , y A , x C , y C , x D , y D are the amplitudes of the non-fault phase compensation current cosine term and the sine term, respectively.
  • the non-fault phase is used to suppress the fault phase short-circuit current
  • the sum of the compensation currents of the thrust fluctuation is zero, and the combined magnetomotive force of the current and the short-circuit fault phase current is zero
  • the short-circuit current for suppressing the fault phase is obtained.
  • Short-circuit compensation current for non-faulty phases that cause thrust fluctuations i′′ A , i′′ C , i′′ D
  • the third part is the mathematical model when the motor does not have short-circuit faults in two adjacent phases.
  • the model of the motor under the short-circuit fault state in the natural coordinate system can be expressed as
  • u A , u C and u D are the phase voltages of the non-faulty phase of the motor; e A , e C and e D are the back EMF of the non-faulty phase of the motor; u Ae , u Ce , and u De are non-faulty of the motor
  • the phase voltages are respectively subtracted from the voltages of the opposite potentials; R is the phase resistance.
  • Step 5 Transform the residual three-phase non-faulty phase currents (i A , i C , i D ) sampled in the natural coordinate system to the current on the two-phase stationary coordinate system using the generalized Clark transform matrix T post (i' ⁇ , i' ⁇ ), and subtract the current and the short-circuit compensation current (i′′ ⁇ , i′′ ⁇ ) to obtain (i ⁇ , i ⁇ ), and use the Parker transformation matrix C 2s/2r to (i ⁇ , i ⁇ ) Transforms the current (i d , i q ) onto the synchronous rotating coordinate system.
  • the remaining three-phase non-faulty phase currents (i A , i C , i D ) sampled on the natural coordinate system, and the short-circuit compensation currents used to suppress the non-faulty phase of the short-circuit fault phase currents causing thrust fluctuations (i " A , i" C , i" D ) is subtracted to obtain (i' A , i' C , i' D ), using the generalized Clark transform matrix T post and the Parker transform matrix C 2s / 2r (i' A , i ' C , i' D ) transforms the feedback current (i d , i q ) onto the synchronous rotating coordinate system.
  • Step 6 Establish a mathematical model of the five-phase permanent magnet embedded fault-tolerant linear motor in the synchronous rotating coordinate system without the adjacent two-phase short-circuit fault state.
  • the motor non-adjacent two-phase short-circuit fault model (19) is transformed into a synchronous rotating coordinate system.
  • the thrust equation of the motor in the fault-tolerant state of the non-adjacent two-phase short-circuit fault is derived from equations (5)-(18).
  • ⁇ m is a permanent magnet flux linkage
  • the five-phase permanent magnet embedded fault-tolerant linear motor of the present invention can output the desired thrust in the non-adjacent two-phase short-circuit fault state.
  • the fourth part the motor non-adjacent two-phase short-circuit fault-tolerant vector control strategy
  • Step 7 design a first-order inertia feedforward voltage compensator, synchronous current command on the rotating coordinate system First-order inertia Compensation voltage
  • the voltage command is implemented using the Parker inverse transform matrix C 2r/2s Transform to a voltage on a two-phase stationary coordinate system
  • Step 8 adopt And C 2r/2s and the mover permanent magnet flux design back EMF observer to observe the non-faulty opposite potential (e A , e C , e D )
  • the short-circuit compensation voltage of the three-phase non-faulty phase is (u′′ A , u′′ C , u′′ D )
  • Step 10 voltage command on the two-phase stationary coordinate system Adding to the short-circuit compensation voltage (u′′ ⁇ , u′′ ⁇ )
  • step 10 using the generalized Clark inverse transformation matrix Voltage command in two-phase stationary coordinate system Transform the voltage command to the natural coordinate system Then, the short-circuit compensation voltages (u′′ A , u′′ C , u′′ D ) of the remaining three-phase non-fault phases are added, and finally the opposite potentials (e A , e C , e D ) of the remaining non-fault phases are respectively Add the desired phase voltage command
  • Step 11 the desired phase voltage command obtained in step 10
  • the CPWM modulation method is used to realize the fault-tolerant vector undisturbed operation of the five-phase permanent magnet embedded fault-tolerant linear motor without adjacent two-phase short-circuit fault.
  • the desired phase voltage of equation (29) or equation (30) is implemented by the voltage source inverter using CPWM modulation based on zero-sequence voltage harmonic injection to realize the five-phase permanent magnet embedded fault-tolerant linear motor in the case of phase B and E-phase short-circuit faults. Undisturbed fault tolerant operation.
  • the high performance non-adjacent two-phase short-circuit fault-tolerant vector control strategy proposed by the present invention is shown in FIGS. 3 and 4.
  • the short-circuit compensation current in step 4 is only set to zero, and the short-circuit compensation voltage in step 9 is set to zero.
  • the fault-tolerant vector control method can make the five-phase permanent magnet
  • the embedded fault-tolerant linear motor operates fault-tolerant without adjacent two-phase open-circuit faults.
  • the control method can make the five-phase permanent magnet embedded error-tolerant linear motor operate fault-tolerant in the case of B-phase open circuit and E-phase short-circuit fault.
  • the control method can make the five-phase permanent magnet embedded fault-tolerant linear motor operate fault-tolerant in the case of B-phase short circuit and E-phase open circuit fault.
  • the five-phase permanent magnet embedded fault-tolerant line shown in Figure 1 is established in Matlab/Simulink.
  • the simulation model of the control system of the motor is simulated by the system.
  • the simulation results of the five-phase permanent magnet embedded fault-tolerant linear motor without adjacent two-phase short-circuit fault-tolerant vector control are obtained.
  • Figure 5 is the phase current waveform of the B-phase and E-phase short-circuit faults.
  • the 0.1s short-circuit fault occurs, the current waveform is distorted, and the fault-tolerant vector control strategy of the present invention is applied for 0.2s, and the current sinusoidality is improved.
  • Figure 6 shows the thrust waveforms of the B-phase and E-phase short-circuit faults.
  • the short-circuit fault occurs at 0.1 s, the output thrust of the motor fluctuates significantly, and the short-circuit fault-tolerant vector control strategy of the present invention is applied for 0.2 s.
  • the output thrust pulsation of the motor is significantly suppressed, and there is almost no pulsation.
  • Figure 7 and Figure 8 show the current and motor output thrust response in the synchronous rotating coordinate system when the thrust command changes stepwise during normal operation of the motor.
  • the thrust response time is 0.2ms.
  • Figure 9 and Figure 10 show the current and motor output thrust response on the synchronous rotating coordinate system when the thrust command is stepped after applying the short-circuit fault-tolerant vector control strategy of the present invention in the case of a short-circuit fault in the B-phase and E-phase of the motor.
  • the time is 0.3ms. Therefore, the five-phase permanent magnet embedded fault-tolerant linear motor non-adjacent two-phase fault-tolerant vector strategy of the present invention enables the motor to have dynamic performance and steady-state performance during normal operation. In addition, the current following performance is good, and the fault-free fault-tolerant operation is realized.
  • the five-phase permanent magnet embedded fault-tolerant linear motor non-adjacent two-phase short-circuit fault-tolerant vector control strategy can ensure not only two adjacent phase faults under the maximum current allowed by the motor drive system ( When the open circuit, short circuit or one-phase open circuit is short-circuited, the output thrust of the motor is consistent with the normal state, and the thrust fluctuation after the failure of the two adjacent phases of the motor can be obviously suppressed. More importantly, the dynamic performance is similar to that before the fault. , steady-state performance and current following accuracy, and suitable for any non-adjacent two-phase fault (open circuit, short circuit or one-phase open-phase one-phase short circuit), strong versatility, no complicated calculation, small CPU overhead, current regulator parameters The setting is simple. Therefore, the present invention has a good application prospect in an electromagnetic active suspension system and the like which have high operational reliability requirements.

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Abstract

一种五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法。先根据这两相开路故障前后磁动势不变以及非故障相电流和为零的原则，以相邻两相电流幅值相等为约束条件，推导出推广克拉克变换矩阵；采用该矩阵的转置估算出反电势；采用内模控制器、一阶惯性前馈电压补偿器、反电势观测器将该类电机在故障状态下的非线性强耦合系统变换为一阶惯性系统。根据非故障相短路补偿电流和两短路故障相电流的合成磁动势为零的原则求出短路补偿电压；将该电压和矢量控制器输出电压叠加。该控制方法不但抑制了电机不相邻两相短路故障导致的电机推力波动，且更为关键的是其动态性能、稳态性能和正常状态下一致，电压源逆变器开关频率恒定。

Description

五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法 技术领域
本发明涉及一种五相永磁电机不相邻两相故障容错控制方法，特别是五相容错永磁直线电机不相邻两相短路容错矢量控制方法。适用于航空航天、电动汽车、深海、医疗器械等对电机的可靠性和动态性能有较高要求的场合。
背景技术
随着社会的发展以及人们生活水平的提高，对汽车驾乘的舒适性和安全稳定性要求越来越高。作为现代汽车的重要组成部分，悬架系统性能对汽车行驶平顺性和操作稳定性等有着极其重要的影响，因此主动悬架系统的研究受到业内高度重视。作为主动电磁悬架系统的核心部件，圆筒直线电机研究受到重视。电机在短路故障状态下的容错性能，直接决定着电磁悬架的可靠性和连续运行的能力。
容错电机在某一相或某两相发生短路故障时，电机仍然具有一定的推力或者转矩输出能力，但是推力或者转矩波动很大，噪声增大，严重恶化系统性能。容错控制的目标是针对不同应用场合对容错电流进行优化，使电机在故障状态下的输出推力或者转矩尽量平滑，并且使电机性能达到或接近故障前的性能。中国发明专利申请号为201510059387.2的专利《一种五相容错永磁电机的短路容错控制方法》针对五相容错表贴式永磁旋转电机，将短路故障对电机转矩的影响分解为两部分：一部分是开路故障对转矩的影响；另一部分是短路电流对转矩影响。针对开路故障，采用故障前后磁动势和不变原则以及故障后电流幅值相等原则，优化剩余非故障相的相电流；针对短路电流引起的转矩波动，采用故障后磁动势为零和铜耗最小原则求出非故障相补偿电流；最后两部分电流相加，求得剩余非故障相的电流指令。根据求得的剩余非故障相电流采用电流滞环控制策略，对五相容错表贴式永磁旋转电机进行控制。该方法用于抑制短路相电流导致转矩波动的剩余非故障相补偿电流的幅值是常数，与电机转速无关，且剩余非故障相的电流之和不为零。同时，该方法不适合两相故障(开路、短路或一相开路一相短路)情况下的容错运行。目前，常用的容错控制方法是：计算出容错电流，然后采用电流滞环策略进行控制。但是，该方法存在开关频率杂乱、噪声大、电机动态性能差等问题，不适合功率较大以及对电机动态性能要求高的场合。中国发明专利申请号为201510661212.9的专利《一种内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方 法》针对五相混合磁材料内嵌式容错直线电机一相短路故障，采用以上相同方法优化剩余非故障电流，然后采用矢量控制策略实现该电机一相短路故障情况的矢量运行。尽管该方法实现了该类电机系统在短路故障状态下的高容错性能、高动态性能、电流良好的跟随性，但该方法无法实现两相故障(开路、短路或一相开路一相短路)故障情况下的容错矢量运行。
发明内容
针对现有电机容错控制技术中存在的不足，根据五相永磁体内嵌式容错直线电机的特性和该类电机不相邻两相故障(开路、短路或一相开路一相短路)特点，本发明目的是克服电机不相邻两相故障(开路、短路或一相开路一相短路)后现有容错策略使用电流滞环控制导致逆变器开关频率杂乱、电机响应速度下降、动态性能差、电流无法精确跟随、噪声严重的缺点、传统电流PI控制由于响应快速性和超调的矛盾引起参数调节困难的问题，以及现有容错矢量控制策略无法实现两相故障情况的容错运行，提出一种用于本发明的五相永磁体内嵌式容错直线电机的不相邻两相短路容错矢量控制方法，实现了反电势的精确估算，降低控制器参数调节难度，实现该类电机系统在不相邻两相故障(开路、短路或一相开路一相短路)状态下的高容错性能、高动态性能、电流良好跟随性，减小CPU开销，实现逆变器开关频率恒定、降低噪声，进而提高本发明的五相永磁体内嵌式容错直线电机不相邻两相故障(开路、短路或一相开路一相短路)状态下的动态性能和可靠性。能解决该类电机相邻两相发生的任意故障(短路、开路、一相短路一相开路)，具有很好通用性。
本发明用于五相永磁体内嵌式容错直线电机的容错矢量控制方法采用如下技术方案：
一种用于五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法，包括以下步骤：
步骤1，建立五相永磁体内嵌式容错直线电机模型；
步骤2，永磁体内嵌式容错直线电机分为A、B、C、D、E这五相，当电机发生B相和E相短路故障时，假设电机B相和E相仅发生开路故障，根据电机故障前后行波磁动势不变原则以及剩余非故障相电流之和为零的约束条件，再由相邻两相C相和D相电流幅值相等作为约束条件，求出B相和E相开路故障后电机容错运行的非故障相电流；
步骤3，根据非故障相电流，求取三个非故障相自然坐标系到两相静止坐标系变换的推广克拉克变换矩阵Tpost、逆变换矩阵
Figure PCTCN2016095627-appb-000001
以及转置矩阵
Figure PCTCN2016095627-appb-000002
步骤4，使用非故障相电流抑制故障相短路电流导致的推力波动，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流，采用推广克拉克变换矩阵Tpost将该电流变换到两相静止坐标系上的短路补偿电流；
步骤5，采用推广克拉克变换矩阵Tpost将在自然坐标系下采样到的剩余三相非故障相电流变换到两相静止坐标系上的电流，并将该电流和步骤4中获得的电流相减得到(iα、iβ)，运用派克变换矩阵C2s/2r将(iα、iβ)变换到同步旋转坐标系上的电流；
或步骤5，将在自然坐标系上采样到的剩余三相非故障相电流，与短路补偿电流相减得到(i′A、i′C、i′D)，采用推广克拉克变换矩阵Tpost和派克变换矩阵C2s/2r将(i′A、i′C、i′D)变换到同步旋转坐标系上的反馈电流；
步骤6，建立五相永磁体内嵌式容错直线电机不相邻两相短路故障状态下在同步旋转坐标系上的数学模型；
步骤7，设计一阶惯性前馈电压补偿器获得前馈补偿电压，同时该电流指令和反馈电流的差值经电流内模控制器得控制电压与前馈补偿电压相加得到同步旋转坐标系上的电压指令，采用派克逆变换矩阵C2r/2s将该电压指令变换到两相静止坐标系上的电压
Figure PCTCN2016095627-appb-000003
Figure PCTCN2016095627-appb-000004
步骤8，采用
Figure PCTCN2016095627-appb-000005
和C2r/2s以及动子永磁磁链设计反电势观测器观测出非故障相反电势，根据非故障相反电势求出故障相反电势；
步骤9，为确保电机输出用于抑制短路电流导致推力波动的非故障相的短路补偿电流，根据B相短路电流和B相反电势的关系以及E相短路电流和E相反电势的关系以及短路补偿电流的数学表达方式，定义剩余三相非故障相的短路补偿电压，采用推广克拉克变换矩阵Tpost将所述补偿电压变换到两相静止坐标系上的短路补偿电压；
步骤10，将两相静止坐标系上的电压指令与短路补偿电压相加得电压指令
Figure PCTCN2016095627-appb-000006
Figure PCTCN2016095627-appb-000007
采用推广克拉克逆变换矩阵
Figure PCTCN2016095627-appb-000008
将电压指令
Figure PCTCN2016095627-appb-000009
变换到自然坐标系上的电 压指令
Figure PCTCN2016095627-appb-000010
再和剩余非故障相的各相反电势分别相加得到期望相电压指令
Figure PCTCN2016095627-appb-000011
或步骤10，采用推广克拉克逆变换矩阵
Figure PCTCN2016095627-appb-000012
将两相静止坐标系下的电压指令
Figure PCTCN2016095627-appb-000013
Figure PCTCN2016095627-appb-000014
变换到自然坐标系上的电压指令
Figure PCTCN2016095627-appb-000015
然后和剩余三相非故障相的短路补偿电压相加，最后再和剩余非故障相的各相反电势分别相加得到期望相电压指令
Figure PCTCN2016095627-appb-000016
Figure PCTCN2016095627-appb-000017
步骤11，将步骤10所得到的期望相电压指令
Figure PCTCN2016095627-appb-000018
经电压源逆变器，采用CPWM调制方法实现五相永磁体内嵌式容错直线电机不相邻两相短路故障后的容错矢量无扰运行。
进一步，当B相和E相仅发生开路故障时，只需将步骤4中的短路补偿电流设为零，步骤9中的短路补偿电压设为零，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在不相邻两相开路故障情况下容错运行。
当B相开路和E相短路故障发生时，只需使步骤4中短路补偿电流表达式中的isc_B＝0，使步骤9中的短路补偿电压表达式中的eB＝0，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在B相开路和E相短路故障情况下容错运行。
当B相短路和E相开路故障发生时，只需使步骤4中短路补偿电流表达式中的isc_E＝0，使步骤9中的短路补偿电压表达式中的eE＝0，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在B相短路和E相开路故障情况下容错运行。
本发明具有以下有益效果：
1、本发明不但能保证该类电机在不相邻两相短路故障情况下容错运行，而且能在不相邻两相发生任意故障(开路、短路或一相开路一相短路)情况下，能使电机容错运行情况下的动态性能、电流跟随性能和正常状态下的性能一致。
2、本发明在保证电机任意不相邻两相故障(开路、短路或一相开路一相短路)前后电机输出推力相等的前提下，不但能有效抑制电机推力波动，而且更为关键的是能使电机容错运行情况下的动态性能、电流跟随性能和正常状态下的性能一致，并且无需复杂的计算，电压源逆变器开关频率恒定、噪声低、CPU开销小。能解决该类电机不相邻两 相发生的任意故障(短路、开路、一相短路一相开路)，具有很好通用性。
3、采用本发明不相邻两相故障容错矢量控制策略后，该类电机在B相和E相故障情况下，容错运行时，其动态性能、稳态性能和电机正常状态下一样，且输出推力几乎没有波动，在电机系统允许的最大电流极限值以下，电磁推力和故障前保持一致，实现了无扰容错运行。
4、由本发明中的剩余非故障相电流矢量推导出的推广克拉克变换矩阵和派克变换矩阵能在不相邻两相故障(开路、短路或一相开路一相短路)状态下将剩余非故障相的稳态电流减去短路补偿电流后按等幅值变换到同步旋转坐标系上，该电流无脉动。而采用传统克拉克变换矩阵及派克变换矩阵在不相邻两相故障状态下只能将剩余非故障相的电流变换到同步旋转坐标系上脉动的电流。
5、推广克拉克变换矩阵和派克变换矩阵相结合实现了不相邻两相故障(开路、短路或一相开路一相短路)状态下剩余非故障相构成的自然坐标系到同步旋转坐标系的变换，为电机不相邻两相故障状态下的容错矢量控制创造了前提条件。
6、推广克拉克变换矩阵的转置矩阵和派克逆变换矩阵和动子永磁磁链相结合实现了该类电机不相邻两相故障(开路、短路或一相开路一相短路)情况下的反电势精确估算，从而实现了该类电机不相邻两相故障(开路、短路或一相开路一相短路)情况下的容错矢量运行。
7、和电流PI控制器相比，电流内模控制器和推广克拉克逆变换矩阵、派克逆变换矩阵、反电势观测器以及一阶惯性前馈电压补偿器相结合将该类电机在不相邻两相故障(开路、短路或一相开路一相短路)状态下的非线性强耦合系统变换为一阶惯性系统，降低了控制器参数整定难度，保证了该类电机系统在不相邻两相故障状态下电流跟随性能、稳态性能、动态性能，使电机动态性能、稳态性能和电机故障前的性能一致，并且能够实现无超调快速响应。
8、推广克拉克变换矩阵和派克变换矩阵以及零序电压谐波注入的CPWM调制相结合，提高了逆变器母线电压利用率，同时减小了容错矢量控制算法的复杂性，降低了CPU开销。
9、不相邻两相故障容错矢量控制策略、反电势估算策略、电流内模控制策略、一阶惯性前馈电压补偿策略、CPWM调制技术与五相永磁体内嵌式容错直线电机相结合，大大提高了该电机在不相邻两相故障(开路、短路或一相开路一相短路)状态下的容错性能、动态性能和稳态性能，节省了CPU开销。和电流滞环控制相比，降低了噪声，降低 了电磁兼容设计难度。进而使得该电机在不相邻两相故障(开路、短路或一相开路一相短路)状态下控制精度高，电流跟随性能好，电机效率高、输出推力响应速度快且推力脉动和故障前一样小，实现了电机系统的在不相邻两相故障(开路、短路或一相开路一相短路)状态下的高可靠性以及高动态性能。
附图说明
图1为本发明实施例五相永磁体内嵌式容错直线电机的结构示意图；
图2为本发明实施例五相永磁体内嵌式容错直线电机矢量控制策略原理图；
图3为本发明实施例五相永磁体内嵌式容错直线电机B相和E相短路容错矢量控制原理图一；
图4为本发明实施例五相永磁体内嵌式容错直线电机B相和E相短路容错矢量控制原理图二；
图5为本发明实施例B相和E相短路故障情况下无容错和容错矢量运行时的相电流波形；
图6为本发明实施例B相和E相短路故障情况下无容错和容错矢量运行时的推力波形；
图7为本发明实施例无故障运行过程中推力指令阶跃时的同步旋转坐标系上的电流波形；
图8为本发明实施例无故障运行过程中推力指令阶跃时的电机输出推力波形；
图9为本发明实施例B相和E相短路容错运行过程中推力指令阶跃时的同步旋转坐标系上的电流波形；
图10为本发明实施例B相和E相短路容错运行过程中推力指令阶跃时的电机输出推力波形；
图中：1.初级；2.次级；3.硅钢片；4.极靴；5.容错齿；6.电枢齿；7.端部齿；8.永磁体；9.绕组线圈。
具体实施方式
下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述。
为了能够更加简单明了地说明本发明的永磁体内嵌式容错直线电机的结构特点和有益效果，下面结合一个具体的五相永磁体内嵌式容错直线电机来进行详细的表述。
步骤1，建立五相永磁体内嵌式容错直线电机模型。
如图1所示，本发明实施例的五相永磁体内嵌式容错直线电机结构示意图，包括初级1、次级2。初级1中包括极靴4、电枢齿6、容错齿5和集中绕组线圈9，且电枢齿6和容错齿5都为10个，次级2上内嵌有稀土永磁体8，初级1和次级2之间存在气隙，初级1和次级2上除永磁体、绕组和极靴之外的部分都是由硅钢片3轴向叠片而成，极靴4由电工纯铁制成，初级1的两个端部齿7是不对称的，且比容错齿和电枢齿宽。
在传统使用正弦波作为调制波的载波脉宽调制(CPWM)方法基础上，在五相正弦调制波中注入c0＝-(max(ui)+min(ui))/2的零序电压谐波(ui是五相正弦调制波每一相函数)的CPWM方法与五相SVPWM方法能获得相同的磁链控制效果。因此本发明采用基于注入零序电压谐波的CPWM方法进行脉宽调制。
五相永磁体内嵌式容错直线电机由电压源逆变器供电，该电机分为A、B、C、D、E五相，采用基于零序电压谐波注入的CPWM技术的矢量控制策略，零序电流控制为零，控制框图见图2所示。电机正常状态稳态运行时，各相绕组电流可表示为
Figure PCTCN2016095627-appb-000019
式中，
Figure PCTCN2016095627-appb-000020
分别是旋转坐标系d轴、q轴的电流指令，θ为电角度
Figure PCTCN2016095627-appb-000021
v直线电机动子运动电速度，τ为极距。
电机产生的行波磁动势(MMF)可表示为
Figure PCTCN2016095627-appb-000022
式中，a＝ej2π/5，N为各相定子绕组的有效匝数。
步骤2，当电机发生B相和E相短路故障时，假设电机B相和E相仅发生开路故障，根据电机故障前后行波磁动势不变原则以及剩余非故障相电流之和为零的约束条件，再由相邻两相C相和D相电流幅值相等作为约束条件，求出B相和E相开路故障后电机容错运行的非故障相电流。
第一部分，当电机不相邻两相发生故障时，假设B相和E相发生短路故障。先使用 电机剩余的非故障相电流补偿短路故障相导致这两相正常推力缺失。此时，假设B相和E相开路，其相电流为零，电机内部的行波磁动势由剩余的三相非故障相绕组产生，可表示为
Figure PCTCN2016095627-appb-000023
为实现电机不相邻两相短路故障后无扰运行，需保持电机故障前后行波磁动势一致，因此需调整剩余非故障相定子电流使电机故障前后行波磁动势的幅值与速度保持不变。于是，令式(2)、式(3)的实部与虚部均相等。
电机绕组采用星形连接，且其中心点与直流母线电压的中心点不相连，因此，绕组相电流之和为零。以相邻两相电流幅值相等为原则优化非故障相电流，假设
Figure PCTCN2016095627-appb-000024
式中，IC和ID分别是C相和D相电流幅值。
由上述约束条件优化非故障相电流，得电机容错运行的相电流指令为
Figure PCTCN2016095627-appb-000025
式(5)采用矩阵形式可表示为
Figure PCTCN2016095627-appb-000026
由式(6)得
Figure PCTCN2016095627-appb-000027
Figure PCTCN2016095627-appb-000028
步骤3，根据非故障相电流，求取三个非故障相自然坐标系到两相静止坐标系变换的两行三列的推广克拉克变换矩阵Tpost、三行两列的逆变换矩阵
Figure PCTCN2016095627-appb-000029
以及转置矩阵
Figure PCTCN2016095627-appb-000030
根据式(8)定义两相静止坐标系到剩余非故障相自然坐标系的变换矩阵为
Figure PCTCN2016095627-appb-000031
由于剩余非故障相电流之和为零，式(9)逆变换矩阵为
Figure PCTCN2016095627-appb-000032
式中，k＝0.386。
由于绕组星形连接，其相电流之和为零，因此去掉式(9)第三行和式(10)第三列，得
Figure PCTCN2016095627-appb-000033
Figure PCTCN2016095627-appb-000034
式(12)的转置矩阵为
Figure PCTCN2016095627-appb-000035
步骤4，使用非故障相电流抑制故障相短路电流导致的推力波动，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″A、i″C、i″D)，采用推广克拉克变换矩阵Tpost将短路补偿电流(i″A、i″C、i″D)变换到两相静止坐标系上的短路补偿电流(i″α、i″β)。
由于零序子空间的电流为零，不需要将其变换到同步旋转坐标系；基波子空间需要进行能量转换，因此将基波子空间的能量转换到同步旋转坐标系。因此定义两相静止坐标系到同步旋转坐标系的变换矩阵C2s/2r及其逆变换矩阵C2r/2s分别为
Figure PCTCN2016095627-appb-000036
Figure PCTCN2016095627-appb-000037
第二部分，在第一部分的基础上，当电机发生相短路故障时，使用非故障相电流抑制短路相电流导致的推力波动。
假设B相的短路电流为isc_B＝Ifcos(ωt-θfB)，E相的短路电流为isc_E＝Ifcos(ωt-θfE)，其中，If是短路电流的幅值，θfB是B相反电势和该相短路电流的夹角，θfE是E相反电势和该相短路电流的夹角；ω＝πv/τ，v直线电机动子运动电速度， τ为极距。
定义A相、C相和D相的补偿电流为：
Figure PCTCN2016095627-appb-000038
其中，xA、yA、xC、yC、xD、yD分别为非故障相补偿电流余弦项和正弦项的幅值。
根据非故障相用于抑制故障相短路电流导致推力波动的补偿电流之和为零、以及这部分电流和短路故障相电流的合成磁动势为零的原则，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″A、i″C、i″D)
Figure PCTCN2016095627-appb-000039
使用推广克拉克变换矩阵Tpost将非故障相补偿电流(i″A、i″C、i″D)变换到两相静止坐标系上的短路补偿电流(i″α、i″β)
Figure PCTCN2016095627-appb-000040
第三部分，电机不相邻两相发生短路故障时的数学模型
由于该容错永磁直线电机的相电感的互感相对自感很小，可忽略不计，假设相电感近似为常数，假设电机反电势为正弦波。反电势矢量角是由每相绕组在空间的位置决定的，因此反电势不能像电流一样使用本发明提出的坐标变换矩阵。因此，为了实现该类容错永磁直线电机在B相和E相短路故障状态下的矢量控制，该电机短路故障状态下在自然坐标系下的模型可表示为
Figure PCTCN2016095627-appb-000041
式中，uA、uC和uD是电机非故障相的相电压；eA、eC和eD是电机非故障相的反电势；uAe、uCe、和uDe是电机非故障相相电压分别减去各相反电势后的电压；R是相电阻。
步骤5，采用推广克拉克变换矩阵Tpost将在自然坐标系下采样到的剩余三相非故障相电流(iA、iC、iD)变换到两相静止坐标系上的电流(i′α、i′β)，并将该电流和短路补偿电流(i″α、i″β)相减得到(iα、iβ)，运用派克变换矩阵C2s/2r将(iα、iβ)变换到同步旋转坐标系上的电流(id、iq)。或者，将在自然坐标系上采样到的剩余三相非故障相电流(iA、iC、iD)，与用于抑制短路故障相电流导致推力波动的非故障相的短路补偿电流(i″A、i″C、i″D)相减得到(i′A、i′C、i′D)，采用推广克拉克变换矩阵Tpost和派克变换矩阵C2s/2r将(i′A、i′C、i′D)变换到同步旋转坐标系上的反馈电流(id、iq)。
步骤6，建立五相永磁体内嵌式容错直线电机不相邻两相短路故障状态下在同步旋转坐标系上的数学模型。
在自然坐标系上电机不相邻两相短路故障模型(19)变换到同步旋转坐标系为
Figure PCTCN2016095627-appb-000042
采用磁共能法，由式(5)-(18)推导出该电机在不相邻两相短路故障容错状态下推力方程
Figure PCTCN2016095627-appb-000043
式中，λm为永磁磁链。
因此，只要在同步旋转坐标系下控制id、iq就能使本发明中的五相永磁体内嵌式容错直线电机在不相邻两相短路故障状态下输出期望的推力。
第四部分，电机不相邻两相短路容错矢量控制策略
步骤7，设计一阶惯性前馈电压补偿器，同步旋转坐标系上的电流指令
Figure PCTCN2016095627-appb-000044
经一阶惯性环节
Figure PCTCN2016095627-appb-000045
得补偿电压
Figure PCTCN2016095627-appb-000046
Figure PCTCN2016095627-appb-000047
电流指令
Figure PCTCN2016095627-appb-000048
和反馈电流(id、iq)的差值经电流内模控制器
Figure PCTCN2016095627-appb-000049
获得控制电压(ud0、uq0)，将该电压与补偿电压相加得到同步旋转坐标系上的电压指令
Figure PCTCN2016095627-appb-000050
Figure PCTCN2016095627-appb-000051
Figure PCTCN2016095627-appb-000052
采用派克逆变换矩阵C2r/2s将该电压指令
Figure PCTCN2016095627-appb-000053
变换到两相静止坐标系上的电压
Figure PCTCN2016095627-appb-000054
步骤8，采用
Figure PCTCN2016095627-appb-000055
和C2r/2s以及动子永磁磁链设计反电势观测器观测出非故障相反电势(eA、eC、eD)
Figure PCTCN2016095627-appb-000056
根据非故障相反电势(eA、eC、eD)求出故障相反电势(eB、eE)
Figure PCTCN2016095627-appb-000057
步骤9，根据B相短路电流iB＝isc_B和B相反电势eB的关系、E相短路电流iE＝isc_E和E相反电势eE的关系以及短路补偿电流的数学表达方式，定义剩余三相非故障相的短路补偿电压为(u″A、u″C、u″D)为
Figure PCTCN2016095627-appb-000058
采用推广克拉克变换矩阵Tpost将(26)变换到两相静止坐标系上的短路补偿电压
Figure PCTCN2016095627-appb-000059
步骤10，两相静止坐标系上的电压指令
Figure PCTCN2016095627-appb-000060
与短路补偿电压(u″α、u″β)相加得
Figure PCTCN2016095627-appb-000061
采用推广克拉克逆变换矩阵
Figure PCTCN2016095627-appb-000062
将电压指令
Figure PCTCN2016095627-appb-000063
变换到自然坐标系上的电压指令
Figure PCTCN2016095627-appb-000064
再和剩余非故障相的各相反电势(eA、eC、eD)分别相加得到期望相电压指令
Figure PCTCN2016095627-appb-000065
Figure PCTCN2016095627-appb-000066
或者步骤10，采用推广克拉克逆变换矩阵
Figure PCTCN2016095627-appb-000067
将两相静止坐标系下的电压指令
Figure PCTCN2016095627-appb-000068
Figure PCTCN2016095627-appb-000069
变换到自然坐标系上的电压指令
Figure PCTCN2016095627-appb-000070
然后和剩余三相非故障相的短路补偿电压(u″A、u″C、u″D)相加，最后再和剩余非故障相的各相反电势(eA、eC、eD)分别相加得到期望相电压指令
Figure PCTCN2016095627-appb-000071
Figure PCTCN2016095627-appb-000072
步骤11，将步骤10所得到的期望相电压指令
Figure PCTCN2016095627-appb-000073
经电压源逆变器，采用CPWM调制方法实现五相永磁体内嵌式容错直线电机不相邻两相短路故障后的容错矢量无扰运行。
式(29)或式(30)期望相电压经电压源逆变器采用基于零序电压谐波注入的CPWM调制实现五相永磁体内嵌式容错直线电机B相和E相短路故障情况下的无扰容错运行。本发明提出的高性能不相邻两相短路故障容错矢量控制策略如图3和图4所示。
当B相和E相仅发生开路故障时，只需将步骤4中的短路补偿电流设为零，步骤9中的短路补偿电压设为零，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在不相邻两相开路故障情况下容错运行。
当B相开路和E相短路故障发生时，只需使步骤4中短路补偿电流表达式中的isc_B＝0，使步骤9中的短路补偿电压表达式中的eB＝0，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在B相开路和E相短路故障情况下容错运行。
当B相短路和E相开路故障发生时，只需使步骤4中短路补偿电流表达式中的isc_E＝0，使步骤9中的短路补偿电压表达式中的eE＝0，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在B相短路和E相开路故障情况下容错运行。
当其它不相邻两相发生故障时，只需将自然坐标系逆时针旋转
Figure PCTCN2016095627-appb-000074
(k＝0、1、2、3、4；B相和E相故障时，k＝0；C相和A相故障时，k＝1；D相和B相故障时，k＝2；E相和C相故障时，k＝3；A相和D相故障时，k＝4))电角度，此时派克变换矩阵及其逆变换矩阵分别为
Figure PCTCN2016095627-appb-000075
Figure PCTCN2016095627-appb-000076
按图2和图3或图4在Matlab/Simulink中建立图1所示五相永磁体内嵌式容错直线 电机的控制系统仿真模型，进行系统仿真，得五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制仿真结果。
图5是B相和E相短路故障下相电流波形，0.1s短路故障发生，电流波形发生畸变，0.2s施加本发明容错矢量控制策略，电流正弦度改善。图6是B相和E相短路故障下推力波形，0.1s时短路故障发生，电机输出推力波动明显，0.2s施加本发明短路容错矢量控制策略，电机输出推力脉动得到明显抑制，几乎没有脉动。图7和图8分别是电机正常运行过程中推力指令发生阶跃变化时的同步旋转坐标系上的电流和电机输出推力响应，推力响应时间为0.2ms。图9和图10是电机B相和E相发生短路故障情况下施加本发明短路容错矢量控制策略后推力指令发生阶跃变化时的同步旋转坐标系上的电流和电机输出推力响应，电机推力响应时间是0.3ms。因此，本发明五相永磁体内嵌式容错直线电机不相邻两相故障容错矢量策略能使电机具有正常运行时的动态性能和稳态性能。另外，电流跟随性能好，实现了无扰容错运行。
从以上所述可知，本发明用于五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制策略在电机驱动系统允许最大电流情况下，不但能保证不相邻两相故障(开路、短路或一相开路一相短路)时电机输出推力和正常状态下一致，而且能明显抑制电机不相邻两相故障后的推力波动，更为关键的是具有和故障前相近的动态性能、稳态性能和电流跟随精度，且适合任何不相邻两相发生故障(开路、短路或一相开路一相短路)的情况，通用性强，无需复杂计算，CPU开销小，电流调节器参数整定简单。因此，本发明在电磁主动悬架系统等对运行可靠性要求高的系统中拥有很好的应用前景。
虽然本发明已以较佳实施例公开如上，但实施例并不是用来限定本发明的。在不脱离本发明之精神和范围内，所做的任何等效变化或润饰，均属于本申请所附权利要求所限定的保护范围。

Claims (5)

  1. 一种五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法，其特征在于，包括以下步骤：
    步骤1，建立五相永磁体内嵌式容错直线电机模型；
    步骤2，永磁体内嵌式容错直线电机分为A、B、C、D、E五相，当电机发生B相和E相短路故障时，假设电机B相和E相仅发生开路故障，根据电机故障前后行波磁动势不变原则以及剩余非故障相电流之和为零的约束条件，再由相邻两相C相和D相电流幅值相等作为约束条件，求出B相和E相开路故障后电机容错运行的非故障相电流
    Figure PCTCN2016095627-appb-100001
    式中，
    Figure PCTCN2016095627-appb-100002
    分别是旋转坐标系下d轴、q轴的电流指令，θ为电角度
    Figure PCTCN2016095627-appb-100003
    v直线电机动子运动电速度，τ为极距。
    步骤3，根据非故障相电流，求取三个非故障相自然坐标系到两相静止坐标系变换的两行三列的推广克拉克变换矩阵Tpost、三行两列的逆变换矩阵
    Figure PCTCN2016095627-appb-100004
    以及转置矩阵
    Figure PCTCN2016095627-appb-100005
    Figure PCTCN2016095627-appb-100006
    Figure PCTCN2016095627-appb-100007
    Figure PCTCN2016095627-appb-100008
    步骤4，使用非故障相电流抑制故障相短路电流导致的推力波动，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″A、i″C、i″D)，采用推广克拉克变换矩阵Tpost将短路补偿电流(i″A、i″C、i″D)变换到两相静止坐标系上的短路补偿电流(i″α、i″β)；
    步骤5，采用推广克拉克变换矩阵Tpost将在自然坐标系下采样到的剩余三相非故障相电流(iA、iC、iD)变换到两相静止坐标系上的电流(i′α、i′β)，并将该电流和步骤4中获得的电流(i″α、i″β)相减得到(iα、iβ)，运用派克变换矩阵C2s/2r将(iα、iβ)变换到同步旋转坐标系上的电流(id、iq)；
    或步骤5，将在自然坐标系上采样到的剩余三相非故障相电流(iA、iC、iD)，与非故障相的短路补偿电流(i″A、i″C、i″D)相减得到(i′A、i′C、i′D)，采用推广克拉克变换矩阵Tpost和派克变换矩阵C2s/2r将(i′A、i′C、i′D)变换到同步旋转坐标系上的反馈电流(id、iq)；
    步骤6，建立五相永磁体内嵌式容错直线电机不相邻两相短路故障状态下在同步旋转坐标系上的数学模型；
    步骤7，设计一阶惯性前馈电压补偿器，同步旋转坐标系上的电流指令
    Figure PCTCN2016095627-appb-100009
    经一阶惯性环节
    Figure PCTCN2016095627-appb-100010
    获得前馈补偿电压
    Figure PCTCN2016095627-appb-100011
    电流指令
    Figure PCTCN2016095627-appb-100012
    和反馈电流(id、iq)的差值经电流内模控制器
    Figure PCTCN2016095627-appb-100013
    得控制电压(ud0、uq0)，将该电压与前馈补偿电压
    Figure PCTCN2016095627-appb-100014
    相加得到同步旋转坐标系上的电压指令
    Figure PCTCN2016095627-appb-100015
    采用派克逆变换矩阵C2r/2s将该电压指令变换到两相静止坐标系上的电压
    Figure PCTCN2016095627-appb-100016
    步骤8，采用
    Figure PCTCN2016095627-appb-100017
    和C2r/2s以及动子永磁磁链设计反电势观测器观测出非故障相反电势(eA、eC、eD)
    Figure PCTCN2016095627-appb-100018
    根据非故障相反电势(eA、eC、eD)求出故障相反电势(eB、eE)
    Figure PCTCN2016095627-appb-100019
    步骤9，为确保电机输出用于抑制短路电流导致推力波动的非故障相的短路补偿电流(i″A、i″C、i″D)，根据B相短路电流iB＝isc_B和B相反电势eB的关系、E相短路电流iE＝isc_E和E相反电势eE的关系以及短路补偿电流的数学表达方式，定义剩余三相非故障相的短路补偿电压为(u″A、u″C、u″D)为
    Figure PCTCN2016095627-appb-100020
    采用推广克拉克变换矩阵Tpost将所述短路补偿电压变换到两相静止坐标系上的短路补偿电压
    Figure PCTCN2016095627-appb-100021
    步骤10，将两相静止坐标系上的电压指令
    Figure PCTCN2016095627-appb-100022
    与短路补偿电压(u″α、u″β)相加得
    Figure PCTCN2016095627-appb-100023
    采用推广克拉克逆变换矩阵
    Figure PCTCN2016095627-appb-100024
    将电压指令
    Figure PCTCN2016095627-appb-100025
    变换到自然坐标系上的电压指令
    Figure PCTCN2016095627-appb-100026
    再和剩余非故障相的各相反电势(eA、eC、eD)分别相加得到期望相电压指令
    Figure PCTCN2016095627-appb-100027
    或步骤10，采用推广克拉克逆变换矩阵
    Figure PCTCN2016095627-appb-100028
    将两相静止坐标系下的电压指令
    Figure PCTCN2016095627-appb-100029
    Figure PCTCN2016095627-appb-100030
    变换到自然坐标系上的电压指令
    Figure PCTCN2016095627-appb-100031
    然后和剩余三相非故障相的短路补偿电压(u″A、u″C、u″D)相加，最后再和剩余非故障相的各相反电势(eA、eC、eD)分别相加得到期望相电压指令
    Figure PCTCN2016095627-appb-100032
    步骤11，将步骤10所得到的期望相电压指令
    Figure PCTCN2016095627-appb-100033
    经电压源逆变器，采用CPWM调制方法实现五相永磁体内嵌式容错直线电机不相邻两相短路故障后的容错矢 量无扰运行。
  2. 根据权利要求1所述五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法，其特征在于，还包括：
    当B相和E相仅发生开路故障时，只需将步骤4中的短路补偿电流设为零，步骤9中的短路补偿电压设为零，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在不相邻两相开路故障情况下容错运行；
    当B相开路和E相短路故障发生时，只需使步骤4中短路补偿电流表达式中的isc_B＝0，使步骤9中的短路补偿电压表达式中的eB＝0，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在B相开路和E相短路故障情况下容错运行；
    当B相短路和E相开路故障发生时，只需使步骤4中短路补偿电流表达式中的isc_E＝0，使步骤9中的短路补偿电压表达式中的eE＝0，该容错矢量控制方法就能使五相永磁体内嵌式容错直线电机在B相短路和E相开路故障情况下容错运行。
  3. 根据权利要求1所述五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法，其特征在于，所述步骤4的具体过程为：
    步骤4.1，假设B相的短路电流为isc_B＝Ifcos(ωt-θfB)，E相的短路电流为isc_E＝Ifcos(ωt-θfE)，其中，If是短路电流的幅值，θfB是B相反电势和该相短路电流的夹角，θfE是E相反电势和该相短路电流的夹角；ω＝πv/τ，v直线电机动子运动电速度，τ为极距；
    步骤4.2，根据非故障相用于抑制故障相短路电流导致推力波动的补偿电流之和为零、以及这部分电流和短路故障相电流的合成磁动势为零的原则，求取非故障相的短路补偿电流(i″A、i″C、i″D)
    Figure PCTCN2016095627-appb-100034
    步骤4.3，使用推广克拉克变换矩阵Tpost将非故障相短路补偿电流(i″A、i″C、i″D)变换到两相静止坐标系上的短路补偿电流(i″α、i″β)
    Figure PCTCN2016095627-appb-100035
  4. 根据权利要求1所述五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控 制方法，其特征在于，所述步骤6的具体过程为：
    步骤6.1，本发明相电感近似为常数Ls，电机相电压减去反电势后，电机B相和E相短路故障后在自然坐标系上的模型表示为
    Figure PCTCN2016095627-appb-100036
    式中，uA、uC和uD是电机非故障相的相电压；eA、eB、eC、eD和eE是电机相反电势；uAe、uCe、和uDe是电机非故障相相电压分别减去各相反电势后的电压；R是相电阻；
    步骤6.2，按步骤5对采样的非故障相电流进行处理，然后采用推广克拉克坐标变换矩阵Tpost和派克变换C2s/2r将自然坐标系上的电机不相邻两相短路故障模型变换到同步旋转坐标系上
    Figure PCTCN2016095627-appb-100037
    步骤6.3，采用磁共能法，由变换矩阵Tpost、
    Figure PCTCN2016095627-appb-100038
    C2s/2r和C2r/2s推导出该电机在不相邻两相短路故障容错状态下的推力方程
    Figure PCTCN2016095627-appb-100039
    式中，λm为永磁磁链。
  5. 根据权利要求1所述五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法，其特征在于，所述不相邻两相故障短路矢量控制方法还适用于五相容错永磁旋转电机控制系统。
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