WO2020147162A1 - 一种五相永磁电机一相短路容错直接转矩控制方法 - Google Patents

一种五相永磁电机一相短路容错直接转矩控制方法 Download PDF

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WO2020147162A1
WO2020147162A1 PCT/CN2019/075503 CN2019075503W WO2020147162A1 WO 2020147162 A1 WO2020147162 A1 WO 2020147162A1 CN 2019075503 W CN2019075503 W CN 2019075503W WO 2020147162 A1 WO2020147162 A1 WO 2020147162A1
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phase
fault
flux
short
tolerant
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PCT/CN2019/075503
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English (en)
French (fr)
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周华伟
周城
陶炜国
徐金辉
张多
刘国海
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江苏大学
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Priority to GB2017834.9A priority Critical patent/GB2587722B/en
Publication of WO2020147162A1 publication Critical patent/WO2020147162A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/0243—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the fault being a broken phase
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/027—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the fault being an over-current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/028—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the motor continuing operation despite the fault condition, e.g. eliminating, compensating for or remedying the fault

Definitions

  • the invention relates to a direct torque fault-tolerant control method for one-phase short-circuit faults of a permanent magnet motor, in particular to a fault-tolerant direct torque control method for a five-phase permanent magnet motor, which can be applied to aerospace, electric vehicles, etc., which is effective for motor reliability and dynamic performance. Strictly required occasions.
  • Phase open-circuit fault proposes a fault-tolerant direct torque method.
  • the open-circuit fault-tolerant method proposes a fault-tolerant algorithm for one-phase open-circuit faults, which is not suitable for short-circuit fault tolerance, and the method uses hysteresis comparison control after one-phase open-circuit fault of the motor , There are problems inherent in traditional direct torque control. Up to now, there is no literature to propose a fault-tolerant control method based on direct torque for one-phase short-circuit faults of five-phase permanent magnet motors.
  • the five-phase permanent magnet motor can ultimately achieve high reliability, high stability and dynamic performance under one-phase short-circuit faults.
  • the control object of the present invention is a five-phase permanent magnet motor.
  • the five phases of the motor are respectively defined as A, B, C, D, and E phases.
  • the fault-tolerant direct torque control method in the case of one-phase short circuit includes the following steps:
  • Step 1 Establish a mathematical model of the stator flux linkage during normal operation of the motor
  • Step 2 When the motor has a short-circuit fault of phase A, the one-phase short-circuit fault-tolerant control strategy can perform short-circuit fault-tolerant control based on the one-phase open-circuit fault-tolerant. Therefore, start with the open-circuit fault-tolerant control.
  • a phase change caused by the open portion is defined as the open flux disturbance flux ⁇ 'A.
  • the non-fault-tolerant open-circuit fault-tolerant current of the motor after the A-phase open-circuit fault is calculated (i' B , i' C, i 'D, i' E).
  • the fault-tolerant transformation matrix T 4/2 and the corresponding inverse transformation matrix for transforming the variables in the natural coordinate system into the ⁇ - ⁇ coordinate system in the case of an open-circuit fault are derived T 2/4 ;
  • Step 3 when a short-circuit fault occurs in phase A, the part of the flux linkage change caused by the short-circuit current i sc of phase A is defined as short-circuit flux linkage disturbance ⁇ ′′ A.
  • the residual non-fault phase compensation flux linkage ( ⁇ compB , ⁇ compC , ⁇ compD , ⁇ compE ) and the compensation current (i comp , i compC , i compD , i compE );
  • Step 4 On the basis of Step 3, use a current sensor to collect the short-circuit fault-tolerant currents (i” B , i " C , i” D , i " E ) of the remaining non-faulty phases after the short-circuit of phase A.
  • Use transformation matrix T 4/2 to transform the open-circuit fault-tolerant current obtained by subtracting the compensation current from the short-circuit fault-tolerant current of the remaining non-fault phase into the current components (i ⁇ , i ⁇ , i y ) in the ⁇ - ⁇ and xy coordinate system,
  • the PI controller is used to control the current i y to zero, and the output of the PI controller is the three-dimensional target voltage
  • Step 5 the remaining non-faulty phase short-circuit fault-tolerant flux linkages ( ⁇ " B , ⁇ " C , ⁇ " D , ⁇ ” E ) subtract the compensation flux linkages ( ⁇ compB , ⁇ compC , ⁇ compD derived from step 3) , ⁇ compE), open-circuit fault-tolerant flux ( ⁇ 'B, ⁇ ' C , ⁇ 'D, ⁇ ' E), T 4/2 converted by the component of flux to flux ⁇ - ⁇ coordinate system ( ⁇ ⁇ , ⁇ ⁇ );
  • Step 5 the T 4/2 of the remaining non-fault-tolerant-phase short circuit fault flux ( ⁇ "B, ⁇ " C , ⁇ "D, ⁇ ” E) conversion components of the short-circuit fault-tolerant flux ⁇ - ⁇ coordinate system ( ⁇ ′′ ⁇ , ⁇ ′′ ⁇ ), and transform the compensation flux linkage of the remaining non-fault phase ( ⁇ compB , ⁇ compC , ⁇ compD , ⁇ compE ) to the compensation flux component ( ⁇ comp ⁇ , ⁇ compE ) in the ⁇ - ⁇ coordinate system comp ⁇ ), and then subtract the compensation flux component ( ⁇ comp ⁇ , ⁇ comp ⁇ ) in the ⁇ - ⁇ coordinate system from the short-circuit fault-tolerant flux component ( ⁇ ′′ ⁇ , ⁇ ′′ ⁇ ) in the ⁇ - ⁇ coordinate system to obtain ⁇ - ⁇
  • Step 6 use the flux observer and torque observer to estimate the motor stator flux amplitude ⁇ s , flux angle ⁇ s and torque T e ;
  • Step 7 torque reference T e and the estimated value for the difference, the difference angle ⁇ is obtained by delta torque PI regulator, and thus the observations stator flux amplitude ⁇ s, given flux value Stator flux angle ⁇ s and torque increment angle ⁇ derive the difference between the given value of flux linkage and the observed stator flux linkage on the ⁇ axis and ⁇ axis ( ⁇ ⁇ , ⁇ ⁇ );
  • Step 8 On the basis of step 7, construct a voltage vector predictor in the ⁇ - ⁇ coordinate system based on the stator voltage equation to predict the components of the given voltage on the ⁇ axis and the ⁇ axis Use T 2/4 to obtain the stator target voltage component Transform to the natural coordinate system to obtain the motor phase voltage command At the same time
  • Step 9 Pass the phase voltage command obtained in step 8 through the voltage source inverter, and adopt the CPWM method to realize the disturbance-free operation of the fault-tolerant direct torque control of the five-phase permanent magnet motor after one-phase short-circuit fault.
  • the fault-tolerant direct torque control method of the present invention can not only realize reliable operation of a permanent magnet motor system containing only permanent magnet torque, but also realize reliable operation of a permanent magnet motor system containing permanent magnet torque and reluctance torque.
  • the fault-tolerant direct torque control strategy extends the fault-tolerant control to the permanent magnet motor system.
  • this fault-tolerant direct torque control strategy can realize the direct torque control based on the maximum torque-current ratio in the case of permanent magnet motor open circuit or short circuit fault, and then make full use of the reluctance torque and effectively improve the permanent magnet motor. Output torque in case of magneto failure.
  • the fault-tolerant direct torque control method of the present invention is different from the traditional fault-tolerant direct torque control.
  • the traditional fault-tolerant direct torque control uses a hysteresis comparator to select the target voltage vector in the switch table.
  • the hysteresis comparator has a voltage discrimination error, resulting in larger thrust pulsation; at the same time, because the switch table query and sector discrimination involve the division of sectors, the calculation of trigonometric functions and irrational functions, the complexity of the program is greatly increased; and the present invention
  • the fault-tolerant direct torque control method adopts the voltage vector prediction method and the pulse width modulation CPWM method based on zero sequence voltage signal injection.
  • the same effect as the space vector pulse width modulation SVPWM can be obtained without identifying sectors and calculations, saving the controller CPU memory Resources, effectively reduce the calculation time of the CPU, at the same time greatly suppress the torque ripple, and improve the torque control accuracy.
  • the non-fault-tolerant error current in the present invention is optimized based on the same stator flux linkage vector before and after the failure, the sum of the non-fault phase currents is zero, and the copper loss minimum principle. Ensure that the same flux vector before and after the fault can achieve the same stator flux trajectory circle, current trajectory circle and the same magnetomotive force before and after the fault.
  • the traditional method generally takes the equal magnetomotive force before and after the fault as the primary condition to optimize the current, and cannot guarantee that the stator flux trajectory obtained on this basis is circular; therefore, this method is more concise and facilitates direct torque control in the motor short circuit Implemented under failure.
  • the fault-tolerant transformation matrix and its inverse matrix used to convert the remaining four-phase variables in the natural coordinate system to the two-phase static coordinate system in the present invention are based on the same stator flux vector before and after the fault, the principle of minimum copper loss, and the non-faulty phase.
  • the sum of current is deduced by the principle of zero, and it is combined with the short-circuit fault-tolerant direct torque strategy to realize the current component and flux component of the motor in the ⁇ - ⁇ coordinate system after the motor one-phase short-circuit fault tolerance.
  • the phase difference is 90 degrees, that is, the current and flux trajectories of the motor in the ⁇ - ⁇ coordinate system before and after the fault are all circles of the same size.
  • the one-phase short-circuit direct torque fault-tolerant control method of the present invention is based on the one-phase open-circuit fault tolerance to derive the residual non-fault phase compensation flux linkage used to suppress the flux linkage disturbance caused by the short-circuit current. Therefore, the method is not only It can realize one-phase short-circuit fault tolerance, while also achieving one-phase open-circuit fault tolerance, and has good dynamic fault tolerance for open-circuit and short-circuit fault tolerance.
  • the one-phase short-circuit direct torque fault-tolerant control method of the present invention derives the compensation flux linkage of the remaining non-faulty phase based on the principle of minimum copper loss, so that it is related to the flux linkage disturbance vector caused by the short-circuit current Combined to zero, not only can eliminate the flux disturbance and torque fluctuation caused by the short-circuit current, but also ensure the same flux circle and current circle before and after the fault.
  • the short-circuit current is a dynamic value that changes with speed
  • the present invention does not need to refine its value. It is only used as a variable, which can improve the robustness and dynamic fault tolerance of one-phase short-circuit fault tolerance of the system, while taking into account High precision and simplicity of short-circuit fault-tolerant control.
  • the fault-tolerant transformation matrix in the present invention realizes the same flux linkage circle and current circle trajectory before and after the fault, which creates the preconditions for direct torque control in the motor fault state; on the other hand, it controls the three-dimensional space current to zero, reducing
  • the copper loss and iron loss of the motor are not only improved, but also the torque ripple caused by the three-dimensional space current is suppressed.
  • the invention combines the fault-tolerant transformation matrix and its inverse matrix, the stator flux linkage vector unchanged before and after the fault, the pulse width modulation CPWM technology based on zero sequence voltage injection, the stator flux observer, the torque observer, and the back-EMF integration method, Not only realizes the disturbance-free operation under the direct torque control after one-phase short-circuit fault, but also improves the utilization rate of the inverter bus voltage, while avoiding the complexity of the traditional SVPWM algorithm; in addition, the CPWM technology adopted by the present invention, this method Concise and clear, highlighting the simple and effective characteristics of direct torque control; in addition, for a type of five-phase permanent magnet motor, the direct torque control method proposed in the present invention can improve the torque control of the motor under short-circuit fault conditions. Accuracy, torque following performance, torque dynamic performance and steady-state performance make the dynamic and steady-state performance of the motor after a fault similar to that before the fault.
  • Fig. 1 is a schematic structural diagram of a five-phase permanent magnet motor according to an embodiment of the present invention
  • FIG. 2 is a schematic diagram of a direct torque control strategy of a five-phase permanent magnet motor according to an embodiment of the present invention
  • Figure 3 is a schematic diagram of a fault-tolerant direct torque control strategy for a five-phase permanent magnet motor A phase short circuit fault according to an embodiment of the present invention
  • Fig. 4 is a phase current waveform of the non-fault-tolerant direct torque control operation under the condition of a phase A from normal to a short-circuit fault according to the embodiment of the present invention
  • Fig. 5 is the torque waveform of the non-fault-tolerant direct torque control operation under the condition of phase A from normal to short-circuit fault according to the embodiment of the present invention
  • Figure 6 is the stator flux trajectory on ⁇ - ⁇ when the non-fault-tolerant direct torque control is running under the condition of phase A from normal to short-circuit fault according to the embodiment of the present invention
  • Fig. 7 is the stator current trajectory on ⁇ - ⁇ when the non-fault-tolerant direct torque control is running from normal to short-circuit fault in the embodiment of the present invention
  • Fig. 8 is a phase current waveform during fault-tolerant direct torque control operation under the condition of phase A from normal to short-circuit fault according to the embodiment of the present invention
  • Fig. 9 is the torque waveform when the fault-tolerant direct torque control is running under the condition of phase A from normal to short-circuit fault according to the embodiment of the present invention.
  • Fig. 10 is the stator flux trajectory on ⁇ - ⁇ when the fault-tolerant direct torque control is running under the condition of phase A from normal to short-circuit fault according to the embodiment of the present invention
  • Fig. 11 is the stator current trajectory on ⁇ - ⁇ during the operation of fault-tolerant direct torque control under the condition of phase A from normal to short-circuit fault according to the embodiment of the present invention
  • FIG. 12 is the output torque waveform of the motor when the torque command is stepped up during the normal operation of the embodiment of the present invention.
  • Fig. 13 is the output torque waveform of the motor when the torque command is stepped up during the operation of the fault-tolerant direct torque control under the A-phase short circuit fault according to the embodiment of the present invention.
  • the schematic diagram of the structure of the five-phase permanent magnet motor of the embodiment of the present invention adopts a 20-slot/22-pole outer rotor structure and a single-layer concentrated winding method; the addition of fault-tolerant teeth makes the magnetic and The thermal coupling is almost zero.
  • the addition of fault-tolerant teeth reduces the coupling between the non-faulty phase and the faulty phase of the motor under fault.
  • the five-phase permanent magnet motor is powered by a voltage source inverter. It is assumed that the five phases of the motor are A, B, C, D, and E.
  • the first part flux disturbance suppression after phase A open circuit
  • Step 1 Establish a mathematical model of the stator flux linkage of the five-phase permanent magnet motor during normal operation.
  • the five-phase permanent magnet motor adopts the direct torque control strategy shown in Figure 2, and uses the transformation matrix of formula (1) to transform the variables in the five-phase natural coordinate system to the ⁇ - ⁇ coordinate system
  • phase currents of the five phases A, B, C, D, and E can be expressed as
  • i A , i B , i C , i D , and i E are the phase currents of phase A, B, C, D, and E respectively, and i ⁇ , i ⁇ are the stator currents on the ⁇ axis and ⁇ axis, respectively Weight.
  • stator flux linkage ( ⁇ A , ⁇ B , ⁇ C , ⁇ D , ⁇ E ) during normal operation of the motor can be expressed as
  • L( ⁇ ) is the inductance matrix of the motor
  • ⁇ is the electrical angle
  • L m 0.2(L d +L q )
  • L ⁇ 0.2(Lq-L d )
  • L d and L q are the motor d, respectively axis and q-axis inductance
  • L E is a, B, C, D, E phase stator inductance
  • L ls motor leakage inductance I 5 ⁇ 5 fifth-order unit Matrix
  • ⁇ m is the amplitude of the permanent magnet flux of the motor.
  • stator flux linkage vector of the motor is
  • the influence of a phase short-circuit fault on the motor system can be regarded as the sum of the influence of the phase open-circuit fault on the motor system and the influence of the phase short-circuit current on the system. Therefore, the present invention first proposes a fault-tolerant direct torque control strategy for open-circuit faults.
  • Step 2 when the A-phase open fault occurs, causing the open phase change portion A is defined as the open flux disturbance flux ⁇ 'A;
  • the stator flux vector is unchanged before and after the fault, the non-fault phase current is zero, and copper the principle of minimum consumption is obtained under the non-fault tolerant operation of the phase open fault tolerant motor current i 'B, i' C, i 'D, i'E; on this basis, the deduced open fault condition the natural coordinates
  • the variables in are transformed to the fault-tolerant transformation matrix T 4/2 and the corresponding inverse matrix T 2/4 in the ⁇ - ⁇ coordinate system. Specifically: when the A phase open circuit fault occurs, the five-phase winding flux is respectively
  • ⁇ 'A [L AB L AC L AD L AE] [i' B i 'C i' D i 'E] T + ⁇ m cos ⁇ (8)
  • ⁇ 'flux DISTURBANCES A is A-phase open-circuit caused
  • ⁇ ' B, ⁇ 'C , ⁇ ' D, ⁇ 'E are compatible dislocation flux remaining non-defective after the A-phase open-circuit
  • i' B , i 'C, i' D , i 'E are compatible with the remaining non-defective after the a-phase open fault current, x 2, y 2, x 3, y 3, x 4, y 4, x 5, y 5 , respectively for the remaining non-fault current coefficient phases B, C, D, E phase inductance matrix L ( ⁇ ) 'is L ( ⁇ ) removing the first row and first column of the new matrix obtained inductance
  • stator flux linkage vector obtained by vector synthesis of the motor stator flux after the A-phase open circuit fault according to the winding space position is
  • Equation (15) can be further expressed as
  • injecting the open-circuit fault-tolerant current of equation (16) can effectively suppress the flux disturbance caused by the open-circuit of phase A, and keep the stator flux vector, winding magnetomotive force and current trajectory unchanged before and after the fault. round.
  • the effective dimension of the motor is reduced from five dimensions to four dimensions, the original transformation matrix T 5/2 is no longer suitable, and the transformation matrix T 5/2 needs to be reduced in order.
  • the system degrees of freedom become three, two degrees of freedom are in the ⁇ - ⁇ plane, and the other degree of freedom is in the xy plane, so the first row and last row of the transformation matrix T 5/2 are deleted, then the transformation The matrix is rewritten as T′ 4/2
  • the current of this phase is 0, and the current i x on the xy plane is related to the current i ⁇ on the ⁇ - ⁇ plane. Therefore, i x does not need to be controlled, only i y needs to be controlled.
  • the third row of the matrix T′ 4/2 is deleted.
  • the third column of the matrix T 1 used to transform the variables in the ⁇ - ⁇ coordinate system to the natural coordinate system is obtained, then the fault-tolerant transformation matrix T 4/2 and its inverse transformation in the case of open-circuit fault tolerance
  • the matrix T 2/4 can be expressed as
  • phase A when an open-circuit fault occurs in phase A, using the above-mentioned fault-tolerant transformation matrix can not only effectively suppress the flux linkage disturbance caused by the open-circuit of phase A, but also achieve the same flux linkage and current trajectory circle as under normal conditions, thereby realizing phase open-circuit failure Disturbance-free operation of direct torque control under the circumstances.
  • Step 3 When a short-circuit fault occurs in phase A, the part of the flux linkage change caused by the short-circuit current i sc of phase A is defined as the short-circuit flux linkage disturbance ⁇ ′′ A ; the remaining non-fault phase compensation flux linkage ( ⁇ compB , ⁇ compC , ⁇ compD , ⁇ compE ), according to the principle of constant stator flux vector before and after the fault, the principle of minimum copper loss or equal amplitude of the remaining non-fault phase compensation current, and the sum of the remaining non-fault phase compensation current is zero Derive the compensation flux linkage and compensation current (i compB , i compC , i compD , i compE ). The specifics are:
  • stator flux linkage shown in equation (27) is vectorized according to the winding space position to obtain the stator flux linkage vector after phase A short-circuit
  • the stator flux vector In order to offset the flux disturbance caused by the short-circuit current, the stator flux vector must be kept unchanged before and after the short-circuit fault.
  • the compensation current of the remaining non-faulty phase after the short-circuit of phase A is defined, which is related to the open-circuit fault-tolerant current of the remaining non-faulty phase and the short-circuit fault-tolerant current (i " B , i" C , I′′ D , i′′ E ) The relationship is
  • i compB , i compC , i compD , i compE are the compensation currents of the remaining non-faulty phases
  • x b , y b , x c , y c , x d , y d , x e , and y e are the remaining The coefficient of the compensation current of the non-faulty phases B, C, D, and E.
  • the compensating flux linkage of the remaining non-fault phase ( ⁇ compB , ⁇ compC , ⁇ compD , ⁇ compE ) is used to offset the flux disturbance caused by the short-circuit current, which is derived from equations (28) and (31)
  • the motor stator flux linkage vector can be further expressed as
  • equation (34) is consistent with equation (11) in the case of open-circuit fault tolerance. Therefore, as long as the flux linkage is compensated to eliminate the flux disturbance caused by the short-circuit current, the stator flux vector in the case of short-circuit fault tolerance is the same as in the case of open-circuit fault.
  • the previous derivation shows that the stator flux linkage under the open circuit condition is equal to the normal condition, so the stator flux linkage vector under the short-circuit fault tolerance condition is also equal to the normal condition.
  • the constant stator flux trajectory circle, current trajectory circle and constant stator winding magnetomotive force are ensured before and after the A-phase short-circuit fault.
  • the relationship between the compensation current of the remaining non-fault phase and the short-circuit current of phase A is obtained. According to the relationship between the phase A short-circuit current and the back EMF, the expression of the phase A short-circuit current can be obtained.
  • I f is the magnitude of the short-circuit current
  • ⁇ sc is the angle between the opposite potential of A and the axis of phase A.
  • the use of the compensation current of formula (38) can effectively eliminate the flux disturbance caused by the short-circuit current and keep the stator flux vector unchanged before and after the fault.
  • Step 4 On the basis of step 3, collect the short-circuit fault-tolerant currents i′′ B , i′′ C , i′′ D , i′′ A of the remaining non-faulty phases after the short-circuit fault of phase A.
  • the open-circuit fault-tolerant current obtained by subtracting the compensation current from the short-circuit fault-tolerant current by the transformation matrix T 4/2 is transformed into the current components in the ⁇ - ⁇ and xy coordinate systems as i ⁇ , i ⁇ , i y , and the controller is used to control The current i y is zero, and the output of the controller is the target voltage in the three-dimensional space
  • the specific process is: Combining equations (16), (29) and (38), using T 4/2 to short-circuit the fault-tolerant currents of the remaining non-fault phases in the natural coordinate system (i′′ B , i′′ C , i′′ D , i′′ E )
  • the current obtained after subtracting the compensation current i compB , i comp
  • i′′ ⁇ , i′′ ⁇ , i′′ y are the short-circuit fault-tolerant current components in the ⁇ - ⁇ and xy coordinate systems after the A phase is short-circuited.
  • Step 5 the remaining non-faulty phase short-circuit fault-tolerant flux ⁇ ” B , ⁇ ′′ C , ⁇ ” D , ⁇ ” E subtract the non-faulty phase compensation flux ⁇ compB , ⁇ compC , ⁇ compD , ⁇ compE , to obtain the open-circuit fault tolerance flux is ⁇ 'B, ⁇ ' C, ⁇ 'D, ⁇ ' E, T 4/2 converted by the component of flux to flux ⁇ - ⁇ coordinate system for ⁇ ⁇ and ⁇ ⁇ ; or step 5 , Transform the remaining non-fault phase short-circuit fault-tolerant flux ⁇ ” B , ⁇ ” C , ⁇ ” D , ⁇ ” E by T 4/2 to the short-circuit fault-tolerant flux component in the ⁇ - ⁇ coordinate system as ⁇ ” ⁇ , ⁇ " ⁇ , and transform the non-fault phase compensation flux ⁇ compB , ⁇ compC , ⁇ compD , ⁇ compE to the compensation flux component in the ⁇
  • the compensation flux of the remaining non-faulty phase eliminates the flux disturbance caused by the short-circuit current of phase A, so that the stator flux vector under the short-circuit fault tolerance condition is the same as that under normal conditions; at the same time, the remaining non-faulty phases are ensured
  • the composite magnetomotive force of the compensation current and the short-circuit current is zero; combined with the fault-tolerant transformation matrix and its inverse matrix, the flux trajectory and the current trajectory in the ⁇ - ⁇ coordinate system before and after the fault are circular, and finally there is no disturbance under the short-circuit fault Fault-tolerant operation.
  • the third part is to realize the disturbance-free operation of one-phase short-circuit fault-tolerant direct torque control
  • Step 6 use the flux linkage observer and torque observer to estimate the amplitude ⁇ s , flux linkage angle ⁇ s and torque T e of the stator flux linkage vector.
  • the specific process is: constructing a flux observer based on a voltage model or a current model.
  • the flux observer constructed according to formula (41) of the present invention observes the stator flux components ( ⁇ ⁇ , ⁇ ⁇ ) in the ⁇ - ⁇ coordinate system for
  • ⁇ s is the amplitude of the stator flux linkage
  • ⁇ s is the stator flux linkage angle
  • P is the number of motor pole pairs.
  • Step 7 torque reference T e and the estimated value for the difference, the difference is obtained by the controller incremental angle Delta] [delta torque, and further in accordance with an observation vector magnitude of the stator flux ⁇ s, given flux value
  • the stator flux angle ⁇ s and the torque increment angle ⁇ derive the given value of the flux linkage and the difference between the stator flux linkage on the ⁇ axis and the ⁇ axis as ⁇ ⁇ , ⁇ ⁇ ; thus, according to the stator voltage equation in ⁇ -Build a voltage vector predictor in the ⁇ coordinate system to predict the components of a given voltage on the ⁇ axis and ⁇ axis as Use T 2/4 to obtain the stator target voltage component Transformed to the natural coordinate system, the motor phase voltage command is At the same time
  • the specific process is: torque setpoint T e and the estimated value for the difference, the difference angle ⁇ is obtained by delta torque PI regulator, and thus the observations stator flux amplitude ⁇ s, given flux value Stator
  • the fault-tolerant transformation matrix T 4/2 is used to transform the remaining non-fault phase voltage equation in the natural coordinate system of equation (47) into the voltage component (u ⁇ , u ⁇ ) in the ⁇ - ⁇ coordinate system
  • T is the controller sampling period.
  • the PI controller is used to control i y to zero, and the output voltage of the PI controller is defined as
  • stator target voltage component Use T 2/4 to formula (49) stator target voltage component And the three-dimensional target voltage Transform to the natural coordinate system to obtain the motor phase voltage command for
  • Step 8 the phase voltage command obtained in step 7 is sent to the voltage source inverter, and the pulse width modulation CPWM method based on zero sequence voltage injection is used to realize the fault-tolerant direct torque control of the five-phase permanent magnet motor after one-phase short-circuit fault.
  • the SVPWM method has the same control effect, so the present invention intends to adopt the CPWM modulation technique based on injecting zero sequence voltage harmonics.
  • phase voltage command of formula (50) Via the voltage source inverter, the pulse width modulation CPWM technology based on zero sequence voltage injection is adopted to realize the fault-tolerant direct torque control of the five-phase permanent magnet motor after a short-circuit fault of one phase.
  • the one-phase short-circuit fault-tolerant direct torque control strategy proposed by the present invention is shown in Figure 3.
  • Figure 4-7 shows the phase current waveform, torque waveform, stator flux linkage and current trajectory on ⁇ - ⁇ when the motor is running without fault tolerance under the condition of phase A from normal to short-circuit fault.
  • a phase short-circuit fault occurs at 0.15s. It can be seen that the current waveform is distorted; the motor torque fluctuates significantly; although the flux amplitude difference is small, the waveform of the stator flux component in the ⁇ - ⁇ coordinate system is still distorted; the current trace fluctuates greatly and is no longer constant Circular trajectory.
  • Figure 8-11 shows the phase current waveform, torque waveform, stator flux linkage and current trajectory in the ⁇ - ⁇ coordinate system during fault-tolerant operation of the motor under the condition of phase A from normal to short-circuit fault.
  • the one-phase short-circuit fault-tolerant direct torque control strategy of the present invention for a five-phase permanent magnet motor can not only ensure the motor output torque and normal state when the motor drive system allows the maximum current It can obviously suppress the torque fluctuation after a short-circuit fault of the motor. More importantly, it has the same dynamic performance, stability performance and torque following accuracy as before the fault, and it is suitable for any phase short-circuit fault. In this case, the versatility is strong, no complicated calculation is required, and the CPU overhead is small. Compared with the fault-tolerant vector control strategy, this invention has the characteristics of simple structure, low CPU overhead, and faster dynamic response speed, which makes it very good in electric vehicles and other systems that require high motor reliability and dynamic performance. Application prospects.

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Abstract

本发明公开了一种五相永磁电机一相短路容错直接转矩控制方法。当A相发生开路故障时，根据故障前后定子磁链矢量不变、非故障相电流之和为零以及铜耗最小的约束条件，推导出电机容错运行的非故障相开路容错电流、容错变换矩阵T 4/2及其逆矩阵T 2/4；当A相发生短路故障时，根据故障前后定子磁链矢量不变、铜耗最小原理或非故障相补偿电流幅值相等、剩余非故障相补偿电流和为零的约束条件，推导出剩余非故障相补偿磁链；结合定子磁链与转矩观测器、给定转矩与定子磁链幅值、坐标变换求出定子相电压指令，该电压结合基于零序电压注入的脉宽调制技术实现电机短路故障前后一致的稳态性能、动态性能，且逆变器开关频率恒定、CPU开销小。

Description

一种五相永磁电机一相短路容错直接转矩控制方法 技术领域
本发明涉及一种永磁电机一相短路故障直接转矩容错控制方法，特别是五相永磁电机容错直接转矩控制方法，能够应用在航空航天、电动汽车等对电机可靠性和动态性能有严格要求的场合。
背景技术
随着社会发展水平的提升，交通工具也在发生着巨大的变革，比如，电动汽车的出现，能够大幅度减少能源消耗以及降低环境污染，与此同时，电动汽车技术标准越来越严谨，电机作为电动车上最重要的零部件之一，其故障状态下的稳态性能、动态性能直接决定着电动车的可靠性，因此有必要保证电机在故障下的容错性能，使其能够拥有正常运行状况下的稳态和动态性能。
在电机发生一相短路故障时，虽然此时电机仍然能输出一定大小的转矩，但是转矩存在着很大的波动，而且故障后的电机运行噪声、损耗都变大，使得电机性能以及使用寿命下降，甚至相短路故障能造成电机驱动系统永久损坏。而当电机发生短路故障时，容错算法的加入能够使电机基本达到故障前的稳态和动态性能，使得故障后的电机输出平滑转矩。中国发明专利申请号为201610540823.2的专利《一种新型的五相永磁电机的短路容错控制方法》针对五相容错表贴式永磁旋转电机提出一种短路容错控制方法。当电机的某一相发生短路故障时，短路电流会产生一个频率和相位都可以确定的转矩脉动。该方法通过改变其他正常相电流的幅值和相位，使所有正常相都能产生一个和短路相相同频率和相位的转矩脉动，且使各相所产生的转矩脉动矢量和为零，同时还必须满足最终合成的平均转矩与正常情况下相等。该方法用于抑制短路相电流导致转矩波动的剩余非故障相补偿电流的幅值是动态变化的，与电机运行工况有关。文献IEEE Transactions on Power Electronics,33(3)：2774-2784，2018“An Experimental Assessment of Open-Phase Fault-Tolerant Virtual-Vector-Based Direct Torque Control in Five-Phase Induction Motor Drives”针对五相感应电机一相开路故障提出了一种容错直接转矩方法，该开路容错方法对一相开路故障提出的容错算法，并不适用于短路容错，而且该方法在电机一相开路故障后，采用滞环比较控制，存在传统直接转矩控制固有的问题。截至目前为止，还没有文献针对五相永磁电机一相短路故障，提出基于直接转矩的容错控制方法。
发明内容
针对现有五相永磁电机直接转矩容错控制技术存在的不足，以及本发明控制对象五相永磁电机的特点和该类电机短路故障特点，在五相永磁电机发生一相短路故障后，没有适用于五相永磁电机的直接转矩控制策略，提出一种用于五相永磁电机的一相短路故障情况下的容错直接转矩控制方法，不仅能够实现该类电机系统在一相短路故障状态下的高容错性能、高动态性能、优良的转矩跟随性能，同时可以减小控制算法的复杂度以及控制器CPU的开销，而且还可以实现逆变器开关频率恒定、电机铜耗最小、电机噪声降低以及简化电磁兼容设计，最终能够实现五相永磁电机一相短路故障下的高可靠性、高稳态和动态性能。
本发明用于五相永磁电机的容错直接转矩控制方法采用如下技术方案：
本发明的控制对象为五相永磁电机，电机的五相分别定义为A、B、C、D、E相，其中一相短路情况下的容错直接转矩控制方法，包括以下步骤：
步骤1，建立电机正常运行时的定子磁链数学模型；
步骤2，当电机发生A相短路故障时，一相短路容错控制策略可以在一相开路容错的基础上进行短路容错控制，因此，先从开路容错控制开始。当A相发生开路故障时，A相开路引起磁链变化部分定义为开路磁链扰动ψ' A。根据电机开路故障前后相同的定子磁链矢量、非故障相电流之和为零、铜耗最小原则，求出A相开路故障后电机容错运行的非故障相开路容错电流(i' B、i' C、i' D、i' E)。在此基础上，根据直接转矩的控制要求，推导出用于将开路故障情况下自然坐标系中的变量变换到α-β坐标系中的容错变换矩阵T 4/2以及对应的逆变换矩阵T 2/4；
步骤3，进一步，当A相发生短路故障时，A相短路电流i sc引起磁链变化部分定义为短路磁链扰动ψ″ A。根据电机短路故障前后定子磁链矢量不变、铜耗最小原理或补偿电流幅值相等原理、以及剩余非故障相补偿电流和为零的约束条件，推导剩余非故障相补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)以及补偿电流(i comp、i compC、i compD、i compE)；
步骤4，在步骤3的基础上，使用电流传感器采集A相短路后剩余非故障相的短路容错电流(i″ B、i″ C、i″ D、i″ E)。采用变换矩阵T 4/2将剩余非故障相的短路容错电流减去补偿电流后得到的开路容错电流变换到α-β和x-y坐标系中的电流分量(i α、i β、i y)，同 时采用PI控制器控制电流i y为零，该PI控制器的输出为三维空间目标电压
Figure PCTCN2019075503-appb-000001
步骤5，进一步，剩余非故障相短路容错磁链(ψ″ B、ψ″ C、ψ″ D、ψ″ E)减去由步骤3推导出的补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)，得到开路容错磁链(ψ' B、ψ' C、ψ' D、ψ' E)，由T 4/2将该磁链变换到α-β坐标系中的磁链分量(ψ α、ψ β)；
或步骤5，由T 4/2将剩余非故障相短路容错磁链(ψ″ B、ψ″ C、ψ″ D、ψ″ E)变换到α-β坐标系中的短路容错磁链分量(ψ″ α、ψ″ β)，以及将剩余非故障相的补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)变换到α-β坐标系中的补偿磁链分量(ψ compα、ψ compβ)，再将α-β坐标系中的短路容错磁链分量(ψ″ α、ψ″ β)减去α-β坐标系中的补偿磁链分量(ψ compα、ψ compβ)得到α-β坐标系中的磁链分量(ψ α、ψ β)；
步骤6，根据步骤4和5，采用磁链观测器和转矩观测器估算出电机定子磁链幅值ψ s、磁链角θ s和转矩T e；
步骤7，转矩给定值
Figure PCTCN2019075503-appb-000002
和估算值T e作差，该差值经PI调节器获得转矩增量角Δδ，进而根据观测的定子磁链幅值ψ s、磁链给定值
Figure PCTCN2019075503-appb-000003
定子磁链角θ s以及转矩增量角Δδ推导出磁链给定值和观测的定子磁链在α轴和β轴上的差值(Δψ α、Δψ β)；
步骤8，在步骤7的基础上，根据定子电压方程，在α-β坐标系中构建电压矢量预测器，预测给定电压在α轴和β轴上的分量
Figure PCTCN2019075503-appb-000004
采用T 2/4将求出的定子目标电压分量
Figure PCTCN2019075503-appb-000005
变换到自然坐标系上，得到电机相电压指令
Figure PCTCN2019075503-appb-000006
同时令
Figure PCTCN2019075503-appb-000007
步骤9，将步骤8所得的相电压指令经电压源逆变器，采用CPWM方式实现五相永磁电机一相短路故障后的容错直接转矩控制的无扰运行。
本发明具有以下有益效果：
1、本发明的容错直接转矩控制方法不仅能实现仅含有永磁转矩的永磁电机系统可靠运行，而且能实现包含永磁转矩和磁阻转矩的永磁电机系统的可靠运行。也就是说，该容错直接转矩控制策略将容错控制拓展到了永磁电机系统。配合最大转矩电流比算法， 该容错直接转矩控制策略能在永磁电机开路或短路故障情况下实现基于最大转矩电流比的直接转矩控制，进而充分利用磁阻转矩，有效提高永磁电机故障情况下的输出转矩。
2、本发明的容错直接转矩控制方法不同于传统的容错直接转矩控制，传统的容错直接转矩控制采用的是通过滞环比较器来选择开关表中的目标电压矢量。滞环比较器存在电压判别误差，导致较大的推力脉动；同时由于开关表查询和扇区判别涉及扇区的划分、三角函数和无理函数的计算，大大增加程序的复杂性；而本发明的容错直接转矩控制方法采用电压矢量预测方法和基于零序电压信号注入的脉宽调制CPWM方法，无需判别扇区和计算就能获得空间矢量脉宽调制SVPWM相同的效果，节省了控制器CPU内存资源，有效减小了CPU的计算时间，同时大大抑制了转矩脉动，提高了转矩控制精度。
3、本发明中非故障相容错电流是根据故障前后相同的定子磁链矢量、非故障相电流之和为零以及铜耗最小原则优化得到的。保证故障前后相同的磁链矢量就能实现故障前后相同的定子磁链轨迹圆、电流轨迹圆以及相同的磁动势。而传统方法一般将故障前后磁动势相等作为首要条件对电流进行优化，无法保证在此基础上获得的定子磁链轨迹是圆形；因此，该方法更加简洁，便于直接转矩控制在电机短路故障下实施。
4、本发明中用于将自然坐标系中剩余四相变量转换到两相静止坐标系的容错变换矩阵及其逆矩阵是基于故障前后相同的定子磁链矢量、铜耗最小原理以及非故障相电流之和为零原则推导出的，其和短路容错直接转矩策略相结合实现了电机一相短路故障容错后电机在α-β坐标系中的电流分量、磁链分量的幅值相等且相位相差90度，也就是故障前后保持电机在α-β坐标系中的电流和磁链轨迹均为同样大小的圆形。另无需再像传统容错直接转矩控制那样设置专门的电压矢量去抑制该三维空间的电流，仅需采用简单的PI控制就能消除该电流，改善了电流的正弦度，降低了电机铜耗、铁耗以及控制算法复杂度。
5、本发明的一相短路直接转矩容错控制方法，是在基于一相开路容错的基础上推导出用于抑制短路电流导致的磁链扰动的剩余非故障相补偿磁链，因此该方法不仅能够实现一相短路容错，同时还能兼顾实现一相开路容错，对于开路、短路容错具有很好的动态容错能力。
6、本发明的一相短路直接转矩容错控制方法，在开路容错的基础上，基于铜耗最小原理推导出剩余非故障相的补偿磁链，从而使得其与短路电流引起的磁链扰动矢量合成为零，不仅能够消除短路电流引起的磁链扰动、转矩波动，同时还保证了故障前后相同的磁链圆和电流圆。
7、由于短路电流是一个随着速度变化的动态值，本发明无需对其进行细化取值，仅作为一个变量，能够提高系统一相短路容错的鲁棒性、动态容错性能，同时兼顾了短路容错控制的高精度、简单性。
8、本发明中的容错变换矩阵实现了故障前后相同的磁链圆和电流圆轨迹，这些为电机故障状态下的直接转矩控制创造了前提条件；另一方面控制三维空间电流为零，减少了电机铜耗和铁耗，不仅提高了电机效率，同时抑制了三维空间电流引起的转矩脉动。本发明将容错变换矩阵及其逆矩阵、故障前后不变的定子磁链矢量、基于零序电压注入的脉宽调制CPWM技术、定子磁链观测器、转矩观测器、反电势积分法相结合，不但实现了一相短路故障后的直接转矩控制下的无扰运行，而且提高了逆变器母线电压利用率，同时避免了传统SVPWM算法的复杂性；另外本发明采用的CPWM技术，该方法简洁明了，突显了直接转矩控制的简单有效的特点；除此之外，对于一类五相永磁电机，本发明提出的直接转矩控制方法均能提高电机短路故障状态下的转矩控制精度、转矩跟随性能、转矩动态性能以及稳态性能，使得电机故障后的动态与稳态性能与故障前相似。
附图说明
图1为本发明实施例五相永磁电机的结构示意图；
图2为本发明实施例五相永磁电机直接转矩控制策略原理图；
图3为本发明实施例五相永磁电机A相短路故障下容错直接转矩控制策略原理图；
图4为本发明实施例A相从正常到短路故障情况下无容错直接转矩控制运行时的相电流波形；
图5为本发明实施例A相从正常到短路故障情况下无容错直接转矩控制运行时的转矩波形；
图6为本发明实施例A相从正常到短路故障情况下无容错直接转矩控制运行时的α-β上的定子磁链轨迹；
图7为本发明实施例A相从正常到短路故障情况下无容错直接转矩控制运行时的α-β上的定子电流轨迹；
图8为本发明实施例A相从正常到短路故障情况下容错直接转矩控制运行时的相电流波形；
图9为本发明实施例A相从正常到短路故障情况下容错直接转矩控制运行时的转矩波形；
图10为本发明实施例A相从正常到短路故障情况下容错直接转矩控制运行时的α-β上的 定子磁链轨迹；
图11为本发明实施例A相从正常到短路故障情况下容错直接转矩控制运行时的α-β上的定子电流轨迹；
图12为本发明实施例正常运行过程中转矩指令阶跃上升时的电机输出转矩波形；
图13为本发明实施例A相短路故障下容错直接转矩控制运行过程中转矩指令阶跃上升时的电机输出转矩波形。
图中：1.定子；2.转子；3.电枢齿；4.容错齿；5.线圈绕组；6.永磁体。
具体实施方式
为了能够更加简单明了地说明本发明的五相永磁电机容错直接转矩控制策略的特点和有益效果，下面结合一个具体的五相永磁电机来进行详细、完整地描述。
如图1所示，本发明实施例五相永磁电机的结构示意图，采用20槽/22极的外转子结构以及单层集中绕组方式；容错齿的加入，使各相绕组之间的磁、热的耦合几乎为零。当电机一相出现故障时，正常相不受故障相的影响，具有较强的容错性能。另外，容错齿的加入，降低了故障下电机非故障相与故障相的耦合。五相永磁电机由电压源逆变器供电，假设该电机五相分别为A、B、C、D、E五相。
第一部分，A相开路后的磁链扰动抑制
步骤1，建立五相永磁电机正常运行时的定子磁链数学模型。具体为：五相永磁电机采用图2所示直接转矩控制策略，采用式(1)变换矩阵将五相自然坐标系上的变量变换到α-β坐标系上
Figure PCTCN2019075503-appb-000008
式中，a＝2π/5。
当电机正常状态稳态运行时，假设电机三维空间电流i x和i y已经控制为零，则A、B、C、D、E五相的相电流可表示为
Figure PCTCN2019075503-appb-000009
式中，i A、i B、i C、i D、i E分别为A、B、C、D、E相的相电流，i α、i β分别是定子电流在α轴和β轴上的分量。
电机正常运行时的定子磁链(ψ A、ψ B、ψ C、ψ D、ψ E)可以表示为
[ψ A ψ B ψ C ψ D ψ E]＝L(θ)[i A i B i C i D i E] T+ψ f  (3)
Figure PCTCN2019075503-appb-000010
Figure PCTCN2019075503-appb-000011
Figure PCTCN2019075503-appb-000012
式中，L(θ)为电机的电感矩阵，θ为电角度，L m＝0.2(L d+L q)，L θ＝0.2(Lq-L d)，L d、L q分别为电机d轴和q轴电感，L A、L B、L C、L D、L E为A、B、C、D、E相的定子电感，L ls为电机漏感，I 5×5为五阶单位矩阵，ψ f为电机永磁体耦合到定子侧的永磁磁链，其可表示为ψ f＝ψ m[cosθ cos(θ-a)cos(θ-2a)cos(θ-3a)cos(θ-4a)] T，ψ m为电机永磁磁链幅值。
进一步，根据各相空间位置，电机定子磁链矢量为
Figure PCTCN2019075503-appb-000013
式中，ε＝e ja。
一相短路故障对电机系统的影响可以看成是该相开路故障对电机系统的影响与该相短路电流对系统的影响之和。因此本发明先提出开路故障的容错直接转矩控制策略。
步骤2，当A相发生开路故障时，A相开路引起磁链变化部分定义为开路磁链扰动ψ' A；根据故障前后定子磁链矢量不变、非故障相电流之和为零、以及铜耗最小原则，求出电机容错运行情况下的非故障相开路容错电流为i' B、i' C、i' D、i' E；在此基础上，推导出将开路故障情况下自然坐标系中的变量变换到α-β坐标系中的容错变换矩阵T 4/2以及对应的逆矩阵T 2/4。具体为：当A相开路故障发生后，五相绕组磁链分别为
ψ' A＝[L AB L AC L AD L AE][i' B i' C i' D i' E] T+ψ mcosθ   (8)
[ψ' B ψ' C ψ' D ψ' E] T＝L(θ)'[i' B i' C i' D i' E] T+ψ' f   (9)
Figure PCTCN2019075503-appb-000014
式中，ψ' A为A相开路后引起的磁链扰动，ψ' B、ψ' C、ψ' D、ψ' E分别为A相开路后的剩余非故障相容错磁链，i' B、i' C、i' D、i' E分别为A相开路后的剩余非故障相容错电流，x 2、y 2、x 3、y 3、x 4、y 4、x 5、y 5分别为剩余非故障相B、C、D、E相的电流系数，电感矩阵L(θ)'为L(θ)去掉第一行以及第一列得到的新电感矩阵，ψ' f为A相开路后电机永磁体耦合到定子侧的永磁磁链，其可表示为ψ' f＝ψ m[cos(θ-a)cos(θ-2a)cos(θ-3a)cos(θ-4a)] T。
进一步，将A相开路故障后的电机定子磁链按照绕组空间位置进行矢量合成得到的定子磁链矢量为
Figure PCTCN2019075503-appb-000015
根据式(7)和(11)，为抑制A相开路后引起的磁链扰动，只要确保开路情况下的定子磁链矢量与正常情况下的一致，即保持故障前后定子磁链矢量的正序与负序部分不变，就能保证故障前后不变的定子磁链轨迹圆和电流轨迹圆以及恒定的定子绕组磁动势。令式(7)和(11)相等，得到
Figure PCTCN2019075503-appb-000016
由于电机绕组采用星形连接且中心点与直流母线中点不连接，故剩余非故障相电流和为零
Figure PCTCN2019075503-appb-000017
设立目标函数
Figure PCTCN2019075503-appb-000018
根据铜耗最小原理，联立式(12)和(13)求解目标函数(14)的最小值，得到
Figure PCTCN2019075503-appb-000019
式(15)可以进一步表示为
Figure PCTCN2019075503-appb-000020
Figure PCTCN2019075503-appb-000021
因此，当A相发生开路故障时，注入式(16)的开路容错电流，能够有效抑制A相开路引起的磁链扰动，保持故障前后不变的定子磁链矢量、绕组磁动势和电流轨迹圆。
进一步，A相开路后，电机有效维数由五维降为四维，原有的变换矩阵T 5/2不再适合，需要对变换矩阵T 5/2进行降阶处理。当A相发生开路故障，系统自由度变为三个，两个自由度在α-β平面，另外一个自由度在x-y平面，因此变换矩阵T 5/2第一行和最后一行删除，则变换矩阵改写为T′ 4/2
Figure PCTCN2019075503-appb-000022
提取矩阵T′ 4/2的前两行，令其为基T 2
Figure PCTCN2019075503-appb-000023
用矩阵T 2将式(9)变换到α-β坐标系中的磁链分量(ψ' α、ψ' β)
Figure PCTCN2019075503-appb-000024
可见，式(20)中α-β坐标系上定子电感、永磁磁链部分的常系数不相等，因此即使电流在α-β坐标系上的轨迹为圆形，但由于磁链方程不再是对称模型，因此定子磁链在α-β坐标系上轨迹不是圆形。为实现故障前后的定子磁链轨迹是相同的圆，α-β坐标系中的电流轨迹将不是圆形。然而直接转矩是建立在电流和磁链圆轨迹基础上的。由此，为实现直接转矩控制，矩阵T 2需要进一步地进行修正。在矩阵T 2的基础上，定义矩阵T 3
Figure PCTCN2019075503-appb-000025
为在α-β坐标系中得到故障前后相同的电流圆和磁链圆轨迹，根据前面分析，定义以下约束条件
Figure PCTCN2019075503-appb-000026
式中，E为二阶单位矩阵，由此求得式(21)中x＝-1。
因此，变换矩阵T′ 4/2可以重新表示为
Figure PCTCN2019075503-appb-000027
由于A相发生开路故障，该相电流为0，x-y平面上的电流i x和α-β平面上的电流i α 相关，因此i x无需控制，仅需控制i y。同时根据x-y平面须与α-β平面正交原则以及剩余非故障相电流和为零的约束条件，删除矩阵T′ 4/2第三行。根据空间正交性原理，得到用于将α-β坐标系中的变量变换到自然坐标系的矩阵T 1的第三列，则开路容错情况下的容错变换矩阵T 4/2及其逆变换矩阵T 2/4可表示为
Figure PCTCN2019075503-appb-000028
Figure PCTCN2019075503-appb-000029
因此，当A相发生开路故障后，使用上述容错变换矩阵，不仅有效抑制A相开路引起的磁链扰动，同时能够实现和正常情况下相同的磁链和电流轨迹圆，进而能够实现相开路故障情况下直接转矩控制的无扰运行。
第二部分，A相短路后的磁链扰动抑制
步骤3，当A相发生短路故障时，A相短路电流i sc引起磁链变化部分定义为短路磁链扰动ψ″ A；定义用来抵消该短路磁链扰动的剩余非故障相补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)，根据故障前后定子磁链矢量不变原则、铜耗最小或剩余非故障相补偿电流幅值相等原理、以及剩余非故障相补偿电流之和为零的约束条件，推导出该补偿磁链以及补偿电流(i compB、i compC、i compD、i compE)。具体为：
在第一部分的基础上，当电机发生相短路故障时，A相电流、电机五相绕组磁链分别为
i″ A＝i sc     (26)
[ψ″ A ψ″ B ψ″ C ψ″ D ψ″ E] T＝L(θ)[i″ A i″ B i″ C i″ D i″ E] T+ψ f    (27)式中，i″ B、i″ C、i″ D、i″ E分别为A相短路后的剩余非故障相的容错电流，ψ″ A为A相短路电流引起的磁链扰动，ψ″ B、ψ″ C、ψ″ D、ψ″ E分别为A相短路后的剩余非故障相容错磁链，i sc为A相短路电流。
将式(27)所示的定子磁链按照绕组空间位置进行矢量合成，得到A相短路后的定子 磁链矢量
Figure PCTCN2019075503-appb-000030
为抵消掉短路电流引起的磁链扰动，需保持短路故障前后不变的定子磁链矢量。根据前面相短路故障和相开路故障对电机系统的影响，定义A相短路后剩余非故障相的补偿电流，其与剩余非故障相的开路容错电流以及短路容错电流(i″ B、i″ C、i″ D、i″ E)关系为
Figure PCTCN2019075503-appb-000031
Figure PCTCN2019075503-appb-000032
式中，i compB、i compC、i compD、i compE分别为剩余非故障相的补偿电流，x b、y b、x c、y c、x d、y d、x e、y e分别为剩余非故障相B、C、D、E相的补偿电流的系数。
定义剩余非故障相的补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)分别为
[ψ compB ψ compC ψ compD ψ compE] T＝L(θ)'[i compB i compC i compD i compE] T   (31)
剩余非故障相的补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)用来抵消短路电流引起的磁链扰动，根据式(28)和式(31)推导出
i sc+εi compB+ε 2i compC+ε 3i compD+ε 4i compE＝0     (32)
根据式(29)和(31)，电机五相磁链可以表示为
Figure PCTCN2019075503-appb-000033
根据式(32)和(33)，电机定子磁链矢量可以进一步表示为
Figure PCTCN2019075503-appb-000034
可见，式(34)和开路容错情况下的式(11)一致，因此只要补偿磁链消除短路电流引起 的磁链扰动，短路容错情况下的定子磁链矢量便开路故障情况下一样，而根据前面推导可知开路情况下的定子磁链与正常情况下相等，于是短路容错情况下的定子磁链矢量也与正常情况下相等。同时确保了A相短路故障前后不变的定子磁链轨迹圆、电流轨迹圆和恒定的定子绕组磁动势。
进一步，求取剩余非故障相的补偿电流与A相短路电流的关系。根据A相短路电流与反电势的关系，可以得到A相短路电流的表达式
i sc＝I fcos(ωt-θ sc)      (35)
式中，I f是短路电流的幅值，θ sc是A相反电势和A相轴线的夹角。
由于电机绕组星形连接，剩余非故障相的补偿电流和应为0，得到
Figure PCTCN2019075503-appb-000035
基于铜耗最小原理或补偿电流幅值相等原理，定义目标函数
Figure PCTCN2019075503-appb-000036
结合式(32)、(35)、(36)和(37)，得到剩余非故障相的补偿电流为
Figure PCTCN2019075503-appb-000037
因此在A相发生短路故障后，采用式(38)的补偿电流能够有效消除掉短路电流引起的磁链扰动，保持故障前后不变的定子磁链矢量。
步骤4，在步骤3的基础上，采集A相短路故障后剩余非故障相的短路容错电流i″ B、i″ C、i″ D、i″ A。采用变换矩阵T 4/2将该短路容错电流减去补偿电流后得到的开路容错电流变换到α-β和x-y坐标系中的电流分量为i α、i β、i y，同时采用控制器控制电流i y为零，该控制器的输出为三维空间目标电压
Figure PCTCN2019075503-appb-000038
具体过程为：结合式(16)、(29)和(38)，采用T 4/2将在自然坐标系上的剩余非故障相短路容错电流(i″ B、i″ C、i″ D、i″ E)减去补偿电流(i compB、i compC、i compD、i compE)后得到的电流变换到α-β和x-y坐标系中
Figure PCTCN2019075503-appb-000039
其中，i″ α、i″ β、i″ y为A相短路后在α-β和x-y坐标系中的短路容错电流分量。
步骤5，剩余非故障相短路容错磁链ψ″ B、ψ″ C、ψ″ D、ψ″ E分别减去非故障相补偿磁链ψ compB、ψ compC、ψ compD、ψ compE，得到开路容错磁链为ψ' B、ψ' C、ψ' D、ψ' E，由T 4/2将该磁链变换到α-β坐标系中的磁链分量为ψ α和ψ β；或步骤5，由T 4/2将剩余非故障相短路容错磁链ψ″ B、ψ″ C、ψ″ D、ψ″ E变换到α-β坐标系中的短路容错磁链分量为ψ″ α、ψ″ β，以及将非故障相的补偿磁链ψ compB、ψ compC、ψ compD、ψ compE变换到α-β坐标系中的补偿磁链分量为ψ compα、ψ compβ，再将短路容错磁链分量ψ″ α、ψ″ β分别减去补偿磁链分量ψ compα、ψ compβ得到α-β坐标系中的磁链分量为ψ α和ψ β。具体过程为：
当A相短路后，剩余非故障相短路容错磁链(ψ″ B、ψ″ C、ψ″ D、ψ″ E)减去补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)，即式(33)减去式(31)，并结合式(38)得到
Figure PCTCN2019075503-appb-000040
可见该磁链和式(9)一致。结合式(39)，用容错变换矩阵T 4/2将式(40)的定子磁链变换到α-β坐标系中的磁链分量(ψ α、ψ β)
Figure PCTCN2019075503-appb-000041
或者，由T 4/2将剩余非故障相短路容错磁链(ψ″ B、ψ″ C、ψ″ D、ψ″ E)变换到α-β坐标系中的短路容错磁链分量(ψ″ α、ψ″ β)
Figure PCTCN2019075503-appb-000042
采用T 4/2将剩余非故障相的补偿磁链(ψ compB、ψ compC、ψ compD、ψ compE)变换到α-β坐标 系中的补偿磁链分量(ψ compα、ψ compβ)
Figure PCTCN2019075503-appb-000043
结合式(39)，将短路容错磁链(ψ″ α、ψ″ β)减去补偿磁链(ψ compα、ψ compβ)得到α-β坐标系中的磁链分量(ψ α、ψ β)如式(41)所示。
当A相发生短路故障时，剩余非故障相的补偿磁链消除A相短路电流带来的磁链扰动，使得短路容错情况下的定子磁链矢量与正常情况下相同；同时确保剩余非故障相的补偿电流与短路电流的合成磁动势为零；结合容错变换矩阵及其逆矩阵，实现故障前后α-β坐标系中的磁链轨迹以及电流轨迹为圆形，最终实现短路故障下无扰容错运行。
第三部分，实现一相短路容错直接转矩控制无扰运行
步骤6，在步骤4和5的基础上，采用磁链观测器和转矩观测器估算出定子磁链矢量的幅值ψ s、磁链角θ s和转矩T e。具体过程为：基于电压模型或者电流模型构建磁链观测器，本发明根据式(41)构建的磁链观测器在α-β坐标系中观测出的定子磁链分量(ψ α、ψ β)为
Figure PCTCN2019075503-appb-000044
式中，ψ s是定子磁链幅值，θ s是定子磁链角。
根据磁共能法，构建容错情况下的转矩观测器，观测出的电机转矩为
Figure PCTCN2019075503-appb-000045
式中，P为电机极对数。
步骤7，转矩给定值
Figure PCTCN2019075503-appb-000046
和估算值T e作差，该差值经控制器获得转矩增量角Δδ，进而根据观测出的定子磁链矢量幅值ψ s、磁链给定值
Figure PCTCN2019075503-appb-000047
定子磁链角θ s以及转矩增量角Δδ推导出磁链给定值和定子磁链在α轴和β轴上的差值为Δψ α、Δψ β；由此，根据定子电压方程在α-β坐标系中构建电压矢量预测器，预测给定电压在α轴和β轴上的分量为
Figure PCTCN2019075503-appb-000048
Figure PCTCN2019075503-appb-000049
采用T 2/4将求出的定子目标电压分量
Figure PCTCN2019075503-appb-000050
变换到自然坐标系上，得到电机相电压指令为
Figure PCTCN2019075503-appb-000051
同时令
Figure PCTCN2019075503-appb-000052
具体过程为：转矩给定值
Figure PCTCN2019075503-appb-000053
和估算值T e作差，该差值经PI调节器获得转矩增量角Δδ，进而根据观测的定子磁链幅值ψ s、磁链给定值
Figure PCTCN2019075503-appb-000054
定子磁链角θ s以及转矩增量角Δδ推导出磁链给定值和估测的定子磁链在α-β轴上的差值(Δψ α、Δψ β)
Figure PCTCN2019075503-appb-000055
当电机A相发生短路故障后，电机绕组中注入式(29)的短路容错电流后，电压方程可以表示为
Figure PCTCN2019075503-appb-000056
采用容错变换矩阵T 4/2将式(47)自然坐标系上剩余非故障相电压方程变换到α-β坐标系中的电压分量(u α、u β)
Figure PCTCN2019075503-appb-000057
式中，u compα＝-0.3998Ri sc＝0.3998(u α-Ri α)＝0.3998dψ α/dt。
根据式(44)和(48)，在α-β坐标系中构建电压矢量预测器，获得该电压矢量在α-轴和β-轴上的定子目标电压分量
Figure PCTCN2019075503-appb-000058
Figure PCTCN2019075503-appb-000059
式中，T为控制器采样周期。
采用PI控制器控制i y为零，定义该PI控制器的输出电压为
Figure PCTCN2019075503-appb-000060
采用T 2/4将式(49)定子目标电压分量
Figure PCTCN2019075503-appb-000061
以及三维空间目标电压
Figure PCTCN2019075503-appb-000062
变换到自然坐标系，得电机相电压指令
Figure PCTCN2019075503-appb-000063
为
Figure PCTCN2019075503-appb-000064
由于A相发生短路故障，故令A相电压指令为
Figure PCTCN2019075503-appb-000065
步骤8，将步骤7所得的相电压指令送电压源逆变器，采用基于零序电压注入的脉宽调制CPWM法实现五相永磁电机一相短路故障后的容错直接转矩控制。具体因为采用基于零序电压谐波信号c 0＝-(max(u i)+min(u i))/2(u i是五相正弦调制波每一相函数)注入的CPWM方法能获得与SVPWM方法相同的控制效果，因此本发明拟采用基于注入零序电压谐波的CPWM调制技术。
式(50)的相电压指令
Figure PCTCN2019075503-appb-000066
经电压源逆变器，且采用基于零序电压注入的脉宽调制CPWM技术实现五相永磁电机一相短路故障后的容错直接转矩控制。本发明提出的一相短路容错直接转矩控制策略如图3所示。
图4-7为A相从正常到短路故障情况下电机无容错运行时的相电流波形、转矩波形、α-β上的定子磁链和电流轨迹，0.15s时A相短路故障发生。可见，电流波形发生畸变；电机转矩波动明显；尽管磁链幅值相差很小，但α-β坐标系中的定子磁链分量波形仍然存在畸变；电流轨迹波动大，且不再是恒定的圆形轨迹。图8-11为A相从正常到短路故障情况下电机容错运行时的相电流波形、转矩波形、α-β坐标系中的定子磁链和电流轨迹。0.15s时A相短路故障发生，立即启动本发明的容错直接转矩控制策略，可见，和故障情况下相比较，电流正弦度明显改善，电机输出转矩脉动得到明显抑制，几乎没有脉动，α-β上定子磁链轨迹与正常运行时几乎一致，故障前后的电流轨迹几乎是相同的圆形轨迹。图12为正常运行过程中转矩指令阶跃上升时的电机输出转矩波形，响应时间为1ms。图13为A相短路容错直接转矩控制运行过程中转矩指令阶跃上升时的电机输出转矩波形，响应时间亦为1ms。可见，采用图3所示的本发明短路容错直接转矩控制策略后，输出转矩几乎没有波动，相电流正弦度较好，故障前后α-β上恒定的电流圆、磁链圆，同时电机的动态性能也没有受到影响，和正常情况下的动态性能相同。
若电机某一相发生短路故障，该相和A相间隔电角度ka，k＝0、1、2、3、4，k＝0 对应着A相短路故障、k＝1对应B相短路故障、k＝2对应C相短路故障、k＝3对应D相短路故障、k＝4对应E相短路故障，将自然坐标系逆时针旋转0.4kπ电角度，使故障前的A相轴线和故障相轴线重合且方向一致，然后，将容错直接转矩控制策略中的θ用θ-0.4kπ、θ s用θ s-0.4kπ代替，此时磁链观测器、α-β坐标系中的磁链差值变为
Figure PCTCN2019075503-appb-000067
Figure PCTCN2019075503-appb-000068
从以上所述可知，本发明用于五相永磁电机的一相短路容错直接转矩控制策略在电机驱动系统允许最大电流情况下，不但能保证一相短路故障时电机输出转矩和正常状态下一致，而且能明显抑制电机一相短路故障后的转矩波动，更为关键的是具有和故障前一样的动态性能、稳定性能和转矩跟随精度，且适用于任何一相发生短路故障的情况，通用性强，无需复杂计算，CPU开销小。该发明和容错矢量控制策略相比，具有结构简单、CPU开销小、动态响应速度更快的特点，使得其在电动汽车等对电机运行可靠性要求高、动态性能要求高的系统中拥有很好的应用前景。
虽然本发明已以较佳实施例公开如上，但实施例并不是用来限定本发明的。在不脱离本发明之精神和范围内，所做的任何等效变化或润饰，均属于本申请所附权利要求所限定的保护范围。
在本说明书的描述中，参考术语“一个实施例”、“一些实施例”、“示意性实施例”、“示例”、“具体示例”、或“一些示例”等的描述意指结合该实施例或示例描述的具体特征、结构、材料或者特点包含于本发明的至少一个实施例或示例中。在本说明书中，对上述术语的示意性表述不一定指的是相同的实施例或示例。而且，描述的具体特征、结构、材料或者特点可以在任何的一个或多个实施例或示例中以合适的方式结合。
尽管已经示出和描述了本发明的实施例，本领域的普通技术人员可以理解：在不脱离本发明的原理和宗旨的情况下可以对这些实施例进行多种变化、修改、替换和变型，本发明的范围由权利要求及其等同物限定。

Claims (8)

  1. 一种五相永磁电机一相短路容错直接转矩控制方法，其特征在于，包括以下步骤：
    步骤1，建立五相永磁电机正常运行时的定子磁链数学模型；
    步骤2，当A相发生开路故障时，A相开路引起磁链变化部分定义为开路磁链扰动ψ' A；根据故障前后定子磁链矢量不变、非故障相电流之和为零、以及铜耗最小原则，求出电机容错运行情况下的非故障相开路容错电流为i' B、i' C、i' D、i' E；在此基础上，推导出将开路故障情况下自然坐标系中的变量变换到α-β坐标系中的容错变换矩阵T 4/2以及对应的逆矩阵T 2/4；
    步骤3，当A相发生短路故障时，A相短路电流i sc引起磁链变化部分定义为短路磁链扰动ψ″ A；定义用来抵消该短路磁链扰动的剩余非故障相补偿磁链ψ compB、ψ compC、ψ compD、ψ compE，根据故障前后定子磁链矢量不变原则、铜耗最小或剩余非故障相补偿电流幅值相等原理、以及剩余非故障相补偿电流之和为零的约束条件，推导出该补偿磁链以及补偿电流i compB、i compC、i compD、i compE；
    步骤4，在步骤3的基础上，采集A相短路故障后剩余非故障相的短路容错电流i″ B、i″ C、i″ D、i″ E。采用变换矩阵T 4/2将该短路容错电流减去补偿电流后得到的开路容错电流变换到α-β和x-y坐标系中的电流分量为i α、i β、i y；同时采用控制器控制电流i y为零，该控制器的输出为三维空间目标电压
    Figure PCTCN2019075503-appb-100001
    步骤5，剩余非故障相短路容错磁链ψ″ B、ψ″ C、ψ″ D、ψ″ E分别减去非故障相补偿磁链ψ compB、ψ compC、ψ compD、ψ compE，得到开路容错磁链为ψ' B、ψ' C、ψ' D、ψ' E，由T 4/2将该磁链变换到α-β坐标系中的磁链分量为ψ α和ψ β；
    或步骤5，由T 4/2将剩余非故障相短路容错磁链ψ″ B、ψ″ C、ψ″ D、ψ″ E变换到α-β坐标系中的短路容错磁链分量为ψ″ α、ψ″ β，以及将非故障相的补偿磁链ψ compB、ψ compC、ψ compD、ψ compE变换到α-β坐标系中的补偿磁链分量为ψ compα、ψ compβ，再将短路容错磁链分量ψ″ α、ψ″ β分别减去补偿磁链分量ψ compα、ψ compβ得到α-β坐标系中的磁链分量为ψ α和ψ β；
    步骤6，在步骤4和5的基础上，采用磁链观测器和转矩观测器估算出定子磁链矢量的幅值ψ s、磁链角θ s和转矩T e；
    步骤7，转矩给定值
    Figure PCTCN2019075503-appb-100002
    和估算值T e作差，该差值经控制器获得转矩增量角Δδ，进而根据观测出的定子磁链矢量幅值ψ s、磁链给定值
    Figure PCTCN2019075503-appb-100003
    定子磁链角θ s以及转矩增量角Δδ推导出磁链给定值和定子磁链在α轴和β轴上的差值为Δψ α、Δψ β，由此，根据定子电压方程在α-β坐标系中构建电压矢量预测器，预测给定电压在α轴和β轴上的分量为
    Figure PCTCN2019075503-appb-100004
    Figure PCTCN2019075503-appb-100005
    采用T 2/4将求出的定子目标电压分量
    Figure PCTCN2019075503-appb-100006
    变换到自然坐标系上，得到电机相电压指令为
    Figure PCTCN2019075503-appb-100007
    同时令A相电压指令
    Figure PCTCN2019075503-appb-100008
    步骤8，将步骤7所得的相电压指令送电压源逆变器，采用基于零序电压注入的脉宽调制CPWM法实现五相永磁电机一相短路故障后的容错直接转矩控制。
  2. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在于，所述步骤2的具体过程为：
    步骤2.1，当A相开路故障发生后，定义A相开路故障引起的磁链扰动为ψ' A＝[L AB L AC L AD L AE][i' B i' C i' D i' E] T+ψ mcosθ，其中，ψ m为永磁磁链幅值，θ为电角度，根据开路故障前后定子磁链矢量不变、非故障相电流之和为零、铜耗最小原则，得到电机容错运行情况下的非故障相开路容错电流i' B、i' C、i' D、i' E以及矩阵T 1
    Figure PCTCN2019075503-appb-100009
    Figure PCTCN2019075503-appb-100010
    式中，i α、i β分别是定子电流在α轴和β轴上的分量；
    步骤2.2，当A相发生开路故障后，系统自由度变为三个，两个自由度在α-β平面，另外一个自由度在x-y平面，因此将正常情况下的变换矩阵改写为T′ 4/2
    Figure PCTCN2019075503-appb-100011
    式中，a＝2π/5；
    步骤2.3，为在α-β坐标系中得到故障前后相同的电流和磁链轨迹圆，在T′ 4/2的基础 上，定义矩阵T 3为
    Figure PCTCN2019075503-appb-100012
    步骤2.4，定义以下约束条件
    Figure PCTCN2019075503-appb-100013
    式中，E为二阶单位矩阵，L m＝0.2(L d+L q)，L θ＝0.2(Lq-L d)，L d、L q分别为电机d轴和q轴电感，ψ' B、ψ' C、ψ' D、ψ' E分别为A相开路后的剩余非故障相容错磁链[ψ' B ψ' C ψ' D ψ' E] T＝L(θ)'[i' B i' C i' D i' E] T+ψ' f，ψ' f为A相开路后电机永磁体耦合到定子侧的永磁磁链，可表示为ψ' f＝ψ m[cos(θ-a) cos(θ-2a) cos(θ-3a) cos(θ-4a)] T，L ls为电机相漏感，L(θ)'为A相开路后将正常运行时电感矩阵L(θ)去掉第一行以及第一列得到的新电感矩阵，再结合步骤2.1所得的开路容错电流表达式，求得步骤2.3矩阵T 3中的x＝-1；
    步骤2.5，由此，变换矩阵T′ 4/2可以重新表示为
    Figure PCTCN2019075503-appb-100014
    步骤2.6，由于A相发生开路故障，x-y平面上的电流i x和α-β平面上的电流i α相关联，因此i x无需控制，删除矩阵T′ 4/2第三行；根据空间正交性原理，推导出用于将α-β坐标系中的变量变换到自然坐标系的矩阵T 1的第三列，则开路情况下的容错变换矩阵T 4/2及其逆变换矩阵T 2/4可表示为
    Figure PCTCN2019075503-appb-100015
    Figure PCTCN2019075503-appb-100016
  3. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在 于，所述步骤3中，定义非故障相补偿磁链ψ compB、ψ compC、ψ compD、ψ compE为：
    [ψ compB ψ compC ψ compD ψ compE] T＝L(θ)'[i compB i compC i compD i compE] T，
    i compB、i compC、i compD、i compE为剩余非故障相补偿电流，根据故障前后定子磁链矢量不变原则，剩余非故障相的补偿磁链用来抵消短路电流引起的磁链扰动，推导出
    i sc+εi compB+ε 2i compC+ε 3i compD+ε 4i compE＝0，
    式中，ε＝e ja；在此基础上，根据铜耗最小或剩余非故障相补偿电流幅值相等原理、以及剩余非故障相补偿电流之和为零的约束条件，推导出非故障相补偿电流
    Figure PCTCN2019075503-appb-100017
  4. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在于，所述步骤4中，该短路容错电流减去补偿电流后得到的开路容错电流在α-β和x-y坐标系中的电流分量i α、i β、i y为：
    Figure PCTCN2019075503-appb-100018
    其中，i″ α、i″ β、i″ y为A相短路后在α‐β和x‐y坐标系中的短路容错电流分量。
  5. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在于，所述步骤5的具体过程为：
    步骤5.1，结合步骤3和4，剩余非故障相短路容错磁链ψ″ B、ψ″ C、ψ″ D、ψ″ E分别减去非故障相的补偿磁链ψ compB、ψ compC、ψ compD、ψ compE得到非故障相磁链，可见该磁链就是开路容错磁链ψ' B、ψ' C、ψ' D、ψ' E
    [ψ' B ψ' C ψ' D ψ' E] T＝L(θ)'[i' B i' C i' D i' E] T+ψ' f；
    步骤5.2，用容错变换矩阵T 4/2将步骤5.1所得的开路容错磁链变换到α-β坐标系中的磁链分量ψ α和ψ β为
    Figure PCTCN2019075503-appb-100019
    或步骤5.1，结合步骤3和4，采用T 4/2将剩余非故障相短路容错磁链ψ″ B、ψ″ C、ψ″ D、ψ″ E变换到α-β坐标系中的短路容错磁链分量ψ″ α、ψ″ β，同时将剩余非故障相的补偿磁链 ψ compB、ψ compC、ψ compD、ψ compE变换到α-β坐标系中的补偿磁链分量ψ compα和ψ compβ，得到
    Figure PCTCN2019075503-appb-100020
    Figure PCTCN2019075503-appb-100021
    或步骤5.2，将短路容错磁链ψ″ α、ψ″ β减去补偿磁链ψ compα、ψ compβ得到α-β坐标系中的磁链分量ψ α和ψ β。
  6. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在于，步骤6中，采用磁链观测器和转矩观测器估算出定子磁链矢量的幅值ψ s、磁链角θ s和转矩T e为：
    Figure PCTCN2019075503-appb-100022
    T e＝2.5p(ψ αi β-ψ βi α)，
    式中，p为电机极对数；
  7. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在于，所述步骤7的具体过程为：
    步骤7.1，当电机A相发生短路故障后，电机绕组中注入步骤4中的短路容错电流后，电压方程可以表示为
    Figure PCTCN2019075503-appb-100023
    式中，R为定子电阻；
    步骤7.2，采用容错变换矩阵T 4/2将自然坐标系上剩余非故障相电压方程变换到α-β坐标系中的电压分量u α和u β
    Figure PCTCN2019075503-appb-100024
    式中，u compα＝-0.3998Ri sc＝0.3998(u α-Ri α)＝0.3998dψ α/dt；
    步骤7.3，在α-β坐标系中，根据Δψ α和Δψ β构建电压矢量预测器，获得该电压矢量在α-β坐标系上的定子目标电压分量
    Figure PCTCN2019075503-appb-100025
    和
    Figure PCTCN2019075503-appb-100026
    Figure PCTCN2019075503-appb-100027
    式中，T为控制器采样周期；
    步骤7.4，采用T 2/4将步骤7.3所得的定子目标电压分量
    Figure PCTCN2019075503-appb-100028
    和
    Figure PCTCN2019075503-appb-100029
    以及步骤4得到的三维空间目标电压
    Figure PCTCN2019075503-appb-100030
    变换到自然坐标系，得电机相电压指令
    Figure PCTCN2019075503-appb-100031
    为
    Figure PCTCN2019075503-appb-100032
    同时令A相电压指令为
    Figure PCTCN2019075503-appb-100033
  8. 根据权利要求1所述五相永磁电机一相短路容错直接转矩控制方法，其特征在于，还包括：若电机某一相发生短路故障，该相和A相间隔电角度ka，k＝0、1、2、3、4，k＝0对应着A相短路故障、k＝1对应B相短路故障、k＝2对应C相短路故障、k＝3对应D相短路故障、k＝4对应E相短路故障，将自然坐标系逆时针旋转0.4kπ电角度，使故障相轴线与故障前的A相轴线重合且方向一致，然后，将容错直接转矩控制策略中的θ用θ-0.4kπ、θ s用θ s-0.4kπ代替，此时磁链观测器、α-β坐标系中的磁链差值变为
    Figure PCTCN2019075503-appb-100034
    Figure PCTCN2019075503-appb-100035
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Cited By (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN113033017A (zh) * 2021-04-14 2021-06-25 中国华能集团清洁能源技术研究院有限公司 一种双转子永磁发电机电磁耦合损耗模拟装置和方法
CN113702766A (zh) * 2021-08-31 2021-11-26 南通大学 一种基于开关管故障诊断的电机主动短路方法
CN114024471A (zh) * 2021-11-08 2022-02-08 西北工业大学 一种基于极坐标系的永磁同步电动机电流滞环控制方法
CN114355190A (zh) * 2022-01-11 2022-04-15 湖南科技大学 一种可切换滑模观测器的双馈电机匝间短路故障检测方法
CN114400945A (zh) * 2021-12-31 2022-04-26 西安理工大学 双三相永磁同步电机缺相容错运行混合控制方法
CN115412005A (zh) * 2022-10-17 2022-11-29 四川大学 三相永磁同步电机系统开路故障无辅助电路容错控制方法
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CN117555238A (zh) * 2023-12-21 2024-02-13 北京嘉海鼎盛科技有限公司 基于动态扰动补偿的测功机断相故障容错控制方法

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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN105245156A (zh) * 2015-10-14 2016-01-13 江苏大学 一种内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法
CN106208891A (zh) * 2016-07-26 2016-12-07 江苏大学 五相永磁体内嵌式容错直线电机不相邻两相短路容错磁场定向控制方法
CN106208871A (zh) * 2016-07-26 2016-12-07 江苏大学 五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法
US9787237B2 (en) * 2014-07-11 2017-10-10 The University Of Akron Fault tolerant control system for multi-phase permanent magnet assisted synchronous reluctance motors
CN108832869A (zh) * 2018-07-12 2018-11-16 哈尔滨工业大学 基于功率不变原则的五相永磁电机短路故障容错控制方法

Family Cites Families (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103501138B (zh) * 2013-10-14 2016-05-04 杭州锐方科技有限公司 一种一相开路时双三相永磁同步电机的故障容错控制系统
CN104682820B (zh) * 2015-02-12 2017-05-03 江苏大学 一种用于五相容错永磁直线电机的容错控制方法
CN105577058B (zh) * 2015-12-28 2018-08-21 江苏大学 基于模糊自抗扰控制器的五相容错永磁电机速度控制方法
CN105958896B (zh) * 2016-04-26 2018-06-26 江苏大学 一种简化的五相永磁电机的短路容错控制方法
JP6769247B2 (ja) * 2016-11-04 2020-10-14 株式会社デンソー 回転電機システム
CN107231111B (zh) * 2017-07-31 2020-03-10 福州大学 五相永磁同步电机高负载能力的直接转矩控制方法
CN108306571B (zh) * 2018-01-11 2019-12-31 江苏大学 一种五相永磁直线电机一相开路容错直接推力控制方法

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US9787237B2 (en) * 2014-07-11 2017-10-10 The University Of Akron Fault tolerant control system for multi-phase permanent magnet assisted synchronous reluctance motors
CN105245156A (zh) * 2015-10-14 2016-01-13 江苏大学 一种内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法
CN106208891A (zh) * 2016-07-26 2016-12-07 江苏大学 五相永磁体内嵌式容错直线电机不相邻两相短路容错磁场定向控制方法
CN106208871A (zh) * 2016-07-26 2016-12-07 江苏大学 五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法
CN108832869A (zh) * 2018-07-12 2018-11-16 哈尔滨工业大学 基于功率不变原则的五相永磁电机短路故障容错控制方法

Cited By (12)

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Publication number Priority date Publication date Assignee Title
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CN113702766A (zh) * 2021-08-31 2021-11-26 南通大学 一种基于开关管故障诊断的电机主动短路方法
CN113702766B (zh) * 2021-08-31 2024-03-29 南通大学 一种基于开关管故障诊断的电机主动短路方法
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CN114355190B (zh) * 2022-01-11 2024-05-10 湖南科技大学 一种可切换滑模观测器的双馈电机匝间短路故障检测方法
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CN117555238A (zh) * 2023-12-21 2024-02-13 北京嘉海鼎盛科技有限公司 基于动态扰动补偿的测功机断相故障容错控制方法
CN117544039A (zh) * 2024-01-10 2024-02-09 泉州装备制造研究所 基于快速搜索的永磁同步电机模型预测电流控制方法
CN117544039B (zh) * 2024-01-10 2024-04-05 泉州装备制造研究所 基于快速搜索的永磁同步电机模型预测电流控制方法

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