WO2017063242A1 - 一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法 - Google Patents

一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法 Download PDF

Info

Publication number
WO2017063242A1
WO2017063242A1 PCT/CN2015/094171 CN2015094171W WO2017063242A1 WO 2017063242 A1 WO2017063242 A1 WO 2017063242A1 CN 2015094171 W CN2015094171 W CN 2015094171W WO 2017063242 A1 WO2017063242 A1 WO 2017063242A1
Authority
WO
WIPO (PCT)
Prior art keywords
phase
fault
short
motor
tolerant
Prior art date
Application number
PCT/CN2015/094171
Other languages
English (en)
French (fr)
Inventor
周华伟
陆震
吉敬华
朱孝勇
赵文祥
刘国海
陈龙
陈前
Original Assignee
江苏大学
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from CN201510662900.7A external-priority patent/CN105207446B/zh
Priority claimed from CN201510661212.9A external-priority patent/CN105245156B/zh
Application filed by 江苏大学 filed Critical 江苏大学
Priority to GB1807620.8A priority Critical patent/GB2559516B/en
Publication of WO2017063242A1 publication Critical patent/WO2017063242A1/zh

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K41/00—Propulsion systems in which a rigid body is moved along a path due to dynamo-electric interaction between the body and a magnetic field travelling along the path
    • H02K41/02—Linear motors; Sectional motors
    • H02K41/03—Synchronous motors; Motors moving step by step; Reluctance motors
    • H02K41/031—Synchronous motors; Motors moving step by step; Reluctance motors of the permanent magnet type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/06—Linear motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/06—Linear motors
    • H02P25/064—Linear motors of the synchronous type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/16—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the circuit arrangement or by the kind of wiring
    • H02P25/22—Multiple windings; Windings for more than three phases
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/02—Providing protection against overload without automatic interruption of supply
    • H02P29/024—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load
    • H02P29/0241—Detecting a fault condition, e.g. short circuit, locked rotor, open circuit or loss of load the fault being an overvoltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02K—DYNAMO-ELECTRIC MACHINES
    • H02K1/00—Details of the magnetic circuit
    • H02K1/02—Details of the magnetic circuit characterised by the magnetic material

Definitions

  • the invention relates to a novel motor and phase short fault fault tolerant control method, in particular to a five-compatible fault permanent magnet linear motor and a phase short fault fault tolerant vector control method thereof. It is suitable for aerospace, electric vehicles and other occasions where high reliability and dynamic performance of the motor are required.
  • Fault-tolerant motor refers to a new type of motor that improves the motor's faulty operation ability by changing the winding mode and the stator tooth structure to realize the electrical isolation, magnetic isolation, thermal isolation and physical isolation between the phases.
  • the embedded motor proposed in the IEEE Transactions on plasma science 39(1):83-86,2011(Magnetic field of a tubular linear motor with special permanent magnet) has no fault tolerance. When a certain phase fails, it cannot be normal. Operation, can not meet the requirements of high reliability and continuous operation.
  • Chinese invention patent application No. 201010120847.5 discloses a three-compatible fault magnetic flux reverse permanent magnet single-sided flat-plate linear motor. Although it introduces isolation teeth, the fault tolerance performance is improved, but the leakage is not solved.
  • the problem of serious magnetic problems is that two arm magnets with opposite magnetization directions are attached to the surface of the armature tooth, which causes a large number of magnetic lines to form a magnetic circuit without passing through the yoke of the mover, causing a short circuit of the magnetic circuit at the tooth end, which is relatively serious. Magnetic flux leakage; In addition, since the permanent magnet is attached to the surface of the armature tooth, the thrust or thrust density of the motor is difficult to increase due to the small mechanical strength between the permanent magnet and the armature tooth.
  • the motor When the fault-tolerant motor has an open circuit or short-circuit fault in a certain phase, the motor still has a certain thrust or torque output capability, but the thrust or torque fluctuates greatly, and the noise increases, which seriously affects the system performance.
  • the goal of fault-tolerant control is to optimize the fault-tolerant current for different applications, so that the output thrust or torque of the motor in the fault state is exhausted. The amount is smooth and the motor performance is at or near the performance before the fault.
  • the Chinese invention patent application number 201510059387.2 patent "a short-circuit fault-tolerant control method for five-compatible short-circuit permanent magnet motor” is directed to a five-compatible misaligned surface-mounted permanent magnet rotating motor, which decomposes the effect of short-circuit fault on motor torque into two parts: One part is the effect of open circuit fault on torque; the other part is the influence of short circuit current on torque.
  • the principle of the magnetomotive force before and after the fault and the principle of the same magnitude of the current after the fault are used to optimize the phase current of the remaining non-faulty phase; for the torque ripple caused by the short-circuit current, the magnetomotive force after the fault is zero and The principle of minimum copper loss is used to find the non-fault phase compensation current; the last two parts of the current are added to obtain the current command of the remaining non-fault phase.
  • the current hysteresis control strategy is adopted to control the five-compatible mis-surface-mounted permanent magnet rotating motor. The method is used to suppress the short-circuit phase current to cause torque fluctuation.
  • the amplitude of the residual non-fault phase compensation current is constant, independent of the motor speed, and the sum of the compensation currents of the remaining non-fault phases is not zero; more importantly, only The short-circuit fault-tolerant current is given, and the simulation is verified by Maxwell. There is no mention of which control strategy is used for control.
  • the commonly used fault-tolerant control method is to calculate the fault-tolerant current and then use the current hysteresis strategy to control.
  • this method has problems such as disordered switching frequency, large noise, and poor dynamic performance of the motor, and is not suitable for occasions where the power is large and the dynamic performance of the motor is high.
  • the present invention proposes a permanent magnet embedded cylindrical linear motor which can save rare earth permanent magnet materials and has good fault tolerance.
  • the motor can reduce the amount of rare earth permanent magnets and reduce the cost of the motor. At the same time, it is more important to improve the fault tolerance of the linear motor on the basis of maintaining the advantages of the conventional permanent magnet embedded motor.
  • the object of the present invention is to overcome the short-circuit fault of the motor phase.
  • the current fault-tolerant strategy uses the current hysteresis control to cause the inverter switching frequency to be disordered, the motor response speed to decrease, the dynamic performance is poor, the current cannot be accurately followed, the noise is serious, and the fault of the existing fault-tolerant control algorithm is complicated.
  • the high-performance short-circuit fault-tolerant vector control method for the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor used in the invention realizes high fault tolerance performance, high dynamic performance and good current following in the short circuit fault state of the motor system , reduce CPU overhead, achieve constant inverter switching frequency, reduce noise, facilitate electromagnetic compatibility design, and improve the dynamic performance and reliability of the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor under short-circuit fault condition of the present invention. Sex.
  • An in-line hybrid magnetic material fault-tolerant cylindrical linear motor comprising primary and secondary, primary length less than secondary length Degree, there is an air gap between the primary and secondary;
  • the primary includes an armature tooth, a fault-tolerant tooth and a coil winding;
  • the primary is evenly distributed with 2*m armature teeth and 2*m fault-tolerant teeth, m is a motor
  • the number of phases is m ⁇ 3;
  • the armature teeth and the fault-tolerant teeth are arranged at intervals, and only one set of disc-shaped coil windings is placed in each of the primary armature slots, and there is no winding on the fault-tolerant teeth;
  • the first armature The concentrated windings placed in the slots on both sides of the teeth and in the slots on both sides of the 2*m+1th armature teeth belong to the same phase, and the inner windings on both sides of the other armature teeth belong to the other phases in turn;
  • the secondary comprises a magnetically permeable material and a permanent magnet; and is placed in an in-line manner between two magnetically permeable materials, each pair of permanent magnets being composed of a mixed magnetic material of a rare earth permanent magnet and a ferrite, and the permanent magnet is used.
  • the axial alternating magnetic charging mode, and the rare earth permanent magnet and the ferrite axial width are equal; each permanent magnet of the same magnetization direction is composed of a permanent magnet material; or each permanent magnet of the same magnetization direction is composed of two
  • the permanent magnet materials are composed of series or parallel; the poles and poles of the permanent magnet are separated by a magnetically permeable material;
  • the armature tooth width w at and the fault tolerance tooth width w ft are equal width, or the armature tooth width w at is greater than or equal to the fault tolerance tooth width w ft ; each of the armature teeth and the fault tolerant tooth have no modulation teeth, Or each of the armature teeth and the fault-tolerant teeth are provided with modulation teeth.
  • the shape of the permanent magnet of each pole is an integral cylinder, or two cylinders inside and outside are nested into a cylinder, or two cylinders up and down (or left and right) are spliced into one cylinder, or n tiles Splicing into a cylinder and n ⁇ 2;
  • the wall thickness of the permanent magnet cylinder is smaller than the wall thickness of the cylinder of the magnetic material, and the inner diameter of the cylinder of the permanent magnet is larger than the inner diameter of the cylinder of the magnetic material, and the outer diameter of the cylinder of the permanent magnet Less than the outer diameter of the cylinder of magnetically permeable material, the permanent magnet cylinder and the cylinder of magnetically permeable material are coaxially mounted;
  • the in-line hybrid magnetic material fault-tolerant linear motor is a single-sided flat plate structure, or a bilateral flat plate structure, or a cylindrical structure, and the motor can be used as a generator or an electric motor.
  • the cylindrical linear motor is a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor, and its short-circuit fault-tolerant vector control method includes The following steps:
  • Step 1 When the motor has A phase short circuit fault, use the motor residual non-fault phase current to compensate the short circuit fault phase, which leads to the lack of normal thrust of the phase. According to the obtained current, the remaining four non-fault phase coordinates are obtained to the two phase stationary coordinates. Transforming the generalized Clark transform matrix T 4s/2s and its inverse transform matrix T 2s/4s , defining the transformation matrix C 2s/2r and its inverse transformation matrix C 2r/2s of the two-phase stationary coordinate system to the synchronous rotating coordinate system;
  • Step 2 Establish a mathematical model of the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor in the phase of the open circuit fault state on the synchronous rotating coordinate system;
  • Step 3 Using the non-fault phase current to suppress the thrust fluctuation caused by the short-circuit fault phase current, and obtain the short-circuit compensation current for the non-fault phase for suppressing the thrust phase fluctuation caused by the fault phase short-circuit current (i′′ B , i′′ C , i′′ D , i′′ E ), transforming the current (i′′ B , i′′ C , i′′ D , i′′ E ) to the short-circuit compensation current (i′′ on the two-phase stationary coordinate system by using the generalized Clark transform matrix T 4s/2s ⁇ , i′′ ⁇ , i′′ z );
  • Step 4 Transform the remaining four-phase non-faulty phase currents (i B , i C , i D , i E ) sampled on the natural coordinate system to the two-phase stationary coordinates using the generalized Clark transform matrix T 4s/2s obtained in step 1.
  • step 4 the remaining four-phase non-faulty phase currents (i B , i C , i D , i E ) sampled on the natural coordinate system, and the short-circuit compensation of the non-faulty phase used to suppress the short-circuit current causing the thrust fluctuation current (i "B, i" C , i "D, i” E) obtained by subtracting (i 'B, i' C , i 'D, i' E), using the generalized Clarke transform matrix T 4s / 2s and Pike Transforming matrix C 2s/2r transforms (i' B , i' C , i' D , i' E ) into a current (i d , i q , i z ) on a rotating coordinate system;
  • Step 5 the current command in the rotating coordinate system And the difference between the feedback currents (i d , i q , i z ) on the rotating coordinate system is obtained by the current regulator to obtain the voltage command on the rotating coordinate system.
  • Step 6 in order for the motor to generate a non-fault phase compensation current (i" B , i" C , i" D , i” E ) for suppressing the thrust fluctuation caused by the short-circuit current, according to the A-phase short-circuit current and the A-side potential
  • the relationship and the mathematical expression of the short-circuit compensation current define the short-circuit compensation voltage (u′′ B , u′′ C , u′′ D , u′′ E ) of the remaining non-faulty phase. Converting the compensation voltage to a short-circuit compensation voltage on a two-phase stationary coordinate system using a generalized Clark transform matrix T 4s/2s
  • Step 7 the voltage command on the two-phase stationary coordinate system Adding to (u" ⁇ , u" ⁇ , u” z )
  • step 7 using the generalized Clark inverse transformation matrix T 2s/4s to apply the voltage command on the two-phase stationary coordinate system Transform the voltage command to the natural coordinate system Then, the compensation voltages (u" B , u" C , u" D , u” E ) of the remaining non-faulty phases are added, and then added to the opposite potentials to obtain the desired phase voltage command.
  • Step 8 the desired phase voltage command obtained in step 7 is implemented by a voltage source inverter using CPWM modulation based on zero sequence voltage harmonic injection to realize a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor after one-phase short-circuit fault The disturbance tolerance vector runs.
  • the permanent magnet of the invention is mounted on the secondary body in an embedded manner, and has a simple structure and high reliability (the permanent magnet does not fall off from the secondary due to excessive thrust, and has good firmness), large thrust, and high thrust density. High efficiency, wide constant power range, and large weak magnetic speed regulation range.
  • the armature winding on the primary of the motor adopts a disk-shaped concentrated winding, which is convenient for winding and has no end winding, which reduces the motor resistance and copper consumption, and can improve the efficiency of the motor. Only one set of windings is placed in the slots on each side of the primary armature teeth, and there are no windings on the fault-tolerant teeth, which physically isolates the phases and phases of the motor, and realizes electrical isolation, thermal isolation and magnetic circuit between phases and phases. Decoupling has good fault tolerance, which makes it have a good application prospect in the field of automotive suspension systems with high reliability requirements. By reducing the tolerance of the motor tolerance tooth gap, the groove area can be increased, thereby improving fault tolerance performance, increasing space utilization and improving efficiency.
  • the combination of the concentrated winding method and the permanent magnet embedded placement method makes the motor structure very compact, the installation is firm, the motor volume is reduced, the power density is increased, and the thrust density is increased.
  • the combination of fault-tolerant teeth and in-line permanent magnets solves the problem of small thrust of fault-tolerant linear motors. Modulating teeth are provided on the armature teeth and the fault-tolerant teeth, and the combination of the modulation teeth and the embedded permanent magnets further increases the thrust density of the motor at low speeds.
  • the motor secondary uses mixed magnetic materials, and a part of ferrite is used to replace a part of rare earth permanent magnets to form four kinds of
  • the same hybrid magnetic material structure greatly reduces the use of rare earth permanent magnets and reduces the cost of the motor.
  • the reduction of the magnetic energy product of the permanent magnet greatly reduces the eddy current loss of the motor, and the efficiency of the motor is improved.
  • the permanent magnet adopts the axial alternating magnetization mode, and is alternately mounted on the secondary with the magnetic conductive material.
  • the wall thickness of the permanent magnet cylinder is smaller than the cylinder wall thickness of the magnetic conductive material, and the inner diameter of the permanent magnet cylinder is larger than the inner diameter of the magnetic conductive material cylinder.
  • the outer diameter of the magnet cylinder is smaller than the outer diameter of the cylinder of the magnetic conductive material, and the permanent magnet cylinder and the cylinder of the magnetic conductive material are coaxially mounted, which greatly reduces the magnetic flux leakage between the adjacent N-pole and S-pole permanent magnets, thereby improving The utilization rate of permanent magnet materials.
  • the combination of permanent magnet embedded structure, mixed magnetic material and the relationship between the size and installation of the permanent magnet cylinder and the magnetic conductive material cylinder greatly reduces the leakage flux of the linear motor, reduces the eddy current loss, and improves the permanent magnet material.
  • the utilization rate reduces the manufacturing cost of the motor, improves the efficiency of the motor, reduces the volume of the motor, and increases the thrust density.
  • the in-line structure of the hybrid magnetic material, the dimensional relationship between the cylinder of the permanent magnet and the cylinder of the magnetically permeable material, and the coaxial mounting method combined with the fault tolerance make the linear motor have low cost, high efficiency, high fault tolerance and high reliability. High thrust density and wide speed range.
  • the invention can not only effectively suppress the motor thrust fluctuation under the premise of ensuring that the output thrust of the motor is equal before and after the short-circuit fault of a certain phase of the motor, but also the dynamic performance and current following performance of the motor under fault-tolerant operation.
  • the performance under normal conditions is consistent, and no complicated calculation is required.
  • the voltage source inverter has a constant switching frequency, low noise, and low CPU overhead. When any one-phase short-circuit fault occurs, the natural coordinate system only needs to rotate counterclockwise by a certain angle to enable The motor is fault-tolerant and the algorithm has certain versatility.
  • the dynamic performance and steady-state performance of the motor in the fault-tolerant operation are the same as in the normal state of the motor, and the output thrust has almost no fluctuation in the motor. Below the maximum current limit allowed by the system, the electromagnetic thrust is consistent with that before the fault, achieving undisturbed fault-tolerant operation.
  • the generalized Clark transform matrix and its inverse transform matrix derived from the residual non-faulty phase current vector in the present invention and the defined Parker transform matrix and its inverse matrix can not only reduce the voltage and current of the remaining non-faulty phase in the short circuit fault state.
  • the resistance and the inductance are transformed into the synchronous rotating coordinate system according to the equal amplitude, and the variables on the synchronous rotating coordinate system can be transformed into the natural coordinate system where the remaining non-faulty phase is located; and the variables of the zero-sequence space can be extracted. For control, it can reduce motor loss and suppress thrust pulsation.
  • the generalized Clark transform matrix and its inverse transform matrix derived from the residual non-fault inverse potential vector and the defined Parker transform matrix and its inverse matrix can only press the voltage, current, resistance and inductance of the remaining non-faulty phase in the short-circuit fault state.
  • the variables on the synchronous rotating coordinate system are transformed to the natural coordinate system where the remaining non-faulty phases are located, the variables of the zero-sequence space cannot be obtained by the transformation.
  • the short-circuit compensation current amplitude is equal or the copper loss is minimum, and the obtained short-circuit compensation current result is consistent, so that the motor
  • the copper consumption per phase is equal, the heat is equal, and the copper consumption is the smallest.
  • the short circuit compensation current combined with the extended Clark transform matrix reduces CPU overhead.
  • the combination of the Clarke transformation matrix and the Parker transformation matrix realizes the transformation from the natural coordinate system composed of the remaining non-fault phases to the synchronous rotating coordinate system in the fault state, which creates the preconditions for fault-tolerant vector control under the motor fault state.
  • the control of the zero-sequence space degree of freedom is realized, the copper loss and iron consumption of the motor are reduced, the motor efficiency is improved, and the motor thrust fluctuation and loss due to the zero-sequence space current are suppressed.
  • the combination of short-circuit compensation current extraction, short-circuit compensation voltage feedforward, generalized Clark transform matrix, Parker transform matrix and CPWM modulation of zero-sequence voltage harmonic injection makes the thrust and flux linkage of the motor phase short-circuit fault state decoupled.
  • the decoupling control of the motor thrust and flux linkage in the synchronous rotating coordinate system under the short-circuit fault condition is realized, the inverter bus voltage utilization rate is improved, and the complexity of the fault-tolerant vector control algorithm is reduced, thus realizing the fault tolerance of the motor It runs and improves the current following performance, motor dynamic performance and steady-state performance under the short-circuit fault condition of the motor, which makes the dynamic performance, steady-state performance of the motor and the performance before the motor failure.
  • the short-circuit fault-tolerant vector control strategy, the CPWM modulation of zero-sequence voltage harmonic injection and the five-phase in-line hybrid magnetic material cylinder linear motor combine to greatly improve the fault-tolerant performance, dynamic performance and stability of the motor under phase short-circuit fault conditions.
  • the performance of the state improves the upper limit speed of the motor, saves CPU overhead, reduces noise, reduces the difficulty of electromagnetic compatibility design, makes the motor have high control precision in phase short-circuit fault state, good current follow performance, high motor efficiency, and output thrust response.
  • the speed is fast and the thrust pulsation is as small as before the fault.
  • the design of the electromagnetic compatibility of the motor system is reduced. It is more suitable for electromagnetic active suspension and other requirements for dynamic performance, stability, control accuracy, reliability, fault tolerance and electromagnetic compatibility of the motor. The occasion.
  • the invention can effectively overcome the shortcomings of the inverter switching frequency caused by the traditional current hysteresis, the motor response speed decreases after the motor failure, the current followability is poor, the noise is serious, and the electromagnetic compatibility design is difficult; especially in the motor phase short circuit fault state Under the fault-tolerant vector control process, the current can be accurately followed, the steady-state performance and the dynamic performance are better than the current hysteresis control, and the high fault tolerance and high dynamic performance of the motor system under the short-circuit fault state are realized.
  • FIG. 1 is a schematic structural view of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention
  • FIG. 2 is a schematic view 1 of an embodiment of a five-phase in-line hybrid magnetic material fault-tolerant flat-plate linear motor according to the present invention
  • FIG. 3 is a second schematic diagram of an embodiment of a five-phase in-line hybrid magnetic material fault-tolerant flat-line linear motor according to the present invention.
  • FIG. 4 is a schematic view showing winding wiring of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention
  • FIG. 5 is a schematic diagram of a hybrid magnetic material permanent magnet of four different structures according to an embodiment of the present invention.
  • FIG. 6 is a schematic diagram of a hybrid magnetic material permanent magnet of four different structures and a modulation tooth arrangement on a fault-tolerant tooth and an armature tooth according to an embodiment of the present invention
  • NdFeB rare earth permanent magnet
  • FIG. 8 is a B-phase armature reaction magnetic field distribution diagram of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention
  • FIG. 9 is a waveform diagram of a B-phase inductance of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention.
  • FIG. 10 is a schematic diagram of a CPWM vector control strategy based on zero-sequence voltage harmonic injection of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention
  • FIG. 11 is a schematic diagram 1 of a short-circuit fault-tolerant vector control strategy for a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention
  • FIG. 12 is a schematic diagram of a short-circuit fault-tolerant vector control strategy for a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor according to an embodiment of the present invention
  • 15 is a phase current waveform of a fault-tolerant operation using a short-circuit fault-tolerant vector control strategy of the present invention in the case of a phase A short-circuit fault according to an embodiment of the present invention
  • 16 is a diagram showing a thrust waveform during a fault-tolerant operation using a short-circuit fault-tolerant vector control strategy of the present invention in the case of a phase A short-circuit fault according to an embodiment of the present invention
  • 17 is a diagram showing an output thrust response of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor during a fault-free step in a fault-free operation process according to an embodiment of the present invention
  • 18 is a diagram showing an output thrust response of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor during phase A short-circuit fault-tolerant operation;
  • FIG. 19 is a phase current waveform of a five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor using a short-circuit fault-tolerant vector control strategy according to an embodiment of the present invention after a short-circuit fault of phase B after a phase-to-phase short-circuit fault recovery .
  • FIG. 20 is a diagram showing the thrust waveform of the fault-tolerant operation of the motor after the short-circuit fault of the B-phase after the recovery of the A-phase short-circuit fault after the A-phase short-circuit fault recovery is performed by using the short-circuit fault-tolerant vector control strategy of the present invention.
  • the in-line hybrid magnetic material fault-tolerant linear motor of the present invention has a cylindrical structure or a flat plate structure (only one primary and secondary; or secondary is located in the middle of two primary; in the case of a flat structure, the above-mentioned rights are involved
  • the cylinders are all changed to a rectangular parallelepiped) and the motor can be used as a generator or an electric motor.
  • a schematic structural view of an in-line hybrid magnetic material fault-tolerant cylindrical linear motor includes a primary 1 and a secondary 2 .
  • the primary 1 includes an armature tooth 3, a fault-tolerant tooth 4, and a coil winding 5, and 10 of the armature teeth 3 and the fault-tolerant teeth 4 are embedded with a rare earth permanent magnet 6 and a ferrite 7 on the secondary 2, the primary 1
  • There is an air gap between the secondary 2 and the secondary 2 and the parts other than the permanent magnet and the winding on the primary 1 and the secondary 2 are made of an inexpensive magnetically permeable material 8, such as an electrical iron, a silicon steel, or a soft magnetic material (such as a slope).
  • FIG. 4 is a schematic diagram of the winding wiring of the embodiment of the present invention.
  • the motor is five-phase, and there are 10 armature teeth 3.
  • the coil winding 5 is wound by a concentrated winding, and ten of the ten armature teeth are placed in the slots on both sides.
  • the disc coil windings are A1 phase, C1 phase, E1 phase, B1 phase, D1 phase, A2 phase, C2 phase, E2 phase, B2 phase, D2 phase, and the winding directions of the respective coils are the same, and A1 and A2 are Phase A is connected in series (or in parallel) to obtain phase A, and the other four phases are available in the same manner.
  • Each permanent magnet on the secondary 2 of the motor is mounted in an embedded manner between two magnetically permeable materials 8 of the secondary 2, and the shape of each permanent magnet on the secondary 2 is a whole cylinder, or two circles inside and outside.
  • the cylinders are nested into a cylinder or two cylinders up and down (or left and right) are connected into a cylinder or n (n ⁇ 2) tiles are assembled into a cylinder, and the permanent magnets are alternately magnetized in the axial direction.
  • the permanent magnet is a hybrid magnetic material, which greatly reduces the cost of the motor; the wall thickness of the permanent magnet cylinder is smaller than that of the magnetic conductive material cylinder The wall thickness is greater than the inner diameter of the cylinder of the permanent magnet, the outer diameter of the cylinder of the permanent magnet is smaller than the outer diameter of the cylinder of the magnetic material, and the cylinder of the permanent magnet and the cylinder of the magnetic material are coaxially mounted.
  • Figure 5 lists four different hybrid magnetic material structures. In Figure 5(a), the NdFeB permanent magnet is placed on the outside of the cylinder, and the ferrite 7 is placed on the inside of the cylinder (or NdFeB is placed).
  • the permanent magnet is placed inside the cylinder, and the ferrite 7 is placed outside the cylinder; in each of the secondary permanent magnets in (b) and (c), the neodymium magnet is composed of NdFeB and ferrite 7 in series; In 5(b), the permanent magnet materials on both sides of the magnetic material on the secondary 2 are the same, not NdFeB or ferrite; and in Figure 5(c), the permanent magnet materials on both sides of the secondary magnetically conductive material are different.
  • NdFeB NdFeB and the other side is ferrite; in Figure 5(d) all ferrites 7 have the same excitation direction, all NdFeB magnetization directions are the same, but their magnetization direction is opposite to that of ferrite 7, ferroniobium
  • the boron permanent magnet and the ferrite permanent magnet are alternately mounted on the secondary.
  • the embodiment of the present invention is characterized by the performance of FIG.
  • FIG. 6 is a schematic diagram of four different structures of mixed magnetic material permanent magnets in the case where the modulating teeth 9 are added to the fault-tolerant teeth and the armature teeth according to an embodiment of the present invention.
  • Fig. 7 is a comparison of the structure of the mixed magnetic material and the counter electromotive force corresponding to the structure of the all-rare-earth permanent magnet. It can be found that the use amount of the rare earth permanent magnet 6 is reduced by 50%, and the counter electromotive force is only decreased by 26%, and The back EMF waveform generated by the full rare earth permanent magnet collapses at the waist, so it is acceptable to use a hybrid magnetic material in the design. In addition, the back electromotive force waveform is sinusoidally symmetrical and is easy to drive in an AC drive mode.
  • Figure 8 is a B-phase armature reaction magnetic field in accordance with an embodiment of the present invention.
  • Fig. 9 is a waveform diagram of a B-phase inductor according to an embodiment of the present invention.
  • phase inductance is a constant.
  • the zero sequence of c 0 -(max(u i )+min(u i ))/2 is injected into the five-phase sinusoidal modulated wave.
  • the CPWM method of voltage harmonics (u i is a five-phase sinusoidal modulation wave per phase function) and the five-phase SVPWM method can obtain the same flux linkage control effect. Therefore, the present invention performs pulse width modulation using a CPWM method based on injection of zero-sequence voltage harmonics.
  • Fig. 10 Five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor is powered by voltage source inverter, based on zero sequence
  • the vector control strategy of CPWM technology for voltage harmonic injection, the control block diagram is shown in Figure 10.
  • the phase winding current can be expressed as
  • the traveling wave magnetomotive force (MMF) generated by the motor can be expressed as
  • phase A has a short circuit fault.
  • the use of the remaining non-faulty phase current of the motor to compensate for the short-circuit fault phase results in the absence of normal thrust in the phase.
  • the phase A current be zero, and the traveling wave magnetomotive force inside the motor is generated by the remaining four-phase non-faulty phase winding, which can be expressed as
  • the traveling wave magnetic potential is consistent before and after the motor phase short-circuit fault. Therefore, the residual non-fault phase stator current needs to be adjusted so that the magnitude and speed of the traveling wave magnetomotive force before and after the motor fault remain unchanged. change. Therefore, the real part and the imaginary part of the equations (2) and (3) are equal.
  • the motor windings are connected in a star shape and their center point is not connected to the center point of the DC bus voltage. Therefore, the sum of the phase currents of the windings is zero. Taking the A phase axis of the short-circuit fault phase as the axis, according to the principle of mirror symmetry,
  • phase current command of the fault-tolerant operation of the motor is obtained by the above constraint and the condition that the amplitude of the non-fault phase current is equal.
  • Equation (5) can be expressed as a matrix
  • the degree of freedom of the system is reduced to three, two of which are located in the fundamental wave subspace, and one degree of freedom is in the zero sequence subspace. Since the electromechanical energy conversion occurs in the fundamental wave subspace, the two degrees of freedom of the fundamental wave subspace need to be controlled according to the motor thrust demand. The degree of freedom of the zero-sequence subspace only increases the loss and thrust pulsations and needs to be controlled to zero. Therefore, in order to realize the fault-tolerant vector control after the fault, the coordinate transformation matrix after the A-phase short-circuit fault needs to be obtained, so the orthogonal T 1 and T 2 should be selected as the basis of the fundamental wave subspace. According to the equation (6) current vector, select
  • the fundamental subspace and the zero sequence subspace must be orthogonal, and the zero sequence current needs to be controlled to zero. Therefore, the vector base Z of the zero sequence subspace needs to satisfy the following conditions:
  • the fundamental subspace requires energy conversion, so the energy of the fundamental subspace is converted to a synchronous rotating coordinate system, and the zero sequence subspace does not need to be transformed into a synchronous rotating coordinate system. Therefore, the transformation matrix C 2s/2r and the inverse transformation matrix C 2r/2s defining the two-phase stationary coordinate system to the synchronous rotating coordinate system are respectively
  • the phase inductance of the fault-tolerant permanent magnet linear motor is relatively small (as shown in Fig. 9), it is negligible, and the amplitude of the self-inductance fluctuates with the secondary position is small, so the phase inductance is approximated as a constant. Therefore, the phase inductance is not affected by the coordinate transformation.
  • the back electromotive force of the motor shown in Fig. 7 has a good sinusoidal degree, and the higher harmonics of the back EMF can be ignored, and the back EMF of the motor is considered to be a sine wave.
  • the back EMF vector angle is determined by the position of each phase winding in space, so the back EMF cannot use the coordinate transformation matrix proposed by the present invention like the current. Therefore, in order to realize the vector control of the fault-tolerant permanent magnet linear motor in the open fault state of the A phase, the model in the natural coordinate system under the open circuit fault state of the motor can be expressed as
  • ⁇ the pole distance
  • v the secondary operating electric speed
  • the thrust equations of the motor under open fault fault tolerance are derived from the transformation matrix equations (14), (15), (19) and (20).
  • the thrust equation of the motor under open fault fault tolerance is derived from the transformation matrix equation (17)-(20).
  • the five-phase in-line hybrid magnetic material in the present invention can be made to be fault-tolerant by controlling i d , i q , i z in the synchronous rotating coordinate system.
  • the cylindrical linear motor outputs the desired thrust in the event of a fault.
  • the non-fault phase current is used to suppress the thrust fluctuation caused by the short-circuit phase current.
  • the motor windings are connected in a star shape, and their center points are not connected to the center point of the DC bus voltage. Therefore, the sum of the compensation currents used to suppress the short-circuit fault phase currents and the thrust fluctuations should be zero.
  • the short-circuit compensation current (i′′ B , i′′ C , i is defined by the vertical line of the A-phase axis of the short-circuit fault phase (the vertical line needs to pass through the center point of the motor winding) as the axis, defining the non-fault phase to suppress the short-circuit fault phase current and causing the thrust fluctuation. " D , i" E )
  • the motor winding is connected in a star shape, and its center point is not connected to the center point of the DC bus voltage.
  • the Lagrange multiplier method is used to solve the minimum value of the objective function formula (31).
  • the third part short-circuit fault-tolerant vector control strategy
  • the current on the two-phase stationary coordinate system of equation (37) is transformed to the current (i d , i q , i z ) on the rotating coordinate system using the Parker transformation matrix C 2s/2r of equation (19).
  • the current of the equation (38) is transformed to the current on the rotating coordinate system (i d , i q by using the equation (14) or (17) to generalize the Clark transform matrix T 4s/2s and the formula (19) the Parker transform matrix C 2s/2r . , i z ).
  • the short-circuit compensation voltage of the remaining four-phase non-fault phase is defined as
  • the expected phase voltage of equation (43) or equation (45) is realized by CPWM modulation based on zero-sequence voltage harmonic injection by voltage source inverter to realize the short-phase fault of phase A of five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor Undisturbed fault tolerant operation. Therefore, the high performance short circuit fault tolerant vector control strategy proposed by the present invention is as shown in FIG. 11 or FIG.
  • the control system simulation model of the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor shown in Fig. 1 is established in Matlab/Simulink, and the system simulation is carried out to obtain five-phase in-line hybrid magnetic Material fault-tolerant cylindrical linear motor short-circuit fault-tolerant vector control simulation results.
  • Figure 13 shows the phase current waveform when the motor is not fault-tolerant in the case of A-phase short-circuit fault, and the current fluctuates significantly.
  • Figure 14 shows the electromagnetic thrust waveform when the motor is not fault-tolerant in the case of A-phase short-circuit fault. The motor thrust fluctuation reaches 34N.
  • Figure 15 shows the phase current waveform of the fault-tolerant operation of the motor after the short-circuit fault-tolerant vector control strategy of the present invention in the case of the A-phase short-circuit fault, and the current fluctuation is reduced, which is consistent with the calculated current of equation (35).
  • Figure 16 shows the output thrust waveform of the motor during fault-tolerant operation of the motor after the fault-tolerant vector control strategy of the present invention in the case of the A-phase short-circuit fault.
  • Fig. 17 is the output thrust response of the thrust command step during the faultless operation of the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor.
  • the response time is 0.6ms.
  • Figure 18 shows the output thrust response of the thrust command step during the fault-tolerant operation of the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor.
  • the response time is also 0.6ms.
  • the motor has the same dynamic performance as the normal state of the motor under fault-tolerant operation in the case of A-phase short-circuit fault, and the output thrust does not fluctuate.
  • the thrust is consistent with the fault before the fault, and the current following performance is good, achieving the fault-free fault-tolerant operation.
  • the axis and the fault phase axis coincide and the direction is the same.
  • ⁇ in C 2s/2r and C 2r/2s is replaced by ⁇ -2k ⁇ /5, ie
  • the in-line hybrid magnetic material fault-tolerant cylindrical linear motor of the present invention adopts a method of mixing magnetic materials, which saves the usage of rare earth permanent magnets and reduces the amount of use of the conventional in-line cylindrical linear motor.
  • Magnetic flux leakage improves the utilization of permanent magnets, greatly reduces the manufacturing cost of the motor, and introduces fault-tolerant teeth, which greatly improves the fault-tolerant performance and reliability of the cylindrical linear motor.
  • the short-circuit fault-tolerant vector control strategy of the five-phase in-line hybrid magnetic material fault-tolerant cylindrical linear motor can ensure the output thrust of the motor is consistent with the normal state when the one-phase short-circuit fault is ensured under the maximum current allowed by the motor drive system. Moreover, it can obviously suppress the thrust fluctuation after the motor phase short-circuit fault, and more importantly, it has the same dynamic performance, stability performance and current following accuracy as before the fault, and is suitable for any short-circuit fault of one phase, and has high versatility and no need Complex calculations, low CPU overhead. It has a good application prospect in systems with high operational reliability requirements such as electromagnetic active suspension systems. Therefore, the present invention has a good application prospect in an electromagnetic active suspension system and the like which have high operational reliability requirements.

Abstract

一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法，包括建立五相内嵌式混合磁材料容错圆筒直线电机模型；使用电机非故障相电流补偿短路故障相导致该相正常推力缺失以及抑制该相短路电流导致的推力波动；通过采取一系列的坐标变换和电压前馈补偿策略得到期望相电压，采用基于零序电压谐波注入的CPWM调制方式实现该电机相短路故障后的容错矢量控制。该方法不但能使该电机在相短路容错运行情况下抑制电机推力波动，而且更为关键的是其动态性能、稳态性能和正常状态下的性能一致，电压源逆变器开关频率恒定、CPU开销小；任何一相短路故障时，自然坐标系只需逆时针旋转一定角度，就能实现电机容错运行。

Description

一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法 技术领域
本发明涉及一种新型电机和相短路故障容错控制方法，特别是五相容错永磁直线电机及其相短路故障容错矢量控制方法。适用于航空航天、电动汽车等对电机的可靠性和动态性能有较高要求的场合。
背景技术
随着社会的发展以及人们生活水平的提高，对汽车驾乘的舒适性和安全稳定性要求越来越高。作为现代汽车的重要组成部分，悬架系统性能对汽车行驶平顺性和操作稳定性等有着极其重要的影响，因此主动悬架系统的研究受到业内高度重视。作为主动电磁悬架系统的核心部件，圆筒直线电机研究受到重视。电机在故障状态下的容错性能，直接决定着电磁悬架的可靠性和连续运行的能力。
容错电机是指通过改变绕组方式和定子齿结构，在空间上实现相与相之间的电隔离、磁隔离、热隔离和物理隔离，从而提高电机带故障运行能力的一类新型电机。文献IEEE Transactions on plasma science 39(1):83-86,2011(Magnetic field of a tubular linear motor with special permanent magnet)中提出的内嵌式电机没有容错能力，当某一相发生故障，便不能正常运行，无法满足高可靠性和连续运行的要求。中国发明专利申请号201010120847.5《一种容错式永磁直线电机》公开了一种三相容错磁通反向永磁单边平板直线电机，虽然其引入了隔离齿，容错性能提高，但没有解决漏磁严重的问题，其电枢齿表面贴装有两块磁化方向相反的永磁体，导致大量的磁力线不经过动子轭部形成磁回路，在齿端造成磁路短路的现象，产生比较严重的漏磁；另外，由于永磁体表贴在电枢齿表面，由于永磁体和电枢齿之间的机械强度较小，电机的推力或者推力密度很难做大。另一方面，近年来，国内外对稀土永磁体的使用量不断递增，而稀土材料储备有限且价格昂贵，所以电机设计过程中应关注如何减少稀土永磁体的用量，降低电机成本。铁氧体永磁材料价格低廉，产量丰富，供应稳定，所以铁氧体材料是稀土永磁体很好的替代品。
容错电机在某一相发生开路或者短路故障时，电机仍然具有一定的推力或者转矩输出能力，但是推力或者转矩波动很大，噪声增大，严重影响系统性能。容错控制的目标是针对不同应用场合对容错电流进行优化，使电机在故障状态下的输出推力或者转矩尽 量平滑，并且使电机性能达到或接近故障前的性能。中国发明专利申请号为201510059387.2的专利《一种五相容错永磁电机的短路容错控制方法》针对五相容错表贴式永磁旋转电机，将短路故障对电机转矩的影响分解为两部分：一部分是开路故障对转矩的影响；另一部分是短路电流对转矩影响。针对开路故障，采用故障前后磁动势和不变原则以及故障后电流幅值相等原则，优化剩余非故障相的相电流；针对短路电流引起的转矩波动，采用故障后磁动势为零和铜耗最小原则求出非故障相补偿电流；最后两部分电流相加，求得剩余非故障相的电流指令。根据求得的剩余非故障相电流采用电流滞环控制策略，对五相容错表贴式永磁旋转电机进行控制。该方法用于抑制短路相电流导致转矩波动的剩余非故障相补偿电流的幅值是常数，与电机转速无关，且剩余非故障相的补偿电流之和不为零；更为主要的是仅仅给出了短路容错电流，用Maxwell进行了仿真验证，没有提及具体使用何种控制策略进行控制。目前，常用的容错控制方法是：计算出容错电流，然后采用电流滞环策略进行控制。但是，该方法存在开关频率杂乱、噪声大、电机动态性能差等问题，不适合功率较大以及对电机动态性能要求高的场合。
发明内容
针对现有直线电机技术的不足，基于原有次级永磁型圆筒直线电机结构，本发明提出一种节省稀土永磁材料且容错性能好的永磁体内嵌式圆筒直线电机。该电机既能降低稀土永磁体的用量，降低电机成本，同时更为关键的是在保持传统永磁体内嵌式电机优点的基础上提高了直线电机的容错性能。
针对现有电机容错控制技术中存在的不足，以及本发明提出的五相内嵌式混合磁材料容错圆筒直线电机的特性和该类电机相短路故障特点，本发明目的是克服电机相短路故障后现有容错策略使用电流滞环控制导致逆变器开关频率杂乱、电机响应速度下降、动态性能差、电流无法精确跟随、噪声严重的缺点以及现有容错控制算法运算复杂的缺陷，提出一种用于本发明的五相内嵌式混合磁材料容错圆筒直线电机的高性能短路容错矢量控制方法，实现该类电机系统在短路故障状态下的高容错性能、高动态性能、电流良好的跟随性，减小CPU开销，实现逆变器开关频率恒定、降低噪声，便于电磁兼容设计，进而提高本发明的五相内嵌式混合磁材料容错圆筒直线电机短路故障状态下的动态性能和可靠性。
本发明的装置采用如下技术方案来实现：
一种内嵌式混合磁材料容错圆筒直线电机，包括初级和次级，初级长度小于次级长 度，初级和次级之间有气隙；所述初级包括电枢齿、容错齿和线圈绕组；所述初级均布2*m个电枢齿和2*m个容错齿，m为电机的相数且m≥3；电枢齿和容错齿交错间隔排列，初级每个电枢齿槽中都只放一套圆盘状线圈绕组，而容错齿上没有绕组；其中，第一个电枢齿两侧的槽内和第2*m+1个电枢齿两侧的槽内放置的集中绕组属于同一相，其余电枢齿两侧的槽内绕组依次属于其他相；
所述次级包括导磁材料和永磁体；采用内嵌方式放置在两块导磁材料之间，每一对永磁体是由稀土永磁体和铁氧体两种混合磁材料组成，永磁体采用轴向交替充磁方式，且稀土永磁体和铁氧体轴向宽度相等；每一个相同充磁方向的永磁体由一种永磁材料组成；或者每一个相同充磁方向的永磁体由两种永磁材料串联或并联组成；永磁体的极与极之间采用导磁材料隔离；
所述电枢齿齿宽wat和容错齿齿宽wft等宽，或电枢齿齿宽wat大于等于容错齿齿宽wft；每一电枢齿以及容错齿上均无调制齿，或者每一电枢齿以及容错齿上均设有调制齿。
进一步，所述内嵌式混合磁材料容错圆筒直线电机采用分数槽结构，极槽关系满足：Ns＝2p±2或者Ns＝2p±1。其中，Ns为初级槽数，p为次级极对数。
进一步，每一极所述永磁体的形状是一个整体圆筒、或内外两个圆筒嵌套成圆筒、或上下(或左右)两个圆筒拼接成一个圆筒、或n块瓦片拼接成一个圆筒且n≥2；永磁体圆筒的壁厚小于导磁材料圆筒的壁厚，且永磁体圆筒的内径大于导磁材料圆筒的内径，永磁体圆筒的外径小于导磁材料圆筒的外径，永磁体圆筒和导磁材料圆筒同轴安装；
进一步，所述内嵌式混合磁材料容错直线电机为单边平板结构、或双边平板结构、或者圆筒型结构，该电机能够作为发电机或者电动机。
本发明的方法的技术方案为：
当电机的相数m＝5时分为A、B、C、D、E五相，该圆筒直线电机为五相内嵌式混合磁材料容错圆筒直线电机，其短路容错矢量控制方法，包括以下步骤：
步骤1，当电机发生A相短路故障时，使用电机剩余非故障相电流补偿短路故障相导致该相正常推力缺失，根据求得的电流求取剩余的四个非故障相坐标到两相静止坐标变换的推广克拉克变换矩阵T4s/2s及其逆变换矩阵T2s/4s，定义两相静止坐标系到同步旋转坐标系的变换矩阵C2s/2r及其逆变换矩阵C2r/2s；
步骤2，建立五相内嵌式混合磁材料容错圆筒直线电机相开路故障状态下在同步旋转坐标系上的数学模型；
步骤3，使用非故障相电流抑制短路故障相电流导致的推力波动，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)，采用推广克拉克变换矩阵T4s/2s将该电流(i″B、i″C、i″D、i″E)变换到两相静止坐标系上的短路补偿电流(i″α、i″β、i″z)；
步骤4，使用步骤1获得的推广克拉克变换矩阵T4s/2s将在自然坐标系上采样的剩余四相非故障相电流(iB、iC、iD、iE)变换到两相静止坐标系上的电流(i′α、i′β、i′z)，并将该电流减去步骤3中获得的电流(i″α、i″β、i″z)得到(iα、iβ、iz)，采用派克变换矩阵C2s/2r将(iα、iβ、iz)变换到旋转坐标系上的电流(id、iq、iz)；
或步骤4，将在自然坐标系上采样到的剩余四相非故障相电流(iB、iC、iD、iE)，同用于抑制短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)相减得到(i′B、i′C、i′D、i′E)，采用推广克拉克变换矩阵T4s/2s和派克变换矩阵C2s/2r将(i′B、i′C、i′D、i′E)变换到旋转坐标系上的电流(id、iq、iz)；
步骤5，将旋转坐标系下的电流指令
Figure PCTCN2015094171-appb-000001
和旋转坐标系上的反馈电流(id、iq、iz)的差值经电流调节器得到旋转坐标系上的电压指令
Figure PCTCN2015094171-appb-000002
采用派克逆变换矩阵C2r/2s将该电压指令变换到两相静止坐标系上的电压
Figure PCTCN2015094171-appb-000003
步骤6，为使该电机产生用于抑制短路电流导致推力波动的非故障相补偿电流(i″B、i″C、i″D、i″E)，根据A相短路电流和A相反电势的关系以及短路补偿电流的数学表达方式，定义剩余非故障相的短路补偿电压(u″B、u″C、u″D、u″E)为
Figure PCTCN2015094171-appb-000004
使用推广克拉克变换矩阵T4s/2s将所述补偿电压变换到两相静止坐标系上的短路补偿电压
Figure PCTCN2015094171-appb-000005
步骤7，将两相静止坐标系上的电压指令
Figure PCTCN2015094171-appb-000006
与(u″α、u″β、u″z)相加得
Figure PCTCN2015094171-appb-000007
使用推广克拉克逆变换矩阵T2s/4s将
Figure PCTCN2015094171-appb-000008
变换到自然坐标系上的电压指令
Figure PCTCN2015094171-appb-000009
再和各相反电势相加得到期望相电压指令
Figure PCTCN2015094171-appb-000010
Figure PCTCN2015094171-appb-000011
或步骤7，采用推广克拉克逆变换矩阵T2s/4s将两相静止坐标系上的电压指令
Figure PCTCN2015094171-appb-000012
Figure PCTCN2015094171-appb-000013
变换到自然坐标系上的电压指令
Figure PCTCN2015094171-appb-000014
然后和剩余非故障相的补偿电压(u″B、u″C、u″D、u″E)相加，再和各相反电势相加得期望相电压指令
Figure PCTCN2015094171-appb-000015
Figure PCTCN2015094171-appb-000016
步骤8，将步骤7所得到的期望相电压指令经电压源逆变器采用基于零序电压谐波注入的CPWM调制实现五相内嵌式混合磁材料容错圆筒直线电机一相短路故障后无扰容错矢量运行。
本发明具有以下有益效果：
1、本发明的永磁体采用内嵌方式安装在次级上，结构简单、可靠性高(永磁体不会因为推力过大而从次级上脱落、牢固性好)、推力大、推力密度高、效率高、恒功率范围宽、有较大的弱磁调速范围。
2、电机初级上电枢绕组采用圆盘状集中绕组，绕线方便且没有端部绕组，减小了电机电阻和铜耗，且能提高电机效率。每个初级电枢齿两侧的槽里只放置一套绕组，容错齿上没有绕组，起到了对电机相与相之间的物理隔离，实现了相与相间的电隔离、热隔离以及磁路解耦，具有很好的容错性能，使其在对可靠性要求比较高的汽车悬架系统等领域中具有很好的应用前景。通过减小电机容错齿齿宽，能够增大槽面积，从而既提高容错性能，又提升空间利用率以及提高效率。另外，集中绕组方式和永磁体内嵌放置方式相结合使电机结构非常紧凑，安装牢固，电机体积减小，功率密度增大，推力密度提高。容错齿和内嵌式永磁体相结合，解决了容错直线电机推力小的问题。在电枢齿和容错齿上设置调制齿，且将调制齿和内嵌式永磁体结合使得电机低速时的推力密度进一步提升。
3、电机次级采用混合磁材料，用一部分铁氧体代替一部分稀土永磁体，构成四种不 同的混合磁材料结构，一方面大大减少了稀土永磁体的使用量，降低了电机的成本，另一方面由于永磁体磁能积降低大大减小了电机的涡流损耗，使得电机的效率提高。永磁体采用轴向交替磁化方式，且和导磁材料交替安装在次级上，永磁体圆筒壁厚小于导磁材料圆筒壁厚，永磁体圆筒内径大于导磁材料圆筒内径，永磁体圆筒外径小于导磁材料圆筒外径，且永磁体圆筒和导磁材料圆筒同轴安装，大大降低了相邻N极和S极永磁体之间的漏磁磁通，提高了永磁材料的利用率。永磁体嵌入式结构、混合磁材料和永磁体圆筒与导磁材料圆筒之间的尺寸关系及安装方式相结合使直线电机的漏磁大大减小，降低了涡流损耗，提高了永磁材料的利用率，降低了电机制造成本，提升了电机效率，减小了电机体积，增加了推力密度。混合磁材料内嵌式结构、永磁体圆筒与导磁材料圆筒之间的尺寸关系及同轴安装方式与容错相结合使得直线电机具有低成本、高效率、高容错性能、高可靠性、高推力密度及宽调速范围。
4、本发明在保证电机某一相短路故障前后电机输出推力相等的前提下，不但能有效抑制电机推力波动，而且更为关键的是能使电机容错运行情况下的动态性能、电流跟随性能和正常状态下的性能一致，并且无需复杂的计算，电压源逆变器开关频率恒定、噪声低、CPU开销小；任何一相短路故障时，自然坐标系只需逆时针旋转一定角度，就能使电机容错运行，算法具有一定的通用性。
5、采用本发明短路容错矢量控制策略后，该类电机在A相短路故障情况下，容错运行时，其动态性能、稳态性能和电机正常状态下一样，且输出推力几乎没有波动，在电机系统允许的最大电流极限值以下，电磁推力和故障前保持一致，实现了无扰容错运行。
6、由本发明中的剩余非故障相电流矢量推导出的推广克拉克变换矩阵及其逆变换矩阵以及定义的帕克变换矩阵和其逆矩阵不但能在短路故障状态下将剩余非故障相的电压、电流、电阻、电感按等幅值变换到同步旋转坐标系上，同时能将同步旋转坐标系上的这些变量变换到剩余非故障相所在的自然坐标系上；而且能将零序空间的变量提取出来，用于控制，能降低电机损耗，抑制推力脉动。采用剩余非故障相反电势矢量推导出的推广克拉克变换矩阵及其逆变换矩阵以及定义的帕克变换矩阵和其逆矩阵只能在短路故障状态下将剩余非故障相的电压、电流、电阻、电感按等幅值变换到同步旋转坐标系上，和将同步旋转坐标系上的这些变量变换到剩余非故障相所在的自然坐标系上，无法通过变换求得零序空间的变量。本发明中在短路相电流和非故障相的短路补偿电流的磁动势和为零的基础上，采用短路补偿电流幅值相等或者铜耗最小原则，求取的短路补偿电流结果一致，使得电机每相铜耗相等，发热相等，且铜耗最小，每相发热均衡，提高 了电机的可靠性。短路补偿电流和推广克拉克变换矩阵相结合，降低了CPU开销。
7、推广克拉克变换矩阵和帕克变换矩阵相结合实现了故障状态下剩余非故障相构成的自然坐标系到同步旋转坐标系的变换，为电机故障状态下的容错矢量控制创造了前提条件，另一方面在故障状态下实现了对零序空间自由度的控制，减小了电机铜耗和铁耗，提高了电机效率，抑制了由于零序空间电流导致的电机推力波动和损耗。短路补偿电流的提取、短路补偿电压的前馈、推广克拉克变换矩阵、帕克变换矩阵以及零序电压谐波注入的CPWM调制相结合使得电机相短路故障状态下的推力和磁链实现了解耦，实现了短路故障状态下在同步旋转坐标系上对电机推力和磁链的解耦控制，提高了逆变器母线电压利用率，同时减小了容错矢量控制算法的复杂性，因此实现了电机容错运行，且提高了电机短路故障状态下的电流跟随性能、电机动态性能、稳态性能，使电机动态性能、稳态性能和电机故障前的性能一致。短路容错矢量控制策略、零序电压谐波注入的CPWM调制与五相内嵌式混合磁材料圆筒直线电机相结合，大大提高了该电机在相短路故障状态下的容错性能、动态性能和稳态性能，提升了电机上限速度，节省了CPU开销，降低了噪声，降低了电磁兼容设计难度，使得该电机在相短路故障状态下控制精度高，电流跟随性能好，电机效率高、输出推力响应速度快且推力脉动和故障前一样小，电机系统电磁兼容设计难度降低，更适合应用于电磁主动悬架等对电机动态性能、稳定性能、控制精度、可靠性、容错性、电磁兼容等要求高的场合。
8、本发明能够有效克服传统电流滞环导致的逆变器开关频率杂乱、电机故障后电机响应速度下降、电流跟随性差、噪声严重、电磁兼容设计难度大的缺点；尤其在电机相短路故障状态下，容错矢量控制过程中电流能够精确跟随、稳态性能和动态性能较电流滞环控制好，实现了电机系统的在短路故障状态下的高容错性以及高动态性能。
附图说明
图1为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机的结构示意图；
图2为本发明的五相内嵌式混合磁材料容错平板直线电机的实施例示意图一；
图3为本发明的五相内嵌式混合磁材料容错平板直线电机的实施例示意图二；
图4为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机的绕组接线示意图；
图5为本发明实施例四种不同结构的混合磁材料永磁体方案图；
图6为本发明实施例四种不同结构的混合磁材料永磁体以及在容错齿和电枢齿上设调制齿方案图；
图7为本发明实施例和与之对应的全稀土永磁体(汝铁硼)容错圆筒直线电机的反电动势波形图；
图8为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机B相电枢反应磁场分布图；
图9为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机B相电感波形图；
图10为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机基于零序电压谐波注入的CPWM矢量控制策略原理图；
图11为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制策略原理图一；
图12为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制策略原理图二；
图13为本发明实施例A相短路故障时电机无容错运行时的相电流波形；
图14为本发明实施例A相短路故障时电机无容错运行时的推力波形；
图15为本发明实施例A相短路故障时使用本发明短路容错矢量控制策略容错运行时的相电流波形；
图16为本发明实施例A相短路故障时使用本发明短路容错矢量控制策略容错运行时的推力波形；
图17为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机无故障运行过程中推力指令阶跃时其输出推力响应图；
图18为本发明实施例五相内嵌式混合磁材料容错圆筒直线电机A相短路容错运行过程中推力指令阶跃时其输出推力响应图；
图19为本发明实施例使用本发明短路容错矢量控制策略的五相内嵌式混合磁材料容错圆筒直线电机在A相短路故障恢复后B相发生短路故障后电机容错运行时的相电流波形。
图20为本发明实施例使用本发明短路容错矢量控制策略的五相内嵌式混合磁材料容错圆筒直线电机在A相短路故障恢复后B相发生短路故障后电机容错运行时推力波形。
图中：1.初级；2.次级；3.电枢齿；4.容错齿；5.线圈绕组；6.稀土永磁体；7.铁氧体；8.导磁材料；9.调制齿。
具体实施方式
下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整 地描述。
为了能够更加简单明了地说明本发明的内嵌式混合磁材料容错圆筒直线电机的结构特点和有益效果，下面结合一个具体的五相内嵌式混合磁材料容错圆筒直线电机来进行详细的表述。
为了更清楚的说明本发明，将本发明内嵌式混合磁材料容错圆筒直线电机的极槽配比具体化，根据上述提出的公式，选择m＝5，即为五相电机，则电机电枢齿3和容错齿4的个数均为2*m＝10个，且Ns＝20，极对数p＝9。电枢齿3齿宽Wat和容错齿4齿宽Wft等宽，在考虑适量扩大槽面积的情况下，可适当减小容错齿的大小，即Wat≥Wft；该实施例取Wat＝Wft，即电枢齿和容错齿齿宽相等；取永磁体轴向宽度Wpm与次级齿宽Wst的比值为Wpm/Wst＝1.2；取初级槽宽Wss和初级齿宽Wat、Wft(Wat＝Wft)的比值为Wss/Wat＝1.72；取初级极距Ts与次级极距Tr的关系为Tr/Ts＝2.2。
本发明的内嵌式混合磁材料容错直线电机为圆筒型结构，或者平板结构(初级和次级均只有一个；或者次级位于两个初级的中间；在平板结构情况下，上述权利中涉及的圆筒均改为长方体)，该电机能够作为发电机或者电动机使用。
如图1所示，本发明实施例的内嵌式混合磁材料容错圆筒直线电机结构示意图，包括初级1、次级2。初级1中包括电枢齿3、容错齿4和线圈绕组5，且电枢齿3和容错齿4都为10个，次级2上内嵌有稀土永磁体6和铁氧体7，初级1和次级2之间存在气隙，初级1和次级2上除永磁体和绕组之外的部分都是由廉价的导磁材料8制成，比如电工铁、硅钢、软磁材料(如坡莫合金)等，实施例中采用的是硅钢。取初级极距Ts与次级极距Tr的关系为Tr/Ts＝2.2。另外，图2和图3为本发明的平板直线电机的实施例示意图。
图4所示本发明实施例的绕组接线示意图，本实施例电机是五相，有10个电枢齿3，线圈绕组5采用集中绕组绕制方式，十个电枢齿两侧槽中放置的圆盘状线圈绕组依次为A1相，C1相，E1相，B1相，D1相，A2相，C2相，E2相，B2相，D2相，而且各个线圈的绕线方向一致，将A1、A2相正向串联(或并联)得A相，其他四相利用同样方式可得。
电机次级2上的每一块永磁体采用嵌入方式安装在次级2的两块导磁材料8之间，次级2上每一极永磁体的形状是整体一个圆筒、或内外两个圆筒嵌套成圆筒或上下(或左右)两个圆筒相接成圆筒或者n(n≥2)块瓦片拼成圆筒，永磁体采用轴向交替充磁方式。永磁体是混合磁材料，大大降低了电机成本；永磁体圆筒壁厚小于导磁材料圆筒 壁厚，且永磁体圆筒的内径大于导磁材料圆筒的内径，永磁体圆筒的外径小于导磁材料圆筒的外径，永磁体圆筒和导磁材料圆筒同轴安装。具体地说，图5列出了四种不同的混合磁材料结构，图5(a)中汝铁硼永磁体放置在圆筒外侧，铁氧体7放置在圆筒内侧(或者将汝铁硼永磁体放置在圆筒内侧，铁氧体7放置在圆筒外侧)；图(b)和(c)中次级上每一极永磁体由汝铁硼和铁氧体7串联组成；但图5(b)中次级2上导磁材料两边的永磁体材料相同，不是汝铁硼就是铁氧体；而图5(c)中次级上导磁材料两边的永磁体材料不相同，一边是汝铁硼，另一边是铁氧体；图5(d)中所有铁氧体7励磁方向相同，所有汝铁硼充磁方向相同，但其充磁方向和铁氧体7相反，汝铁硼永磁体和铁氧体永磁体交替依次安装在次级上。本发明实施例以图5(d)进行性能说明，即由稀土(汝铁硼)永磁体6和铁氧体永磁体7的极性构成电机次级的一对极，且两者的轴向宽度相等，次级上每个极上的永磁体是整体一个圆筒，永磁体圆筒壁厚是次级上导磁材料圆筒壁厚的0.75倍，永磁体圆筒内径大于导磁材料内径，且永磁体圆筒和导磁材料圆筒是同轴安装。图6为本发明实施例在容错齿和电枢齿上加调制齿9的情况下四种不同结构的混合磁材料永磁体方案图。
图7为采用混合磁材料的结构和采用全稀土永磁体结构所对应的反电动势的比较，可以发现，稀土永磁体6的使用量降低了50％，而反电动势仅下降了26％，且由全稀土永磁体产生的反电势波形在腰部发生了塌陷，因此设计中采用混合磁材料是可接受的。另外，反电动势波形正弦对称，易于交流驱动方式进行驱动。图8为本发明实施例的B相电枢反应磁场。由于初级1中引入了容错齿4，线圈上的磁通大部分通过电枢齿3和电枢齿3两侧的容错齿4构成回路，只有很少的磁通与其他线圈交链，实现电枢绕组的空间物理隔离，使得电机的各相绕组相互独立，相与相之间磁路独立，避免了相间短路的发生，实现了热隔离、电隔离以及磁路解耦，提高了电机的容错性能。图9为本发明实施例的B相电感波形图。可知电机相与相之间的互感大大降低了，仅为自感的1.0％，说明电机具有很好的相间独立性，即电机有很好的容错性能，且自感的波动较小，可以认为相电感是常数。
在传统使用正弦波作为调制波的载波脉宽调制(CPWM)方法基础上，在五相正弦调制波中注入c0＝-(max(ui)+min(ui))/2的零序电压谐波(ui是五相正弦调制波每一相函数)的CPWM方法与五相SVPWM方法能获得相同的磁链控制效果。因此本发明采用基于注入零序电压谐波的CPWM方法进行脉宽调制。
图10五相内嵌式混合磁材料容错圆筒直线电机由电压源逆变器供电，采用基于零序 电压谐波注入的CPWM技术的矢量控制策略，控制框图见图10所示。电机正常状态稳态运行时，各相绕组电流可表示为
Figure PCTCN2015094171-appb-000017
式中，
Figure PCTCN2015094171-appb-000018
分别是旋转坐标系d轴、q轴的电流指令。
电机产生的行波磁动势(MMF)可表示为
Figure PCTCN2015094171-appb-000019
式中，α＝ej2π/5，N为各相定子绕组的有效匝数。
第一部分，当电机发生相短路故障时，假设A相发生短路故障。先使用电机剩余的非故障相电流补偿短路故障相导致该相正常推力缺失。此时，设A相电流为零，电机内部的行波磁动势由剩余的四相非故障相绕组产生，可表示为
Figure PCTCN2015094171-appb-000020
为实现电机相短路故障后无扰运行，需保持电机相短路故障前后行波磁动势一致，因此需调整剩余非故障相定子电流使电机故障前后行波磁动势的幅值与速度保持不变。于是，令式(2)、式(3)的实部与虚部均相等。
电机绕组采用星形连接，且其中心点与直流母线电压的中心点不相连，因此，绕组相电流之和为零。以短路故障相A相轴线为轴，根据镜像对称原理，设
Figure PCTCN2015094171-appb-000021
由上述约束条件以及非故障相电流幅值相等的条件，得电机容错运行的相电流指令为
Figure PCTCN2015094171-appb-000022
式(5)采用矩阵形式可表示为
Figure PCTCN2015094171-appb-000023
在A相发生开路故障后，系统自由度降为三个，其中两个自由度位于基波子空间，一个自由度位于零序子空间。由于机电能量转换发生在基波子空间，基波子空间的两自由度需要根据电机推力需求进行控制。零序子空间的自由度只会增加损耗和推力脉动，需要控制为零。因此为实现故障后的容错矢量控制，需获得A相短路故障后的坐标变换矩阵，因此需选择正交的T1和T2作为基波子空间的基。根据式(6)电流矢量，选择
Figure PCTCN2015094171-appb-000024
Figure PCTCN2015094171-appb-000025
或者，根据绕组空间分布，选择
Figure PCTCN2015094171-appb-000026
Figure PCTCN2015094171-appb-000027
基波子空间和零序子空间必须正交，且零序电流需控制为零，因此零序子空间的矢量基Z需满足如下条件：
Figure PCTCN2015094171-appb-000028
考虑相电流和为零的约束条件(4)，由式(7)、式(8)和式(11)求得从自然坐标系到两相静止坐标系的推广克拉克变换矩阵T4s/2s为
Figure PCTCN2015094171-appb-000029
其逆变换矩阵为
Figure PCTCN2015094171-appb-000030
由于绕组星形连接，其相电流之和为零，式(12)第四行将自然坐标系上的相电流变换到零序空间的电流为零，因此去掉式(12)第四行和式(13)第四列，得
Figure PCTCN2015094171-appb-000031
Figure PCTCN2015094171-appb-000032
或者，将式(9)和(10)代入式(11)，求得零序空间的矢量Z为
Z＝[z z z z]   (16)
式中，z为常数。若取z＝0.4522，从自然坐标系到两相静止坐标系的推广克拉克变换矩阵T4s/2s为
Figure PCTCN2015094171-appb-000033
其逆变换矩阵T2s/4s为
Figure PCTCN2015094171-appb-000034
由于绕组星形连接，其相电流之和为零，式(17)第三行将自然坐标系上的电流变换到零序空间的电流为零，因此可去掉式(17)第三行和式(18)第三列。但为了与式(14)和式(15)在后期公式推导过程一致，便于公式推导，此处暂不去除。
基波子空间需要进行能量转换，因此将基波子空间的能量转换到同步旋转坐标系，零序子空间不需要变换到同步旋转坐标系。因此定义两相静止坐标系到同步旋转坐标系的变换矩阵C2s/2r及其逆变换矩阵C2r/2s分别为
Figure PCTCN2015094171-appb-000035
Figure PCTCN2015094171-appb-000036
由于该容错永磁直线电机的相电感的互感相对自感很小(如图9所示)，可忽略不计，且自感随次级位置波动的幅值较小，因此将相电感近似为常数，于是相电感不受坐标变换的影响。图7所示该电机的反电势，正弦度较好，可忽略该反电势的高次谐波，认为电机反电势为正弦波。反电势矢量角是有每相绕组在空间的位置决定的，因此反电势不能像电流一样使用本发明提出的坐标变换矩阵。因此，为了实现该类容错永磁直线电机在A相开路故障状态下的矢量控制，该电机开路故障状态下在自然坐标系下的模型可表示为
Figure PCTCN2015094171-appb-000037
采用坐标变换矩阵T4s/2s式(14)或式(17)和C2s/2r式(19)将式(21)变换到同步旋转坐标系
Figure PCTCN2015094171-appb-000038
式中ω＝πv/τ＝2πf，τ为极距，v是次级运行电速度。
采用磁共能方法，由变换矩阵式(14)、式(15)、式(19)和式(20)推导出该电机在开路故障容错状态下的推力方程
Figure PCTCN2015094171-appb-000039
式中，λm为永磁磁链，θ为电角度θ＝∫ωdt。
或者，采用磁共能方法，由变换矩阵式(17)-式(20)推导出该电机在开路故障容错状态下的推力方程
Figure PCTCN2015094171-appb-000040
因此，根据式(22)以及式(23)或式(24)，只要在同步旋转坐标系下控制id、iq、iz就能使本发明中的五相内嵌式混合磁材料容错圆筒直线电机在故障状态下输出期望的推力。
第二部分，在第一部分的基础上，当电机发生相短路故障时，使用非故障相电流抑制短路相电流导致的推力波动。
假设A相的短路电流为isc＝Ifcos(ωt-θf)，其中，If是短路电流的幅值，θf是A相反电势和A相短路电流的夹角。
方法一：
电机绕组采用星形连接，且其中心点与直流母线电压的中心点不相连，因此，非故障相用于抑制短路故障相电流导致推力波动的补偿电流之和应为零。以短路故障相A相轴线的垂线(该垂线需经过电机绕组中心点)为轴，定义非故障相抑制短路故障相电流导致推力波动的短路补偿电流(i″B、i″C、i″D、i″E)
Figure PCTCN2015094171-appb-000041
根据非故障相用于抑制短路故障相电流导致推力波动的补偿电流幅值相等的原则，式(25)进一步定义为
Figure PCTCN2015094171-appb-000042
其中，xb、yb分别为这部分补偿电流余弦项和正弦项的幅值，ω＝πv/τ＝2πf，v直线电机动子电速度，τ为极距。
根据非故障相用于抑制故障相短路电流导致电机推力波动的补偿电流和短路故障相电流的合成磁动势为零的原则，即MMF＝Nisc+αNi″B+α2Ni″C+α3Ni″D+α4Ni″E＝0，由式(26)解得
Figure PCTCN2015094171-appb-000043
将式(27)代入式(26)求得用于抑制短路相电流导致电机推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)
Figure PCTCN2015094171-appb-000044
方法二：
假设用于抑制A相短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)
Figure PCTCN2015094171-appb-000045
其中，xb、xc、xd、xe、yb、yc、yd、ye分别为这部分补偿电流的余弦项和正弦项的幅值，ω＝πv/τ＝2πf，v直线电机动子电速度，τ为极距。
电机绕组采用星形连接，且其中心点与直流母线电压的中心点不相连，因此，
i″B+i″C+i″D+i″E＝0   (30)
根据非故障相用于抑制短路故障相电流导致电机推力波动的短路补偿电流和短路故障相电流的合成磁动势为零的原则，即MMF＝Nisc+αNi″B+α2Ni″C+α3Ni″D+α4Ni″E＝0，以及基于这部分补偿电流产生的铜耗最小原理，设计目标函数
f(xb,yb,xc,yc,xd,yd,xe,ye)＝(xb 2+yb 2)+(xc 2+yc 2)+(xd 2+yd 2)+(xe 2+ye 2)   (31)
结合以上约束条件，采用拉格朗日乘数法求解目标函数式(31)的最小值，从而求得
Figure PCTCN2015094171-appb-000046
Figure PCTCN2015094171-appb-000047
将式(32)和式(33)代入式(29)，求得用于抑制短路相电流导致电机推力波动的非故障相的短路补偿电流为
Figure PCTCN2015094171-appb-000048
因此，A相短路故障后，无论是基于非故障相补偿电流幅值相等还是基于铜耗最小原理，电机要获得和故障前一样的推力，结合式(5)和式(28)或者式(34)，其合成电流指令为
Figure PCTCN2015094171-appb-000049
第三部分，短路容错矢量控制策略
使用式(14)或式(17)推广克拉克变换矩阵T4s/2s将式(28)或式(34)变换到两相静止坐标系
Figure PCTCN2015094171-appb-000050
使用式(14)或式(17)推广克拉克变换矩阵T4s/2s将在自然坐标系下采样的剩余四相非 故障相电流(iB、iC、iD、iE)变换到两相静止坐标系上的电流(i′α、i′β、iz′)，将该电流减去式(36)，得
Figure PCTCN2015094171-appb-000051
采用式(19)派克变换矩阵C2s/2r将式(37)两相静止坐标系上的电流变换到旋转坐标系上的电流(id、iq、iz)。
或者，将在自然坐标系下采样到的剩余四相非故障相电流(iB、iC、iD、iE)减去式(28)或式(34)电流，即
Figure PCTCN2015094171-appb-000052
采用式(14)或式(17)推广克拉克变换矩阵T4s/2s和式(19)派克变换矩阵C2s/2r将式(38)电流变换到旋转坐标系上的电流(id、iq、iz)。
Figure PCTCN2015094171-appb-000053
将旋转坐标系上的电流指令
Figure PCTCN2015094171-appb-000054
和反馈电流(id、iq、iz)的差值经电流PI调节器得到旋转坐标系上的电压指令
Figure PCTCN2015094171-appb-000055
采用式(20)派克逆变换矩阵C2r/2s将该电压指令变换到两相静止坐标系上的电压指令
Figure PCTCN2015094171-appb-000056
为获得式(28)或式(34)推导出的补偿电流，根据A相短路电流和A相反电势的关系，以及短路补偿电流的数学表达方式，定义剩余四相非故障相的短路补偿电压为
Figure PCTCN2015094171-appb-000057
使用式(14)或式(17)推广克拉克变换矩阵T4s/2s将式(40)变换到两相静止坐标系，得
Figure PCTCN2015094171-appb-000058
将两相静止坐标系上的电压指令
Figure PCTCN2015094171-appb-000059
和式(41)相加，得
Figure PCTCN2015094171-appb-000060
采用式(15)或式(18)推广克拉克逆变换矩阵T2s/4s将式(42)电压指令变换到自然坐标系上的电压指令
Figure PCTCN2015094171-appb-000061
再和各相反电势相加得期望相电压指令
Figure PCTCN2015094171-appb-000062
Figure PCTCN2015094171-appb-000063
Figure PCTCN2015094171-appb-000064
或者，采用式(15)或式(18)推广克拉克逆变换矩阵T2s/4s将两相静止坐标系上的电压指令
Figure PCTCN2015094171-appb-000065
变换到自然坐标系上的电压指令
Figure PCTCN2015094171-appb-000066
Figure PCTCN2015094171-appb-000067
将式(44)电压和式(40)短路补偿电压相加，再和各相反电势相加得期望相电压指令
Figure PCTCN2015094171-appb-000068
Figure PCTCN2015094171-appb-000069
式(43)或式(45)期望相电压经电压源逆变器采用基于零序电压谐波注入的CPWM调制实现五相内嵌式混合磁材料容错圆筒直线电机A相短路故障情况下的无扰容错运行。因此本发明提出的高性能短路容错矢量控制策略如图11或图12所示。
按图10和图11或图12在Matlab/Simulink中建立图1所示五相内嵌式混合磁材料容错圆筒直线电机的控制系统仿真模型，进行系统仿真，得五相内嵌式混合磁材料容错圆筒直线电机短路故障容错矢量控制仿真结果。
图13是A相短路故障情况下电机无容错运行时的相电流波形，电流波动明显。图14是A相短路故障情况下电机无容错运行时的电磁推力波形，电机推力波动达到34N。图15为A相短路故障情况下采用本发明短路容错矢量控制策略后电机容错运行时的相电流波形，电流波动减小，和式(35)计算电流吻合。图16为A相短路故障时采用本发明容错矢量控制策略后电机容错运行时电机输出推力波形，电机短路容错运行后和故障前一样几乎没有推力波动，推力波动得到有效抑制。图17为五相内嵌式混合磁材料容错圆筒直线电机无故障运行过程中推力指令阶跃时输出推力响应，响应时间为0.6ms。图18为五相内嵌式混合磁材料容错圆筒直线电机A相短路故障后电机容错运行过程中推力指令阶跃时输出推力响应，响应时间同样为0.6ms。可见，采用图11或图12所示的本发明短路容错矢量控制策略后，电机在A相短路故障情况下，容错运行时，其动态性能和电机正常状态下一样，且输出推力没有波动，电磁推力和故障前保持一致，电流跟随性能好，实现了无扰容错运行。
若电机任何一相发生短路故障，该相和A相间隔电角度2kπ/5(k＝0、1、2、3、4，A相故障时，k＝0；B相故障时，k＝1；C相故障时，k＝2；D相故障时，k＝3；E相故障时，k＝4)，只需将自然坐标系逆时针旋转2kπ/5电角度，使故障前的A相轴线和故障相轴线重合且方向一致。然后将C2s/2r和C2r/2s中的θ用θ-2kπ/5代替，即
Figure PCTCN2015094171-appb-000070
Figure PCTCN2015094171-appb-000071
以B相故障开路为例，只需将自然坐标系逆时针旋转2π/5，即，令式(46)和式(47)中k＝1。图19和图20为使用本发明容错矢量控制策略后的五相内嵌式混合磁材料容错圆筒直线电机在A相短路故障0.06s后恢复正常，再过0.06s后B相发生短路故障，在整个过程中电机的相电流波形及输出推力波形，在短路情况下的推力和正常时的一样，几乎没有波动。可见本发明短路容错矢量控制策略适合电机五相中任何一相发生短路故障的情况，避免了复杂计算，节省了CPU开销，通用性强。
从以上所述可知，本发明内嵌式混合磁材料容错圆筒直线电机采用混合磁材料的方法，和传统内嵌式圆筒直线电机相比，节省了稀土永磁体的使用量，减小了漏磁磁通，提高了永磁体利用率，大大降低电机的制作成本，同时引入了容错齿，大大提高了圆筒直线电机的容错性能和可靠性。
本发明用于五相内嵌式混合磁材料容错圆筒直线电机的短路容错矢量控制策略在电机驱动系统允许最大电流情况下，不但能保证一相短路故障时电机输出推力和正常状态下一致，而且能明显抑制电机相短路故障后的推力波动，更为关键的是具有和故障前一样的动态性能、稳定性能和电流跟随精度，且适合任何一相发生短路故障的情况，通用性强，无需复杂计算，CPU开销小。使得其在电磁主动悬架系统等对运行可靠性要求高的系统中拥有很好的应用前景。因此，本发明在电磁主动悬架系统等对运行可靠性要求高的系统中拥有很好的应用前景。
虽然本发明已以较佳实施例公开如上，但实施例并不是用来限定本发明的。在不脱离本发明之精神和范围内，所做的任何等效变化或润饰，均属于本申请所附权利要求所限定的保护范围。

Claims (10)

  1. 一种内嵌式混合磁材料容错圆筒直线电机，其特征在于，包括初级(1)和次级(2)，初级(1)长度小于次级(2)长度，初级(1)和次级(2)之间有气隙；
    所述初级(1)包括电枢齿(3)、容错齿(4)和线圈绕组(5)；所述初级(1)均布2*m个电枢齿(3)和2*m个容错齿(4)，m为电机的相数且m≥3；电枢齿(3)和容错齿(4)交替间隔排列，初级(1)上每个电枢齿两侧的槽中都只放置一套线圈绕组(5)，而容错齿(4)上没有绕组；其中，第一个电枢齿(3)两侧的槽内和第2*m+1个电枢齿(3)两侧的槽内放置的集中绕组属于同一相，其余电枢齿(3)两侧槽内的绕组依次属于其他相；
    所述次级(2)包括导磁材料(8)和永磁体；永磁体采用内嵌方式放置在两块导磁材料之间，每一对永磁体是由稀土永磁体(6)和铁氧体(7)两种混合磁材料组成，永磁体采用轴向交替充磁方式，且稀土永磁体(6)和铁氧体(7)轴向宽度相等；每一个相同充磁方向的永磁体由一种永磁材料组成，或者每一个相同充磁方向的永磁体由两种永磁材料串联或并联组成；永磁体的极与极之间用导磁材料(8)隔离；
    所述电枢齿(3)齿宽wat和容错齿(4)齿宽wft等宽，或电枢齿(3)齿宽wat大于等于容错齿(4)齿宽wft；每一电枢齿以及容错齿上均无调制齿(9)，或者每一电枢齿以及容错齿上均设有调制齿(9)。
  2. 根据权利要求1所述的一种内嵌式混合磁材料容错圆筒直线电机，其特征在于，所述内嵌式混合磁材料容错圆筒直线电机采用分数槽结构，极槽关系满足：Ns＝2p±2或者Ns＝2p±1，Ns为初级槽数，p为次级极对数。
  3. 根据权利要求1所述的一种内嵌式混合磁材料容错圆筒直线电机，其特征在于，每一极所述永磁体的形状是一个整体圆筒、或内外两个圆筒嵌套成圆筒、或左右两个圆筒拼接成一个圆筒、或n块瓦片拼接成一个圆筒且n≥2；永磁体圆筒的壁厚小于导磁材料圆筒的壁厚，且永磁体圆筒的内径大于导磁材料圆筒的内径，永磁体圆筒的外径小于导磁材料圆筒的外径，永磁体圆筒和导磁材料圆筒同轴安装。
  4. 根据权利要求1所述的一种内嵌式混合磁材料容错圆筒直线电机，其特征在于，所述内嵌式混合磁材料容错直线电机为单边平板结构、或双边平板结构、或者圆筒型结构，该电机能够作为发电机或者电动机。
  5. 一种由权利要求1所述的内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法，其特征在于，当电机的相数m＝5时分为A、B、C、D、E五相，短路容错矢量控 制方法包括以下步骤：
    步骤1，当电机发生A相短路故障时，使用电机剩余的非故障相电流补偿短路故障相导致该相正常推力缺失，进而求取剩余的四个非故障相坐标到两相静止坐标变换的推广克拉克变换矩阵T4s/2s及其逆变换矩阵T2s/4s，定义两相静止坐标系到同步旋转坐标系的变换矩阵C2s/2r及其逆变换矩阵C2r/2s；
    步骤2，建立五相内嵌式混合磁材料容错圆筒直线电机相开路故障状态下在同步旋转坐标系上的数学模型；
    步骤3，使用非故障相电流抑制故障相短路电流导致的推力波动，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)，采用推广克拉克变换矩阵T4s/2s将电流(i″B、i″C、i″D、i″E)变换到两相静止坐标系上的短路补偿电流(i″α、i″β、i″z)；
    步骤4，采用步骤1获得的推广克拉克变换矩阵T4s/2s将在自然坐标系下采样到的剩余四相非故障相电流(iB、iC、iD、iE)变换到两相静止坐标系上的电流(i′α、i′β、i′z)，并将该电流和步骤3中获得的电流(i″α、i″β、i″z)相减得到(iα、iβ、iz)，运用派克变换矩阵C2s/2r将(iα、iβ、iz)变换到同步旋转坐标系上的电流(id、iq、iz)；
    或步骤4，将在自然坐标系上采样到的剩余四相非故障相电流(iB、iC、iD、iE)，与用于抑制短路故障相电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)相减得到(i′B、i′C、i′D、i′E)，采用推广克拉克变换矩阵T4s/2s和派克变换矩阵C2s/2r将(i′B、i′C、i′D、i′E)变换到同步旋转坐标系上的反馈电流(id、iq、iz)；
    步骤5，将同步旋转坐标系上的电流指令
    Figure PCTCN2015094171-appb-100001
    和反馈电流(id、iq、iz)的差值经电流调节器得到同步旋转坐标系上的电压指令
    Figure PCTCN2015094171-appb-100002
    采用派克逆变换矩阵C2r/2s将该电压指令变换到两相静止坐标系上的电压
    Figure PCTCN2015094171-appb-100003
    步骤6，为确保电机输出用于抑制短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)，根据A相短路电流iA＝isc和A相反电势eA的关系以及短路补偿电流的数学表达方式，定义剩余四相非故障相的短路补偿电压为(u″B、u″C、u″D、u″E) 为
    Figure PCTCN2015094171-appb-100004
    采用推广克拉克变换矩阵T4s/2s将所述补偿电压变换到两相静止坐标系上的短路补偿电压
    Figure PCTCN2015094171-appb-100005
    步骤7，将两相静止坐标系上的电压指令
    Figure PCTCN2015094171-appb-100006
    与短路补偿电压(u″α、u″β、u″z)相加得
    Figure PCTCN2015094171-appb-100007
    采用推广克拉克逆变换矩阵T2s/4s将电压指令
    Figure PCTCN2015094171-appb-100008
    变换到自然坐标系上的电压指令
    Figure PCTCN2015094171-appb-100009
    再和剩余非故障相的各相反电势分别相加得到期望相电压指令
    Figure PCTCN2015094171-appb-100010
    或步骤7，采用推广克拉克逆变换矩阵T2s/4s将两相静止坐标系下的电压指令
    Figure PCTCN2015094171-appb-100011
    Figure PCTCN2015094171-appb-100012
    变换到自然坐标系上的电压指令
    Figure PCTCN2015094171-appb-100013
    然后和剩余四相非故障相的短路补偿电压(u″B、u″C、u″D、u″E)相加，最后再和剩余非故障相的各相反电势分别相加得到期望相电压指令
    Figure PCTCN2015094171-appb-100014
    步骤8，将步骤7所得到的期望相电压指令
    Figure PCTCN2015094171-appb-100015
    经电压源逆变器，采用基于零序电压谐波注入的CPWM调制方法实现五相内嵌式混合磁材料容错圆筒直线电机一相短路故障后的无扰容错运行。
  6. 根据权利要求5所述的内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法，其特征在于，所述步骤1的具体过程为：
    步骤1.1，当电机A相发生短路故障时，假设电机A相仅发生开路故障，根据电机故障前后行波磁场幅值和速度不变以及剩余非故障相电流幅值相等的原则，由绕组星形连接且无中线和母线电压中点相连的电机相电流之和等于零的约束条件，以短路故障相A相轴线为对称轴，采用镜像对称原理优化非故障相电流，求出电机A相开路故障容错运行的相电流指令；
    步骤1.2，根据空间机电能量转换原理以及电机容错运行的电流矢量以及反电势矢量，在基波子空间选择两个正交相量基T1、T2，在零序子空间选择一个相量基Z，基波子空间和零序子空间必须正交，且零序电流需为零，因此T1、T2和Z需满足如下条件：
    Figure PCTCN2015094171-appb-100016
    由Z和T1、T2求得剩余的四个非故障相坐标到两相静止坐标变换的推广克拉克变换矩阵T4s/2s和逆变换矩阵T2s/4s
    Figure PCTCN2015094171-appb-100017
    Figure PCTCN2015094171-appb-100018
    或者，
    Figure PCTCN2015094171-appb-100019
    Figure PCTCN2015094171-appb-100020
    步骤1.3，为将基波子空间的能量转换到同步旋转坐标系，定义两相静止坐标系到同步旋转坐标系的变换矩阵C2s/2r及其逆变换矩阵C2r/2s分别为
    Figure PCTCN2015094171-appb-100021
    Figure PCTCN2015094171-appb-100022
  7. 根据权利要求5所述的内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法，其特征在于，所述步骤2的具体过程为：
    步骤2.1，由于本发明电机互感远小于自感，且自感波动的幅值很小，因此相电感可近似为常数Ls，电机相电压减去反电势后，电机开路故障后在自然坐标系上的模型表示为
    Figure PCTCN2015094171-appb-100023
    式中，uB、uC、uD和uE是电机非故障相的相电压；eB、eC、eD和eD是电机非故障相的反电势；uBe、uCe、uDe和uEe是电机非故障相相电压分别减去各相反电势后的电压；R是相电阻。
    步骤2.2，采用坐标变换矩阵T4s/2s和C2s/2r将自然坐标系上的电机开路故障模型变换到同步旋转坐标系上
    Figure PCTCN2015094171-appb-100024
    式中，ω＝πv/τ＝2πf，τ为极距，v是次级运行电速度；
    步骤2.3，采用磁共能法，由变换矩阵T4s/2s、T2s/4s、C2s/2r和C2r/2s推导出该电机在开路故障容错状态下的推力方程
    Figure PCTCN2015094171-appb-100025
    或者，
    Figure PCTCN2015094171-appb-100026
    式中，λm为永磁磁链，θ为电角度θ＝∫ωdt。
  8. 根据权利要求5所述的内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法，其特征在于，所述步骤3的具体过程为：
    步骤3.1，假设A相的短路电流为isc＝If cos(ωt-θf)，其中，If是短路电流的幅值，θf是A相反电势和该相短路电流的夹角；ω＝πv/τ＝2πf，v直线电机动子运动电速度，τ为极距；
    步骤3.2，根据非故障相用于抑制故障相短路电流导致推力波动的补偿电流幅值相等和这部分电流之和为零、以及这部分电流和短路故障相电流的合成磁动势为零的原则，以短路故障相A相轴线的垂线(该垂线需经过电机绕组中心点)为对称轴，求取用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)
    Figure PCTCN2015094171-appb-100027
    或步骤3.2，根据非故障相用于抑制故障相短路电流导致推力波动的补偿电流和短路故障相电流的合成磁动势为零、以及这部分电流之和为零的原则，基于铜耗最小原理，采用拉格朗日乘数法，求得用于抑制故障相短路电流导致推力波动的非故障相的短路补偿电流(i″B、i″C、i″D、i″E)
    Figure PCTCN2015094171-appb-100028
    步骤3.4，使用推广克拉克变换矩阵T4s/2s将用于抑制故障相短路电流导致推力波动的非故障相补偿电流(i″B、i″C、i″D、i″E)变换到两相静止坐标系上的短路补偿电流(i″α、i″β、i″z)
    Figure PCTCN2015094171-appb-100029
  9. 根据权利要求5所述的内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法，其特征在于，还包括：当其它某一相发生短路故障时，只需将自然坐标系逆时针旋转
    Figure PCTCN2015094171-appb-100030
    (k＝0、1、2、3、4；A相故障时，k＝0；B相故障时，k＝1；C相故障时，k＝2；D相故障时，k＝3；E相故障时，k＝4)电角度，使电机故障后短路故障相所在的轴线与短路故障前电机A相所在自然坐标系上的轴线重合且方向一致，此时派克变换矩阵及其逆变换矩阵分别为
    Figure PCTCN2015094171-appb-100031
    Figure PCTCN2015094171-appb-100032
  10. 根据权利要求5所述的内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法，其特征在于，所述短路容错矢量控制方法还适用于五相容错永磁旋转电机控制系统。
PCT/CN2015/094171 2015-10-14 2015-11-10 一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法 WO2017063242A1 (zh)

Priority Applications (1)

Application Number Priority Date Filing Date Title
GB1807620.8A GB2559516B (en) 2015-10-14 2015-11-10 Interior composite magnetic material fault-tolerant tubular linear motor, and short-circuit fault-tolerant vector control method thereof

Applications Claiming Priority (4)

Application Number Priority Date Filing Date Title
CN201510662900.7A CN105207446B (zh) 2015-10-14 2015-10-14 一种内嵌式混合磁材料容错圆筒直线电机
CN201510661212.9 2015-10-14
CN201510661212.9A CN105245156B (zh) 2015-10-14 2015-10-14 一种内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法
CN201510662900.7 2015-10-14

Publications (1)

Publication Number Publication Date
WO2017063242A1 true WO2017063242A1 (zh) 2017-04-20

Family

ID=58518282

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2015/094171 WO2017063242A1 (zh) 2015-10-14 2015-11-10 一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法

Country Status (2)

Country Link
GB (1) GB2559516B (zh)
WO (1) WO2017063242A1 (zh)

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN107124084A (zh) * 2017-06-09 2017-09-01 浙江理工大学 一种永磁直线同步电机非均匀混合永磁体励磁拓扑结构
CN111740577A (zh) * 2020-04-29 2020-10-02 南京理工大学 基于故障电压空间矢量重构的两电平pwm整流器容错控制方法
CN114362472A (zh) * 2021-12-28 2022-04-15 中国人民解放军海军工程大学 具备段间磁场补偿绕组的可分段拼接模块化电机及其拼接补偿方法
CN114513106A (zh) * 2022-01-24 2022-05-17 国网河南省电力公司电力科学研究院 一种基于pwm思想的永磁同步直线电机磁极结构设计方法
CN115313941A (zh) * 2022-08-22 2022-11-08 沈阳工业大学 一种基于机理数据混合模型直线电机推力波动抑制方法

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20120242173A1 (en) * 2011-03-21 2012-09-27 Automotive Research & Testing Center Linear actuating device for vehicle use
CN103779991A (zh) * 2014-01-16 2014-05-07 江苏大学 一种并联型混合磁材料电机
CN104052234A (zh) * 2014-06-10 2014-09-17 江苏大学 一种车辆电磁悬架驱动用五相圆筒型容错永磁直线作动器
CN104682820A (zh) * 2015-02-12 2015-06-03 江苏大学 一种用于五相容错永磁直线电机的容错控制方法
CN104682807A (zh) * 2015-02-04 2015-06-03 江苏大学 一种五相容错永磁电机的短路容错控制方法
CN104767353A (zh) * 2015-04-29 2015-07-08 哈尔滨工业大学 高度模块化的圆筒型多相永磁直线电机

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20120242173A1 (en) * 2011-03-21 2012-09-27 Automotive Research & Testing Center Linear actuating device for vehicle use
CN103779991A (zh) * 2014-01-16 2014-05-07 江苏大学 一种并联型混合磁材料电机
CN104052234A (zh) * 2014-06-10 2014-09-17 江苏大学 一种车辆电磁悬架驱动用五相圆筒型容错永磁直线作动器
CN104682807A (zh) * 2015-02-04 2015-06-03 江苏大学 一种五相容错永磁电机的短路容错控制方法
CN104682820A (zh) * 2015-02-12 2015-06-03 江苏大学 一种用于五相容错永磁直线电机的容错控制方法
CN104767353A (zh) * 2015-04-29 2015-07-08 哈尔滨工业大学 高度模块化的圆筒型多相永磁直线电机

Cited By (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN107124084A (zh) * 2017-06-09 2017-09-01 浙江理工大学 一种永磁直线同步电机非均匀混合永磁体励磁拓扑结构
CN107124084B (zh) * 2017-06-09 2023-07-25 浙江理工大学 一种永磁直线同步电机非均匀混合永磁体励磁拓扑结构
CN111740577A (zh) * 2020-04-29 2020-10-02 南京理工大学 基于故障电压空间矢量重构的两电平pwm整流器容错控制方法
CN111740577B (zh) * 2020-04-29 2022-08-16 南京理工大学 基于故障电压空间矢量重构的两电平pwm整流器容错控制方法
CN114362472A (zh) * 2021-12-28 2022-04-15 中国人民解放军海军工程大学 具备段间磁场补偿绕组的可分段拼接模块化电机及其拼接补偿方法
CN114513106A (zh) * 2022-01-24 2022-05-17 国网河南省电力公司电力科学研究院 一种基于pwm思想的永磁同步直线电机磁极结构设计方法
CN114513106B (zh) * 2022-01-24 2024-01-30 国网河南省电力公司电力科学研究院 一种基于pwm思想的永磁同步直线电机磁极结构设计方法
CN115313941A (zh) * 2022-08-22 2022-11-08 沈阳工业大学 一种基于机理数据混合模型直线电机推力波动抑制方法
CN115313941B (zh) * 2022-08-22 2024-04-26 沈阳工业大学 一种基于机理数据混合模型直线电机推力波动抑制方法

Also Published As

Publication number Publication date
GB2559516A8 (en) 2018-10-03
GB2559516A (en) 2018-08-08
GB201807620D0 (en) 2018-06-27
GB2559516B (en) 2022-08-31

Similar Documents

Publication Publication Date Title
US11159113B2 (en) Fault-tolerant direct thrust-force control method for five-phase permanent-magnet linear motor with one open-circuit phase fault
Zhao et al. Design and analysis of a linear permanent-magnet vernier machine with improved force density
WO2018018672A1 (zh) 五相永磁体内嵌式容错直线电机不相邻两相短路容错矢量控制方法
WO2018018671A1 (zh) 电磁悬架容错永磁游标圆筒电机及相邻两相短路容错矢量控制方法
CN105207446B (zh) 一种内嵌式混合磁材料容错圆筒直线电机
CN105245156B (zh) 一种内嵌式混合磁材料容错圆筒直线电机短路容错矢量控制方法
Xu et al. Pole optimization and thrust ripple suppression of new Halbach consequent-pole PMLSM for ropeless elevator propulsion
WO2017063242A1 (zh) 一种内嵌式混合磁材料容错圆筒直线电机及其短路容错矢量控制方法
CN106026839B (zh) 容错永磁游标圆筒电机相邻两相短路容错矢量控制方法
Ebadpour et al. Modeling and synchronized control of dual parallel brushless direct current motors with single inverter
Hua et al. Comparative study of high torque density spoke-type pm in-wheel motors for special vehicle traction applications
Zhao et al. Comparative study of modular-stator and conventional outer-rotor flux-switching permanent-magnet motors
Cai et al. Analysis of synergistic stator permanent magnet machine with the synergies of flux-switching and flux-reversal effects
WO2023284283A1 (zh) 广域运行的交替极型永磁辅助同步磁阻电机
CN106100496B (zh) 五相永磁体内嵌式容错直线电机相邻两相故障容错矢量控制方法
Yu et al. Dual three-phase flux-modulated switched reluctance motor drive with maximum torque per ampere strategy
Zhao et al. A novel separated hybrid-magnetic-circuit variable flux memory machine
Zhang et al. Rediscovery of permanent magnet flux-switching machines applied in EV/HEVs: Summary of new topologies and control strategies
CN106059442B (zh) 五相永磁容错直线电机相邻两相开路容错矢量控制方法
CN103337993B (zh) 一种正弦波电感的开关磁阻电机控制方法
CN115395854A (zh) 一种基于主动式无位置传感器策略的永磁容错轮毂电机及其驱动、设计方法
Bian et al. Mechanism investigation of ring type winding in linear permanent magnet vernier machine for improving force density
Qu et al. Milestones, hotspots and trends in the development of electric machines
Liu et al. Terminal voltage oriented control of excitation winding for new AC-excited hybrid excitation generator
CN110289744B (zh) 单绕组宽转子齿结构无轴承开关磁阻电机及发电控制方法

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 15906124

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

ENP Entry into the national phase

Ref document number: 201807620

Country of ref document: GB

Kind code of ref document: A

Free format text: PCT FILING DATE = 20151110

WWE Wipo information: entry into national phase

Ref document number: 1807620.8

Country of ref document: GB

122 Ep: pct application non-entry in european phase

Ref document number: 15906124

Country of ref document: EP

Kind code of ref document: A1