GB2559516A - Interior composite magnetic material fault-tolerant tubular linear motor, and short-circuit fault-tolerant vector control method thereof - Google Patents

Abstract

An interior composite magnetic material fault-tolerant tubular linear motor, and short-circuit fault-tolerant vector control method thereof. The method comprises: establishing a five-phase interior composite magnetic material fault-tolerant tubular linear motor model; using a non-failure phase current of the motor to perform compensation for a short-circuit failure phase resulting in a normal thrust deficiency in the phase, and to suppress thrust fluctuation; and obtaining, by employing a series of coordinate transformations and voltage feed forward compensation policies, a desired phase voltage, and employing a CPWM modulation method based on zero-sequence voltage harmonic injection to realize fault-tolerant vector control after a phase short-circuit failure of the motor. The method can suppress motor thrust fluctuation when the motor is in phase short-circuit fault-tolerant operation, and more importantly, the method has the advantages of dynamic state and steady state performances consistent with those in a normal state, a constant switching frequency of a voltage source inverter, and a small CPU overhead. Moreover, upon a short-circuit failure of any phase, the natural coordinate system only has to be rotated counterclockwise by a certain angle to realize fault-tolerant operation of the motor.

Description

(56) Documents Cited:

CN 104767353 A CN 104682807 A CN 103779991 A (58) Field of Search:

INT CL H02K, H02P

H02P 25/064 (2016.01)

CN 104682820 A CN 104052234 A US 20120242173 A1 (86) International Application Data:

PCT/CN2015/094171 Zh 10.11.2015 (87) International Publication Data:

WO2017/063242 Zh 20.04.2017 (71) Applicant(s):

Jiangsu University

301 Xuefu Road Zhenjiang, Jiangsu 212013, China (72) Inventor(s):

HuawerZhou Zhen Lu Jinghua Ji Xiaoyong Zhu Wenxiang Zhao Guohai Liu Long Chen Qian Chen (continued on next page) (54) Title of the Invention: Interior composite magnetic material fault-tolerant tubular linear motor, and shortcircuit fault-tolerant vector control method thereof

Abstract Title: Interior composite magnetic material fault-tolerant tubular linear motor, and short-circuit faulttolerant vector control method thereof (57) An interior composite magnetic material fault-tolerant tubular linear motor, and short-circuit fault-tolerant vector control method thereof. The method comprises: establishing a five-phase interior composite magnetic material fault-tolerant tubular linear motor model; using a non-failure phase current of the motor to perform compensation for a short-circuit failure phase resulting in a normal thrust deficiency in the phase, and to suppress thrust fluctuation; and obtaining, by employing a series of coordinate transformations and voltage feed forward compensation policies, a desired phase voltage, and employing a CPWM modulation method based on zero-sequence voltage harmonic injection to realize fault-tolerant vector control after a phase short-circuit failure of the motor. The method can suppress motor thrust fluctuation when the motor is in phase short-circuit fault-tolerant operation, and more importantly, the method has the advantages of dynamic state and steady state performances consistent with those in a normal state, a constant switching frequency of a voltage source inverter, and a small CPU overhead.

Moreover, upon a short-circuit failure of any phase, the natural coordinate system only has to be rotated counterclockwise by a certain angle to realize fault-tolerant operation of the motor.

l?l II

This international application has entered the national phase early

GB 2559516 A continuation (74) Agent and/or Address for Service:

Chapman IP

Kings Park House, 22 Kings Park Road, Southampton, Hampshire, SO15 2AT, United Kingdom

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INTERIOR COMPOSITE MAGNETIC MATERIAL FAULT-TOLERANT

TUBULAR LINEAR MOTOR, AND SHORT-CIRCUIT FAULT-TOLERANT

VECTOR CONTROL METHOD THEREOF

TECHNICAL FIELD

The invention relates to a new type of motor and fault-tolerant control method for short-circuit phase fault. Especially, it relates to the five-phase fault-tolerant permanent magnet (PM) linear motor and its FT-FOC method in short-circuit phase fault. The invention has good application prospect where it requires high reliability and good dynamic performance, such as electromagnetic active suspension system, electric vehicle and aerospace field.

BACKGROUND ART

With the development of the society and improvement of the living standard, people are taking more care of vehicle riding comfort and handing stability. Because the performance of suspension system have great influence on riding stability and reliability, the research on active suspension system is drawn much more attention. As the core component of the suspension system, the research on tubular linear motor has been paid more attention. In addition, the fault-tolerant capability of the motor directly determines the reliability of electromagnetic suspension.

The fault-tolerant motor is that primary (stator) phase windings are isolated each other in electric/magnetic/thermal by changing the winding mode and primary tooth structure. Then, the motor can inherently offer the fault-tolerance. The article, “magnetic field of a tubular linear motor with special permanent magnet, IEEE Transactions on plasma science 39(1): 83-86, 2011”, proposed an interior PM motor without fault-tolerance. That is to say, when a fault occurs in some coils, the motor could not operate properly. The application number 201010120847.5 of the China National Invention Patent - “a fault-tolerant permanent magnet linear motor” publishes a three phase fault-tolerant flux-reversal one-side flat linear motor. Although the introduction of the fault tolerant teeth (FTT) to offer fault-tolerance, the magnetic flux leakages of the motor are serious. There are two pieces surface-mounted PMs, which have different magnetization direction, on the armature teeth. A large number of flux lines form return circuits without across the secondary (mover) yoke, hence it results in serious flux leakages. In addition, the mechanical strength between the PMs and armature teeth is weak, and the force density of the motor is low due to the SPMs. Moreover, the constantly increasing use of expensive rare earth PM on both domestic and abroad in recent years motor design process should focus on that how to reduce the use of rare earth PM and the motor cost. Ferrite has many advantages such as low prices, rich production, stable supply and so on. That is to say, ferrite is a good substitute for rare earth PM.

Although the fault-tolerant motor still has a certain thrust capacity when a fault occurs in some coils, the motor thrust ripple and noise are large, which seriously affect the system performances. The goal of fault-tolerant control is to optimize the fault-tolerant currents for different applications, and to make the output thrust of the motor smooth as far as possible in fault condition and to have almost the same performance as that in healthy condition. The application number 201510059387.2 of the China National Invention Patent -“A short-circuit fault-tolerant control method for five-phase fault-tolerant PM motor”, is aimed at five-phase fault-tolerant SPM motor, the impact on motor torque caused by short-circuit phase fault is decomposed into two parts. The one is the impact on torque caused by open-circuit phase fault, and the other is the impact on torque caused by short-circuit current. For open-circuit fault, based on the principles of equal magnetomotive force (MMF) pre- and post-fault, and the same remained healthy current amplitude in post-fault condition, the remained phase currents can be optimized.. For short-circuit current, based on the principle that MMF of the compensation currents and short-circuit current is zero in post-fault and the minimum copper loss principle, the compensation currents can be worked out. Then, adding the two parts of the currents to work out the remained healthy phase currents. The method can restrain the torque ripple caused by the short circuit current. However, the healthy phase compensation current amplitude, which is not associated with motor speed, is constant. In addition, the compensation current of the healthy phase is not zero. It is worth mentioning that, the short-circuit fault-tolerant currents are presented only and Maxwell simulated results is used to verify. However, it doesn't mention to use what kind of control strategy. Now, common method of fault-tolerant control can be divided into two steps: firstly, the fault-tolerant currents are calculated; secondly, the current hysteresis PWM control strategy is adopted. However, this method has some problems, such as scrambled switching frequency, high noise and poor dynamic performance. Therefore, the method is not suitable for large power and high dynamic performance of motor.

SUMMARY OF THE INVENTION

Aimed at the shortcomings of the existing linear motor technology and based on the original mover-permanent-magnet linear motor structure, this invention proposes an interior PM linear motor which can save the rare earth PM material and have good faulttolerant performance. The motor can reduce the amount of rare earth PMs and reduce motor cost. At the same time, the motor not only maintains the advantages of the traditional interior PM motor, but also improves the fault-tolerant performance of the motor.

Aimed at the existing shortcomings of fault tolerant control technology for the motor, and based on the characteristics of the proposed motor and the features of short-circuit phase fault, this invention proposes a high-performance FT-FOC method for the described motor in short-circuit phase fault. It can overcome many disadvantages that produced by using current hysteresis control method in the existing fault-tolerant strategy, such as the disorder inverter switching frequency, low motor response speed, poor dynamic performance, inaccurate currents following, severely noise and so on. Also, the proposed control method avoids complication of the existing fault-tolerant control methods. The proposed method can make the motor system has high faulttolerant performance, high dynamic performances, good current following performance, and low CPU usage under fault condition. In addition, it has many advantages, such as, constant inverter switching frequency, decreased noise, and convenient in the design of electromagnetic compatibility. Finally, the reliability and dynamic performance of the proposed five-phase IPM-FTTL motor with hybrid magnetic material under shortcircuit phase fault condition are improved.

The five-phase interior PM fault-tolerant tubular linear (IPM-FTTL) motor with hybrid magnetic material includes primary and secondary. The length of primary is shorter than secondary. There is air gap between primary and secondary. The described primary contains armature teeth, fault-tolerant teeth (FTT) and windings. The number of the armature teeth and FTT are 2m, respectively, and m is the phase number of the machine and m>3. The FFT are interleaved with the armature teeth, and the armature teeth and FTT are staggered arrangement. Only a single-layer winding is in one slot and the slots on both sides of armature teeth contain coil turns from a single phase. That is to say, the windings in the first slot and the number (2m+l) th slot belongs to one phase, and so on.

The described secondary includes permeability magnetic materials and PMs. The interior PMs are placed between two pieces of permeability magnetic materials, and then the permeability magnetic materials can isolate the each pole of PMs. Each pair of PMs is consists of rare earth PMs and ferrite magnetic materials, and both of them are axial magnetization alternatively. In addition, the axial width of rare earth PMs and ferrite are equal. Pieces of PM material with same magnetization direction consist of one PM material or two kinds of PM material connected with series or parallel.

The width of described the armature teeth w_at and FTT Wf_t are equal, but the width of the armature teeth w_at can also be greater than the FTT Wf_t sometimes. Both in the armature teeth and FTT have no modulation teeth, or have some modulation teeth.

The described IPM-FTTL motor with hybrid magnetic material employs fractional-slot construction. The relationship between the number of primary slots and the secondary pole pairs should meet the condition of N_s=2p±2 or N_s=2p±l, where N_s is the number of primary slots, and p is the secondary pole pairs.

The shape of the described each pole PM can be one whole tubular or two level of nesting tubulars or connection of two tubulars or spliced into a tubular by n pieces of tiles, n>2. The radial width of PM is shorter than the width of permeability magnetic materials. In addition, the inner diameter of the PM is greater than the inner diameter of the permeability magnetic material and the outer diameter of the PM is shorter than the outer diameter of permeability magnetic materials. It is worth being mentioned that the PM tubulars and the permeability magnetic material tubulars are mounted coaxially.

The described IPM-FTTL motor with hybrid magnetic material can be the oneside flat structure or double-side flat structure or tubular structure. In addition, the machine can be used as generator or motor.

The following is the control scheme of the invention:

The proposed motor has five phase such as phase-A, phase-B, phase-C, phase-D, and phase-E. The FT-FOC method for the proposed motor in short-circuit phase fault includes the following steps:

(1) The remained healthy phase currents are used to compensate for the lack of normal force caused by short-circuit phase fault. According to the remained healthy phase currents, the generalized Clark transformation matrix Γ_4ί/2ί and its inverse transformation matrix /]_s/4s can be obtained, with which the remained four healthy phases in the natural stationary frame can be transformed into the stationary frame. Meanwhile, the generalized Park transformation matrix C_2i/2r and its inverse transformation matrix C_2j._/2, are defined.

(2) Establishing the motor mathematical model in the synchronous rotating frame under the open-circuit phase fault condition.

(3) The remained healthy phase compensation currents are used to restrain thrust ripple caused by short-circuit fault phase current. Then, the healthy phase compensation currents ,/' ,/' ,/' ) can be obtained,. Finally, the compensation currents (/'' ,z , ί'ή , ) are transformed to the compensation currents (/, ί_β , i ) in stationary frame by the generalized Clark transformation matrix Γ_4ί/2ί.

(4) The remained four healthy phase currents (z_B, i_c , i_D, i_E ) can be transformed to the currents (ζ,ζ^ ,zj) in stationary frame by the generalized Clark transformation matrix Γ_4ί/2ϊ, then, the currents (i_a , ΐ_β , i_z) can be obtained by using the current (i'_a , ΐ'_β , i'_z) to subtract the current (/, z, i ) worked out in step (3) , finally, the currents (z_a, ΐ_β , i_z) can be transformed to the currents (i_d > i > i_z) in the synchronous rotating frame by using the Clark transformation matrix C_2i/2r.

(4’) The currents (i'_B ,i'_c ,i'_D ,i_E ) can be obtained by using the remained four healthy phase currents (i_B, i_c , i_D, i_E ) to minus healthy phase compensation currents (i_B z , i_D , i_E ), then, by using the generalized Clark transformation matrix 7]_s._/2s. and

Park transformation matrix C_2s/2r, the currents (i'_B , i'_c , i'_D , i'_E ) can be transformed to the currents (z_rf > i > z'_z) in synchronous rotating frame.

(5) By using the current regulators, the voltage references (u_d , u*, zz*) can be obtained with the differences between the current references (i*_d , i*, i*) and the feedback currents (z_rf, i, i_z) in synchronous rotating frame. Then, the voltage references (zz*, zz*, zz* ) can be transformed into the stationary frame by using the Park inverse transformation matrix C_2r/2s. Thus, the voltage references (zz*, zz* , zz*) in stationary frame are obtained.

(6) To make the remained healthy phases of the described motor generate the compensation currents (i_B , z , i_E , i_E ), which are used to suppress thrust ripple caused by short-circuit current. Based on the mathematical expression of phase-A short-circuit current, phase-A back electromotive force (EMF) and short-circuit compensation current, the compensation voltages of the remained healthy phases (zz ,zz ,zz ,zz ) can be defined as:

u” = 0.44Te_A zzg = -0.447^ ^< zz = -0.447^

X = 0.447e,

Based on generalized Clark transformation matrix T_As/2s , the compensation voltages in the natural stationary frame can be transformed into the stationary frame, thus the compensation voltages in the stationary frame are:

u”_a = 0.3998^ < u”p = 0 u” = 0 (7) Making the voltage references (u*_a,u*_p ,u*) add compensation voltages (w, u'p , u) in stationary frame, it can be obtained:

^u*a =^u*_a + 0.3998^ < Up =Up //. = //

Then, the voltage references ( u_B , u_c , u_D , u*_E ) can be obtained by using the generalized Clark inverse transformation matrix Γ_2ί/4ί to transform the voltage references (u*_a*,u**,u*) into the natural stationary frame, then adding the phase back EMFs, the expect phase voltage references (u* , , u* , u*) are obtained.

(7’) The voltage references (u_B ,u_c ,u*_D,u_E) can be obtained by using the generalized Clark transformation matrix 7_4j/2j to transform the voltage references ( u*_a Up ,u*) into the voltages (u*_B,u*_c , //ζ, w* ) in the natural coordinate system. Then, adding the remained healthy phase compensation voltages (u_B > u_c , u_D , u_E ), the expect phase voltage references (u* , , u* , u*) are obtained.

(8) The expect phase voltage references (u*, // , u*, u*) are transmitted to voltage source inverter, which is adopted CPWM technique based on zero-sequence voltage harmonic injection, then the FT-FOC for the described five-phase IPM-FTTL motor with hybrid magnetic material is realized when short-circuit phase fault occurs.

The invention has the following beneficial effects:

(1) The proposed motor in the invention adopts spoke-type PMs, which are embedded in the mover. This construction has many advantages such as simple structure, high reliability (the PMs will not be broken off from the mover), high force ability, high efficiency, wide constant power range and wide flux-weakening speed control range.

(2) This motor adopts single layer concentrated and discoid windings which make the motor have no end winding and be convenient to wind the windings. In addition, it can reduce the motor resistance and copper loss and can improve motor efficiency. Only a single-layer winding is in one slot and the slots on both sides of armature teeth contain coil turns from a single phase. That is to say, the FTT have no winding. Thus, the adjacent phase windings of the primary are essentially isolated, leading to the merit of phase decoupling and electric/magnetic/thermal isolation. This merit is highly desirable to enable fault-tolerant operation, which make the motor have a good application prospect on reliability requirement areas such as the suspension system. It can increase the slot area and improve space utilization by reduce the motor FTT width. Additionally, the combination of the concentrated winding and interior PM has many advantages such as compact motor structure, small motor size, high power density, high force density and so on. The combination of the interior PMs and FTT solves the problem that the fault tolerant linear motor has low force ability. It can further improve the thrust density of motor at low speed when the motor has some modulation teeth in the armature teeth and FTT.

(3) The secondary adopts the hybrid magnetic material, that is to say, the motor uses the ferrite to replace some rare earth PMs. It can not only greatly reduce the use of rare earth PMs and the cost of the motor, but also greatly reduce the eddy current loss caused by lower magnetic energy product and improve the motor efficiency. The PMs are axial magnetized alternatively, and are mounted with the permeability magnetic materials in the mover alternatively. In addition, the radial width of PMs is less than the permeability magnetic materials. The inner diameter of the PMs is greater than that of the permeability magnetic material and the outer diameter of the PMs is less than that of the permeability magnetic materials. It is worth being mentioned that the PM tubular and the permeability magnetic material tubular are mounted coaxially. Thus, it greatly reduces the magnetic flux leakage between the adjacent N pole and S pole of PMs, and improves the utilization rate of the PMs. The combination of the embedded structure , hybrid magnetic material, the dimension relation between PM tubular and permeability magnetic material tubular and their installation can greatly decrease magnetic flux leakage, reduce the eddy current loss, improve the utilization rate of the PMs, reduce the motor manufacturing costs, improve the motor efficiency, reduce the motor volume and increase the thrust density. The combination of the interior PMs, the dimension relationship between PMs and permeability magnetic materials, and fault-tolerance make the motor have many advantages such as low cost, high efficiency, high fault tolerance performance, high reliability, high thrust density and wide speed range.

(4) On the premise of ensuring the motor output constant thrust before and after one phase short-circuit fault, the invention not only can effectively restrain the motor thrust ripple, but also can even make the motor dynamic performance and current tracking performance under fault-tolerant condition in agreement with that under healthy condition. In addition, it also has following advantages: constant voltage source inverter switching frequency, low noise and low CPU overhead. The algorithm has certain universal property, that is to say, making the natural stationary frame counterclockwise a certain angle to ensure the motor operate in fault-tolerant condition when one phase is short-circuit fault.

(5) When a short-circuit fault occurs in phase-A, the invented FT-FOC method makes the proposed motor have the same dynamic performance and steady-state performance as that in healthy condition, and have almost no thrust ripple. In addition, when the current is less than the maximum current limit, it can make the electromagnetic thrust accord with that in healthy condtion. Therefore, it achieves faulttolerant operation.

(6) By using Clark transformation matrix, which is deduced from the remained healthy phase current vector, and its inverse transformation matrix and defined Park transformation matrix, the remained healthy phase voltage, current, resistance and inductance can be transformed with equal amplitude into the synchronous rotating frame under fault condition. It can also transform the variables in the synchronous rotating frame into the natural stationary frame. Meanwhile, it can extract the zero sequence variables to reduce the motor loss and restrain the thrust ripple. By using Clark transformation matrix, which is deduced from the remained healthy phase back-EMF vector, and its inverse transformation matrix and defined Park transformation matrix, the remained healthy phase voltage, current, resistance and inductance can be transformed with amplitude into the synchronous rotating frame under fault condition, and transform the variables from the synchronous rotating frame into the natural stationary frame, however, it can’t extract the zero sequence variables by transformation. Base on the sum of the MMF of the short-circuit current and the remained healthy phase compensation currents, which is zero, it can obtained the same compensation currents with the principle of equal short-circuit compensation current amplitude or the principle of minimum copper loss, which makes each phase have the same copper loss. At the same time, it can improve the motor reliability. The combination of the short-circuit compensation currents and the generalized Clark transformation matrix reduces the CPU overhead.

(7) The combination of the generalized Clark transformation matrix and Park transformation matrix achieves the transformation of the remained healthy phase from the natural stationary frame into the synchronous rotating frame. It creates prerequisites of the FT-FOC for the motor under short-circuit phase fault condition. The other, it can realize the control of the degree of freedom in the zero sequence space under fault condition, reduce the motor copper loss and iron loss, improve the motor efficiency, and suppress the motor thrust ripple and copper loss caused by the zero sequence current. The combination of extraction of short-circuit compensation current, the shortcircuit compensate voltage, the generalized Clark transformation matrix, the Park transformation matrix and CPWM modulation based on zero sequence voltage harmonic injection realize decoupling of the thrust and flux linkage under short-circuit phase fault station in the synchronous rotating frame, increase the utilization ratio of the inverter bus voltage, and reduce the complexity of FT-FOC algorithm. Thus, the motor has a good fault tolerant ability. It can improve the current tracking performance, dynamic performance and steady-state performance of the motor under fault condition and make the motor have the same dynamic performance and steady-state performance as that in healthy condition. The combination of FT-FOC strategy in short-circuit fault, CPWM modulation based on zero sequence voltage harmonic injection, and the five10 phase IPM-FTTL motor with hybrid magnetic material greatly improves the faulttolerant performance, dynamic performance and steady-state performance, increases the upper limit of motor speed, saves the CPU overhead, reduces the noise, reduces the design difficulty of electromagnetic compatibility in short-circuit phase fault condition. In addition, it also has many other advantages such as high control precision, good current tracking performance, and the high motor efficiency, fast response of thrust force and low thrust ripple. Thus, the system is suitable for application, where requires high electromagnetic compatibility, good dynamic performance, stable performance, precise control, reliability, and fault tolerance.

(8) The invention can effectively overcome many shortcomings such as the disorderly inverter switching frequency caused by traditional current hysteresis control, decreasing response speed, poor current following performance, serious noise, and design difficulty of electromagnetic compatibility. When the motor is with short-circuit phase fault, the current can be accurately followed with FT-FOC method. In addition, the steady performance and dynamic performance are better than the performance of current hysteresis control. Thus, the motor system has high fault-tolerance and high dynamic performance under short-circuit fault condition.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention are described in relation to the following schematic drawings in which:

Fig. 1 shows the structure diagram of five-phase IPM-FTTL motor with hybrid magnetic material of the present invention.

Fig. 2 shows the structure diagram of five-phase interior PM fault-tolerant one-side flat linear motor with hybrid magnetic material of the present invention.

Fig. 3 shows the structure diagram of five-phase interior PM fault-tolerant two-side flat linear motor with hybrid magnetic material of the present invention.

Fig. 4 shows the winding distribution diagram of five-phase IPM-FTTL motor with hybrid magnetic material of the present invention.

Fig. 5 shows four different structures of hybrid magnetic materials of IPM-FTTL motor of the present invention.

Fig. 6 shows four different structures of hybrid magnetic materials of IPM-FTTL motor when both the armature teeth and FTT have modulation teeth of the present invention.

Fig. 7 shows the back-EMF waveform of IPM-FTTL motor with all rare earth PM or hybrid magnetic material alternatively of the present invention.

Fig. 8 shows phase-B armature reaction magnetic field distribution of IPM-FTTL motor with hybrid magnetic material of the present invention.

Fig. 9 shows phase-B inductance waveform of IPM-FTTL motor with hybrid magnetic material of the present invention.

Fig. 10 shows principle diagram of field-oriented control method adopted CPWM technique based on zero sequence voltage harmonic injection for the IPM-FTTL motor with hybrid magnetic material of the present invention.

Fig. 11 shows one principle diagram of FT-FOC method for IPM-FTTL motor with hybrid magnetic material in short-circuit phase fault of the present invention.

Fig. 12 shows the other principle diagram of FT-FOC method for the IPM-FTTL motor with hybrid magnetic material in short-circuit phase fault of the present invention.

Fig. 13 shows phase current waveform of the IPM-FTTL motor without fault-tolerant operation when phase-A is with short-circuit of the present invention.

Fig. 14 shows the force waveform of the IPM-FTTL motor without fault-tolerant operation when phase-A is with short-circuit of the present invention.

Fig. 15 shows phase current waveform of the IPM-FTTL motor with FT-FOC method when phase-A is with short-circuit of the present invention.

Fig. 16 shows the force waveform of the IPM-FTTL motor with FT-FOC method when phase-A is with short-circuit of the present invention.

Fig. 17 shows the force waveform of the IPM-FTTL motor in healthy when the force reference steps of the present invention.

Fig. 18 shows the force waveform of the IPM-FTTL motor with FT-FOC under shortcircuit fault condition when the force reference steps of the present invention.

Fig. 19 shows phase current waveform ofthe IPM-FTTL motor with FT-FOC method when phase-A is recovered from short-circuit fault and phase-B is with short-circuit fault of the present invention.

Fig. 20 shows the force waveform of the IPM-FTTL motor with FT-FOC method when phase-A is recovered from short-circuit fault and phase-B is with short-circuit fault of the present invention.

In the Figures, 1 represents primary, 2 represents secondary, 3 represents armature teeth, 4 represents FTT, 5 represents winding, 6 represents rare earth PM, 7 represents ferrite PM, 8 represents permeability magnetic materials, and 9 represents modulation teeth.

DETAILED DESCRIPTION

The following is detailed description about technical solutions of the present invention.

In order to illustrate the structure characteristics and beneficial effects of the fivephase IPM-FTTL motor more simply and clearly, a specific five-phase IPM-FTTL motor with hybrid magnetic material and its FT-FOC method in short-circuit fault is stated in the following.

In order to state this invention more clearly, we choose the number of the motor phase is m=5. Thus, the number of the armature teeth and FTT are 10 respectively. In addition, the number of slots is N_s=20 and the number of PM pole pairs is p=9. In general, the width of the armature teeth Wat and the width of FTT Wft is equal. Sometimes we can reduce the width of the FTT appropriately when consider expanding the area of slot, namely, W.-n>Wn. In the description, the width of the armature teeth and FTT is equal, namely, W_at=Wft. Meantime, the ratio of the axial width of PM W_pm and the axial width of secondary teeth W_st are chosen as W_pm/Wst=1.2 and the ratio of the primary slot width W_ss and the primary teeth width W_at> Wft ( W_at=Wft) is Wss/W_at=1.72. In addition, the ratio of the primary pole pitch T_s and the pole pitch T_r can be chosen as T_r/T_s=2.2.

The topology of the IPM-FTTL motor with hybrid magnetic material can be tubular or flat. The motor can be one-side flat structure or double-side flat structure. In addition, the machine can be used as generator or motor.

Fig. 1 shows the structure diagram of five-phase IPM-FTTL motor with hybrid magnetic material of the present invention. The motor includes primary and secondary. The primary includes armature teeth, FTT and windings. The both number of the armature teeth and FTT are 10. Each pair of PMs consists of rare earth PMs and ferrite magnetic materials. There is air gap between the primary and secondary. The primary armature and secondary iron core consist of cheap permeability magnetic materials such as electric iron, silicon steel, soft magnetic materials (such as permalloy) and so on. In the present invention, the silicon steel is adopted. Additionally, Fig. 2 shows the structure diagram of the five-phase interior PM fault-tolerant one-side flat linear motor with hybrid magnetic material of the present invention. The ratio of the primary pole pitch T_s and the pole pitch T_r is chosen as 7//7/=2.2. Fig. 3 shows the structure diagram of the five-phase interior PM fault-tolerant both-side flat linear motor with hybrid magnetic material of the present invention.

Fig. 4 shows the winding distribution diagram of five-phase IPM-FTTL motor with hybrid magnetic material of the present invention. This motor adopts single layer concentrated windings. The disc coil windings are placed in the slots on both sides of armature teeth, and the coil windings in the slots are phase-Al, phase-Cl, phase-El, phase-Bl, phase-Dl, phase-A2, phase-C2, phase-E2, phase-B2, phase-D2, in turn. The direction of winding is same. Phase-Acan be obtained by connection of phase-Al and phase-A2 in series or parallel, and other phase can be obtained in the same way.

The interior PMs are placed between two pieces of permeability magnetic materials. The shape of the PMs under each pole can be one whole tubular or two level of nesting tubulars or connection of two tubulars or spliced into a tubular by n pieces of tiles, n>2. Each pair of PMs is consists of rare earth PM materials and ferrite magnetic materials, and all of them are axial magnetization alternately. The usage of ferrite magnetic materials can decrease the motor cost greatly. In addition, the axial width of rare earth PMs and ferrite magnetic materials is equal. Each magnetization direction of PM consists of one PM material or two kinds of PM material connection in series or parallel. The inner diameter of the PM tubular is greater than the inner diameter of the permeability magnetic material tubular and the outer diameter of the PM tubular is shorter than the outer diameter of permeability magnetic material tubular, and the PM tubulars and the permeability magnetic material tubulars are mounted coaxially. The rare earth PM is NdFeB in the present invention. Fig. 5 shows four different structures of hybrid magnetic materials of the motor in the present invention. In Fig. 5(a), the NdFeB and ferrite magnetic materials are parallel nesting in the outside and inside of tubular. It can be seen that each pole of PM consists of NdFeB and ferrite connection in series in Fig. 5(b) and Fig. 5(c). The difference is the PM materials on the both sides of the permeability magnetic material tubular are same in Fig. 5(b), while they are different in Fig. 5(c). It can be seen from Fig. 5(d) that the NdFeB and ferrite are used alternatively and the materials of the same magnetized direction PMs are same, the magnetized direction of NdFeB and ferrite is different. The performance of the structure shown in Fig. 5(d) is only described in the present invention. In Fig. 5(d), the axial width of rare earth PMs and ferrite is equal and each pole PMs are composed of the NdFeB and ferrite, the shape of each pole PMs is one whole tubular. The thickness of the PM tubular is 0.75 times of the thickness of the permeability magnetic material tubular. The inner diameter of the PM tubular is greater than the inner diameter of the permeability magnetic material tubular. It is worth being mentioned that the PM tubular and the permeability magnetic material tubular are mounted coaxially. Fig. 6 shows four different structures of hybrid magnetic materials of the present invention when both the armature teeth and FTT have modulation teeth.

Fig. 7 shows the back-EMF waveform of IPM-FTTL motor with all rare earth PM or hybrid magnetic material alternatively of the present invention. It can be seen that the usage of rare earth PM is reduced by 50% with the back-EMF decreased by 26%. The results are acceptable and satisfactory to save the motor cost. In addition, the sinusoidal and symmetrical feasibility ofbackEMF waveforms is easy to AC drive. Fig. 8 shows phase-B armature reaction magnetic field distribution of IPM-FTTL motor with hybrid magnetic material of the present invention. Due to the introduction of FTT in the primary, the fluxes mainly go through the adjacent FTT, rather than through the adjacent armature teeth. Namely, the magnetic fields of two adjacent phases are independent from each other, which is due to the FTT decouple the motor phases. Thus, the adjacent phase windings of the primary are essentially isolated, leading to the merit of phase decoupling. Fig. 9 shows the inductance waveform of IPM-FTTL motor with hybrid magnetic material of the present invention. It can be seen that the ratio of mutualinductance to self-inductance is only 1.0%, which indicates the motor is essentially phases decoupling. In addition, the motor has good fault-tolerant capability and low self-inductance ripple. That is to say, the phase inductance is constant approximately.

The CPWM method based on zero sequence harmonic voltage co=-(max(w,)+min(wz))/2 injection has the same flux control effect as the five-phase SVPWM method. Thus, the invention adopts the CPWM method based on zero sequence harmonic voltage injection to modulate the pulse width.

Fig. 10 shows the principle diagram of field-oriented control adopted CPWM technique based on zero sequence voltage harmonic injection for the IPM-FTTL motor with hybrid magnetic material of the present invention. When the motor works under healthy and steady conditions, each phase current can be expressed as:

z* = -z* sin(0) + i*_d cos(0) z* = -z* sin(0 - 2π / 5) + z* cos(0 - 2π / 5) < 4 =-z* sin(0-4π/5) + z* cos(0-4π/5) (1) ζ'ζ ⁼ sin(0 -6π 15)+z* cos(0 - 6π 15) z* = -z* sin(0 - 8 π / 5) + z* cos(0 - 8 π / 5) where i*_d > i* are rZ-axis current reference and i/-axis current reference in the synchronous rotating frame, respectively.

The travelling-wave MMF of the motor can be expressed as:

E

MMF = Y MMF h (2) = Ni_A +aNi_B +a²Nic +a³NiD + <ΕΝί_Ε where a = , and N is the effective number of the each primary phase winding turns.

Part One

Phase-A open-circuit is supposed when it is with short-circuit fault. Firstly, the remained healthy phase currents are used to compensate for the lack of force caused by open-circuit phase fault. Meantime, the phase-A current is supposed as zero. The travelling-wave MMF, which is produced by the remained four healthy phase winding, can be expressed as:

MMF = ^MMF_t i=B (3) =αΝζ + a ²Ni*c + a ³Ni*D + a ^Νζ.

To realize fault-tolerant operation under open-circuit phase fault condition, the traveling-wave MMF should be constant before and after fault. Thus, the remained healthy phase currents need to be adjusted to keep the traveling-wave MMF amplitude and speed be constant pre- and post-fault. Thus, the real part and imaginary part of equation (1) and equation (2) should be equal, respectively.

The windings are star-connected, and its neutral point is not connected with the dc bus-voltage midpoint. Thus, the sum of phase currents is zero. According to the mirror symmetry about the fault phase-A, the remained healthy phase currents can be assumed as:

a (0)=4(-0) < A (0) = A (-^) (4)

A +A +A + A - ⁰

According to the aforementioned constraint requirements and the condition that the amplitude of the healthy phase current is equal, the remained phase current references under fault-tolerant condition can be achieved:

A = ° z* = 1.3 82(-z* sin(0 - π / 5) + i*_d cos(0 - π / 5)) <4=1.3 82(-z* sin(0 - 4π / 5) + i*_d cos(0 - 4π / 5)) i*_D = 1.3 82(-z* sin(0 - 6π 15) + z* cos(0 - 6π / 5)) z* = 1.3 82(-z* sin(0 - 9π / 5) + z* cos(0 -9π / 5)) (5)

Equation (5) can be expressed in the form of matrix as:

(6)

1.382

1 cos—π 5	. 1 sm—π 5
4 cos—π 5	. 4 sm—π 5	COS#	-sin#
6 cos—π 5	. 6 sin—π 5	sin#	cos#
9 cos—π 5	. 9 sin —π 5 J

When phase-A open-circuit phase fault occurs, The five-phase IPM-FTTL motor loses one degree of freedom, and then it remains only three degrees of freedom. Two of them are located in the fundamental subspace and another one is located in the zero sequence subspace. The electromechanical energy conversion is taken place in the fundamental subspace. Thus, the degrees of freedom located in the fundamental sub space can be controlled according to the requirements of the motor thrust force. The degree of freedom located in the zero sequence subspace has to be controlled to zero because it only increases loss and thrust ripple. Thus, to realize FT-FOC, it is necessary to achieve transformation matrix after phase-A occurring short-circuit fault. That is to say, the two orthonormal basis Ti and T2 in the fundamental subspace can be selected as:

^Ti ^T2

6 9 cos —π cos—π cos—π cos —π

5 5 5 . 1 . 4 . 6 . 9 sm —π sm —π sm—π sm—π

5 5 5 (7) (8)

Or, according to winding space distribution, the two orthonormal basis T\ and Ti can be chosen as:

^Ti ^T2

4 6 8 cos —π cos—π cos—π cos—π

5 5 5 . 2 . 4 . 6 . 8 sm —π sm—π sm—π sm —π

5 5 5 (9) (10)

It is necessary to keep the fundamental subspace and the zero sequence subspace be orthogonal, and to ensure the zero sequence current be equal zero, the basis Z in the zero sequence sub space must meet the following requirements:

(11)

X =r₂Z^J =0

Considering the constraint requirement (4), and by using (7), (8), and (11), the generalized Clark transformation matrix T_4s/2i used to transform the natural stationary frame into stationary frame can be calculated as:

cos-π /3.618 cos—π /3.618 cos—π /3.618 cos — π /3.618

T ¹4s/2s sin-π/1.91 sin —π/1.91 sin —π/1.91 sin —π/1.91 • ² A sin —π /5 5 I 0.181 • ⁸ A sin-π /5 5 / 0.181 • ¹² A sin — π /5 5 /

0.181 • ¹⁸ A sin — π /5 5 /

0.181 (12)

Its inverse transformation matrix Γ_2ί/4ί can be expressed as

X =1-382

1	. 1	. 2
cos—π	sin —π	sin—π
5	5	5
4	. 4	. 8
cos—π	sin—π	sin—π
5	5	5
6	. 6	. 12
cos—π	sin —π	sin—π
5	5	5
9	. 9	. 18
cos—π	sin—π	sin — π
L 5	5	5

(13)

The sum of the phase currents is zero because the windings are star-connected. Thus, the current transformed from the natural stationary frame by the fourth row of (12) is zero in the zero sequence subspace, when the fourth row of (12) and (13) is removed, then the transformation matrix 7)_s/2s and its inverse transformation matrix

Γ_2ί74ί. can be expressed as:

(14) τ

J- Λ

4,/2, cosj7r/3.618 cos ^π^3.618 cos ^π/^.όΐδ cos ^3.618 sin-π /1.91 sin —π/1.91 sin —π/1.91 sin —π/1.91 • ² A sin —π /5 5 / • ⁸ A sin-π /5 /

• ¹² A sin — π /5 5 / • ¹⁸ A sin — π /5 /

/,4,=1-382

1	. 1	. 2
cos—π	sin—π	sin —π
5	5	5
4	. 4	. 8
cos—π	sin —π	sin —π
5	5	5
6	. 6	. 12
cos—π	sin—π	sin — π
5	5	5
9	. 9	. 18
cos—π	sin —π	sin—π
L 5	5	5 J

(15)

Or, by using (9)-(11), the basis Z in the zero sequence subspace can be expressed as:

Z = [z z z z] (16) where the z is constant. When z is selected as 0.4522, the generalized Clark transformation matrixT_4s/2s can be expressed as:

4 6

T =^slls 5

cos—π 5	cos—π 5	cos—π 5	cos—π 5
. 2	. 4	. 6	. 8
sin —π	sin—π	sin—π	sin—π
5	5	5	5
0.4522	0.4522	0.4522	0.4522

(17)

Thus, its inverse transformation matrix Γ_2ί74ί. can be expressed as:

¹ · ¹ 1 cos—π sin—π 1

A„4, =1-382 . 4 cos—π sin—π 1 ⁶ · ⁶ 1 cos—π sin—π 1

5 ⁹ · ⁹ 1 cos—π sin—π 1

5 (18)

The sum of the phase currents is zero because the windings are star-connected.

Similarly, the current transformed from the natural stationary frame by the third row of (17) is zero in the zero sequence subspace, thus the third row of (17) and (18) can be removed. However, they will not be eliminated in order to deduce the formula easily.

The electromechanical energy conversion is taken place in the fundamental subspace, and there is no energy conversion in the zero-sequence subspace. Therefore, the fundamental sub space has transformed into the synchronous rotating frame. Thus, the Park transformation matrix C_2i72r from stationary frame into the synchronous rotating frame and its inverse transformation matrix C_2r/2i can be expressed as:

	COS#	sin#	0’
c ^2sl2r	-sin#	cos#	0
	0	0	1
	cos#	-sin#	0’
c ^2r/2s	sin#	cos#	0
	0	0	1

(19) (20)

It can be seen from Fig.9 that, compared with the self-inductance, the mutualinductance is very low, the self-inductance has little fluctuation, hence the mutualinductance can be ignored, and the self-inductance L_s can be regard as constant.

Therefore, the phase self-inductance is affected by the transformation matrix. It can be seen from Fig.7 that the waveform of back-EMF is sinusoidal. However, the backEMFs can’t be transformed by using the proposed transformation matrix, because the phasor angle of the back-EMF is determined by the space location of each winding. In order to realize FT-FOC when the motor is open-circuit, the model in the natural coordinate system can be expressed as:

^UCe ~^UC ^eC — ¹¹ ii,.· ¹¹ ii Ci R?D + 4 di_B dt di_c dt di_D dt r> · τ / ^UEe ~^UE + ~7~ dt (21)

By using the transformation matrix Γ_4ί/2ί , namely (14) or (17), and the transformation matrix C_2s/2r , transformation (19), the equation (21) can be transformed into synchronous rotating frame:

^Ude /e ⁼ⁱq^R+Ls^- + ^^Ls^ld u = iR + L.

ze z dt di_ (22) dt where ω = πν)τ = 2π/, τ is pole pitch and v is the electrical velocity of the secondary.

By employing the magnetic co-energy method, and using the transformation matrix (14), (15), (19) and (20), the force equation under fault-tolerant condition can be deduced.

p. dL_s p dA._m ^s 8Θ ^{s s} 8Θ = 2.5-,-/.+0.955-,.4 cos» τ τ (23) where λ_ηι is the PM flux linkage, and θ = Jok// is the electrical angle.

In addition, by employing the magnetic co-energy method, and using the transformation matrix (17)-(20), the force equation under fault-tolerant condition can be deduced.

π il _T dL _T c\_m = 2.5-/,^+1.382^+/^,,() τ τ (24)

Thus, according to (22) and (23) or (24), if the i_d , i and i_z are controlled well in synchronous rotating frame, the desired force can be outputted when the motor is fault.

Part Two

On the basis of the part one, the remained healthy phase compensation currents can be used to restrain the thrust ripple caused by short-circuit current.

Assuming that the short-circuit current of phase-A is i_sc = I_f cos(®/-0,), where

7/is the amplitude of the short-circuit current, 6_f is the angle between the back-EMF and short-circuit current of phase A.

The first method:

The phase windings of the motor are star-connected and have no neural point line connected with the bus-voltage midpoint. Thus, the sum of the compensation currents, which are used to restrain the thrust ripple caused by short-circuit current, is zero. Based on the mirror symmetry about the axis which passes through the neutral point of the motor windings and is perpendicular to the axis of phase-A, the phase compensation currents (/j , z , z , z ) which are used to restrain the thrust ripple caused by short circuit current can be defined as:

X=~>c < i_E = -i_D (25) ζ·;+^+ζ+ζ; = ο

According to the principle of the amplitude of compensation currents is equal, the compensation currents can be further defined as:

z = x_b cosat + y_b sin co/ z = —x_b cosat-y_b sinat < (26) z = —x_b cosat - y_b sin at z = x_b cosat + y_b sinat where Xh, yb are the amplitude of the cosine and sine term of the current respectively, a = πν/τ = 2π/, v is the electrical velocity of the secondary and τ is pole pitch.

According to that the MMFs of the phase compensation currents and short-fault phase currents are zero, which can be expressed as MMF = Ni_sc + aNi, + a¹Ni +a³NiD + a‘ⁱNiE = 0, the Xb> yb can be calculated as:

x_h = -0.447/, cos0, (27) y_b = -0.447/^ sinri,

Then, by using (26) and (27), the compensation currents (z^ > z , i_D , z ) can be obtained:

(28) i”_B = -0.447/^ cos^t-3_f) = -0.447z_s( i_c = 0.447/_z cos(ot-0_f) = 0.447z_w i_D = 0.4471_f cos(ot-0_f) = 0.447i_sc i_E = -0.4471_f cos(ot-e_f) = -0.447i_sc

The second method:

Assuming that the compensation currents (z i_E ) as:

i_B = x_b coso t + y_b sinω t i = x_c coso t + y_c sino t i_D = x_d coso t + y_d sino t i_E = x_e coso t + y_e sino t (29) where Xh, x_c> xa> x_e> yt>> y_c> ya> y_e are the amplitude of the cosine and sine term of the currents, respectively, ω = πν/τ = 2π/ , v is the electrical velocity of the secondary and τ is pole pitch.

The windings are star-connected, and its neutral point is not connected with the midpoint of dc-bus voltage. Thus,

(30)

According to MMF = Ni_sc + aNi'_B +a²Nic + a³NiE + a⁴Ni_E = 0 , and the principle of copper loss minimum, the objective function can be designed as:

The Lagrange multiplier method is adopted to solve the minimum of the objective function based on the above constraints. Then, the results can be achieved:

x_b = -0.4471_f cos6_f x_c - 0.447/^ cos0_y x_d - 0.4471_f cos9_f x_e = -0.447/^ cos0_y (32) (33) y_b = -0.447/^ sin y_c = 0.447/^ sin0_z y_d = 0.447/^ sin Q_f y_e = -0.447/^ sinO,

Then, by using (29), (32) and (33) the compensation currents can be achieved:

i”_B = -0.44!I_f cos(ot-0_f) - -0.447/ i_c = 0.447/_z cos(ot-e_f) = 0.447/_c i_D = 0.447/_y cos(ot-e_f) = 0.447/_c i_E = -Q.44T I_f cos(cot-e_f) = -0.447/ (34)

Therefore, based on the same amplitude of the phase compensation currents or the minimum of copper loss minimum, to output the same thrust force in short-circuit phase-fault as that in the healthy condition, the remained phase synthetic current references can be achieved by using (5) and (28) or (34).

C = ° i*_B = 1.382(/ sin(0 - π / 5) + i*_d cos(0 - π / 5)) - 0.447i_sc < i*c = 1.382(-z* sin(0 -4π 15) + i*_d cos(0 -4π 15)) + 0.44Ίi_sc (35) i** = 1.382(/ sin(0 - 6π 15) + z* cos(0 - 6π / 5)) + 0.447i_sc i** = 1.382(/ sin(0 - 9π / 5) + z* cos(0 - 9π / 5)) - 0.447i_sc

Part Three

The FT-FOC strategy in short-circuit phase fault will be given.

By using the transformation matrix Γ_4ί/2ί defined by (14) or (17), (28) or (34) can be transformed into the synchronous rotating frame.

i”_a = -0.3998/_f cos(fifr-0_z) = -0.3998/

(36)

By using generalized Clark transformation matrix Γ_4ί/2ί defined by (14) or (17), the remained four healthy phase currents (i_B ,i_c ,i_D ,i_E ) can be transformed into the currents (i'_a , ϊ_β ,i'_z) in the stationary frame. And then, by using the currents (i'_a ,ΐ'_β , i'_z) to subtract the currents (z, ί_β , i_z ) defined by (36), the currents (z_a, i_p , i_z) in the stationary frame can be expressed as:

i =j’ -i” =i' + 0.3998/, i -i = z a a a a <^ιβ ~^ιβ~(g ^— (3 (37)

Finally, by using the Park transformation matrix C_ls/lr defined by (19), the currents (z_a , i_p , i_z) shown in (37) can be transformed to the currents (z_rf , i, z_z) in the synchronous rotating frame.

Or, by using the remained four healthy phase currents (z_B, i_c , i_D, i_E ) subtract the remained healthy phase compensation currents (/ ,/' ,/ ,/' ) defined by (28) or (34), the currents (i_B ,i'_c ,i'_D ,i'_E ) can be expressed as:

ύ ^— ύ ^— ύ ^— ύ + 0.447/jc ^zc ^{= z}c ^{— z}c ⁼ ύ ^— θ'447/γ ^{— — —} ύ ^— 0.447/^ ό· ^— ό ^— ύ ^— ό + θ·447ζ_ίε (38)

Then, by using the generalized Clark transformation matrix Γ_4ί/2ί defined by (14) or (17) and Park transformation matrix C_2i/2r defined by (19), the currents (i'_B , i'_c , i'_D , i'_E ) can be transformed to the currents (i_d > z > z_z) in the rotating coordinate system.

	i'c l'D
	= C T ^2s/2r14s/2s
z
z		Je_

(39)

The voltage references (u_d, u_q, u_z) can be obtained by the current regulators with the differences between the current references (u ,/*,/*) and the feedback currents (i_d , i , i_z) in the synchronous rotating frame. Then, by using the Park inverse transformation matrix C_2r/2i defined by (20), the voltage references (u*_d, u, u_z) can be transformed to the voltage references ( u_a , u_p , u_z) in the stationary frame.

In order to make the remained healthy phases of the motor generate compensation currents (i_B , z , i_E , i_E ) defined by (28) and (34), based on the mathematical expression of phase-A short-circuit current, phase-A back-EMF and short-circuit compensation current, the compensation voltages of the remained healthy phases (u_B , u_c ,u”_D ,u_E ) can be defined as:

'u”_B = 0.447^ uⁿc = -0.447^ ' _{U D} = -0.447^ u_E = 0.447e₄ (40)

Based on generalized Clark transformation matrix Γ_4ί/2ί defined by (14) and (17), the compensation voltages defined by (40) can be transformed into the stationary frame:

<-	= 0.3998e
	= 0
u =	= 0

(41)

Making the voltage references (u*_a, u_p , u_z) add the compensation voltages (w, , u') defined by (41) in the stationary frame, it can be obtained:

>C =^u*a +0.3998^ < u* = u_p (42) // = it

Then, by using the generalized Clark transformation matrix 7_4j/2j defined by (15) or (18), the voltage references (u*_a*,u**,u*) defined by (42) can be transformed to the voltage references (u_B,u_c , ζζζ, m* ) in the natural stationary frame. Finally, making them add the phase back EMFs, the expect phase voltage references (m*\ // > u* , w“) can be obtained.

2Ί

u*	= 0
**	*
uB	— uB+eB
<u*	= uc +ec	(43)
uD	— uD + eD
uE	= uE+eE

Or, by using the generalized Clark transformation matrix /j_{s 2s} defined by (15) or (18), the voltage references (u_a ,u*_p ,u*) in the stationary frame can be transformed into the natural stationary frame, then, the voltage references (u*_B, u_c, u_D, u_E) can be obtained as

UB		Γ * Ί
		U
		a
uc	- T 12sl4s	Up
uD		*
*		Λ.
uE _

(44)

Then, making the voltage references (u_B, u_c , u_D , u_E ) defined by (44) add the remained healthy phase compensation voltages (u_B ,u_c ,u_D ,u_E ) defined by (40) and add the phase back EMFs, the expect phase voltage references (m*\ u*> u* > w“) can be obtained.

u_A = 0 u* =u_B+u_B+e_B =u_B+ 0.447^ + e_B < m** = u_c+ u_c + e_c = u_c- 0.447^ + e_c (45) u_E = u_E + u_E + e_E = u_E — 0.447^ + e_E u* =u*_E+u_E+e_E =u_E + 0.447^ + e_E

The expect phase voltage references (u* > u** > u** > u_E ) defined by (43) or (45) are transmitted to the voltage source inverter, which is adopted CPWM technique based on zero sequence voltage harmonic injection, then the FT-FOC for the five-phase IPMFTTL motor with hybrid magnetic material is realized when short-circuit phase fault occurs, and the motor fault-tolerant operation is achieved. Fig. 11 and Fig. 12 show two principle diagrams of FT-FOC strategy for the five-phase IPM-FTTL motor with hybrid magnetic material.

Then, according to Fig. 10 and Fig. 11 or Fig. 12, the simulation model of the control system based on the five-phase IPM-FTTL motor with hybrid magnetic material is established in Matlab/Simulink. The simulated results were used to evaluate the feasibility of the FT-FOC method in short-circuit fault.

Fig. 13 shows phase current waveform of the IPM-FTTL motor without faulttolerant operation when phase A is with short-circuit of the present invention. It can be seen that the current ripple is greatly. Fig. 14 shows the force waveform of the IPMFTTL motor without fault-tolerant operation when phase-A is with short circuit. The thrust force ripple is large and the difference between the maximum and minimum value of thrust force reaches 34N. Fig. 15 shows phase current waveform of the IPM-FTTL motor with IPM-FTTL when phase-A is with short-circuit. The current ripple are decreased, that is to say, the results are consistent with the calculation results of (35). Fig 16 shows the force waveform of the IPM-FTTL motor with FT-FOC method when phase-A is with short-circuit. It can be seen that the force ripple is restrained and also is the same as that in healthy condition. Fig. 17 shows the force waveform of the IPMFTTL motor in healthy when the force reference steps. The response time is 0.6ms. However, the force response time in short-circuit fault-tolerant condition is also 0.6ms, which can be seen from Fig. 18. It can be concluded that the dynamic performance of the five-phase IPM-FTTL motor with FT-FOC strategy in fault-tolerant condition is the same as that in healthy condition. Meantime, with the FT-FOC method, it has less force ripple and good current tracking performance in short-circuit phase fault.

If short-circuit fault occurs in any phase of the motor, the electrical degree difference between it and phase-A is . When the fault phase is phase-A, B, C, D, and E, the k is 0, 1, 2, 3, and 4 respectively. So, the natural stationary frame should be rotated electrical degrees counterclockwise to make the axis of fault phase overlap with the axis of phase-A in healthy. Then, the variable Θ should be replaced by Θ - in the matrix C_2sl2r and C_2rl2s. Namely, (46)

Take phase-B as an example, when phase-B is with short-circuit fault, the natural stationary frame just have to be rotated electrical degrees counterclockwise, namely, k=\ in (46) and (47). Fig. 19 and Fig. 20 show phase current and force waveform of the IPM-FTTL motor with FT-FOC method when phase A is recovered from short-circuit fault and phase-B is with short-circuit fault. It can be seen that the thrust force is the same as that in healthy condition. It can be concluded that the FTFOC strategy proposed in this invention is appropriate for any the short-circuit phase fault. In addition, the strategy avoids the complex calculation and saves the CPU overhead.

It can be noted from the aforementioned statement that, compared with conventional IPM tubular linear motor, the five-phase IPM-FTTL motor with hybrid magnetic material can save the usage of the rare earth PMs, raise the utilization ratio of the PMs, decrease the cost, and reduce the leakage magnetic flux. In addition, due to the introduction of the FTT, the adjacent phase windings of the primary are essentially isolated, and it improves the fault-tolerant capability and reliability of the motor.

When the current is less than the maximum current limit, the FT-FOC method not only can make the electromagnetic thrust accord with that in healthy condition, but also can restrain the thrust force ripple. More importantly, the dynamic performance, stability performance, and current tracking performance are in consistent with that in healthy condition. In addition, the strategy is appropriate for any short-circuit phase fault, it avoids the complex calculation and saves the CPU overhead. The invention has good application prospect where requires the high reliability such as electromagnetic active suspension system.

Although the present invention has been made public as above implement example, the example is not used to limit the invention. Any equivalent change or retouching within the spirit and field of the present invention belongs to the protective range of the invention.

Claims (10)

Claims

1. A five-phase interior permanent magnet fault-tolerant tubular linear (IPM-FTTL) motor with hybrid magnetic material contains primary and secondary, the length of primary is shorter than secondary, and there is air gap between primary and secondary;

the described primary includes armature teeth, fault-tolerant teeth (FTT) and windings, the 2*m armature teeth and 2*m fault-tolerant teeth are distributed evenly in the primary, m is the phase number of the machine and m>3, the FTT are interleaved with the armature teeth, the armature teeth and FTT are staggered in arrangement, every slot contains only one single-layer winding, and the slots on both sides of armature teeth contain coil turns from a single phase, but there is no winding wound the FTT, the windings in the 1st slot and the (2*m+l)th slot belongs to one phase, and so on;

the described secondary includes permeability magnetic materials and permanent magnets (PMs), the interior PMs are placed between two pieces of permeability magnetic materials, and then the permeability magnetic materials can isolate the each pole of PMs, each pair of PMs is consists of rare earth PM materials and ferrite magnetic materials, and both of them are axial magnetization alternately. Additionally, the axial width of rare earth PMs and ferrite magnetic materials is equal, the pieces of PM with same direction of magnetization consist of same PM material, or connect with two kinds of PM material in series or parallel;

the width of described the armature teeth wat and FTT wp is equal, but the width of the armature teeth wat can be greater than FTT wp sometimes, both the armature teeth and FTT have no modulation teeth, or have some modulation teeth.

2. An IPM-FTTL motor with hybrid magnetic material as claimed in claim 1, wherein the described motor employs fractional-slot construction, the relationship between the number of primary slots and the secondary pole pairs should meet the condition of Ns=2p±2 or Ns=2p±l, where Ns is the number of primary slots, andp is the secondary pole pairs.

3. An IPM-FTTL motor with hybrid magnetic material as claimed in claim 1, wherein the shape of described PM under each pole can be one whole tubular or two level of nesting tubulars or connection of two tubulars or spliced into a tubular by n pieces of tiles, n>2, the radial width of PM is shorter than the width of permeability magnetic materials, additionally, the inner diameter of the PM tubular is greater than the inner diameter of the permeability magnetic material tubular and the outer diameter of the PM tubular is shorter than the outer diameter of permeability magnetic material tubular, and the PM tubulars and the permeability magnetic material tubulars are mounted coaxially.

4. An IPM-FTTL motor with hybrid magnetic material as claimed in claim 1, wherein the described motor can be the one-side flat structure or double-side flat structure or tubular structure, additionally, the machine can be used as generator or motor.

5. A fault-tolerant field-oriented control (FT-FOC) method for five-phase IPMFTTL motor with hybrid magnetic material as claimed in claim 1 in short-circuit phase fault, wherein the described motor has five phases, such as phase-A, phase-B, phase-C, phase-D, phase-E; the FT-FOC method in short-circuit phase fault condition includes the following steps:

(1) the remained healthy phase currents are used to compensate for the lack of normal force caused by short-circuit phase fault, according to the remained healthy phase currents, the generalized Clark transformation matrix T4j/2j and its inverse transformation matrix T2s/iis can be obtained, with which the remained four healthy phases in the natural stationary frame can be transformed into the stationary frame, meanwhile, the generalized Park transformation matrix C2s/2r and its inverse transformation matrix C2r/2s are defined;

(2) establishing the motor mathematical model in the synchronous rotating frame in the open-circuit phase fault condition;

(3) the remained healthy phase compensation currents are used to restrain thrust ripple caused by short-circuit fault phase current, the healthy phase compensation currents!/'' ,/ ,/ ,/ ) can be obtained, then the compensation currents (/'' ,/ ,/ , iE ) are transformed to the compensation currents (/, ip , i ) in stationary frame by the generalized Clark transformation matrix T\s!2s

(4) the remained four healthy phase currents (iB, ic , iD, iE ) can be transformed to the currents (i'af'p ,i'2) in stationary frame by the generalized Clark transformation matrix 7 js./2s., then, the currents (ia , ΐβ , iz) can be obtained by using the current (i'a , ϊβ , i'z ) to subtract the current (z, ίβ , i ) worked out in step (3) , finally, the currents (ζα, ίβ , z'z) can be transformed to the currents (id > i > zz) in the synchronous rotating frame by using the Clark transformation matrix C2z/2r;

(4’) the currents (i'B , i'c , i'D , i'E ) can be obtained by using the remained four healthy phase currents (zB,zc,zD,z£) to minus remained healthy phase compensation currents (iB ,i'c ,iD ,iE ), then, by using the generalized Clark transformation matrix 7js./2s. and

Park transformation matrix C2z/2r, the currents (i'B ,i'c ,i'D ,i'E ) can be transformed to the currents (id > i > z'z) in synchronous rotating frame;

(5) by using the current regulators, the voltage references (zz* , zz*, zz*) can be obtained with the differences between the current references (id , i*, i*) and the feedback currents (Jd ,iq,if in synchronous rotating frame, then, the voltage references (zz* ,zz*, zz* ) can be transformed into the stationary frame by using the Park inverse transformation matrix C2r/2s, thus, the voltage references (ua , zz* , zz*) in stationary frame are obtained;

(6) to make the remained healthy phases of the described motor generate the compensation currents (iB , z , iE , iE ), which are used to suppress thrust ripple caused by short-circuit phase current, based on the mathematical expression of phase-A shortcircuit current, phase-A back electromotive force (EMF) and short-circuit compensation currents, the compensation voltages of the remained healthy phases (zz , uc ,uD ,uE ) can be defined as:

/=0.447^ w =-0.447^ < uD = -0.447^ ’ uE = 0.447^ based on generalized Clark transformation matrix Γ4ί/2ί , the compensation voltages in the natural stationary frame can be transformed into the stationary frame, thus, the compensation voltages in the stationary frame are:

/ = 0.3998^ < w = 0 ;

< = 0

(7) making the voltage references («*/,«*) add compensation voltages (w, / , u') in stationary frame, it can be obtained :

Ua=Ua +0.3998^ < ΙΙβ —Up u** = w* then, the voltage references (uB, uc, uD, uE) can be obtained by using the generalized Clark inverse transformation matrix 7 ]s./4s. to transform the voltage references ( u*, u*, u*) into the natural stationary frame, then adding the phase back EMFs, the expect phase voltage references (u* > > u* > u*) are obtained;

(7’) the voltage references ( uB , u*c , uD , uE ) can be obtained by using the generalized Clark transformation matrix Γ4ί/2ί to transform the voltages references (ua , u*p , uz) into in the natural stationary frame, then, adding the remained healthy phase compensation voltages ( uB > uc > uD > uE ) , the expect phase voltage references (u** > u** > u** > uE ) are obtained;

(8) the expect phase voltage references (uB > uc > uD > uE ) are transmitted to voltage source inverter, which is adopted CPWM technique based on zero-sequence voltage harmonic injection, then the FT-FOC for the described five-phase IPM-FTTL motor with hybrid magnetic material is realized when phase-A short-circuit fault occurs.

6. A FT-FOC method for five-phase IPM-FTTL motor with hybrid magnetic material in short-circuit phase fault as claimed in claim 5, the detailed processes of the described step (1) are:

(1.1) the phase windings of the described motor are star-connected and having no neural point line connected with the bus-voltage midpoint, supposing an open-circuit phase fault occurs in phase-A when phase-A short-circuit phase fault occurs, according to the constant traveling-wave constant magnetomotive force before and after the fault and the equality of the remained four healthy phase currents amplitude, and based on the constraint condition that the sum of the remained healthy phase currents is zero, the remained healthy phase current references can be calculated by inter-relating on the basis of the spatial symmetry about the fault phase-A;

(1.2) the two orthonormal basis Ti and T2 are selected in the fundamental subspace and the basis Z in the zero sequence subspace according to the electromechanical energy conversion principle, the fault-tolerant current vector and back EMF vector; it is necessary to keep the fundamental subspace and the zero sequence subspace orthogonal, and to ensure that the zero sequence current is zero, therefore, the vector base of zero sequence subspace must meet the following requirements:

Ί[ΖΤ = < zzc }e _

based on basis Τι, T2 and Z, the generalized Clark transformation matrix Γ4ί/2ί used to transform the natural stationary frame into the stationary frame can be worked out:

cos-π/3.618 cos—π/3.618 cos—π/3.618 cos —π/3.618 τ

/1

45/25 είη^π /ΐ.91 • 2 A sin —π /5 5 I

0.181 βΐη^π/1.91 • 8 A sm-π /5 5 / 0.181 sin ^π /.91 • 12 A sin — π /5 5 /

0.181 sin ^π /.91 • 18 A sin — π /5 5 I

0.181 or:

cos-π/3.618 cos^ /3.618 cos —π/3.618 cos —π/3.618

T

J- Λ

4,/2, sin-π/1.91 sin—π/1.91 sin-π/1.91 sin-π/1.91 • 2 A sin —π /5 5 / sin-π /5 5 / sin — π/5 sin — π/5 and then its inverse transformation matrix can be expressed as:

^/4,=1-382

1 . 1 . 2 cos—π sin —π sin—π 5 5 5 4 . 4 . 8 cos—π sin—π sin—π 5 5 5 6 . 6 . 12 cos—π sin —π sin—π 5 5 5 9 . 9 . 18 cos—π sin—π sin — π L 5 5 5

or:

^/4,=1-382

1 . 1 . 2 cos—π sin—π sin—π 5 5 5 4 . 4 . 8 cos—π sin —π sin —π 5 5 5 6 . 6 . 12 cos—π sin—π sin — π 5 5 5 9 . 9 . 18 cos—π sin —π sin—π L 5 5 5 J

(1.3) to transform the energy conversion in the fundamental subspace into the synchronous rotating frame, the Park transformation matrix C2i72r used to transform the stationary frame into the synchronous rotating frame and its inverse matrix C2r/2s can be defined as:

2s/2r

2r/2i

COS#

-sin# sin# cos#

COS# sin#

-sin# cos#

7. A FT-FOC method for five-phase IPM-FTTF motor with hybrid magnetic material in short-circuit phase fault as claimed in claim 5, the detailed processes of the described step (2) are:

(2.1) compared with the self-inductance, the mutual-inductance is very low, so the mutual-inductance can be ignored, the self-inductance has little fluctuation, hence the self-inductance can be regard as constant Ls, the phase voltages subtract the phase back EMFs, respectively, then the model in the natural stationary frame under the opencircuit fault condition can be expressed as:

U ~ d — + A, diB dt dic dt diD dt diE dt where ua, ub, uc, ud, ue are the remained healthy phase voltages, ee, ec, eD, es are the remained healthy phase back EMFs, uee, uce, UDe, use are the results of the phase voltages subtracting the phase back EMFs, respectively;

(2.2) by using the transformation matrix Γ4ί/2ί and C2i/2r, the motor model under open-circuit fault condition can be transformed into synchronous rotating frame:

Ude = idR+LS^-^Lsiqur =^+4 u = iR + L.

ze z dt di_ dt where ω=πν/τ=2π/ , τ is pole pitch and v is the electrical velocity of the secondary;

(2.3) by using the magnetic co-energy method and the transformation matrix T\s/2s,

T2s!lis, C2s!2r and C2r!2s, when the motor is under fault-tolerant condition, the torque can be expressed as

F = 2.5-zλη + 0.955-z'A cos0 τ τ or:

F = 2.5-z +1.382-ZX sin0 τ τ where ληι is the PM flux linkage, and Θ = Joz/t is the electrical angle.

8. A FT-FOC method for five-phase IPM-FTTF motor with hybrid magnetic material in short-circuit phase fault as claimed in claim 5, the detailed processes of the described step (3) are:

(3.1) assuming that the short-circuit current of phase-A is isc = I f cvs.(a>t -Θ , where If is the amplitude of the short-circuit current, 6f is the angle between the back-EMF and short-circuit current of phase-A;

(3.2) the sum of the compensation currents used to restrain the thrust ripple caused by short-circuit current is zero, and the magnetomotive force sum of the compensation current and short-circuit phase current is zero; based on the mirror symmetry about the axis which passes through the neutral point of the motor windings and is perpendicular to the axis of phase-A, the compensation currents used to restrain the thrust ripple caused by short-circuit current can be calculated as:

-0.447/

SC

0.447/

SC

0.447/ ’

SC

-0.447/

SC (3.2’) according to MMF = Nisc + aNi'B +a2Nic + a'NiD + afNiE = 0 i'B + 4 + i'D +iE = 0 and the principle of copper loss minimum, the Lagrange multiplier method is adopted to solve the compensation currents (iB > / > iE > iE ) :

7;=-o.447z;c i=0-447z;c ^'c=0-447z;c ’ z=-0.447zjc (3.3) by using the generalized Clark transformation matrix F4j/2j , the compensation currents (iB > / > iE > iE ) used to restrain the thrust ripple caused by short-circuit current can be transformed into the synchronous rotating frame, then the compensation currents (/ > ίβ > / ) in the stationary frame can be expressed as:

C=-0.3998z;c </;=o z>0

9. A FT-FOC method for five-phase IPM-FTTL motor with hybrid magnetic material in short-circuit phase fault as claimed in claim 5, if short-circuit phase fault occurs in any phase, the electrical degree difference between it and phase-A is 2^7^/, when the fault phase is phase-A, B, C, D, or E, the k is 0, 1, 2, 3, or 4, respectively, hence the natural stationary frame should be rotated electrical degrees counterclockwise to make the axis of phase-A in the healthy condition overlap with the axis of fault phase, then, the variable Θ should be replaced by θ-2^ττ/ in the matrix C2s/2r and C2r/2s, namely,

2s/2r

10. A FT-FOC method for five-phase IPM-FTTL motor with hybrid magnetic material in short-circuit phase fault as claimed in claim 5, the described FT-FOC strategy used in short-circuit phase fault condition is also appropriate for five-phase fault-tolerant permanent-magnet motor control system.