WO2022134769A1 - 一种永磁辅助同步磁阻电机振荡抑制的控制方法 - Google Patents

一种永磁辅助同步磁阻电机振荡抑制的控制方法 Download PDF

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WO2022134769A1
WO2022134769A1 PCT/CN2021/124380 CN2021124380W WO2022134769A1 WO 2022134769 A1 WO2022134769 A1 WO 2022134769A1 CN 2021124380 W CN2021124380 W CN 2021124380W WO 2022134769 A1 WO2022134769 A1 WO 2022134769A1
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current
pass filter
voltage
current loop
controller
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PCT/CN2021/124380
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English (en)
French (fr)
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詹哲军
张瑞峰
苏鹏程
张宇龙
于森林
丁志勇
杨高兴
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中车永济电机有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/08—Reluctance motors
    • H02P25/098—Arrangements for reducing torque ripple
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

Definitions

  • the present disclosure relates to a motor oscillation suppression method in an urban rail permanent magnet traction drive system, in particular to an active damping compensation method in the oscillation suppression, in particular to a control method for the permanent magnet assisted synchronous reluctance motor oscillation suppression.
  • the urban rail permanent magnet traction drive system is mostly powered by the DC traction network.
  • the DC side voltage will continue to oscillate, which will cause the DC side current and the output torque of the motor to oscillate. It affects the stability and comfort of vehicle operation. In severe cases, it will lead to overvoltage and overcurrent faults in the converter, triggering the Traction Control Unit (TCU) to block the pulse and causing the urban rail vehicle to lose traction.
  • TCU Traction Control Unit
  • the oscillation mechanism of the permanent magnet-assisted synchronous reluctance traction system for urban rail is analyzed. Assuming that the controller bandwidth of the permanent magnet traction system is infinite, the inverter is a continuous energy conversion system, and the output torque of the traction motor can perfectly follow the command value, then The inverter-motor system can be regarded as an ideal constant power load.
  • the simplified constant power load model of the system is shown in Figure 1.
  • the grid-side voltage E w supplies power to the vehicle traction inverter
  • R represents the line
  • the sum of resistance and inductance resistance, L is the filter inductance, C is the support capacitor, u dc is the DC bus voltage input to the inverter side, Z m is the equivalent impedance of the inverter plus the motor.
  • the inverter-motor system presents a negative impedance characteristic Y
  • the front-end damping coefficient of the DC side of the converter and the overall damping coefficient of the transmission system are:
  • the basic condition for system stability is that the system damping coefficient is positive, so the system stability criterion under the ideal model is determined as
  • P 0 and u dc are the steady-state output power and the DC bus voltage input to the inverter side, respectively.
  • the voltage equation of the permanent magnet-assisted synchronous reluctance motor in the d-q coordinate system can be expressed as:
  • ud and u q are the stator voltages of the d and q axes
  • R s is the stator resistance
  • ⁇ r is the electrical angular velocity of the motor rotor
  • L d and L q are the inductances of the d and q axes of the motor, respectively
  • id and i q are d
  • q-axis stator current ⁇ f is the permanent magnet flux linkage
  • the vehicle often gives the vehicle traction command through the handle, and then distributes it to each motor.
  • the block diagram of the permanent magnet-assisted synchronous reluctance motor vector control algorithm is shown in Figure 2.
  • the permanent magnet-assisted synchronous reluctance motor measures the rotor position ⁇ of the motor through the resolver, and the electrical angular velocity ⁇ r of the motor is obtained after ⁇ is differentiated.
  • the current converter passes the measured motor currents i a and i b through Clark transformation and Park transformation to obtain the currents id and i q under the dq coordinate.
  • the given motor torque T e * is distributed to obtain the given currents id * and i q * through the MTPA look-up module (Maximum Torque Per Ample, MTPA), which is based on the calibrated maximum torque-current ratio algorithm is calculated.
  • MTPA Maximum Torque Per Ample
  • i d *, i q *, id , i q , ⁇ r and u dc are the inputs of the current loop controller
  • ud * and u q * are the outputs of the current loop controller.
  • ud *, uq *, ⁇ , ⁇ r and bus u dc are input into the segment PWM modulation module to generate PWM pulses to the inverter in part 6.
  • the present disclosure performs active damping compensation on the basis of the vector control algorithm block diagram of the permanent magnet-assisted synchronous reluctance motor shown in FIG. 2 .
  • the present disclosure aims to solve the technical problem of the DC side oscillation caused by the impedance mismatch of the urban rail permanent magnet traction system, and provides a control method for the oscillation suppression of the permanent magnet assisted synchronous reluctance motor.
  • the present disclosure solves its technical problems by adopting a voltage q-axis active damping compensation method, specifically: a control method for the oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, using a current loop controller, and the input signal of the current loop controller is i d *, i q *, d -axis stator current id , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * It is the given motor torque T e * through the maximum torque-current ratio control distribution to obtain the given current, and active damping compensation is performed on the output signal u q * of the current loop controller.
  • the current loop controller includes a current decoupling controller and a stable Compensator; id *, i q *, id , i q and ⁇ r are processed by the current decoupling controller to obtain u d1 * and u q1 * ;
  • the stable compensator includes a high-pass filter, a first low-pass filter , a second low-pass filter and an adder, the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1 , and the DC bus voltage u dc is processed by the second low-pass filter.
  • the voltage u 2 is obtained, u 1 and u 2 are added by the adder to obtain the voltage u 3 , and the voltage u 3 is brought into the formula
  • the result of multiplying u d1 * and 1 is the output ud * of the current loop controller
  • the result of multiplying u q1 * and ⁇ u q is the current loop control output u q * after active damping compensation.
  • the present disclosure solves its technical problems by adopting a voltage d-axis active damping compensation method, specifically: a control method for oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, which adopts a current loop controller, and the input signal of the current loop controller is: id *, i q *, d -axis stator current id , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * is the motor given torque T e *
  • the given current is obtained through the control distribution of the maximum torque to current ratio, and the output signal ud * of the current loop controller is actively damped and compensated.
  • the current loop controller includes the current decoupling controller and Stable compensator; id *, i q *, id , i q and ⁇ r are processed by the current decoupling controller to obtain u d1 * and u q1 * ; the stable compensator includes a high-pass filter, a first low-pass filter The DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1 , and the DC bus voltage u dc is processed by the second low-pass filter.
  • the voltage u 2 is obtained, u 1 and u 2 are added by the adder to obtain the voltage u 3 , and the voltage u 3 is brought into the formula
  • the output of the stable compensator ⁇ u d , u d1 * and ⁇ u d are multiplied, which is the result of multiplying the output ud *, u q1 * and 1 of the current loop controller after active damping compensation. is the output u q * of the current loop controller.
  • the present disclosure solves its technical problems by adopting a current q-axis active damping compensation method, specifically: a control method for oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, using a current loop controller, and the input signal of the current loop controller is: id *, i q *, d -axis stator current id , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * is the motor given torque T e * The given current is obtained through the maximum torque-current ratio control distribution, and active damping compensation is performed on the input signal i q * of the current loop controller.
  • the current loop controller includes a current decoupling controller and A stable compensator; the stable compensator includes a high-pass filter, a first low-pass filter, a second low-pass filter and an adder, and the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance
  • the voltage u 1 and the DC bus voltage u dc are processed by the second low-pass filter to obtain the voltage u 2
  • u 1 and u 2 are added by the adder to obtain the voltage u 3
  • the voltage u 3 is brought into the formula
  • the output of the stable compensator ⁇ i q , i d * and 1 are multiplied to obtain the input i d1 * of the current decoupling controller, i q * and ⁇ i q are multiplied to obtain the input i of the current decoupling controller q1 * ; i d1 *, i q1 *, id , i q and ⁇ r are
  • the present disclosure solves its technical problems and also adopts the current d-axis active damping compensation method, specifically: a control method for oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, which adopts a current loop controller, and the input signal of the current loop controller is: id *, i q *, d -axis stator current id , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * is the motor given torque T e *
  • the given current is obtained through the control distribution of the maximum torque to current ratio, and the input signal i d * of the current loop controller is actively damped and compensated.
  • the current loop controller includes the current decoupling controller and A stable compensator; the stable compensator includes a high-pass filter, a first low-pass filter, a second low-pass filter and an adder, and the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance
  • the voltage u 1 and the DC bus voltage u dc are processed by the second low-pass filter to obtain the voltage u 2 , u 1 and u 2 are added by the adder to obtain the voltage u 3 , and the voltage u 3 is brought into the formula
  • the output ⁇ id of the stable compensator, id * and ⁇ id are multiplied to obtain the input i d1 * of the current decoupling controller, i q * and 1 are multiplied to obtain the input i of the current decoupling controller q1 * ; i d1 *, i q1 *, id , i q and ⁇ r are processed in the current
  • the present disclosure also adopts the torque active damping compensation method to solve the technical problem, specifically: a control method for the vibration suppression of the permanent magnet assisted synchronous reluctance motor, which adopts the MTPA look-up module, and the input signal of the MTPA look-up module is the motor
  • active damping compensation is performed on the motor given torque T e * through a stable compensator, which includes a high-pass filter, a first low-pass filter, a second low-pass filter and an adder
  • the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1
  • the DC bus voltage u dc is processed by the second low-pass filter to obtain the voltages u 2 , u 1 and u 2
  • the voltage u 3 is obtained by adding the adder, and the voltage u 3 is brought into the formula
  • the output ⁇ T e of the stable compensator is obtained by calculation, and ⁇ T e is
  • the segmented PWM modulation module adopts a segmented modulation method combining multiple modulation methods.
  • segmented modulation The methods are as follows: that is, the asynchronous modulation method is used when [0 ⁇ f 0 ), the 15-frequency synchronous modulation method is used when [f 0 ⁇ f 1 ), and the 12-frequency synchronous modulation method is used when [f 1 ⁇ f 2 ).
  • [f 2 ⁇ f 3 ) adopt the 9-frequency synchronous modulation method
  • [f 3 ⁇ f 4 ) adopt the 7-frequency synchronous modulation method
  • [f 4 ⁇ f 5 ) adopt the 5-frequency synchronous modulation method
  • [f 5 ⁇ f 6 ) the frequency division synchronous modulation method
  • [f 6 ⁇ f 7 ] adopts the square wave modulation method; where f 0 is one-fifteenth of the switching frequency in the asynchronous modulation stage, and f 1 is the maximum allowable power device.
  • f 2 is one-twelfth of the maximum allowable switching frequency of the power device
  • f 3 is one-ninth of the maximum allowable switching frequency of the power device
  • f 4 is the maximum allowable switching frequency of the power device.
  • One-seventh of the switching frequency of the power device f5 is one - fifth of the maximum allowable switching frequency of the power device
  • f6 is one-third of the maximum allowable switching frequency of the power device
  • f7 is the highest frequency of the motor.
  • f 1 to f 6 are the frequencies that "the motor can only run up to", which can be entered in advance under the conditions allowed by the switching frequency and the system.
  • the maximum switching frequency of the traction inverter is limited by heat dissipation conditions and is often only a few hundred Hz, while the maximum operating frequency of the traction motor can reach about 300 Hz, such as within the entire speed regulation range.
  • a segmented modulation method combining a variety of modulation methods is used in PWM modulation.
  • the segmented modulation method employs different modulation strategies at different motor frequencies.
  • the harmonic distribution of different modulation strategies is different, and the system oscillation is also related to the harmonic distribution. When the system oscillates, the current modulation method can be switched to the next modulation method to change the harmonic distribution to alleviate the oscillation. , which makes the oscillation suppression effect of the control method better.
  • the present disclosure is based on the original vector control method of the permanent magnet assisted synchronous reluctance motor, and then adopts the voltage q-axis active damping compensation method, the voltage d-axis active damping compensation method, the current q-axis active damping compensation method, and the current d-axis active damping method.
  • the compensation method, the torque active damping compensation method and the modulation variation method solve the technical problem of the DC side oscillation caused by the impedance mismatch of the urban rail permanent magnet traction system, and improve the stability of the urban rail permanent magnet traction system under the control of the original vector control method. It can realize the stability of the system without increasing the hardware cost of the system.
  • FIG. 1 is a schematic structural diagram of the constant power load model of the traction drive system described in the background art of the disclosure.
  • FIG. 2 is a block diagram of the vector control of the permanent magnet assisted synchronous reluctance motor described in the background art of the disclosure.
  • FIG. 3 is a control block diagram of the current loop controller described in Embodiment 1 of the present disclosure.
  • FIG. 4 is a control block diagram of the current decoupling controller described in Embodiment 1 of the present disclosure.
  • FIG. 5 is a control block diagram of the current loop controller described in Embodiment 2 of the present disclosure.
  • FIG. 6 is a control block diagram of the current decoupling controller described in Embodiment 2 of the present disclosure.
  • FIG. 7 is a control block diagram of the current loop controller described in Embodiment 3 of the present disclosure.
  • FIG. 8 is a control block diagram of the current decoupling controller described in Embodiment 3 of the present disclosure.
  • FIG. 9 is a control block diagram of the current loop controller described in Embodiment 4 of the present disclosure.
  • FIG. 10 is a control block diagram of the current decoupling controller described in Embodiment 4 of the present disclosure.
  • FIG. 11 is a control block diagram of adding torque compensation described in Embodiment 5 of the present disclosure.
  • FIG. 12 is a modulation block diagram of the segmented PWM modulation module of the present disclosure.
  • FIGS. 1-12 a control method for suppressing oscillation of a permanent magnet-assisted synchronous reluctance motor described in the present disclosure will be described in detail.
  • Embodiment 1 A method for controlling oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, as shown in Figure 3, a current loop controller is used, and the input signals of the current loop controller are id*, iq *, and d axes Stator current i d , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * are the given motor torque T e * The given current is obtained through the maximum torque-current ratio control distribution, and active damping compensation is performed on the output signal u q * of the current loop controller.
  • the current loop controller includes a current decoupling controller and a stable compensator; i d *, i q *, id , i q and ⁇ r are processed by the current decoupling controller to obtain u d1 * and u q1 * ;
  • the stable compensator includes a high-pass filter, a first low-pass filter, a second low-pass filter and an addition
  • the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1
  • the DC bus voltage u dc is processed by the second low-pass filter to obtain the voltages u 2 , u 1 and u 2
  • the voltage u 3 is obtained by adding the adder, and the voltage u 3 is brought into the formula
  • the output ⁇ u q of the stable compensator is obtained by calculation.
  • the specific calculation process of the stable compensator is as follows:
  • u dc_HPF is the value of the bus voltage u dc after high-pass filtering
  • G HPF (s) is the transfer function of the high-pass filter
  • ⁇ H is equal to 2 ⁇ f HPF
  • f HPF is the cut-off frequency of the high-pass filter
  • u dc_PF is the filter The value after the back voltage u dc_HPF passes through the first low-pass filter
  • G LPF1 (s) is the transfer function of the first low-pass filter
  • ⁇ L1 is equal to 2 ⁇ f LPF1
  • f LPF1 is the cut-off frequency of the first low-pass filter
  • G LPF2 (s) is the transfer function of the second low-pass filter
  • ⁇ L2 is equal to 2 ⁇ f LPF2
  • f LPF2 is the cut-off frequency of the second low-pass filter
  • ⁇ is the bus voltage compensation coefficient
  • ⁇ takes a value around 1, n is the order, and n is generally 2 to 4;
  • the control block diagram of the current decoupling controller is shown in FIG. 4 .
  • the outputs of the coupled controller are u d1 * and u q1 * , and the calculation process of u d1 * and u q1 * is shown in the following formulas:
  • G PId (s) is the transfer function of the current loop d-axis PI regulator
  • G PIq (s) is the transfer function of the current loop q-axis PI regulator, respectively
  • k is the control coefficient. Adjusting k can increase the stability of the system.
  • ⁇ f is the permanent magnet flux linkage.
  • Embodiment 2 A method for controlling oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, as shown in Figure 5, a current loop controller is used, and the input signals of the current loop controller are id*, iq *, and d axes Stator current i d , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * are the given motor torque T e * The given current is obtained through the maximum torque-current ratio control distribution, and active damping compensation is performed on the output signal ud* of the current loop controller.
  • the current loop controller includes a current decoupling controller and a stable compensator ; i d * , i q *, id , i q and ⁇ r are processed by the current decoupling controller to obtain u d1 * and u q1 * ; the stable compensator includes a high-pass filter, a first low-pass filter, a second low-pass filter and an addition
  • the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1
  • the DC bus voltage u dc is processed by the second low-pass filter to obtain the voltages u 2 , u 1 and u 2
  • the voltage u 3 is obtained by adding the adder, and the voltage u 3 is brought into the formula After calculation, the output ⁇ ud of the stable compensator is obtained.
  • the specific calculation process of the stable compensator is as follows:
  • u dc_HPF is the value of the bus voltage u dc after high-pass filtering
  • G HPF (s) is the transfer function of the high-pass filter
  • ⁇ H is equal to 2 ⁇ f HPF
  • f HPF is the cut-off frequency of the high-pass filter
  • u dc_PF is the filter The value after the back voltage u dc_HPF passes through the first low-pass filter
  • G LPF1 (s) is the transfer function of the first low-pass filter
  • ⁇ L1 is equal to 2 ⁇ f LPF1
  • f LPF1 is the cut-off frequency of the first low-pass filter
  • G LPF2 (s) is the transfer function of the second low-pass filter
  • ⁇ L2 is equal to 2 ⁇ f LPF2
  • f LPF2 is the cut-off frequency of the second low-pass filter
  • ⁇ is the bus voltage compensation coefficient
  • ⁇ takes a value around 1, n is the order, and n is generally 2 to 4;
  • the control block diagram of the current decoupling controller is shown in FIG. 6 .
  • the outputs of the coupled controller are u d1 * and u q1 * , and the calculation process of u d1 * and u q1 * is shown in the following formulas:
  • G PId (s) is the transfer function of the current loop d-axis PI regulator
  • G PIq (s) is the transfer function of the current loop q-axis PI regulator, respectively
  • k is the control coefficient. Adjusting k can increase the stability of the system.
  • ⁇ f is the permanent magnet flux linkage.
  • Embodiment 3 A method for controlling oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, as shown in Figure 7, a current loop controller is used, and the input signals of the current loop controller are id*, iq *, and d axes Stator current i d , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * are the given motor torque T e * The given current is obtained through the maximum torque-current ratio control distribution, and the active damping compensation is performed on the input signal i q * of the current loop controller.
  • the current loop controller includes a current decoupling controller and a stabilization compensator;
  • the stabilization compensator includes a high-pass filter device, the first low-pass filter, the second low-pass filter and the adder, the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1 , and the DC bus voltage u dc passes through
  • the voltage u 2 is obtained after processing by the second low-pass filter, u 1 and u 2 are added by the adder to obtain the voltage u 3 , and the voltage u 3 is brought into the formula
  • the specific calculation process of the stable compensator is:
  • u dc_HPF is the value of the bus voltage u dc after high-pass filtering
  • G HPF (s) is the transfer function of the high-pass filter
  • ⁇ H is equal to 2 ⁇ f HPF
  • f HPF is the cut-off frequency of the high-pass filter
  • u dc_PF is the filter The value after the back voltage u dc_HPF passes through the first low-pass filter
  • G LPF1 (s) is the transfer function of the first low-pass filter
  • ⁇ L1 is equal to 2 ⁇ f LPF1
  • f LPF1 is the cut-off frequency of the first low-pass filter
  • G LPF2 (s) is the transfer function of the second low-pass filter
  • ⁇ L2 is equal to 2 ⁇ f LPF2
  • f LPF2 is the cut-off frequency of the second low-pass filter
  • ⁇ is the bus voltage compensation coefficient
  • ⁇ takes a value around 1, n is the order, and n is generally 2 to 4;
  • the input i d1 * of the current decoupling controller is obtained after id * and 1 are multiplied, and the input i q1 * of the current decoupling controller is obtained after the multiplication of i q * and ⁇ i q , which is expressed as:
  • i d1 *, i q1 *, id , i q and ⁇ r are processed as input quantities in the current decoupling controller and then output ud * and u q * , the output quantities ud * and u of the current decoupling controller q * is the output of the current loop controller.
  • the calculation process of the current decoupling controller is:
  • G PId (s) is the transfer function of the current loop d-axis PI regulator
  • G PIq (s) is the transfer function of the current loop q-axis PI regulator, respectively
  • k is the control coefficient. Adjusting k can increase the stability of the system.
  • ⁇ f is the permanent magnet flux linkage.
  • Embodiment 4 A method for controlling oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, as shown in FIG. 9 , a current loop controller is used, and the input signals of the current loop controller are id*, iq *, and d axes Stator current i d , q-axis stator current i q and rotor electrical angular velocity ⁇ r , the output signals of the current loop controller are ud * and u q * , id * and i q * are the given motor torque T e * The given current is obtained through the maximum torque-current ratio control distribution, and active damping compensation is performed on the input signal i d * of the current loop controller.
  • the current loop controller includes a current decoupling controller and a stabilization compensator;
  • the stabilization compensator includes a high-pass filter device, the first low-pass filter, the second low-pass filter and the adder, the DC bus voltage u dc is processed by the high-pass filter and the first low-pass filter to obtain the disturbance voltage u 1 , and the DC bus voltage u dc passes through
  • the voltage u 2 is obtained after processing by the second low-pass filter, u 1 and u 2 are added by the adder to obtain the voltage u 3 , and the voltage u 3 is brought into the formula After calculation, the output ⁇ id of the stable compensator is obtained.
  • the specific calculation process of the stable compensator is:
  • u dc_HPF is the value of the bus voltage u dc after high-pass filtering
  • G HPF (s) is the transfer function of the high-pass filter
  • ⁇ H is equal to 2 ⁇ f HPF
  • f HPF is the cut-off frequency of the high-pass filter
  • u dc_PF is the filter The value after the back voltage u dc_HPF passes through the first low-pass filter
  • G LPF1 (s) is the transfer function of the first low-pass filter
  • ⁇ L1 is equal to 2 ⁇ f LPF1
  • f LPF1 is the cut-off frequency of the first low-pass filter
  • G LPF2 (s) is the transfer function of the second low-pass filter
  • ⁇ L2 is equal to 2 ⁇ f LPF2
  • f LPF2 is the cut-off frequency of the second low-pass filter
  • ⁇ is the bus voltage compensation coefficient
  • ⁇ takes a value around 1, n is the order, and n is generally 2 to 4;
  • the input i d1 * of the current decoupling controller is obtained after i d * and ⁇ id are multiplied, and the input i q1 * of the current decoupling controller is obtained after the multiplication of i q * and 1, which is expressed as:
  • i d1 *, i q1 *, id , i q and ⁇ r are processed as input quantities in the current decoupling controller and then output ud * and u q * , the output quantities ud * and u of the current decoupling controller q * is the output of the current loop controller.
  • the calculation process of the current decoupling controller is:
  • G PId (s) is the transfer function of the current loop d-axis PI regulator
  • G PIq (s) is the transfer function of the current loop q-axis PI regulator, respectively
  • k is the control coefficient. Adjusting k can increase the stability of the system.
  • ⁇ f is the permanent magnet flux linkage.
  • Embodiment 5 A method for controlling oscillation suppression of a permanent magnet-assisted synchronous reluctance motor, as shown in FIG. 11 , an MTPA look-up table module is used, and the input signal of the MTPA look-up table module is the motor given torque T e * ,
  • the stable compensator performs active damping compensation for the motor given torque T e * ,
  • the stable compensator includes a high-pass filter, a first low-pass filter, a second low-pass filter and an adder, and the DC bus voltage u dc passes through the high-pass filter in turn
  • the filter and the first low-pass filter are processed to obtain the disturbance voltage u 1
  • the DC bus voltage u dc is processed by the second low-pass filter to obtain the voltage u 2
  • u 1 and u 2 are added by the adder to obtain the voltage u 3
  • the voltage u 3 is brought into the formula
  • the output ⁇ T e of the stable compensator is obtained, and the specific calculation
  • u dc_HPF is the value of the bus voltage u dc after high-pass filtering
  • G HPF (s) is the transfer function of the high-pass filter
  • ⁇ H is equal to 2 ⁇ f HPF
  • f HPF is the cut-off frequency of the high-pass filter
  • u dc_PF is the filter The value after the back voltage u dc_HPF passes through the first low-pass filter
  • G LPF1 (s) is the transfer function of the first low-pass filter
  • ⁇ L1 is equal to 2 ⁇ f LPF1
  • f LPF1 is the cut-off frequency of the first low-pass filter
  • G LPF2 (s) is the transfer function of the second low-pass filter
  • ⁇ L2 is equal to 2 ⁇ f LPF2
  • f LPF2 is the cut-off frequency of the second low-pass filter
  • ⁇ is the bus voltage compensation coefficient
  • ⁇ takes a value around 1, n is the order, and n is generally 2 to 4;
  • the segment modulation method that is, the asynchronous modulation method is used when [0 ⁇ f 0 ), the 15-frequency synchronous modulation method is used when [f 0 ⁇ f 1 ), and the 12-frequency synchronous modulation method is used when [f 1 ⁇ f 2 ).
  • [f 2 ⁇ f 3 ) adopt the 9-frequency synchronous modulation method
  • [f 3 ⁇ f 4 ) adopt the 7-frequency synchronous modulation method
  • [f 4 ⁇ f 5 ) adopt the 5-frequency synchronous modulation method
  • [f 5 ⁇ f 6 ) the frequency division synchronous modulation method
  • [f 6 ⁇ f 7 ] adopts the square wave modulation method, where f 0 is one-fifteenth of the switching frequency in the asynchronous modulation stage, and f 1 is the maximum allowable power device.
  • f 2 is one-twelfth of the maximum allowable switching frequency of the power device
  • f 3 is one-ninth of the maximum allowable switching frequency of the power device
  • f 4 is the maximum allowable switching frequency of the power device.
  • One-seventh of the switching frequency of the power device f5 is one - fifth of the maximum allowable switching frequency of the power device
  • f6 is one-third of the maximum allowable switching frequency of the power device
  • f7 is the highest frequency of the motor.
  • the maximum allowable switching frequency of the power device is 600Hz
  • the switching frequency in the asynchronous modulation stage is 450Hz
  • f1 ⁇ f 6 is the frequency that the "motor can only run up to", which can be entered in advance under the conditions of the switching frequency and the system permitting.
  • the maximum switching frequency of the traction inverter is limited by heat dissipation conditions and is often only a few hundred Hz, while the maximum operating frequency of the traction motor can reach about 300 Hz.
  • the internal use of asynchronous modulation the carrier ratio changes in a large range, and the motor runs in a high frequency band, the voltage utilization rate is low, and the control performance is poor. Therefore, a segmented modulation method combining a variety of modulation methods is used in PWM modulation.
  • the segmented modulation method employs different modulation strategies at different motor frequencies.
  • the harmonic distribution of different modulation strategies is different, and the system oscillation is also related to the harmonic distribution.
  • the current modulation method can be switched to the next modulation method to change the harmonic distribution to alleviate the oscillation. , which makes the oscillation suppression effect of the control method better.
  • the modulation algorithm can be switched to the 9-frequency synchronous modulation method to change the harmonic distribution;
  • the modulation algorithm can be switched to the 7-frequency synchronous modulation method to change the harmonic distribution;
  • the motor runs in the 7-frequency synchronous modulation method, the system oscillates. Switch the modulation algorithm to the divide-by-5 synchronous modulation method to change the harmonic distribution.

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Abstract

本公开涉及城轨永磁牵引传动系统中电机振荡抑制方法，具体振荡抑制中的主动阻尼补偿方法，具体为一种永磁辅助同步磁阻电机振荡抑制的控制方法，解决了背景技术中的技术问题。本公开是在永磁辅助同步磁阻电机原有矢量控制方法的基础上再通过电压q轴主动阻尼补偿法、电压d轴主动阻尼补偿法、电流q轴主动阻尼补偿法、电流d轴主动阻尼补偿法、转矩主动阻尼补偿法以及调制变化法解决了城轨永磁牵引系统阻抗不匹配引起的直流侧振荡的技术问题，提升了在原有矢量控制方法控制下城轨永磁牵引系统的稳定性，而且在实现系统稳定性的同时，不增加系统的硬件成本。

Description

一种永磁辅助同步磁阻电机振荡抑制的控制方法
相关申请的交叉引用
本公开基于申请号为202011566465.5、申请日为2020年12月25日的中国专利申请提出，并要求该中国专利申请的优先权，该中国专利申请的全部内容在此以引入方式并入本申请。
技术领域
本公开涉及城轨永磁牵引传动系统中电机振荡抑制方法，具体振荡抑制中的主动阻尼补偿方法，具体为一种永磁辅助同步磁阻电机振荡抑制的控制方法。
背景技术
城轨永磁牵引传动系统多采用直流牵引网供电，当系统的牵引功率增大至一定程度时，其直流侧电压会出现持续振荡，进而导致直流侧电流及电机的输出转矩均发生振荡，影响车辆运行的稳定性和舒适度，严重时会导致变流器发生过压和过流故障，触发牵引控制单元(Traction Control Unit，TCU)封锁脉冲，使城轨车辆丧失牵引力。
该不稳定现象是由直流侧LC滤波环节输出阻抗与逆变器-电机系统输入阻抗不匹配造成的，可以通过改变逆变器直流侧输出阻抗和逆变器-电机系统输入阻抗两个方面进行抑制，分别被称为被动阻尼补偿与主动阻尼补偿。在工程上，常采用在直流侧串入电阻、加大支撑电容等被动阻尼补偿方式，但串入电阻会增加功率损耗，降低系统效率，而增大支撑电容则又会受到变流器重量和空间的限制。
对城轨永磁辅助同步磁阻牵引系统振荡机理进行分析，假设永磁牵引系统的控制器带宽无限大，逆变器为连续能量转换系统，牵引电机的输出转矩能完美跟随指令值，则逆变器-电机系统可视为理想的恒功率负载，该系统简化后的恒功率负载模型如图1所示，图1中网侧电压E w为车辆牵引逆变器供电，R表 示为线路电阻和电感电阻的和，L是滤波电感，C是支撑电容，u dc是输入到逆变器侧的直流母线电压，Z m是逆变器加电机的等效阻抗。
在图1所示理想的恒功率负载模型中，逆变器-电机系统呈现负阻抗特性Y，变流器直流侧前端阻尼系数以及传动系统整体阻尼系数为：
Figure PCTCN2021124380-appb-000001
系统稳定的基本条件是系统阻尼系数为正，从而确定理想模型下的系统稳定判据为
Figure PCTCN2021124380-appb-000002
其中P 0、u dc分别是稳态输出功率与输入到逆变器侧的直流母线电压。由上式不稳定判据可知，随着永磁牵引系统功率的增大，系统逐渐失稳，这和试验时出现的现象是一致的，可以确定逆变器直流侧参数的取值和逆变器-电机系统呈现的负阻抗特性是该类不稳定现象的根本原因。本公开就是通过主动阻尼补偿方式对永磁牵引系统阻抗不匹配问题进行研究。
永磁辅助同步磁阻电机在d-q坐标系下的电压方程可表示为：
Figure PCTCN2021124380-appb-000003
式中u d、u q为d、q轴定子电压，R s是定子电阻，ω r是电机转子电角速度，L d、L q分别是电机d轴、q轴电感，i d、i q为d、q轴定子电流，ψ f是永磁体磁链；
永磁辅助同步磁阻电机的电磁转矩方程可表示为：T e＝n p [ψ fi q+(L d-L q)i di q]，式中：T e为电机电磁转矩，n p为电机极对数。
车辆往往通过手柄给定整车牵引力指令，进而分配到每个电机上。永磁辅助同步磁阻电机矢量控制算法框图如图2所示。如图2所示，第①部分，永磁辅助同步磁阻电机通过旋转变压器测得电机转子位置θ，θ经微分后得到电机电角速度ω r。第②部分，电流变换器将测量得到的电机电流i a、i b经过Clark变换 和Park变换，得到d-q坐标下的电流i d和i q。第③部分，电机给定转矩T e *通过MTPA查表模块(Maximum Torque Per Ample，MTPA)后分配得到给定电流i d *和i q *，该模块是按照标定的最大转矩电流比算法进行计算的。第④部分，i d*、i q*、i d、i q、ω r和u dc作为电流环控制器的输入，u d*和u q*是电流环控制器的输出。第⑤部分，u d*、u q*、θ、ω r和母线u dc输入到分段PWM调制模块中以产生PWM脉冲到第⑥部分逆变器中。本公开是在图2所示的永磁辅助同步磁阻电机矢量控制算法框图的基础上进行主动阻尼补偿的。
发明内容
有鉴于此，本公开旨在解决城轨永磁牵引系统阻抗不匹配引起的直流侧振荡的技术问题，提供了一种永磁辅助同步磁阻电机振荡抑制的控制方法。
本公开解决其技术问题采用了电压q轴主动阻尼补偿法，具体是：一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输出信号u q *进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；i d*、i q*、i d、i q和ω r经过电流解耦控制器处理后得到u d1 *和u q1 *；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000004
后，计算得到稳定补偿器的输出Δu q，u d1 *和1相乘后的结果即为电流环控制器的输出u d*，u q1 *和Δu q相乘后的结果即为电流环控制器经过主动阻尼补偿后的输出u q *。
本公开解决其技术问题还采用了电压d轴主动阻尼补偿法，具体是：一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控 制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输出信号u d*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；i d*、i q*、i d、i q和ω r经过电流解耦控制器处理后得到u d1 *和u q1 *；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000005
后，计算得到稳定补偿器的输出Δu d，u d1 *和Δu d相乘后的结果即为电流环控制器经过主动阻尼补偿后的输出u d*，u q1 *和1相乘后的结果即为电流环控制器的输出u q *。
本公开解决其技术问题还采用了电流q轴主动阻尼补偿法，具体是：一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输入信号i q*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000006
后，计算得到稳定补偿器的输出Δi q，i d *和1相乘后得到电流解耦控制器的输入i d1*，i q *和Δi q相乘后得到电流解耦控制器的输入i q1 *；i d1*、i q1*、i d、i q和ω r作为输入量在电流解耦控制器中处理后输出u d *和u q *，电流解耦控制器的输出量u d *和u q *即为电流环控制器的输出。
本公开解决其技术问题还采用了电流d轴主动阻尼补偿法，具体是：一种 永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输入信号i d*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000007
后，计算得到稳定补偿器的输出Δi d，i d *和Δi d相乘后得到电流解耦控制器的输入i d1*，i q *和1相乘后得到电流解耦控制器的输入i q1 *；i d1*、i q1*、i d、i q和ω r作为输入量在电流解耦控制器中处理后输出u d *和u q *，电流解耦控制器的输出量u d *和u q *即为电流环控制器的输出。
本公开解决其技术问题还采用了转矩主动阻尼补偿法，具体是：一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了MTPA查表模块，MTPA查表模块的输入信号为电机给定转矩T e *，通过稳定补偿器对电机给定转矩T e *进行主动阻尼补偿，稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000008
后，计算得到稳定补偿器的输出ΔT e，将ΔT e加到电机给定转矩T e *上实现抑制振荡。
进一步的，包括分段PWM调制模块，分段PWM调制模块采用多种调制方法相结合的分段调制方法，当控制系统发生振荡时，将当前的调制方法切换至下一调制方法；分段调制方法分别为：即在[0～f 0)时采用异步调制法，[f 0～f 1)时采用15分频同步调制法，[f 1～f 2)时采用12分频同步调制法，[f 2～f 3)时采用9分频同步调制法，[f 3～f 4)时采用7分频同步调制法，[f 4～f 5)时采用5分频同 步调制法，[f 5～f 6)时采用3分频同步调制法，[f 6～f 7]采用方波调制方法；其中f 0为异步调制阶段开关频率的十五分之一，f 1为功率器件最高允许的开关频率的十五分之一，f 2为功率器件最高允许的开关频率的十二分之一，f 3为功率器件最高允许的开关频率的九分之一，f 4为功率器件最高允许的开关频率的七分之一，f 5为功率器件最高允许的开关频率的五分之一，f 6为功率器件最高允许的开关频率的三分之一，f 7为电机的最高频率。这里的f 1～f 6是“电机最高只能运行到”的频率，在开关频率和系统允许的条件下是可以提前进入的。在城轨地铁等大功率牵引系统中，牵引逆变器的最高开关频率受到散热条件的限制往往只有几百赫兹，而牵引电机的运行频率最高可以达到300赫兹左右，如在整个调速范围内采用异步调制，载波比变化的范围大，且电机运行在高频段，电压利用率低，控制性能不佳。因此在采用PWM调制采用多种调制方法相结合的分段调制方法。分段调制方法在不同的电机频率下采用不同的调制策略。不同的调制策略的谐波分布是不一样的，系统振荡与谐波分布也是有关的，当系统发生振荡时，可通过切换当前调制方式至下一调制方法以改变谐波分布情况从而来缓解振荡，这使所述控制方法的抑制振荡的效果更好。
本公开是在永磁辅助同步磁阻电机原有矢量控制方法的基础上再通过电压q轴主动阻尼补偿法、电压d轴主动阻尼补偿法、电流q轴主动阻尼补偿法、电流d轴主动阻尼补偿法、转矩主动阻尼补偿法和调制变化法解决了城轨永磁牵引系统阻抗不匹配引起的直流侧振荡的技术问题，提升了在原有矢量控制方法控制下城轨永磁牵引系统的稳定性，而且在实现系统稳定性的同时，不增加系统的硬件成本。
附图说明
图1为本公开背景技术中所述牵引传动系统恒功率负载模型的结构示意图。
图2为本公开背景技术中所述永磁辅助同步磁阻电机矢量控制框图。
图3为本公开实施例1中所述电流环控制器的控制框图。
图4为本公开实施例1中所述电流解耦控制器的控制框图。
图5为本公开实施例2中所述电流环控制器的控制框图。
图6为本公开实施例2中所述电流解耦控制器的控制框图。
图7为本公开实施例3中所述电流环控制器的控制框图。
图8为本公开实施例3中所述电流解耦控制器的控制框图。
图9为本公开实施例4中所述电流环控制器的控制框图。
图10为本公开实施例4中所述电流解耦控制器的控制框图。
图11为本公开实施例5中所述添加转矩补偿的控制框图。
图12为本公开所述分段PWM调制模块的调制框图。
具体实施方式
参照图1-图12，对本公开所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法进行详细说明。
实施例1：一种永磁辅助同步磁阻电机振荡抑制的控制方法，如图3所示，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输出信号u q *进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；i d*、i q*、i d、i q和ω r经过电流解耦控制器处理后得到u d1 *和u q1 *；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000009
后，计算得到稳定补偿器的输出Δu q，稳定补偿器的具体计算过程如下：
Figure PCTCN2021124380-appb-000010
Figure PCTCN2021124380-appb-000011
u 1＝u dc_PF
Figure PCTCN2021124380-appb-000012
u 3＝u 2+u 1
Figure PCTCN2021124380-appb-000013
式中，u dc_HPF为母线电压u dc经过高通滤波后的值，G HPF(s)是高通滤波器的传递函数，ω H等于2πf HPF，f HPF为高通滤波器的截止频率；u dc_PF为滤波后电压u dc_HPF经过第一低通滤波器后的值，G LPF1(s)是第一低通滤波器的传递函数，ω L1等于2πf LPF1，f LPF1为第一低通滤波器的截止频率；G LPF2(s)是第二低通滤波器的传递函数，ω L2等于2πf LPF2，f LPF2为第二低通滤波器的截止频率；λ为母线电压补偿系数，λ取1左右的值，n为阶数，n一般为2～4；
u d1 *和1相乘后的结果即为电流环控制器的输出u d*，u q1 *和Δu q相乘后的结果即为电流环控制器经过主动阻尼补偿后的输出u q *，则电流环控制器的输出表示为：
Figure PCTCN2021124380-appb-000014
进一步的，作为本公开实施例1中所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法的一种具体实施方式，电流解耦控制器的控制框图如图4所示，电流解耦控制器的输出为u d1 *和u q1 *，u d1 *和u q1 *的计算过程如下列公式所示：
Figure PCTCN2021124380-appb-000015
式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，调节k能够增加系统的稳定性，ψ f为永磁体磁链。
实施例2：一种永磁辅助同步磁阻电机振荡抑制的控制方法，如图5所示， 采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输出信号u d*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；i d*、i q*、i d、i q和ω r经过电流解耦控制器处理后得到u d1 *和u q1 *；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000016
后，计算得到稳定补偿器的输出Δu d，稳定补偿器的具体计算过程如下：
Figure PCTCN2021124380-appb-000017
Figure PCTCN2021124380-appb-000018
u 1＝u dc_PF
Figure PCTCN2021124380-appb-000019
u 3＝u 2+u 1
Figure PCTCN2021124380-appb-000020
式中，u dc_HPF为母线电压u dc经过高通滤波后的值，G HPF(s)是高通滤波器的传递函数，ω H等于2πf HPF，f HPF为高通滤波器的截止频率；u dc_PF为滤波后电压u dc_HPF经过第一低通滤波器后的值，G LPF1(s)是第一低通滤波器的传递函数，ω L1等于2πf LPF1，f LPF1为第一低通滤波器的截止频率；G LPF2(s)是第二低通滤波器的传递函数，ω L2等于2πf LPF2，f LPF2为第二低通滤波器的截止频率；λ为母线电压补偿系数，λ取1左右的值，n为阶数，n一般为2～4；
u d1 *和Δu d相乘后的结果即为电流环控制器经过主动阻尼补偿后的输出u d*，u q1 *和1相乘后的结果即为电流环控制器的输出u q *，则电流环控制器的输出表 示为：
Figure PCTCN2021124380-appb-000021
进一步的，作为本公开实施例2中所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法的一种具体实施方式，电流解耦控制器的控制框图如图6所示，电流解耦控制器的输出为u d1 *和u q1 *，u d1 *和u q1 *的计算过程如下列公式所示：
Figure PCTCN2021124380-appb-000022
式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，调节k能够增加系统的稳定性，ψ f为永磁体磁链。
实施例3：一种永磁辅助同步磁阻电机振荡抑制的控制方法，如图7所示，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输入信号i q*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000023
后，计算得到稳定补偿器的输出Δi q，则稳定补偿器的具体计算过程为：
Figure PCTCN2021124380-appb-000024
Figure PCTCN2021124380-appb-000025
u 1＝u dc_PF
Figure PCTCN2021124380-appb-000026
u 3＝u 2+u 1
Figure PCTCN2021124380-appb-000027
式中，u dc_HPF为母线电压u dc经过高通滤波后的值，G HPF(s)是高通滤波器的传递函数，ω H等于2πf HPF，f HPF为高通滤波器的截止频率；u dc_PF为滤波后电压u dc_HPF经过第一低通滤波器后的值，G LPF1(s)是第一低通滤波器的传递函数，ω L1等于2πf LPF1，f LPF1为第一低通滤波器的截止频率；G LPF2(s)是第二低通滤波器的传递函数，ω L2等于2πf LPF2，f LPF2为第二低通滤波器的截止频率；λ为母线电压补偿系数，λ取1左右的值，n为阶数，n一般为2～4；
i d *和1相乘后得到电流解耦控制器的输入i d1*，i q *和Δi q相乘后得到电流解耦控制器的输入i q1 *，则用公式表示为：
Figure PCTCN2021124380-appb-000028
i d1*、i q1*、i d、i q和ω r作为输入量在电流解耦控制器中处理后输出u d *和u q *，电流解耦控制器的输出量u d *和u q *即为电流环控制器的输出。
进一步的，作为本公开实施例3中所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法的一种具体实施方式，如图8所示，电流解耦控制器的计算过程为：
Figure PCTCN2021124380-appb-000029
式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流 环q轴PI调节器的传递函数，k为控制系数，调节k能够增加系统的稳定性，ψ f为永磁体磁链。
实施例4：一种永磁辅助同步磁阻电机振荡抑制的控制方法，如图9所示，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，对电流环控制器的输入信号i d*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000030
后，计算得到稳定补偿器的输出Δi d，稳定补偿器的具体计算过程为：
Figure PCTCN2021124380-appb-000031
Figure PCTCN2021124380-appb-000032
u 1＝u dc_PF
Figure PCTCN2021124380-appb-000033
u 3＝u 2+u 1
Figure PCTCN2021124380-appb-000034
式中，u dc_HPF为母线电压u dc经过高通滤波后的值，G HPF(s)是高通滤波器的传递函数，ω H等于2πf HPF，f HPF为高通滤波器的截止频率；u dc_PF为滤波后电压u dc_HPF经过第一低通滤波器后的值，G LPF1(s)是第一低通滤波器的传递函数，ω L1等于2πf LPF1，f LPF1为第一低通滤波器的截止频率；G LPF2(s)是第二低通滤波器的传递函数，ω L2等于2πf LPF2，f LPF2为第二低通滤波器的截止频率；λ为母线电压补偿系数，λ取1左右的值，n为阶数，n一般为2～4；
i d *和Δi d相乘后得到电流解耦控制器的输入i d1*，i q *和1相乘后得到电流解耦控制器的输入i q1 *，则用公式表达为：
Figure PCTCN2021124380-appb-000035
i d1*、i q1*、i d、i q和ω r作为输入量在电流解耦控制器中处理后输出u d *和u q *，电流解耦控制器的输出量u d *和u q *即为电流环控制器的输出。
进一步的，作为本公开实施例4中所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法的一种具体实施方式，如图10所示，电流解耦控制器的计算过程为：
Figure PCTCN2021124380-appb-000036
式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，调节k能够增加系统的稳定性，ψ f为永磁体磁链。
实施例5：一种永磁辅助同步磁阻电机振荡抑制的控制方法，如图11所示，采用了MTPA查表模块，MTPA查表模块的输入信号为电机给定转矩T e *，通过稳定补偿器对电机给定转矩T e *进行主动阻尼补偿，稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
Figure PCTCN2021124380-appb-000037
后，计算得到稳定补偿器的输出ΔT e，ΔT e的具体计算过程为：
Figure PCTCN2021124380-appb-000038
Figure PCTCN2021124380-appb-000039
u 1＝u dc_PF
Figure PCTCN2021124380-appb-000040
u 3＝u 2+u 1
Figure PCTCN2021124380-appb-000041
式中，u dc_HPF为母线电压u dc经过高通滤波后的值，G HPF(s)是高通滤波器的传递函数，ω H等于2πf HPF，f HPF为高通滤波器的截止频率；u dc_PF为滤波后电压u dc_HPF经过第一低通滤波器后的值，G LPF1(s)是第一低通滤波器的传递函数，ω L1等于2πf LPF1，f LPF1为第一低通滤波器的截止频率；G LPF2(s)是第二低通滤波器的传递函数，ω L2等于2πf LPF2，f LPF2为第二低通滤波器的截止频率；λ为母线电压补偿系数，λ取1左右的值，n为阶数，n一般为2～4；
将ΔT e加到电机给定转矩T e *上实现抑制振荡。进一步的，作为本公开所述的一种永磁辅助同步磁阻电机的稳定性控制方法的一种具体实施方式，如图12所示，分段PWM调制模块采用多种调制方法相结合的分段调制方法，即在[0～f 0)时采用异步调制法，[f 0～f 1)时采用15分频同步调制法，[f 1～f 2)时采用12分频同步调制法，[f 2～f 3)时采用9分频同步调制法，[f 3～f 4)时采用7分频同步调制法，[f 4～f 5)时采用5分频同步调制法，[f 5～f 6)时采用3分频同步调制法，[f 6～f 7]采用方波调制方法，其中f 0为异步调制阶段开关频率的十五分之一，f 1为功率器件最高允许的开关频率的十五分之一，f 2为功率器件最高允许的开关频率的十二分之一，f 3为功率器件最高允许的开关频率的九分之一，f 4为功率器件最高允许的开关频率的七分之一，f 5为功率器件最高允许的开关频率的五分之一，f 6为功率器件最高允许的开关频率的三分之一，f 7为电机的最高频率。本实施例中，如图12所示，功率器件最高允许的开关频率为600Hz，异步调制阶段开关频率为450Hz，那么f 0＝450Hz/15＝30Hz，f 1＝600Hz/15＝40Hz，f 2＝600Hz /12＝50Hz，f 3＝600Hz/9＝66.66Hz，f 4＝600Hz/7＝85.71Hz，f 5＝600Hz/5＝120Hz，f 6＝600Hz/3＝200Hz，这里的f 1～f 6是“电机最高只能运行到”的频率，在开关频率和系统允许的条件下是可以提前进入的。在城轨、地铁等大功率牵引系统中，牵引逆变器的最高开关频率受到散热条件的限制往往只有几百赫兹，而牵引电机的运行频率最高可以达到300赫兹左右，如在整个调速范围内采用异步调制，载波比变化的范围大，且电机运行在高频段，电压利用率低，控制性能不佳。因此在采用PWM调制采用多种调制方法相结合的分段调制方法。分段调制方法在不同的电机频率下采用不同的调制策略。不同的调制策略的谐波分布是不一样的，系统振荡与谐波分布也是有关的，当系统发生振荡时，可通过切换当前调制方式至下一调制方法以改变谐波分布情况从而来缓解振荡，这使所述控制方法的抑制振荡的效果更好。比如具体实施例中，当电机运行在12分频同步调制方法时控制系统发生振荡，这时可将调制算法切换到9分频同步调制方法，以改变谐波分布情况；当电机运行在9分频同步调制方法时控制系统发生振荡，这时可将调制算法切换到7分频同步调制方法，以改变谐波分布情况；当电机运行在7分频同步调制方法时系统发生振荡，这时可将调制算法切换到5分频同步调制方法，以改变谐波分布情况。
以上具体结构是对本公开的较佳实施例进行了具体说明，但本公开创造并不限于所述实施例，熟悉本领域的技术人员在不违背本公开精神的前提下还可做出种种的等同变形或者替换，这些等同的变形或替换均包含在本申请权利要求所限定的范围内。

Claims (10)

  1. 一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，其中，对电流环控制器的输出信号u q *进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；i d*、i q*、i d、i q和ω r经过电流解耦控制器处理后得到u d1 *和u q1 *；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
    Figure PCTCN2021124380-appb-100001
    后，计算得到稳定补偿器的输出Δu q，u d1 *和1相乘后的结果即为电流环控制器的输出u d*，u q1 *和Δu q相乘后的结果即为电流环控制器经过主动阻尼补偿后的输出u q *。
  2. 根据权利要求1所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法，其中，电流解耦控制器的计算过程如下列公式所示：
    Figure PCTCN2021124380-appb-100002
    式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，ψ f为永磁体磁链。
  3. 一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，其中，对电流环控制器的输出信号u d*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；i d*、i q*、i d、i q和ω r经过电流解耦控制器处理后得到u d1 *和u q1 *；稳定补偿器包 括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
    Figure PCTCN2021124380-appb-100003
    后，计算得到稳定补偿器的输出Δu d，u d1 *和Δu d相乘后的结果即为电流环控制器经过主动阻尼补偿后的输出u d*，u q1 *和1相乘后的结果即为电流环控制器的输出u q *。
  4. 根据权利要求3所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法，其中，电流解耦控制器的计算过程如下列公式所示：
    Figure PCTCN2021124380-appb-100004
    式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，ψ f为永磁体磁链。
  5. 一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，其中，对电流环控制器的输入信号i q*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
    Figure PCTCN2021124380-appb-100005
    后，计算得到稳定补偿器的输出Δi q，i d *和1相乘后得到电流解耦控制器的输入i d1*，i q *和Δi q相乘后得到电流解耦控制器的输入i q1 *；i d1*、i q1*、i d、i q和ω r作为输入量在电流解耦控制器中处理后输出u d *和u q *，电流解耦控制器的输出u d *和u q *即为电流环控制器的输出。
  6. 根据权利要求5所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法， 其中，电流解耦控制器的计算过程如下列公式所示：
    Figure PCTCN2021124380-appb-100006
    式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，ψ f为永磁体磁链。
  7. 一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了电流环控制器，电流环控制器的输入信号为i d*、i q*、d轴定子电流i d、q轴定子电流i q和转子电角速度ω r，电流环控制器的输出信号为u d*和u q *，i d*和i q*是电机给定转矩T e *通过最大转矩电流比控制分配得到给定电流，其中，对电流环控制器的输入信号i d*进行主动阻尼补偿，电流环控制器包括电流解耦控制器和稳定补偿器；稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
    Figure PCTCN2021124380-appb-100007
    后，计算得到稳定补偿器的输出Δi d，i d *和Δi d相乘后得到电流解耦控制器的输入i d1*，i q *和1相乘后得到电流解耦控制器的输入i q1 *；i d1*、i q1*、i d、i q和ω r作为输入量在电流解耦控制器中处理后输出u d *和u q *，电流解耦控制器的输出u d *和u q *即为电流环控制器的输出。
  8. 根据权利要求7所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法，其中，电流解耦控制器的计算过程如下列公式所示：
    Figure PCTCN2021124380-appb-100008
    式中，G PId(s)是电流环d轴PI调节器的传递函数，G PIq(s)分别为电流环q轴PI调节器的传递函数，k为控制系数，ψ f为永磁体磁链。
  9. 一种永磁辅助同步磁阻电机振荡抑制的控制方法，采用了MTPA查表模块，MTPA查表模块的输入信号为电机给定转矩T e *，其中，通过稳定补偿器 对电机给定转矩T e *进行主动阻尼补偿，稳定补偿器包括高通滤波器、第一低通滤波器、第二低通滤波器和加法器，直流母线电压u dc依次经过高通滤波器和第一低通滤波器处理后得到扰动电压u 1，直流母线电压u dc经过第二低通滤波器处理后得到电压u 2，u 1和u 2经过加法器相加得到电压u 3，电压u 3带入公式
    Figure PCTCN2021124380-appb-100009
    后，计算得到稳定补偿器的输出ΔT e，将ΔT e加到电机给定转矩T e *上实现抑制振荡。
  10. 根据权利要求9所述的一种永磁辅助同步磁阻电机振荡抑制的控制方法，包括分段PWM调制模块，其中，分段PWM调制模块采用多种调制方法相结合的分段调制方法，当控制系统发生振荡时，将当前的调制方法切换至下一调制方法；分段调制方法分别为：在[0～f 0)时采用异步调制法，[f 0～f 1)时采用15分频同步调制法，[f 1～f 2)时采用12分频同步调制法，[f 2～f 3)时采用9分频同步调制法，[f 3～f 4)时采用7分频同步调制法，[f 4～f 5)时采用5分频同步调制法，[f 5～f 6)时采用3分频同步调制法，[f 6～f 7]采用方波调制方法；其中f 0为异步调制阶段开关频率的十五分之一，f 1为功率器件最高允许的开关频率的十五分之一，f 2为功率器件最高允许的开关频率的十二分之一，f 3为功率器件最高允许的开关频率的九分之一，f 4为功率器件最高允许的开关频率的七分之一，f 5为功率器件最高允许的开关频率的五分之一，f 6为功率器件最高允许的开关频率的三分之一，f 7为电机的最高频率。
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