WO2021254536A2 - 谐振变换器及其同步整流控制方法 - Google Patents

谐振变换器及其同步整流控制方法 Download PDF

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Publication number
WO2021254536A2
WO2021254536A2 PCT/CN2021/111126 CN2021111126W WO2021254536A2 WO 2021254536 A2 WO2021254536 A2 WO 2021254536A2 CN 2021111126 W CN2021111126 W CN 2021111126W WO 2021254536 A2 WO2021254536 A2 WO 2021254536A2
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Prior art keywords
current
parameter
synchronous rectification
resonant converter
limit value
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PCT/CN2021/111126
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English (en)
French (fr)
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WO2021254536A3 (zh
Inventor
王雷
童文平
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深圳市正浩创新科技股份有限公司
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Priority to EP21826699.7A priority Critical patent/EP4318914A4/en
Priority to JP2021567068A priority patent/JP7133105B2/ja
Priority to KR1020227005779A priority patent/KR102628891B1/ko
Publication of WO2021254536A2 publication Critical patent/WO2021254536A2/zh
Publication of WO2021254536A3 publication Critical patent/WO2021254536A3/zh
Priority to US18/476,814 priority patent/US20240022177A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0032—Control circuits allowing low power mode operation, e.g. in standby mode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the application relates to the field of converters, and in particular to a resonant converter and a synchronous rectification control method thereof.
  • the method of adjusting the bus voltage function is usually used to solve the problem of excessive derating of the Vds voltage of the secondary side synchronous rectification power tube in the steady state, but it cannot solve the problem of the Vds voltage of the secondary side synchronous rectification power tube in the dynamic process.
  • the issue of derating in the Super League For example, the power of the resonant converter is increased from 7.5KW to 10KW, the output rated current is increased from 75A to 100A, and the current limit is 110% (110A). Under the condition of full load and no load, the voltage spike of the secondary side synchronous rectifier power tube Up to 290V, the problem of over-derating is serious.
  • a resonant converter and a synchronous rectification control method thereof are provided.
  • a synchronous rectification control method for a resonant converter includes: obtaining the output current of the resonant converter; if the output current is greater than a first current threshold, obtaining a first parameter; if the output current is less than a second current threshold, obtaining a second parameter, A current threshold is greater than the second current threshold, the first parameter is greater than the second parameter; the first current hysteresis is established according to the first parameter and the second current hysteresis is established according to the second parameter, the parameters of the first current hysteresis It is the first parameter, and the parameter of the second current hysteresis is the second parameter; the first current hysteresis and the second current hysteresis are used to control the synchronous rectification of the resonant converter under load-cutting conditions.
  • a synchronous rectification control method of a resonant converter includes:
  • the output current is less than the second current threshold, a second parameter is obtained, the first current threshold is greater than the second current threshold, and the first parameter is greater than the second parameter;
  • a first current hysteresis is established according to the first parameter and a second current hysteresis is established according to the second parameter, the parameters of the first current hysteresis are the first parameters, and the second current hysteresis is The parameter is the second parameter; wherein, the first parameter includes a first lower limit value and a first upper limit value, the second parameter includes a second upper limit value and a second lower limit value, and the first parameter The lower limit value is less than the second current threshold;
  • the first current hysteresis and the second current hysteresis are used to control the secondary side synchronous rectification power tube of the resonant converter in sequence; when the output current is less than or equal to the first lower limit, Then control the secondary side synchronous rectification power tube to turn off; when the secondary side synchronous rectification power tube is turned off for greater than the time threshold, control the secondary side synchronous rectification power tube to turn on; switching from the first current hysteresis loop to the second current hysteresis loop has Delay.
  • the secondary side synchronous rectifier power tube will be turned off first, and then turned on after a preset time delay, and enters the control of the second current hysteresis; If the output current is less than or equal to the second lower limit value, the secondary side synchronous rectification power tube is controlled to be turned off;
  • the second current hysteresis and the first current hysteresis are used to control the secondary synchronous rectification power tube of the resonant converter in sequence;
  • the secondary side synchronous rectifier power tube is controlled to turn on; as the output current changes, if the output current is greater than or equal to the first
  • the upper limit value still controls the opening of the secondary side synchronous rectifier power tube.
  • a resonant converter that performs synchronous rectification control by the above-mentioned synchronous rectification control method.
  • Figure 1a is a schematic diagram of a simple circuit structure of the resonant converter of the present application.
  • Figure 1b is a schematic diagram of the detailed circuit structure of the resonant converter of the present application.
  • FIG. 2 is a schematic flowchart of an embodiment of a synchronous rectification control method for a resonant converter of the present application.
  • FIG. 3 is a schematic diagram of hysteresis curves of the first current hysteresis and the second current hysteresis of the present application.
  • step S205 is a schematic diagram of a specific flow of step S205 in the synchronous rectification control method of the resonant converter in the embodiment of FIG. 2.
  • FIG. 5 is a schematic diagram of the hysteresis curve of the first current hysteresis loop and the second current hysteresis loop and the switching state of the secondary side synchronous rectification power tube under the load cut condition of the resonant converter of the present application.
  • FIG. 6 is a schematic diagram of the hysteresis curve of the first current hysteresis loop and the second current hysteresis loop and the switching state of the secondary synchronous rectifier power tube under the non-load-cutting condition of the resonant converter of the present application.
  • first and second in this application are only used for descriptive purposes, and cannot be understood as indicating or implying relative importance or implicitly indicating the number of indicated technical features.
  • plural means at least two, such as two, three, etc., unless specifically defined otherwise.
  • the terms “including” and “having” and any variations thereof are intended to cover non-exclusive inclusions. For example, a process, method, system, product, or device that includes a series of steps or units is not limited to the listed steps or units, but optionally includes unlisted steps or units, or optionally also includes Other steps or units inherent in these processes, methods, products or equipment.
  • the resonant converter is a resonant circuit that achieves a constant output voltage by controlling the switching frequency (adjusting the frequency). Its advantages are: to realize the zero voltage switch (Zero Voltage Switch, ZVS) of the primary side main MOS switch (Q 1 -Q 4 ) and the zero current turn off (Zero Current Switch) of the secondary side synchronous rectifier diodes (D 1 , D 2) Switch, ZCS), through the soft switching technology, the switching loss can be reduced, and the efficiency and power density of the resonant converter can be improved.
  • ZVS Zero voltage switch
  • ZCS Zero Current Switch
  • the synchronous rectifier diodes (D 1 , D 2 ) in Figure 1a are the parasitic diodes of the secondary-side synchronous rectification power tube.
  • Figure 1a omits the secondary-side synchronous rectification power tube and only shows the parasitic diodes (D 1 , D 2 ) and Parasitic capacitance (C1, C2).
  • the resonant converter provided by this embodiment includes a switching network (not marked in the figure), a resonant network (not marked in the figure), a center-tapped transformer T, a secondary synchronous rectifier power tube Q5 and its internal parasitic diode D1 and Its parasitic capacitance C1, secondary side synchronous rectifier power tube Q6 and its internal parasitic diode D2 and its parasitic capacitance C2, output filter capacitance Co, load R, etc.; among them, the switch network consists of the main MOS switch (Q 1 -Q 4 ) and its Internal parasitic diode (not marked in the figure) and parasitic capacitance (not marked in the figure); the resonant network is composed of resonant capacitor Cr, series resonant inductance Lr and parallel resonant inductance Lm.
  • the resonant converter may also be a half-bridge resonant converter or the like.
  • the resonant converter of the present application can adopt the following synchronous rectification control method to realize synchronous rectification control, so as to reduce voltage spikes and reduce power consumption.
  • Fig. 2 is a schematic flow chart of an embodiment of a synchronous rectification control method for a resonant converter of the present application
  • Fig. 3 is the first embodiment of the present application.
  • the synchronous rectification control method of the resonant converter of this embodiment specifically includes the following steps:
  • Step S201 Obtain the output current of the resonant converter.
  • the output current of the resonant converter refers to the output current of the secondary side synchronous rectification power tube; this output current can be obtained through a current acquisition circuit.
  • Step S202 If the output current is greater than the first current threshold I 1 , obtain the first parameter.
  • the output current is compared with the first current threshold I 1. If the output current is greater than the first current threshold I 1 , the first parameter is obtained.
  • the method for obtaining the first parameter will be described below; wherein, the first parameter includes the first parameter Lower limit value A 1 and first upper limit value B 1 .
  • Step S203 If the output current is less than the second current threshold I 2 , a second parameter is obtained.
  • the first current threshold I 1 is greater than the second current threshold I 2 , and the first parameter is greater than the second parameter.
  • the output current is compared with the second current threshold I 2. If the output current is less than the second current threshold I 2 , the second parameter is obtained.
  • the method for obtaining the second parameter will be introduced below; where the second parameter includes the first Two lower limit value A 2 and second upper limit value B 2 .
  • Step S204 A first current hysteresis Z1 is established according to the first parameter and a second current hysteresis Z2 is established according to the second parameter.
  • the parameter of the first current hysteresis Z1 is the first parameter
  • the parameter of the second current hysteresis Z2 is the first parameter. Two parameters.
  • step S202 to step S204 can establish a current double hysteresis loop for the synchronous rectification control of the secondary side synchronous rectification power tube.
  • the first current threshold I 1 of this embodiment is greater than the second current threshold I 2
  • the first parameter is greater than the second parameter, that is, the output current corresponding to the first current hysteresis Z1 is greater than the output current corresponding to the second current hysteresis Z2.
  • the first lower limit value A 1 of the first current hysteresis Z1 in this embodiment is smaller than the second current threshold value I 2 , which can avoid the blind zone of the secondary side synchronous rectification power tube turning on.
  • the assignment of the first parameter of the first current hysteresis Z1 and the second parameter of the second current hysteresis Z2 can be realized by the following procedure:
  • I o is the output current of the secondary side synchronous rectification power tube
  • a and B are the current values for controlling the secondary side synchronous rectification power tube switch.
  • the above-mentioned assignment sequence of the first parameter of the first current hysteresis Z1 and the second parameter of the second current hysteresis Z2 can avoid interruption of insertion.
  • Step S205 Use the first current hysteresis Z1 and the second current hysteresis Z2 to control the synchronous rectification of the resonant converter under load cut conditions.
  • the first current hysteresis Z1 and the second current hysteresis Z2 are successively used to control the operation of the secondary synchronous rectifier power tube of the resonant converter.
  • the method of the embodiment of FIG. 4 can be used to implement synchronous rectification of the resonant converter in switching from full load to no-load operation.
  • the method of this embodiment includes step S401 to step S403.
  • Step S401 It is determined that the resonant converter is switched from full load to no load.
  • Step S402 When the output current is less than or equal to the first lower limit value A 1 , control the secondary side synchronous rectification power tube to turn off.
  • Step S403 when the off time of the secondary side synchronous rectification power tube is greater than the time threshold, control the secondary side synchronous rectification power tube to turn on.
  • the time threshold is less than or equal to the switching judgment period of the first current hysteresis Z1 and the second current hysteresis Z2, that is, the difference between the first lower limit A 1 and the second upper limit B 2.
  • the switching judgment duration should be greater than or equal to twice the current switching duration.
  • the switching judgment time can be 5ms, and the switching time of the output current from 100A to 0A can be 500 ⁇ s.
  • Step S404 When the output current is less than or equal to the second lower limit value A 2 , the secondary side synchronous rectification power tube is controlled to be turned off.
  • the resonant converter When the output current drops rapidly (switching from full load to no-load), the resonant converter first works in the first current hysteresis Z1 to realize the rapid turn-off of the secondary side synchronous rectifier power tube. After about 5ms, the resonant converter works again in the first current hysteresis. Two current hysteresis Z2.
  • the resonant converter When the full load is switched to no load, the resonant converter first works in the first current hysteresis Z1 to realize the rapid shutdown of the secondary side synchronous rectifier power tube, which can reduce the voltage spike of the secondary side synchronous rectifier power tube; after a delay, the secondary side synchronizes The rectifier power tube is turned on again, and the synchronous rectification is controlled through the second current hysteresis Z2, which can delay the exit of the synchronous rectification, and thus can reduce the power consumption.
  • the second current hysteresis Z2 and the first current hysteresis Z1 are used in sequence to control the operation of the secondary synchronous rectifier power tube of the resonant converter.
  • the secondary side synchronous rectifier power tube is controlled to turn on; as the output current changes, if the output current If it is greater than or equal to the first upper limit value B 1 , the secondary side synchronous rectification power tube is still controlled to be turned on.
  • the resonant converter When the output current rises rapidly (no-load switching to full load), the resonant converter first works in the second current hysteresis Z2, about 5ms later, the resonant converter then works in the first current hysteresis Z1.
  • the synchronous rectification control method of the resonant converter of this embodiment can also be used for synchronous rectification control under non-load-cutting conditions.
  • the second current hysteresis is adopted.
  • Z2 controls the secondary side synchronous rectification power tube of the resonant converter to work.
  • the secondary side synchronous rectifier power tube is controlled to be turned off.
  • the secondary side synchronous rectification power tube is controlled to turn on.
  • the output current should be obtained in real time, and the real-time output current should be used for the above judgment and control.
  • the sampling and judging time of the output current can be shortened to increase the exit speed of the secondary side synchronous rectification drive during load cut to reduce voltage spikes; for example, the sampling and judging time of the output current can be shortened from 160 ⁇ s to 10 ⁇ s.
  • the current judgment threshold of the output current can be increased to increase the exit speed of the secondary side synchronous rectification drive during load cut, so as to reduce voltage spikes.
  • the present application can verify the feasibility of the current double hysteresis (the first current hysteresis Z1 and the second current hysteresis Z2) of the present application and determine the parameters of the current double hysteresis by the following methods.
  • the verification environment is:
  • Probe model Tektronix TPP0101 10X voltage probe, TEK P5200A 50MHz isolation probe, TEK TCPA300 current test probe, TEK TCPA303 current test probe;
  • Module motherboard R752A072M11PCB: V8.0; PFC_MOS: TK39N60W; PFC_DIO: APT30DQ120; DCDC_MOS: TK39N60W5; SR_MOS: IRFP4768 (IR company), the breakdown voltage of the device is 250V (working in the extreme transient area, need to meet the avalanche derating requirements ).
  • the sampling and judging duration of the output current is 10 ⁇ s, and the current double hysteresis parameters are determined.
  • the selected components have a derating design, and the derating design is exceeded during use, which is called over-derating.
  • Test conditions (1) Full load 99.3V/102A (into the current limit) cut no load, this condition is the worst working condition in the test; (2) 100V/81.6A cut no load, this is only working in the first Working condition of two hysteresis Z2.
  • Test method Cycle the load for 50 times, and take the maximum value of the voltage spike of the synchronous rectifier power tube on the secondary side. The test results are as follows:
  • the maximum value of the voltage spike is 247V, which does not exceed the device derating, but when the 100V/81.6A is cut no-load, the maximum value of the voltage spike is 279V, which is beyond device breakdown
  • the voltage (250V) is relatively large. Because under this working condition, working in the second current hysteresis Z2, the current judgment threshold is small, so the parameters of the second current hysteresis Z2 are increased by 13A (to ensure light load efficiency, the current cannot be raised too high), In order to improve the voltage spike of the secondary synchronous rectifier power tube when only working in the second current hysteresis Z2.
  • test results are as follows: (1) fully loaded with 99.3V/102A (into the current limit) cut to no load, (2) loaded with 100V/96A cut to no load, (3) loaded with 100V/81.6A cut to no load, (4) With load 100V/75A cut no load.
  • the test results are as follows:
  • Test conditions (1)(2) work in the current double hysteresis loop, which can realize the rapid shutdown of the secondary side synchronous rectifier power tube, the maximum voltage stress tested is 249V, and the full load device is derated; test conditions (3)(5 ) Only working in the second current hysteresis Z2, the tested maximum voltage stress is 257V, which exceeds the breakdown voltage of the device by 7V, but the probability of an over-derated spike is about one-fifth, and there is only one pulse.
  • Vds voltage spike is 261V (the wire is connected in series to measure the current, which is larger than the actual), and the avalanche period is 14ns. Due to the voltage spike Occurs after the output current Io drops to zero, so the drain current is very small, here is 1A.
  • the current stabilization accuracy of the resonant converter under the current limiting conditions of 20%, 25%, 50%, and 100% is tested.
  • the test results meet the requirements of 1% at each current limit point.
  • the load regulation rate of the resonant converter under the output 100V, 81V, 70V, 50V operating conditions was tested respectively.
  • the test results meet that the DC output voltage and the output current have negative monotonicity under different load conditions, and the difference between the output voltage setting value and the output voltage setting value should be ⁇ ⁇ 0.5% of the output voltage setting value.
  • the synchronous rectification is controlled by the current double hysteresis loop, which can accelerate the exit speed of the secondary side synchronous rectifier power tube when the full load is switched to no load, without affecting the turning on and off of the secondary side synchronous rectifier power tube when the resonant converter is working normally.
  • This solves the problem that the voltage stress of the secondary side synchronous rectifier power tube is excessively derated when the load is switched to no load.
  • the maximum voltage stress of the rectifier power tube exceeds the breakdown voltage of 7V, but the voltage spike duration is very short, the current flowing through the device is small, and the avalanche energy is very low, which meets the device derating requirements; after testing, after adding the current double hysteresis control, The current stabilization accuracy and load regulation rate of the resonant converter meet the requirements.
  • the present application further proposes a computer storage medium on which program instructions are stored.
  • the program instructions are executed by a processor, the synchronous rectification control method of the resonant converter is realized.
  • the computer storage medium of this embodiment can be, but is not limited to, a U disk, an SD card, a PD optical drive, a mobile hard disk, a large-capacity floppy drive, a flash memory, a multimedia memory card, a server, and the like.
  • the synchronous rectification control method of the resonant converter of the present application includes: obtaining the output current of the resonant converter; if the output current is greater than the first current threshold, obtain the first parameter; if the output current is less than the second current threshold, obtain The second parameter, the first current threshold is greater than the second current threshold, and the first parameter is greater than the second parameter; based on the first parameter and the second parameter, establish the first current hysteresis loop and the second current hysteresis loop, the first current hysteresis loop
  • the parameter is the first parameter
  • the parameter of the second current hysteresis is the second parameter; the first current hysteresis and the second current hysteresis are used to control the synchronous rectification of the resonant converter under load-cutting conditions.
  • the present application establishes the first current hysteresis loop and the second current hysteresis loop, and uses the first current hysteresis loop and the second current hysteresis loop to control the synchronous rectification of the resonant converter under load-cutting conditions, and the first
  • the first parameter of the current hysteresis is greater than the second parameter of the second current hysteresis, so the voltage spike of the secondary synchronous rectifier power tube of the resonant converter can be reduced through the first current hysteresis, and the resonance can be reduced through the second current hysteresis.
  • the power consumption of the converter therefore, the present application can reduce the peak voltage and reduce the power consumption.
  • this application also provides a storage device storing program data.
  • the program data can be executed to implement the method of the foregoing embodiment, and the storage device may be, for example, a U disk, an optical disk, a server, and the like. That is to say, this application can be embodied in the form of a software product, which includes several instructions to make an intelligent terminal execute all or part of the steps of the method described in each embodiment.
  • first and second are only used for descriptive purposes, and cannot be understood as indicating or implying relative importance or implicitly indicating the number of indicated technical features. Therefore, the features defined with “first” and “second” may explicitly or implicitly include at least one of the features. In the description of the present application, "a plurality of” means at least two, such as two, three, etc., unless specifically defined otherwise.
  • a "computer-readable medium” can be any device that can contain, store, communicate, propagate, or transmit a program for use by an instruction execution system, device, or device or in combination with these instruction execution systems, devices, or devices.
  • computer-readable media include the following: electrical connections (electronic devices) with one or more wiring, portable computer disk cases (magnetic devices), random access memory (RAM), Read only memory (ROM), erasable and editable read only memory (EPROM or flash memory), fiber optic devices, and portable compact disk read only memory (CDROM).
  • the computer-readable medium may even be paper or other suitable medium on which the program can be printed, because it can be used for example by optically scanning the paper or other medium, followed by editing, interpretation or other suitable media if necessary. The program is processed in a way to obtain the program electronically and then stored in the computer memory.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Rectifiers (AREA)

Abstract

一种谐振变换器的同步整流控制方法包括：获取谐振变换器的输出电流；若输出电流大于第一电流阈值，得到第一参数；若输出电流小于第二电流阈值，得到第二参数，第一电流阈值大于第二电流阈值，第一参数大于第二参数；根据第一参数建立第一电流滞环及根据第二参数建立第二电流滞环，第一电流滞环的参数为第一参数，第二电流滞环的参数为第二参数；采用第一电流滞环及第二电流滞环控制谐振变换器在切载工况下的同步整流。

Description

谐振变换器及其同步整流控制方法 技术领域
本申请涉及变换器领域，特别是涉及一种谐振变换器及其同步整流控制方法。
背景技术
这里的陈述仅提供与本申请有关的背景信息，而不必然地构成示例性技术。
在谐振变换器技术中，通常通过母线电压函数调整的方法解决副边同步整流功率管的Vds电压在稳态时超降额的问题，但是不能解决副边同步整流功率管的Vds电压在动态过程中超降额的问题。例如，谐振变换器的功率由7.5KW提升到10KW，其输出额定电流由75A提升到100A，限流110％(110A)，在满载切空载工况下，副边同步整流功率管的电压尖峰最高可达290V，超降额的问题严重。
为改善超降额的问题，现有技术通常提高电流判断阈值(同步整流快速退出)，但会增加功耗。
发明内容
根据本申请的各种实施例，提供一种谐振变换器及其同步整流控制方法。
一种谐振变换器的同步整流控制方法，包括：获取谐振变换器的输出电流；若输出电流大于第一电流阈值，得到第一参数；若输出电流小于第二电流阈值，得到第二参数，第一电流阈值大于第二电流阈值，第一参数大于第二参数；根据所述第一参数建立第一电流滞环及根据所述 第二参数建立第二电流滞环，第一电流滞环的参数为第一参数，第二电流滞环的参数为第二参数；采用第一电流滞环及第二电流滞环控制谐振变换器在切载工况下的同步整流。
一种谐振变换器的同步整流控制方法，包括：
获取所述谐振变换器的输出电流；
若所述输出电流大于第一电流阈值，得到第一参数；
若所述输出电流小于第二电流阈值，得到第二参数，所述第一电流阈值大于所述第二电流阈值，所述第一参数大于所述第二参数；
根据所述第一参数建立第一电流滞环及根据所述第二参数建立第二电流滞环，所述第一电流滞环的参数为所述第一参数，所述第二电流滞环的参数为所述第二参数；其中，所述第一参数包括第一下限值和第一上限值，所述第二参数包括第二上限值和第二下限值，其中所述第一下限值小于所述第二电流阈值；
采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器在切载工况下的同步整流；
若谐振变换器由满载切换到空载，则依次采用第一电流滞环及第二电流滞环控制谐振变换器的副边同步整流功率管工作；在输出电流小于或等于第一下限值，则控制副边同步整流功率管关断；在副边同步整流功率管的关断时间大于时间阈值，则控制副边同步整流功率管开通；从第一电流滞环切换到第二电流滞环具有延时，因此在区域第一下限值与第二上限值中，副边同步整流功率管会先关断，然后延时预设时长后再开通，进入第二电流滞环的控制；在输出电流小于或等于第二下限值，则控制副边同步整流功率管关断；
若谐振变换器由空载切换到满载，则依次采用第二电流滞环及第一电流滞环控制谐振变换器的副边同步整流功率管工作；
谐振变换器由空载切换到满载工况时，若输出电流大于或等于第二上限值，则控制副边同步整流功率管开通；随着输出电流的变化，若输出电流大于或等于第一上限值，仍然控制副边同步整流功率管开通。
一种谐振变换器，该谐振变换器通过上述同步整流控制方法进行同 步整流控制。
本申请的一个或多个实施例的细节在下面的附图和描述中提出。本申请的其他特征、目的和优点将从说明书、附图以及权利要求书变得明显。
附图说明
为了更清楚地说明本申请实施例或现有技术中的技术方案，下面将对实施例中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他的附图。
图1a是本申请谐振变换器的简易电路结构示意图。
图1b是本申请谐振变换器的详细电路结构示意图。
图2是本申请谐振变换器的同步整流控制方法一实施例的流程示意图。
图3是本申请第一电流滞环与第二电流滞环的滞环曲线示意图。
图4是图2实施例谐振变换器的同步整流控制方法中步骤S205的一具体流程示意图。
图5是本申请谐振变换器在切载工况下第一电流滞环与第二电流滞环的滞环曲线及副边同步整流功率管开关状态的示意图。
图6是本申请谐振变换器在非切载工况下第一电流滞环与第二电流滞环的滞环曲线及副边同步整流功率管开关状态的示意图。
具体实施方式
下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本申请一部分实施例，而不是全部的实施例。基于本申请中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，均属于本申请保护的范围。
本申请中的术语“第一”、“第二”仅用于描述目的，而不能理解为指示或暗示相对重要性或者隐含指明所指示的技术特征的数量。本申请 的描述中，“多个”的含义是至少两个，例如两个，三个等，除非另有明确具体的限定。此外，术语“包括”和“具有”以及它们任何变形，意图在于覆盖不排他的包含。例如包含了一系列步骤或单元的过程、方法、系统、产品或设备没有限定于已列出的步骤或单元，而是可选地还包括没有列出的步骤或单元，或可选地还包括对于这些过程、方法、产品或设备固有的其它步骤或单元。
本申请首先提出一种谐振变换器，如图1a所示，谐振变换器是一种通过控制开关频率(调节频率)来实现输出电压恒定的谐振电路。它的优点是：实现原边主MOS开关(Q 1-Q 4)的零电压开通(Zero Voltage Switch，ZVS)和副边同步整流二极管(D 1、D 2)的零电流关断(Zero Current Switch，ZCS)，通过软开关技术，可以降低开关损耗，提高谐振变换器的效率和功率密度。图1a中的同步整流二极管(D 1、D 2)为副边同步整流功率管的寄生二极管，图1a中省略了副边同步整流功率管仅仅示意出了寄生二极管(D 1、D 2)和寄生电容(C1、C2)。
如图1b所示，本实施例提供的谐振变换器包括开关网络(图未标)、谐振网络(图未标)、中心抽头变压器T、副边同步整流功率管Q5及其内部寄生二极管D1及其寄生电容C1、副边同步整流功率管Q6及其内部寄生二极管D2及其寄生电容C2、输出滤波电容Co、负载R等；其中，开关网络由主MOS开关(Q 1-Q 4)及其内部寄生二极管(图未标)、寄生电容(图未标)组成；谐振网络由谐振电容Cr、串联谐振电感Lr及并联谐振电感Lm组成。
在其它实施例中，谐振变换器还可以是半桥式谐振变换器等。
本申请的谐振变换器可以采用下述同步整流控制方法实现同步整流控制，以降低电压尖峰，且降低功耗。
本申请进一步提出一种谐振变换器的同步整流控制方法，如图2及图3所示，图2是本申请谐振变换器的同步整流控制方法一实施例的流程示意图；图3是本申请第一电流滞环与第二电流滞环的滞环曲线示意图。本实施例谐振变换器的同步整流控制方法具体包括以下步骤：
步骤S201：获取谐振变换器的输出电流。
谐振变换器的输出电流是指副边同步整流功率管的输出电流；可以通过电流采集电路获取该输出电流。
步骤S202：若输出电流大于第一电流阈值I 1，得到第一参数。
将输出电流与第一电流阈值I 1进行比较，若输出电流大于第一电流阈值I 1，则获取第一参数，关于第一参数的获取方法将在下文进行介绍；其中，第一参数包括第一下限值A 1和第一上限值B 1。
步骤S203：若输出电流小于第二电流阈值I 2，得到第二参数，第一电流阈值I 1大于第二电流阈值I 2，第一参数大于第二参数。
将输出电流与第二电流阈值I 2进行比较，若输出电流小于第二电流阈值I 2，则获取第二参数，关于第二参数的获取方法将在下文进行介绍；其中，第二参数包括第二下限值A 2和第二上限值B 2。
步骤S204：根据第一参数建立第一电流滞环Z1及根据第二参数建立第二电流滞环Z2，第一电流滞环Z1的参数为第一参数，第二电流滞环Z2的参数为第二参数。
本实施例可以通过步骤S202至步骤S204为副边同步整流功率管的同步整流控制建立电流双滞环。且本实施例的第一电流阈值I 1大于第二电流阈值I 2，第一参数大于第二参数，即第一电流滞环Z1对应的输出电流大于第二电流滞环Z2对应的输出电流。
可选地，本实施例的第一电流滞环Z1的第一下限值A 1小于第二电流阈值I 2，能够避免出现副边同步整流功率管开通盲区。
本实施例可以通过下述程序实现第一电流滞环Z1的第一参数及第二电流滞环Z2的第二参数的赋值：
Figure PCTCN2021111126-appb-000001
Figure PCTCN2021111126-appb-000002
其中，I o为副边同步整流功率管的输出电流，A、B为控制副边同步整流功率管开关的电流值。
本实施例第一电流滞环Z1的第一参数及第二电流滞环Z2的第二参数的上述赋值顺序能够避免插入中断。
步骤S205：采用第一电流滞环Z1及第二电流滞环Z2控制谐振变换器在切载工况下的同步整流。
在一应用场景中，若谐振变换器由满载切换到空载，则依次采用第一电流滞环Z1及第二电流滞环Z2控制谐振变换器的副边同步整流功率管工作。
具体地，基于图3实施例的电流双滞环的滞流曲线，可以采用图4实施例的方法实现谐振变换器由满载切换到空载工况下的同步整流。本实施例的方法包括步骤S401至步骤S403。
步骤S401：判定谐振变换器由满载切换到空载。
若检测到输出电流快速下降，则判定谐振变换器由满载切换到空载。
步骤S402：在输出电流小于或等于第一下限值A 1，则控制副边同步整流功率管关断。
如图5所示，若谐振变换器由满载切换到空载，则获取副边同步整流功率管的输出电流；在输出电流小于或等于第一下限值A 1，控制副边同步整流功率管关断。
步骤S403：在副边同步整流功率管的关断时间大于时间阈值，则控制副边同步整流功率管开通。
从第一电流滞环Z1切换到第二电流滞环Z2具有延时，因此在区域A 1-B 2中，副边同步整流功率管会先关断，然后延时预设时长后再开通，进入第二电流滞环Z2的控制。
时间阈值小于或等于第一电流滞环Z1与第二电流滞环Z2切换判断时长，即第一下限值A 1与第二上限值B 2之间的差值。且该切换判断时长 应大于或者等于两倍的电流切换时长。
在一应用场景中，切换判断时长可以为5ms，输出电流从100A到0A的切换时长可以为500μs。
步骤S404：在输出电流小于或等于第二下限值A 2，则控制副边同步整流功率管关断。
获取副边同步整流功率管的输出电流，在输出电流小于或等于第二下限值A 2，控制副边同步整流功率管关断。
当输出电流快速下降(满载切换到空载)时，谐振变换器先工作在第一电流滞环Z1，实现副边同步整流功率管的快速关断，约5ms后，谐振变换器再工作在第二电流滞环Z2。
满载切空载时，谐振变换器先工作在第一电流滞环Z1，实现副边同步整流功率管的快速关断，能够降低副边同步整流功率管的电压尖峰；延时后，副边同步整流功率管再次开通，通过第二电流滞环Z2控制同步整流，能够延迟同步整流退出，因此能够降低功耗。
在另一应用场景中，若谐振变换器由空载切换到满载，则依次采用第二电流滞环Z2及第一电流滞环Z1控制谐振变换器的副边同步整流功率管工作。
具体地，谐振变换器由空载切换到满载工况时，若输出电流大于或等于第二上限值B 2，则控制副边同步整流功率管开通；随着输出电流的变化，若输出电流大于或等于第一上限值B 1，仍然控制副边同步整流功率管开通。
当输出电流快速上升(空载切换到满载)时，谐振变换器先工作在第二电流滞环Z2，约5ms后，谐振变换器再工作在第一电流滞环Z1。
可选地，本实施例的谐振变换器的同步整流控制方法还可以用于非切载工况下的同步整流控制，例如，若谐振变换器处于非切载工况，采用第二电流滞环Z2控制谐振变换器的副边同步整流功率管工作。
具体地，在非切载工况(输出电流缓慢变化)下，如图6所示，若输出电流小于或者等于第二下限值A 2，则控制副边同步整流功率管关断，若输出电流大于或者等于第二上限值B 2，则控制副边同步整流功率管开 通。
在同步整流控制过程中，应实时获取输出电流，用实时的输出电流进行上述判断、控制。
本实施例可以通过缩短输出电流的采样判断时长，来提高切载时的副边同步整流驱动退出的速度，以降低电压尖峰；例如，可以将输出电流的采样判断时长从160μs缩短为10μs。
在其它实施例中，还可以通过提高输出电流的电流判断阈值，来提高切载时的副边同步整流驱动退出的速度，以降低电压尖峰。
本申请可以通过下述方法验证本申请电流双滞环(第一电流滞环Z1及第二电流滞环Z2)的可行性及确定电流双滞环的参数。
验证环境为：
示波器型号：安捷伦DSO3034A 350MHz；
探头型号：Tektronix TPP0101 10X电压探头、TEK P5200A 50MHz隔离探头、TEK TCPA300电流测试探头、TEK TCPA303电流测试探头；
模块主板：R752A072M11PCB：V8.0；PFC_MOS：TK39N60W；PFC_DIO：APT30DQ120；DCDC_MOS：TK39N60W5；SR_MOS：IRFP4768(IR公司)，器件的击穿电压为250V(工作在极限瞬态区，需满足雪崩降额要求)。
输出电流的采样判断时长为10μs的电流双滞环的参数确定。
首先确定第一下限值A 1和第一上限值B 1值，即副边同步整流功率管开通或者关断的电流判断值提高到多少满足电压应力不超降额的要求。设置不同第一参数(B 1、A 1)下满载(99.3V/102.5A，限流模式)切空载进行测试；第一种测试参数：B 1＝70A、A 1＝65A；第二种测试参数：B 1＝80A、A 1＝75A；通道CH1为副边同步整流功率管的电压波形；通道CH2为副边同步整流功率管的输出电流。在本申请中，所选用的元器件具有降额设计，在使用过程中超出其降额设计，称为超降额。
在B 1＝70A、A 1＝65A下，副边同步整流功率管的电压应力最高达275V，不满足要求。在B 1＝80A、A 1＝75A下，副边同步整流功率管的电压应力最高达245V，满足要求。因此确定参数B 1＝80A、A 1＝75A。
为保证不出现副边同步整流功率管开通盲区，需要满足：A 1<I 2，因 此设置I 2＝78A；I 1比I 2大即可，这里可以配置I 1＝I 2+5A＝83A。其中，副边同步整流功率管退出的电流判断阈值可以设置为：A 1-5A＝70A(Z1)；A 2-5A＝10A(Z2)，其具体数值可以根据实际工况进行调整。
为了进一步评价电流双滞环的可行性，对副边同步整流功率管的电压应力、稳流精度和负载调整率重新进行了测试。
a)满载切空载时，副边同步整流功率管的电压应力。测试工况：(1)满载99.3V/102A(进入限流)切空载，这种工况是测试中最恶劣工况；(2)100V/81.6A切空载，这是仅工作在第二滞环Z2的工况。测试方法：循环切载50次，取副边同步整流功率管电压尖峰的最大值。测试结果如下：
测试工况 Q5 Q6
99.3V/102A切空载 245V 243V
100V/81.6A切空载 279V 261V
其中，在满载99.3V/102A切空载时，电压尖峰的最大值为247V，不超过器件降额，但是在100V/81.6A切空载时，电压尖峰的最大值为279V，超器件击穿电压(250V)较大。因为在该工况下，工作在第二电流滞环Z2中，电流判断阈值较小，因此将第二电流滞环Z2的参数提高13A(为保证轻载效率，电流不能提的太高)，以改善仅工作在第二电流滞环Z2时的副边同步整流功率管的电压尖峰。新的电流双滞环参数为：A 2＝23A、B 2＝28A、A 1＝75A、B 1＝80A、I 2＝78A、I 1＝83A。
重新测试新的电流环滞环参数下的切载工况下的电压应力。测试工况为：(1)满载99.3V/102A(进入限流)切空载、(2)带载100V/96A切空载、(3)带载100V/81.6A切空载、(4)带载100V/75A切空载。测试结果如下：
测试工况 Q5 Q6
99.3V/102A切空载 249V 249V
100V/96A切空载 241V 237V
100V/81.6A切空载 253V 245V
100V/75A切空载 247/257V 239V
测试工况(1)(2)工作在电流双滞环，能实现副边同步整流功率管的快速关断，测试的最大电压应力为249V，满载器件降额；测试工况(3)(5)仅工作在第二电流滞环Z2中，测试的最大电压应力为257V，超出器件击穿电压7V，但是超降额的尖峰出现的概率大约五十分之一，且只有一个脉冲。
为了评估谐振变换器工作的可靠性，从雪崩击穿的角度进行评价，测得Vds电压尖峰为261V(为测电流串了导线，比实际偏大点)，雪崩期间时间为14ns，由于电压尖峰发生在输出电流Io下降到零后，因此漏极电流很小，这里取1A。
雪崩能量为雪崩能量为：E AR＝1.8μJ，远低于器件手册给出的770mJ。因此在工况(3)(4)中，虽然Vds电压尖峰超过了器件的击穿电压，但是工作在极限瞬时区，雪崩能量很小，符合器件降额要求。
b)稳流精度
测试了谐振变换器在20％、25％、50％、100％限流条件下的稳流精度，测试结果满足在各个限流点下，稳流精度满足1％的要求。
c)负载调整率
分别测试了输出100V、81V、70V、50V工况下谐振变换器的负载调整率，测试结果满足不同负载情况下直流输出电压与输出电流具有负单调性，且与输出电压整定值的差应≤±0.5％的输出电压整定值。
d)采用电流双滞环时，当从大电流(>A 1)切载到(B 2～A 1区间)时，同步整流会先关断载重新开通。
实验测试结果表明，如理论分析，设定SR驱动波形及输出电流Io从大电流(>A 1)切载到(B 2～A 1区间)时，驱动会关断后开通，原因是从第一电流滞环Z1切换到第二电流滞环Z2需要延时，延时约300ms(主要是滞环切换判断的电流是显示电流，这里对速度并无太高要求)这种情形对谐振变换器的正常工作没有影响，这种情况同时说明电流双滞环是起作用的。
本申请通过电流双滞环控制同步整流，能够加速满载切空载时副边同步整流功率管退出的速度，而不影响谐振变换器正常工作时副边同步 整流功率管的开通及关断，解决了在满载切空载时副边同步整流功率管电压应力超降额的问题。电流双滞环的参数为：A 2＝23A、B 2＝28A、A 1＝75A、B 1＝80A、I 2＝78A、I 1＝83A；虽然非满载切空载工况，副边同步整流功率管电压应力最大超过击穿电压7V，但该电压尖峰持续时间很短，流过器件电流小，雪崩能量很低，满足器件降额要求；经测试，通过加入电流双滞环控制后，谐振变换器的稳流精度、负载调整率都满足要求。
本申请进一步提出一种计算机存储介质，计算机存储介质其上存储有程序指令，程序指令被处理器执行时实现上述谐振变换器的同步整流控制方法。
本实施例计算机存储介质可以是但不局限于U盘、SD卡、PD光驱、移动硬盘、大容量软驱、闪存、多媒体记忆卡、服务器等。
区别于现有技术，本申请谐振变换器的同步整流控制方法包括：获取谐振变换器的输出电流；若输出电流大于第一电流阈值，得到第一参数；若输出电流小于第二电流阈值，得到第二参数，第一电流阈值大于第二电流阈值，第一参数大于第二参数；基于第一参数和第二参数，建立第一电流滞环及第二电流滞环，第一电流滞环的参数为第一参数，第二电流滞环的参数为第二参数；采用第一电流滞环及第二电流滞环控制谐振变换器在切载工况下的同步整流。通过这种方式，本申请建立第一电流滞环及第二电流滞环，并采用第一电流滞环及第二电流滞环控制谐振变换器在切载工况下的同步整流，且第一电流滞环的第一参数大于第二电流滞环的第二参数，因此能够通过第一电流滞环降低谐振变换器的副边同步整流功率管的电压尖峰，并通过第二电流滞环降低谐振变换器的功耗，因此本申请能够降低尖峰电压，且降低功耗。
另外，上述功能如果以软件功能的形式实现并作为独立产品销售或使用时，可存储在一个移动终端可读取存储介质中，即，本申请还提供一种存储有程序数据的存储装置，所述程序数据能够被执行以实现上述实施例的方法，该存储装置可以为如U盘、光盘、服务器等。也就是说，本申请可以以软件产品的形式体现出来，其包括若干指令用以使得一台智能终端执行各个实施例所述方法的全部或部分步骤。
在本申请的描述中，参考术语“一个实施例”、“一些实施例”、“示例”、“具体示例”、或“一些示例”等的描述意指结合该实施例或示例描述的具体特征、结构、材料或者特点包含于本申请的至少一个实施例或示例中。在本说明书中，对上述术语的示意性表述不必须针对的是相同的实施例或示例。而且，描述的具体特征、结构、材料或者特点可以在任一个或多个实施例或示例中以合适的方式结合。此外，在不相互矛盾的情况下，本领域的技术人员可以将本说明书中描述的不同实施例或示例以及不同实施例或示例的特征进行结合和组合。
此外，术语“第一”、“第二”仅用于描述目的，而不能理解为指示或暗示相对重要性或者隐含指明所指示的技术特征的数量。由此，限定有“第一”、“第二”的特征可以明示或者隐含地包括至少一个该特征。在本申请的描述中，“多个”的含义是至少两个，例如两个，三个等，除非另有明确具体的限定。
流程图中或在此以其他方式描述的任何过程或方法描述可以被理解为，表示包括一个或更多个用于实现特定逻辑功能或过程的步骤的可执行指令的代码的模块、片段或部分，并且本申请的优选实施方式的范围包括另外的实现，其中可以不按所示出或讨论的顺序，包括根据所涉及的功能按基本同时的方式或按相反的顺序，来执行功能，这应被本申请的实施例所属技术领域的技术人员所理解。
在流程图中表示或在此以其他方式描述的逻辑和/或步骤，例如，可以被认为是用于实现逻辑功能的可执行指令的定序列表，可以具体实现在任何计算机可读介质中，以供指令执行系统、装置或设备(可以是个人计算机，服务器，网络设备或其他可以从指令执行系统、装置或设备取指令并执行指令的系统)使用，或结合这些指令执行系统、装置或设备而使用。就本说明书而言，"计算机可读介质"可以是任何可以包含、存储、通信、传播或传输程序以供指令执行系统、装置或设备或结合这些指令执行系统、装置或设备而使用的装置。计算机可读介质的更具体的示例(非穷尽性列表)包括以下：具有一个或多个布线的电连接部(电子装置)，便携式计算机盘盒(磁装置)，随机存取存储器(RAM)，只 读存储器(ROM)，可擦除可编辑只读存储器(EPROM或闪速存储器)，光纤装置，以及便携式光盘只读存储器(CDROM)。另外，计算机可读介质甚至可以是可在其上打印所述程序的纸或其他合适的介质，因为可以例如通过对纸或其他介质进行光学扫描，接着进行编辑、解译或必要时以其他合适方式进行处理来以电子方式获得所述程序，然后将其存储在计算机存储器中。
以上所述仅为本申请的实施方式，并非因此限制本申请的专利范围，凡是利用本申请说明书及附图内容所作的等效结构或等效流程变换，或直接或间接运用在其他相关的技术领域，均同理包括在本申请的专利保护范围。

Claims (15)

  1. 一种谐振变换器的同步整流控制方法，包括：
    获取所述谐振变换器的输出电流；
    若所述输出电流大于第一电流阈值，得到第一参数；
    若所述输出电流小于第二电流阈值，得到第二参数，所述第一电流阈值大于所述第二电流阈值，所述第一参数大于所述第二参数；
    根据所述第一参数建立第一电流滞环及根据所述第二参数建立第二电流滞环，所述第一电流滞环的参数为所述第一参数，所述第二电流滞环的参数为所述第二参数；
    采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器在切载工况下的同步整流；
    其中，所述第一参数包括第一下限值和第一上限值，所述第二参数包括第二上限值和第二下限值，其中所述第一下限值小于所述第二电流阈值。
  2. 根据权利要求1所述的同步整流控制方法，其中，采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器在切载工况下的同步整流包括：
    若所述谐振变换器由满载切换到空载，则依次采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器的副边同步整流功率管工作。
  3. 根据权利要求2所述的同步整流控制方法，其中，所述依次采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器的副边同步整流功率管工作包括：
    若所述输出电流小于或等于所述第一下限值，则控制所述副边同步整流功率管关断；
    若所述副边同步整流功率管的关断时间大于时间阈值，则控制所述副边同步整流功率管开通；
    若所述输出电流小于或等于所述第二下限值，则控制所述副边同步 整流功率管关断。
  4. 根据权利要求3所述的同步整流控制方法，其中，所述时间阈值小于或等于所述第一下限值与所述第二上限值之间的差值所对应的时间。
  5. 根据权利要求1所述的同步整流控制方法，其特征在于，采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器在切载工况下的同步整流进一步包括：
    若所述谐振变换器由空载切换到满载，则依次采用所述第二电流滞环及所述第一电流滞环控制所述谐振变换器的副边同步整流功率管工作。
  6. 根据权利要求1所述的同步整流控制方法，其中，所述同步整流控制方法进一步包括：
    若所述谐振变换器处于非切载工况，则采用所述第二电流滞环控制所述谐振变换器的副边同步整流功率管工作。
  7. 根据权利要求1所述的同步整流控制方法，其中，所述同步整流控制方法进一步包括：
    缩短所述输出电流的采样判断时长。
  8. 根据权利要求3所述的同步整流控制方法，其中，所述同步整流控制方法进一步包括：
    若所述输出电流在第一下限值与第二上限值的区域中，副边同步整流功率管先关断，延时预设时长后再开通，进入第二电流滞环的控制。
  9. 根据权利要求5所述的同步整流控制方法，其中，所述同步整流控制方法还包括：
    所述谐振变换器由空载切换到满载工况时，若所述谐振变换器的输出电流大于或等于第二上限值，则控制副边同步整流功率管开通；根据所述输出电流的变化，若输出电流大于或等于第一上限值，仍然控制副边同步整流功率管开通。
  10. 一种谐振变换器的同步整流控制方法，包括：
    获取所述谐振变换器的输出电流；
    若所述输出电流大于第一电流阈值，得到第一参数；
    若所述输出电流小于第二电流阈值，得到第二参数，所述第一电流阈值大于所述第二电流阈值，所述第一参数大于所述第二参数；
    根据所述第一参数建立第一电流滞环及根据所述第二参数建立第二电流滞环，所述第一电流滞环的参数为所述第一参数，所述第二电流滞环的参数为所述第二参数；其中，所述第一参数包括第一下限值和第一上限值，所述第二参数包括第二上限值和第二下限值，其中所述第一下限值小于所述第二电流阈值；
    采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器在切载工况下的同步整流；
    若谐振变换器由满载切换到空载，则依次采用第一电流滞环及第二电流滞环控制谐振变换器的副边同步整流功率管工作；在输出电流小于或等于第一下限值，则控制副边同步整流功率管关断；在副边同步整流功率管的关断时间大于时间阈值，则控制副边同步整流功率管开通；从第一电流滞环切换到第二电流滞环具有延时，因此在区域第一下限值与第二上限值中，副边同步整流功率管会先关断，然后延时预设时长后再开通，进入第二电流滞环的控制；在输出电流小于或等于第二下限值，则控制副边同步整流功率管关断；
    若谐振变换器由空载切换到满载，则依次采用第二电流滞环及第一电流滞环控制谐振变换器的副边同步整流功率管工作；
    谐振变换器由空载切换到满载工况时，若输出电流大于或等于第二上限值，则控制副边同步整流功率管开通；随着输出电流的变化，若输出电流大于或等于第一上限值，仍然控制副边同步整流功率管开通。
  11. 根据权利要求10所述的同步整流控制方法，其中，所述时间阈值小于或等于所述第一下限值与所述第二上限值之间的差值所对应的时间。
  12. 根据权利要求10所述的同步整流控制方法，其中，所述同步整流控制方法进一步包括：
    若所述谐振变换器处于非切载工况，则采用所述第二电流滞环控制 所述谐振变换器的副边同步整流功率管工作。
  13. 根据权利要求10所述的同步整流控制方法，其中，所述同步整流控制方法进一步包括：
    缩短所述输出电流的采样判断时长。
  14. 一种谐振变换器，通过同步整流控制方法进行同步整流控制，所述同步整流控制方法，包括：
    获取所述谐振变换器的输出电流；
    若所述输出电流大于第一电流阈值，得到第一参数；
    若所述输出电流小于第二电流阈值，得到第二参数，所述第一电流阈值大于所述第二电流阈值，所述第一参数大于所述第二参数；
    根据所述第一参数建立第一电流滞环及根据所述第二参数建立第二电流滞环，所述第一电流滞环的参数为所述第一参数，所述第二电流滞环的参数为所述第二参数；
    采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器在切载工况下的同步整流；
    其中，所述第一参数包括第一下限值和第一上限值，所述第二参数包括第二上限值和第二下限值，其中所述第一下限值小于所述第二电流阈值。
  15. 一种计算机存储介质，计算机存储介质其上存储有程序指令，程序指令被处理器执行时实现谐振变换器的同步整流控制方法；
    所述同步整流控制方法，包括：
    获取所述谐振变换器的输出电流；
    若所述输出电流大于第一电流阈值，得到第一参数；
    若所述输出电流小于第二电流阈值，得到第二参数，所述第一电流阈值大于所述第二电流阈值，所述第一参数大于所述第二参数；
    根据所述第一参数建立第一电流滞环及根据所述第二参数建立第二电流滞环，所述第一电流滞环的参数为所述第一参数，所述第二电流滞环的参数为所述第二参数；
    采用所述第一电流滞环及所述第二电流滞环控制所述谐振变换器 在切载工况下的同步整流；
    其中，所述第一参数包括第一下限值和第一上限值，所述第二参数包括第二上限值和第二下限值，其中所述第一下限值小于所述第二电流阈值。
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