WO2020253846A1 - 高功率密度单相级联h桥整流器、控制方法及控制系统 - Google Patents

高功率密度单相级联h桥整流器、控制方法及控制系统 Download PDF

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WO2020253846A1
WO2020253846A1 PCT/CN2020/097207 CN2020097207W WO2020253846A1 WO 2020253846 A1 WO2020253846 A1 WO 2020253846A1 CN 2020097207 W CN2020097207 W CN 2020097207W WO 2020253846 A1 WO2020253846 A1 WO 2020253846A1
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voltage
power
unit
bridge rectifier
decoupling
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PCT/CN2020/097207
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English (en)
French (fr)
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杜春水
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山东大学
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Priority to US17/271,370 priority Critical patent/US11621651B2/en
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H02M7/487—Neutral point clamped inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/0074—Plural converter units whose inputs are connected in series
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H02M1/15—Arrangements for reducing ripples from dc input or output using active elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0083—Converters characterised by their input or output configuration
    • H02M1/009—Converters characterised by their input or output configuration having two or more independently controlled outputs
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H02M7/4835—Converters with outputs that each can have more than two voltages levels comprising two or more cells, each including a switchable capacitor, the capacitors having a nominal charge voltage which corresponds to a given fraction of the input voltage, and the capacitors being selectively connected in series to determine the instantaneous output voltage

Definitions

  • the present disclosure belongs to the field of cascaded multilevel converters, and in particular relates to a high power density single-phase cascaded H-bridge rectifier, a control method and a control system.
  • the first aspect of the present disclosure provides a high-power density single-phase cascaded H-bridge rectifier, which uses an independent active power decoupling circuit for the power unit of the cascaded multilevel converter.
  • Sub-power pulsation provides a path.
  • the number of filter capacitors can theoretically be reduced by 10 times under the same conversion power and the same DC voltage ripple, which can effectively reduce system volume and weight, and increase system power density and reliability .
  • a high-power density single-phase cascaded H-bridge rectifier including:
  • each power conversion unit includes an H-bridge power unit connected in parallel, a decoupling unit and a DC side equivalent load; each decoupling unit is an independent A step-down active power decoupling circuit, the decoupling unit is used to buffer the secondary ripple power to reduce the capacity of the DC bus capacitor.
  • the decoupling unit includes a series-connected power module
  • the series-connected power module is composed of two series-connected switching power elements, and both ends of the series-connected power module are respectively connected to the positive and negative bus bars of the DC side of the corresponding power conversion unit
  • the middle connection point of the series power module is connected with a series inductance and capacitance loop, and the other end of the capacitor is connected to the negative bus on the DC side of the corresponding power conversion unit.
  • each power conversion unit further includes: a DC filter capacitor connected in parallel with an equivalent load on the DC side, and the DC filter capacitor is used to eliminate high-order harmonics.
  • the equivalent load on the DC side is a resistive element, a DC/DC converter or a capacitive element.
  • the second aspect of the present disclosure provides a high-power density single-phase cascaded H-bridge rectifier control method, which uses a comprehensive control strategy of DC-side voltage equalization and active power decoupling, while realizing cascaded units DC side output voltage balance and secondary ripple voltage ripple suppression.
  • a high power density single-phase cascaded H-bridge rectifier control method includes:
  • the closed loop controls the grid-side current to generate the average modulation signal factor
  • the average value of the total system voltage is compared with the voltage of each power unit, and the difference is normalized and multiplied by the grid-side voltage phase as the deviation modulation signal factor. At the same time, the voltage balance and the suppression of the secondary pulsation of the DC bus voltage are realized;
  • the final modulation signal of each power conversion unit is compared with the carrier, and the single-pole frequency multiplication carrier phase-shift modulation algorithm is used to generate the driving signal of the switch tube of each power conversion unit;
  • Each power conversion unit extracts the secondary ripple current on the DC bus in real time, and calculates the duty cycle of the power switch in the decoupling unit online to realize the transfer of the secondary ripple power on the DC bus to the decoupling unit.
  • the process of generating the operating voltage of each power conversion unit and the total voltage of the H-bridge rectifier is:
  • the actual output voltage of the DC side of each cascaded unit is filtered and compared with the voltage reference value. Each difference is adjusted by PI to obtain the corresponding operating control voltage amplitude.
  • the operation control voltage of each unit is added in real time to obtain the total voltage of the H-bridge rectifier.
  • the process of generating the average modulation signal factor is:
  • the total voltage of the H-bridge rectifier is divided by the grid-side input voltage amplitude, and then multiplied by the grid-side voltage phase-locked loop output phase as the grid-side current reference value;
  • the actual detected instantaneous value of the grid current is compared with the grid-side current reference value, and then the grid-side filtering inductor current is closed-loop without static error tracking, and the average modulation signal factor is output.
  • the third aspect of the present disclosure provides a high-power density single-phase cascaded H-bridge rectifier control system, which uses a comprehensive control strategy of DC side voltage equalization and active power decoupling, while realizing cascaded units DC side output voltage balance and secondary ripple voltage ripple suppression.
  • a high-power density single-phase cascaded H-bridge rectifier control system includes:
  • DC bus voltage control module which is used for closed-loop control of the DC bus voltage of each power conversion unit, generating the operating voltage of each power conversion unit and the total voltage of the H-bridge rectifier;
  • Unit power factor rectifier module which is used to take the total voltage of the H-bridge rectifier as a given value in the outer loop, control the grid-side current in a closed loop, and generate an average modulation signal factor;
  • Voltage balance module which is used to compare the average value of the total voltage of the system with the voltage of each power unit, and the difference is standardized and multiplied by the grid-side voltage phase as the deviation modulation signal factor, while achieving voltage balance and DC bus Suppression of secondary voltage ripple;
  • Modulation signal generation module which is used to superimpose the deviation modulation signal factor of each power conversion unit with the average modulation signal factor to generate the final modulation signal of each power conversion unit;
  • Drive signal generation module which is used to compare the final modulation signal of each power conversion unit with the carrier wave, and generate the drive signal of the switch tube of each power conversion unit by adopting a single-pole frequency multiplication carrier phase-shift modulation algorithm;
  • Active power decoupling control module which is used to extract the secondary ripple current on the DC bus of each power conversion unit in real time, and calculate the duty cycle of the power switch tube in the decoupling unit online to achieve the secondary ripple on the DC bus The transfer of power to the decoupling unit.
  • the process of generating the operating voltage of each power conversion unit and the total voltage of the H-bridge rectifier is:
  • the actual output voltage of the DC side of each cascaded unit is filtered and compared with the voltage reference value. Each difference is adjusted by PI to obtain the corresponding operating control voltage amplitude.
  • the operation control voltage of each unit is added in real time to obtain the total voltage of the H-bridge rectifier.
  • the process of generating the average modulation signal factor is:
  • the total voltage of the H-bridge rectifier is divided by the grid-side input voltage amplitude, and then multiplied by the grid-side voltage phase-locked loop output phase as the grid-side current reference value;
  • the actual detected instantaneous value of the grid current is compared with the grid-side current reference value, and then the grid-side filtering inductor current is closed-loop without static error tracking, and the average modulation signal factor is output.
  • a buck active power decoupling unit is added to the DC side of the single-phase cascaded H-bridge multilevel converter, which is beneficial to reduce the capacitor voltage.
  • Control use the integrated control strategy of DC side voltage equalization and active power decoupling, and achieve DC side output voltage equalization and suppression of secondary ripple voltage ripple between cascaded units at the same time; use adaptive frequency selector to extract in real time
  • the secondary ripple current in the DC bus is also involved in the calculation of the duty cycle of the switch tube of the bridge arm of the decoupling unit.
  • the control is simple and the secondary ripple voltage ripple suppression effect on the DC side is obvious.
  • the power switch in the decoupling unit works in a current discontinuous mode, which is especially suitable for a new generation of SiC or GaN devices.
  • the power switch turns on with zero current and has no turn-on loss; the turn-off speed is extremely fast, and the turn-off loss is approximately Therefore, the current discontinuous mode can effectively increase the switching frequency of the decoupling unit, reduce the inductance and its loss, thereby increasing the power density.
  • FIG. 1 is a topological diagram of a high power density single-phase cascaded H-bridge multilevel converter provided by an embodiment of the disclosure
  • FIG. 2(a) is a Buck working stage of the Buck type active power decoupling topology provided by an embodiment of the disclosure
  • Fig. 2(b) is the Boost working stage of the Buck-type active power decoupling topology provided by the embodiment of the disclosure
  • FIG. 3(a) is a schematic diagram of the decoupling inductor current in the charging phase of the decoupling capacitor provided by an embodiment of the disclosure
  • FIG. 3(b) is a schematic diagram of the decoupling inductor current during the discharge phase of the decoupling capacitor provided by an embodiment of the disclosure
  • FIG. 4 is a block diagram of a high-power density single-phase cascaded H-bridge multilevel converter control system using the cascade connection of two modules as an example of the embodiment of the disclosure;
  • Figure 5(a) is a grid-side voltage and current waveform diagram provided by an embodiment of the disclosure.
  • Figure 5(b) is a waveform diagram of the DC side output voltage and the decoupling capacitor voltage provided by an embodiment of the disclosure
  • FIG. 6 is a relevant waveform diagram of Buck-type active power decoupling provided by an embodiment of the disclosure.
  • FIG. 7(a) is a waveform diagram of the output voltages on the DC side of the two modules before and after the load mutation provided by the embodiments of the disclosure;
  • FIG. 7(b) is an enlarged view of the output voltage waveforms of the two modules on the DC side before and after the load mutation provided by the embodiments of the disclosure;
  • FIG. 8 is a waveform diagram of the output voltage on the DC side during voltage equalization control after a sudden change in the system provided by an embodiment of the disclosure.
  • FIG. 9(a) is a waveform diagram of the output voltage on the DC side when an unbalanced load is started according to an embodiment of the disclosure
  • FIG. 9(b) is an enlarged view of the output voltage waveform of the DC side when the load is unbalanced in the embodiment of the disclosure.
  • FIG. 10 is a schematic diagram of a single-phase carrier phase-shifted frequency multiplication modulation algorithm provided by an embodiment of the disclosure.
  • the high power density single-phase cascaded H-bridge multilevel converter topology in this embodiment is composed of an AC power supply, an AC side filter inductor, and N cascaded power conversion sub-modules.
  • N is a positive integer greater than or equal to 2.
  • the topology diagram is shown in Figure 1.
  • the AC power supply is connected to the input side of the two cascaded power conversion modules through an AC side filter inductor.
  • the power conversion sub-module includes a full bridge Circuit, power decoupling circuit, DC side supporting capacitor, DC side equivalent load, in which the output of the full bridge circuit is connected to the DC side through a power decoupling circuit, and the power decoupling circuit is used to suppress the secondary ripple power of the DC bus , Reduce the capacity of DC side supporting capacitor.
  • a full bridge circuit with a diode reverse power switch S i1 -S i4 (i 1,2 ) composed of the collector and emitter wherein S i1 S i2 is connected to the arm A i configuration, S i3 emission of and collector connected Si4 B i constituting the bridge arm, midway between the input arm of the full bridge circuit, S i1 common collector connected to the S i3, S i2 and S i4 common emitter connection, common collector The connection point and the common emitter connection point are used as the output terminals of the full bridge circuit.
  • the power decoupling circuit adopts Buck type active power decoupling. It consists of two power switch tubes S i5 and S i6 with reverse diodes, decoupling inductor L ri and decoupling capacitor C si .
  • the emitter of S i5 is connected to
  • the collector of Si6 is connected to form a decoupling bridge arm, and the decoupling inductor and the decoupling capacitor are connected in series to connect the midpoint of the decoupling bridge arm to the common ground.
  • DC side equivalent load capacitance C i and the equivalent load resistance R i is supported by a DC bus connected in parallel.
  • i comp1, i comp2 rectifying unit 1 respectively, the output current of the rectifying unit, for extracting the ripple of the secondary current
  • u cs1, u cs2 are decoupling capacitors C s1
  • the voltage across C s2, i Lr1, i Lr2 is the current flowing through the decoupling inductors L r1 and L r2
  • u dc1 and u dc2 are the DC side output voltages.
  • Buck type active power decoupling control and rectifier unit control are not coupled and can be controlled independently.
  • the real-time extraction scheme of secondary ripple current based on adaptive frequency selector is adopted, and the duty cycle of decoupling bridge arm switches is distributed in real time through calculation.
  • This independent control makes the addition of active power decoupling control not increase the complexity of system control, and through the cooperation with voltage equalization control, the system has both high reliability and high power density.
  • the power decoupling circuit compensates for the DC bus.
  • the principle of secondary ripple power is as follows:
  • the decoupling capacitor is used as an energy storage element to store all the ripple energy.
  • the decoupling inductor is only responsible for energy transfer.
  • the working mode of the Buck type active power decoupling topology is shown in the figure. 2(a)- Figure 2(b) shows that in the process of transferring the secondary ripple power of the DC bus, the decoupling unit switches between two working modes to realize the two-way communication with the ripple energy in the DC bus.
  • the unit operates in the decoupling mode Buck, S 16 is in the oFF state, the energy stored in the decoupling achieved ripple capacitor by turning on and off of the S 15, in S 15 During the turn-on period, the DC bus simultaneously charges the decoupling inductor and the decoupling capacitor. After S 15 is turned off, the decoupling inductor continues to charge the decoupling capacitor. Through this process, the DC bus secondary ripple energy is all stored to the decoupling capacitor.
  • the coupling capacitor as shown in Figure 2(b), when the decoupling unit works in Boost mode, S 15 is in the off state.
  • the decoupling capacitor charges the decoupling inductor and turns off at S 16 After that, the decoupling capacitor and the decoupling inductor release energy to the DC bus at the same time. Through this process, the ripple energy in the decoupling capacitor is all compensated to the DC bus.
  • U dc is the voltage across the DC bus of the decoupling circuit of the first power sub-module
  • U cs is the voltage across the decoupling capacitor C s of the decoupling circuit of the first power sub-module
  • L r is the decoupling inductance of the decoupling circuit of the first power sub-module.
  • the secondary ripple current i ripple on the DC bus can be regarded as a constant value, as shown in Figure 3(a).
  • the total amount of secondary ripple current flowing through the switching tube S 15 can be shaded Part area representation:
  • Ts is the switching period
  • the on-duty ratio of S 16 in each switching cycle can be expressed as:
  • t 1 is the decoupling inductor current rise time
  • t 2 is the decoupling inductor current fall time
  • Buck type active power decoupling bridge arm switch tube duty cycle allocation requires accurate secondary ripple reference current setting, combined with the good frequency selection characteristics of the adaptive filter, the adaptive filter transfer function form is transformed, It can become a frequency selector for the extraction of specific frequency components in the signal. Its transfer function is as follows:
  • ⁇ is an adaptive filter parameter
  • k is a constant coefficient
  • ⁇ is a frequency
  • the actual output voltage of the DC side of each cascaded unit is filtered and compared with the voltage reference value. Each difference is adjusted by PI to obtain the corresponding operating control voltage amplitude.
  • the closed loop controls the grid-side current to generate the average modulation signal factor; the process of generating the average modulation signal factor is:
  • the total voltage of the H-bridge rectifier is divided by the grid-side input voltage amplitude, and then multiplied by the grid-side voltage phase-locked loop output phase as the grid-side current reference value;
  • the average value of the total system voltage is compared with the voltage of each power unit, and the difference is normalized and multiplied by the grid-side voltage phase as the deviation modulation signal factor. At the same time, the voltage balance and the suppression of the secondary pulsation of the DC bus voltage are realized;
  • the final modulation signal of each power conversion unit is compared with the carrier, and the single-pole frequency multiplication carrier phase-shift modulation algorithm is used to generate the driving signal of the switch tube of each power conversion unit;
  • Each power conversion unit extracts the secondary ripple current on the DC bus in real time, and calculates the duty cycle of the power switch in the decoupling unit online to realize the transfer of the secondary ripple power on the DC bus to the decoupling unit.
  • Single-phase cascaded H-bridge rectifier independent active power decoupling control strategy can simultaneously realize unit power factor rectification, active power decoupling control and DC side voltage equalization control.
  • the control strategy is controlled by unit power factor rectifier module and DC bus voltage.
  • Module, voltage equalization module, modulation signal generation module, active power decoupling module and drive signal generation module are composed, and its control block diagram is shown in Figure 4.
  • the unit power factor rectifier module generates the average modulation signal factor of the system through closed-loop control of the grid-side current; the voltage control module generates the operating power of each module and the total voltage of the system through the closed-loop control of the DC bus voltage of the cascade unit, which is the unit power factor
  • the rectifier module and the voltage balance control module provide the outer loop setting; the voltage balance module generates the deviation modulation signal factor of the cascaded unit.
  • the voltage balance control is the basis for the stable operation of the system.
  • the modulation signal generation module generates the final modulation signal by superimposing the cascade module unit deviation modulation signal factor on the system average modulation signal factor;
  • the drive signal generation module uses a single-phase carrier phase shifting frequency multiplication modulation algorithm , The modulation signal is compared with the carrier to generate the drive signal of the switch tube.
  • the single-phase carrier phase-shifting and frequency-multiplying modulation algorithm its working principle is shown in Figure 10.
  • Three cascaded units share a modulating wave, the three-unit carrier lags by 60° sequentially, such as N units lags behind by 180°/N (N is the number of cascaded units); the two bridge arms of the same unit have a carrier difference of 180° ° (or inverted).
  • the switching frequency of the device is f s
  • the equivalent frequency of the grid-side filter is 2*N*f s , which is beneficial to reduce the reactor and improve the quality of grid-connected current.
  • Figure 5 (a)- Figure 5 (b) is a single-phase two-module cascaded H-bridge rectifier active power decoupling simulation analysis diagram, where Figure 5 (a) is the grid-side voltage and current waveforms, Figure 5 (b) is DC side output voltage and decoupling capacitor voltage waveform.
  • the decoupling circuit is in working condition before 0.8 seconds, and the decoupling circuit is disconnected after 0.8 seconds.
  • the DC side voltage fluctuation range is 792 ⁇ 806V, which is about ⁇ 1% of the given value of DC side bus voltage.
  • AC input current THD 3.64%.
  • Figure 6 is the relevant parameter waveform diagram of Buck type active power decoupling loop working process, which more intuitively shows the process of decoupling loop ripple power.
  • Figure 7 (a)- Figure 7 (b) are the simulation waveforms when the system starts when the DC side load is balanced and the load changes suddenly. In 0.4 seconds, the load resistance R1 changes to 80 ⁇ , and R2 changes to 300 ⁇ .
  • Figure 7(a) shows the output voltage waveforms of the two modules on the DC side during the simulation.
  • Figure 7(b) shows the DC side output voltage waveforms of the two modules after the DC side load balance and load mutation. Before the load mutation, the DC side voltage fluctuation range of the two modules is 800 ⁇ 6V, and the voltage fluctuation suppression effect is good.
  • the u dc1 voltage fluctuation range is ⁇ 6V
  • the u dc2 voltage fluctuation range is ⁇ 9V
  • Figure 8 shows the output voltage waveform on the DC side when there is no voltage equalization control after a sudden load change.
  • the simulation results show that the power decoupling function can still be realized when the cascade module has no voltage equalization control, which illustrates the independence of the active function decoupling control strategy based on the secondary ripple current extraction based on the real-time distribution of the duty cycle. And effectiveness.
  • it will undoubtedly increase the voltage stress of the DC side supporting capacitor, and the peak value of the inductor current and the capacitance fluctuation range of the decoupling unit will also change, which is not conducive to the stable operation of the system.
  • Figure 9(a)- Figure 9(b) are the simulation analysis of active power decoupling when the system starts under unbalanced load conditions.
  • Figure 9(a) is the output voltage waveform of the DC side when the load is unbalanced
  • Figure 9(b) It is 0.4 second to 0.5 second zoomed in waveform. It can be seen from the figure that the ripple voltage suppression effect is good.
  • the system can simultaneously suppress the DC side ripple voltage suppression, and the voltage equalization control of the various levels of the unit, so that the system can achieve Safe, reliable, stable operation with high power density.

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Abstract

本发明提供了一种高功率密度单相级联H桥整流器、控制方法及控制系统。其中，高功率密度单相级联H桥整流器，包括：交流网侧滤波电感和至少两个级联的功率变换单元；每个功率变换单元包括并联连接的H桥功率单元、解耦单元和直流侧等效负载；每个解耦单元均为一个独立式降压型有源功率解耦回路，所述解耦单元用于缓冲二次纹波功率，以减小直流母线电容的容量。

Description

高功率密度单相级联H桥整流器、控制方法及控制系统 技术领域
本公开属于级联式多电平变换器领域，尤其涉及一种高功率密度单相级联H桥整流器、控制方法及控制系统。
背景技术
本部分的陈述仅仅是提供了与本公开相关的背景技术信息，不必然构成在先技术。
级联式多电平电力电子变压器(PET，Power Electronic Transformer)功率变换器，直流侧存在二倍频的功率脉动，为降低直流母线电压纹波，可采用增加滤波电容或者无源LC二次滤波网络的方式，但发明人发现，这样增加了系统重量和体积，尤其是LC滤波网络在谐波电流作用下非线性磁芯特性畸变，易导致系统不稳定。
增加电容固然可以降低电压纹波，但发明人发现，其级联式多电平电力电子变压器的体积和成本都将提高，不利于机载级联式多电平PET功率密度的提升。在系统直流电压脉动和系统成本与体积之间的矛盾无法调和的情况，不得不提升功率开关模块的电压裕量，这样势必会降低系统可靠性和功率变换能力。
发明内容
为了解决上述问题，本公开的第一个方面提供一种高功率密度单相级联H桥整流器，其将独立式有源功率解耦电路用于级联多电平变换器功率单元，为二次功率脉动提供通路，相比纯增加电容方式，在相同变换功率和相同直流电压纹波下，滤波电容数量理论上可降低10倍，可以有效降低系统体积和重量， 增加系统功率密度和可靠性。
为了实现上述目的，本公开采用如下技术方案：
一种高功率密度单相级联H桥整流器，包括：
交流网侧滤波电感和至少两个级联的功率变换单元；每个功率变换单元包括并联连接的H桥功率单元、解耦单元和直流侧等效负载；每个解耦单元均为一个独立式降压型有源功率解耦回路，所述解耦单元用于缓冲二次纹波功率，以减小直流母线电容的容量。
作为一种实施方式，所述解耦单元包括串联功率模块，所述串联功率模块由两只串联的开关功率元件构成，串联功率模块的两端分别与相应功率变换单元直流侧的正负母线相连，串联功率模块的中间连接点接有串联的电感和电容回路，电容的另一端接到相应功率变换单元直流侧的负母线上。
作为一种实施方式，每个功率变换单元还包括：与直流侧等效负载并联连接的直流滤波电容，所述直流滤波电容用于消除高次谐波。
作为一种实施方式，所述直流侧等效负载为电阻元件、DC/DC变换器或者电容元件。
为了解决上述问题，本公开的第二个方面提供一种高功率密度单相级联H桥整流器的控制方法，其运用直流侧电压均衡和有源功率解耦综合控制策略，同时实现级联单元间直流侧输出电压均衡和二次纹波电压脉动抑制。
为了实现上述目的，本公开采用如下技术方案：
一种高功率密度单相级联H桥整流器的控制方法，包括：
闭环控制各个功率变换单元直流母线电压，生成各个功率变换单元的运行电压以及H桥整流器的总电压；
将H桥整流器的总电压作为外环给定值，闭环控制网侧电流，生成平均调制信号因子；
将系统总电压平均值分别与各功率单元电压比较，差值标幺化后与网侧电压相位相乘后的结果作为偏差调制信号因子，同时实现电压均衡与直流母线电压二次脉动的抑制；
将各个功率变换单元的偏差调制信号因子与平均调制信号因子叠加，生成各个功率变换单元的最终调制信号；
将各个功率变换单元的最终调制信号与载波比较，采用单极倍频载波移相调制算法产生各个功率变换单元的开关管的驱动信号；
实时各个功率变换单元的提取直流母线上的二次纹波电流，在线计算解耦单元中功率开关管占空比，以实现直流母线上二次纹波功率向解耦单元的转移。
作为一种实施方式，生成各个功率变换单元的运行电压以及H桥整流器的总电压的过程为：
为实现各个级联单元直流侧母线电压稳定平衡，将各级联单元直流侧实际输出电压滤波后与电压参考值比较，各个差值均经PI调节得到对应的相应运行控制电压幅值，在系统运行过程中，实时将各单元运行控制电压相加得到H桥整流器的总电压。
作为一种实施方式，生成平均调制信号因子的过程为：
H桥整流器的总电压除以网侧输入电压幅值标幺化，再与网侧电压锁相环输出相位相乘，作为网侧电流参考值；
将实际检测的电网电流瞬时值与网侧电流参考值比较，再经网侧滤波电感电流闭环无静差跟踪，输出平均调制信号因子。
为了解决上述问题，本公开的第三个方面提供一种高功率密度单相级联H桥整流器的控制系统，其运用直流侧电压均衡和有源功率解耦综合控制策略，同时实现级联单元间直流侧输出电压均衡和二次纹波电压脉动抑制。
为了实现上述目的，本公开采用如下技术方案：
一种高功率密度单相级联H桥整流器的控制系统，包括：
直流母线电压控制模块，其用于闭环控制各个功率变换单元直流母线电压，生成各个功率变换单元的运行电压以及H桥整流器的总电压；
单位功率因数整流模块，其用于将H桥整流器的总电压作为外环给定值，闭环控制网侧电流，生成平均调制信号因子；
电压均衡模块，其用于将系统总电压平均值分别与各功率单元电压比较，差值标幺化后与网侧电压相位相乘后的结果作为偏差调制信号因子，同时实现电压均衡与直流母线电压二次脉动的抑制；
调制信号生成模块，其用于将各个功率变换单元的偏差调制信号因子与平均调制信号因子叠加，生成各个功率变换单元的最终调制信号；
驱动信号生成模块，其用于将各个功率变换单元的最终调制信号与载波比较，采用单极倍频载波移相调制算法产生各个功率变换单元的开关管的驱动信号；
有源功率解耦控制模块，其用于实时各个功率变换单元的提取直流母线上的二次纹波电流，在线计算解耦单元中功率开关管占空比，以实现直流母线上二次纹波功率向解耦单元的转移。
作为一种实施方式，在所述直流母线电压控制模块中，生成各个功率变换单元的运行电压以及H桥整流器的总电压的过程为：
为实现各个级联单元直流侧母线电压稳定平衡，将各级联单元直流侧实际输出电压滤波后与电压参考值比较，各个差值均经PI调节得到对应的相应运行控制电压幅值，在系统运行过程中，实时将各单元运行控制电压相加得到H桥整流器的总电压。
作为一种实施方式，在所述单位功率因数整流模块中，生成平均调制信号因子的过程为：
H桥整流器的总电压除以网侧输入电压幅值标幺化，再与网侧电压锁相环输出相位相乘，作为网侧电流参考值；
将实际检测的电网电流瞬时值与网侧电流参考值比较，再经网侧滤波电感电流闭环无静差跟踪，输出平均调制信号因子。
本公开的有益效果是：
(1)拓扑方面，在单相级联型H桥多电平变换器的直流侧增加了一个降压(Buck)型有源功率解耦单元，有利于降低电容电压。
(2)控制方面：运用直流侧电压均衡和有源功率解耦综合控制策略，同时实现级联单元间直流侧输出电压均衡和二次纹波电压脉动抑制；运用自适应选频器，实时提取直流母线中的二次纹波电流，并参与到解耦单元桥臂开关管占空比的计算中，控制简单且直流侧二次纹波电压脉动抑制效果明显。
(3)所述解耦单元中的功率开关管工作在电流断续方式，尤其适合新一代SiC或GaN器件，功率开关零电流开通，无开通损耗；关断速度极快，关断损耗近似为零，因此电流断续模式可有效提高解耦单元的开关频率，减小电感及其损耗，从而提高功率密度。
附图说明
构成本公开的一部分的说明书附图用来提供对本公开的进一步理解，本公开的示意性实施例及其说明用于解释本公开，并不构成对本公开的不当限定。
图1为本公开实施例提供的高功率密度单相级联H桥多电平变换器拓扑图；
图2(a)为本公开实施例提供的Buck型有源功率解耦拓扑的Buck工作阶段；
图2(b)为本公开实施例提供的Buck型有源功率解耦拓扑的Boost工作阶段；
图3(a)为本公开实施例提供的解耦电容充电阶段解耦电感电流示意图；
图3(b)为本公开实施例提供的解耦电容放电阶段解耦电感电流示意图；
图4为本公开实施例提供的两模块级联为例的高功率密度单相级联H桥多电平变换器控制系统框图；
图5(a)为本公开实施例提供的网侧电压、电流波形图；
图5(b)为本公开实施例提供的直流侧输出电压和解耦电容电压波形图；
图6为本公开实施例提供的Buck型有源功率解耦相关波形图；
图7(a)为本公开实施例提供的负载突变前后两模块直流侧输出电压波形图；
图7(b)为本公开实施例提供的负载突变前后两模块直流侧输出电压波形放大图；
图8为本公开实施例提供的系统突变后切除电压均衡控制时直流侧输出电压波形图；
图9(a)为本公开实施例提供的负载不平衡启动时直流侧输出电压波形图；
图9(b)为本公开实施例提供的负载不平衡启动时直流侧输出电压波形放大图；
图10为本公开实施例提供的单相载波移相倍频调制算法原理图。
具体实施方式
下面结合附图与实施例对本公开作进一步说明。
应该指出，以下详细说明都是例示性的，旨在对本公开提供进一步的说明。除非另有指明，本文使用的所有技术和科学术语具有与本公开所属技术领域的普通技术人员通常理解的相同含义。
需要注意的是，这里所使用的术语仅是为了描述具体实施方式，而非意图限制根据本公开的示例性实施方式。如在这里所使用的，除非上下文另外明确指出，否则单数形式也意图包括复数形式，此外，还应当理解的是，当在本说明书中使用术语“包含”和/或“包括”时，其指明存在特征、步骤、操作、器件、组件和/或它们的组合。
本实施例所述高功率密度单相级联H桥多电平变换器拓扑由交流电源，交流侧滤波电感，N个级联的功率变换子模块构成。其中，N为大于或等于2的正整数。
为便于分析，以两模块级联为例，其拓扑结构图如图1所示，交流电源经过交流侧滤波电感连接到两个级联的功率变换模块的输入侧，功率变换子模块包括全桥电路，功率解耦电路，直流侧支撑电容，直流侧等效负载，其中全桥电路的输出经过功率解耦电路接到直流侧，所述功率解耦回路用于抑制直流母线二次纹波功率，减小直流侧支撑电容容量。
全桥电路由并有反接二极管的功率开关管S i1-S i4(i＝1,2)组成，其中S i1的 发射极与S i2的集电极相连构成A i桥臂，S i3的发射极与Si4的集电极相连构成B i桥臂，两桥臂的中点为全桥电路的输入端，S i1与S i3共集电极连接，S i2与S i4共发射极连接，共集电极连接点和共发射极连接点作为全桥电路的输出端。
功率解耦电路采用Buck型有源功率解耦，由两个并有反接二极管的功率开关管S i5，S i6，解耦电感L ri和解耦电容C si组成，S i5的发射极与S i6的集电极相连构成解耦桥臂，解耦电感与解耦电容串联，连接解耦桥臂中点与公共地。
直流侧等效负载由直流母线支撑电容C i与等效负载电阻R i并联组成。
i comp1、i comp2分别为整流单元1、整流单元2的输出电流，用于二次纹波电流的提取，u cs1、u cs2分别为解耦电容C s1、C s2两端电压，i Lr1、i Lr2为流过解耦电感L r1、L r2的电流，u dc1、u dc2为直流侧输出电压。
Buck型有源功率解耦控制与整流单元控制无耦合，可以独立控制，采用基于自适应选频器的二次纹波电流实时提取方案，通过运算实时分配解耦桥臂开关管占空比。这种独立式的控制使得有源功率解耦控制的加入没有增加系统控制上的复杂性，且通过和电压均衡控制的配合使得系统高可靠性和高功率密度兼备，功率解耦电路补偿直流母线二次纹波功率的原理如下：
以第一个功率子模块的解耦电路为例，解耦电容作为储能元件储存全部纹波能量，解耦电感只负责能量的传递，Buck型有源功率解耦拓扑的工作模态如图2(a)-图2(b)所示，在进行直流母线二次纹波功率的转移过程中，解耦单元在两种工作模态中转换，以实现和直流母线中纹波能量的双向交互，如图2(a)所示，解耦单元工作在Buck模式，S 16处于关断状态，通过S 15的导通和关断实现纹波能量在解耦电容中的储存，在S 15导通期间，直流母线同时给解耦电感和解耦电容充电，在S 15关断后，解耦电感继续给解耦电容充电，通过这一过程，直 流母线二次纹波能量全部储存到解耦电容中，如图2(b)所示，解耦单元工作在Boost模式时，S 15处于关断状态，在S 16导通期间，解耦电容给解耦电感充电，在S 16关断后，解耦电容和解耦电感同时将能量释放到直流母线上，通过这一过程，解耦电容中的纹波能量全部补偿到直流母线上。
有源功率解耦模块开关管占空比计算：
在Buck工作阶段，S 15导通时，解耦电感充电，电流上升斜率为：
Figure PCTCN2020097207-appb-000001
S 15关断时，解耦电感放电，电流下降斜率为：
Figure PCTCN2020097207-appb-000002
在Boost工作阶段，S 16导通时，解耦电感充电，电流上升斜率为：
Figure PCTCN2020097207-appb-000003
S 16关断时，解耦电感放电，电流下降斜率为：
Figure PCTCN2020097207-appb-000004
其中，U dc为第一个功率子模块的解耦电路的直流母线两端的电压；
U cs为第一个功率子模块的解耦电路的解耦电容C s两端的电压；
L r为第一个功率子模块的解耦电路的解耦电感。
一个开关周期内，直流母线上二次纹波电流i ripple可看作恒值，如图3(a)，一个开关周期内，流过开关管S 15的二次纹波电流总量可以用阴影部分面积表示：
Figure PCTCN2020097207-appb-000005
则在Buck工作阶段，每个开关周期内S 15的导通占空比可表示为：
Figure PCTCN2020097207-appb-000006
如图3(b)，同理，在Boost工作阶段，一个开关周期内流过开关管S 15的二次纹波电流总量可以用阴影部分面积表示：
Figure PCTCN2020097207-appb-000007
Ts为开关周期；
图3(b)中t 1和t 2之间的关系可表示为：
t 1Boost_up＝t 2Boost_down      (8)
则有：
Figure PCTCN2020097207-appb-000008
将式(9)代入(7)可得：
Figure PCTCN2020097207-appb-000009
则Boost工作阶段，每个开关周期内S 16的导通占空比可表示为：
Figure PCTCN2020097207-appb-000010
其中，t 1为解耦电感电流上升时间，t 2为解耦电感电流下降时间。
Buck型有源功率解耦桥臂开关管占空比的分配需要精确的二次纹波参考电流给定，结合自适应滤波器良好的选频特性，将自适应滤波器传递函数形式加以变换，即可成为选频器，用于信号中特定频率分量的提取，其传递函数如下：
Figure PCTCN2020097207-appb-000011
其中，ξ为自适应滤波器参数，k为常系数，ω为频率。
本实施例的一种高功率密度单相级联H桥整流器的控制方法原理为：
闭环控制各个功率变换单元直流母线电压，生成各个功率变换单元的运行电压以及H桥整流器的总电压；
其中，生成各个功率变换单元的运行电压以及H桥整流器的总电压的过程为：
为实现各个级联单元直流侧母线电压稳定平衡，将各级联单元直流侧实际输出电压滤波后与电压参考值比较，各个差值均经PI调节得到对应的相应运行控制电压幅值，在系统运行过程中，实时将各单元运行控制电压相加得到H桥整流器的总电压；
将H桥整流器的总电压作为外环给定值，闭环控制网侧电流，生成平均调制信号因子；其中，生成平均调制信号因子的过程为：
H桥整流器的总电压除以网侧输入电压幅值标幺化，再与网侧电压锁相环输出相位相乘，作为网侧电流参考值；
将实际检测的电网电流瞬时值与网侧电流参考值比较，再经网侧滤波电感电流闭环无静差跟踪，输出平均调制信号因子；
将系统总电压平均值分别与各功率单元电压比较，差值标幺化后与网侧电压相位相乘后的结果作为偏差调制信号因子，同时实现电压均衡与直流母线电压二次脉动的抑制；
将各个功率变换单元的偏差调制信号因子与平均调制信号因子叠加，生成各个功率变换单元的最终调制信号；
将各个功率变换单元的最终调制信号与载波比较，采用单极倍频载波移相调制算法产生各个功率变换单元的开关管的驱动信号；
实时各个功率变换单元的提取直流母线上的二次纹波电流，在线计算解耦 单元中功率开关管占空比，以实现直流母线上二次纹波功率向解耦单元的转移。
单相级联H桥整流器独立式有源功率解耦控制策略能够同时实现单位功率因数整流、有源功率解耦控制和直流侧电压均衡控制，控制策略由单位功率因数整流模块、直流母线电压控制模块、电压均衡模块、调制信号生成模块、有源功率解耦模块以及驱动信号生成模块组成，其控制框图如图4所示。单位功率因数整流模块通过对网侧电流的闭环控制生成系统的平均调制信号因子；电压控制模块，通过对级联单元直流母线电压的闭环控制生成各模块运行功率以及系统总电压，为单位功率因数整流模块和电压均衡控制模块提供外环给定；电压均衡模块生成级联单元的偏差调制信号因子，电压均衡控制是系统稳定运行的基础，与有源功率解耦控制模块配合同时实现电压均衡与直流母线电压二次脉动的抑制；调制信号生成模块通过将级联模块单元偏差调制信号因子叠加到系统平均调制信号因子上生成最终调制信号；驱动信号生成模块通过单相载波移相倍频调制算法，将调制信号与载波比较产生开关管的驱动信号。
单相载波移相倍频调制算法，其工作原理如图10所示，以三单元级联H桥整流器载波移相倍频调制为例。三个级联单元共用一个调制波，三单元载波依次滞后60°，如N个单元则依次滞后180°/N(N为级联单元数量)；同一个单元左右两个桥臂采用载波相差180°(或者反相)。如此，若器件开关频率为f s，则网侧滤波器等效频率为2*N*f s，有利于减小电抗器，提高并网电流质量。
经上述控制，可实现单位功率因数整流，直流侧输出电压均衡控制以及有源功率解耦。
在MATLAB/Simulink中进行仿真实验，验证所提拓扑及控制方案的有效性，仿真参数如表1所示。
表1
Figure PCTCN2020097207-appb-000012
图5(a)-图5(b)为单相两模块级联H桥整流器有源功率解耦仿真分析图，其中图5(a)为网侧电压、电流波形，图5(b)为直流侧输出电压和解耦电容电压波形。在0.8秒前解耦回路处于工作状态，0.8秒后断开解耦回路。解耦时，直流侧电压波动范围792～806V，约为直流侧母线电压给定值的±1％，交流输入电流THD＝3.64％，切除解耦回路后直流母线纹波电压范围增大到719～878V，交流输入电流THD＝4.85％。若想使无功率解耦时的纹波电压波动范围保持±1％，则需要直流母线支撑电容的容值为1000uF。所提拓扑及控制方案极大程度地抑制了二次纹波功率，减小了直流母线支撑电容。
图6为Buck型有源功率解耦回路工作过程的相关参数波形图，更直观的显 示了解耦回路纹波功率解耦的过程。
图7(a)-图7(b)为系统在直流侧负载平衡时启动，负载发生突变时的仿真波形。在0.4秒，负载电阻R1突变为80Ω，R2突变为300Ω。图7(a)为仿真过程中两模块直流侧输出电压波形。图7(b)分别为直流侧负载均衡和负载突变之后，两模块直流侧输出电压波形。在负载突变前两模块直流侧电压波动范围为800±6V，电压波动抑制效果良好，在负载突变之后，u dc1电压波动范围为±6V，u dc2电压波动范围为±9V，说明负载突变之后，在电压均衡控制和有源功率解耦的综合控制下，能够实现低电压脉动和电压均衡输出。
图8为负载突变之后无电压均衡控制作用时，直流侧输出电压波形。仿真效果表明在级联模块无电压均衡控制时，功率解耦功能仍能实现，说明了所采用的基于二次纹波电流提取的实时分配占空比的有源功能解耦控制策略的独立性和有效性。但是，未加电压均衡控制策略，无疑会增加直流侧支撑电容电压应力，解耦单元电感电流峰值和电容波动范围也会发生变化，不利于系统的稳定运行。
图9(a)-图9(b)为系统在负载不平衡条件下启动时有源功率解耦仿真分析，图9(a)为负载不平衡启动直流侧输出电压波形，图9(b)为0.4秒到0.5秒的放大波形。从图中可以得出纹波电压抑制效果良好，经电压均衡和有源功率解耦综合控制，系统能够同时抑制直流侧纹波电压抑制，和各级联单元的电压均衡控制，使得系统能够实现安全、可靠、高功率密度稳定运行。
上述仿真实验结果表明，本实施例所述具备独立式有源功率解耦的单相级联H桥整流器及其控制策略在保证系统单位功率因数以及直流母线电压平衡的条件下，较好地实现了有源功率解耦功能，有效的抑制了直流侧纹波功率，减 小了直流侧支撑电容的容值，提升了系统的可靠性，提高了系统的功率密度。
以上所述仅为本公开的优选实施例而已，并不用于限制本公开，对于本领域的技术人员来说，本公开可以有各种更改和变化。凡在本公开的精神和原则之内，所作的任何修改、等同替换、改进等，均应包含在本公开的保护范围之内。

Claims (9)

  1. 一种高功率密度单相级联H桥整流器的控制方法，所述的一种高功率密度单相级联H桥整流器包括：交流网侧滤波电感和至少两个级联的功率变换单元；每个功率变换单元包括并联连接的H桥功率单元、解耦单元和直流侧等效负载；每个解耦单元均为一个独立式降压型有源功率解耦回路，所述解耦单元用于缓冲二次纹波功率，以减小直流母线电容的容量；其特征在于，
    闭环控制各个功率变换单元直流母线电压，生成各个功率变换单元的运行电压以及H桥整流器的总电压；
    将H桥整流器的总电压作为外环给定值，闭环控制网侧电流，生成平均调制信号因子；
    将系统总电压平均值分别与各功率单元电压比较，差值标幺化后与网侧电压相位相乘后的结果作为偏差调制信号因子，同时实现电压均衡与直流母线电压二次脉动的抑制；
    将各个功率变换单元的偏差调制信号因子与平均调制信号因子叠加，生成各个功率变换单元的最终调制信号；
    将各个功率变换单元的最终调制信号与载波比较，采用单极倍频载波移相调制算法产生各个功率变换单元的开关管的驱动信号；
    实时各个功率变换单元的提取直流母线上的二次纹波电流，在线计算解耦单元中功率开关管占空比，以实现直流母线上二次纹波功率向解耦单元的转移。
  2. 如权利要求1所述的一种高功率密度单相级联H桥整流器的控制方法，其特征在于，所述解耦单元包括串联功率模块，所述串联功率模块由两只串联的开关功率元件构成，串联功率模块的两端分别与相应功率变换单元直流侧的正负母线相连，串联功率模块的中间连接点接有串联的电感和电容回路，电容 的另一端接到相应功率变换单元直流侧的负母线上。
  3. 如权利要求1所述的一种高功率密度单相级联H桥整流器控制方法，其特征在于，每个功率变换单元还包括：与直流侧等效负载并联连接的直流滤波电容，所述直流滤波电容用于消除高次谐波。
  4. 如权利要求1所述的一种高功率密度单相级联H桥整流器的控制方法，其特征在于，所述直流侧等效负载为电阻元件、DC/DC变换器或者电容元件。
  5. 如权利要求1所述的一种高功率密度单相级联H桥整流器的控制方法，其特征在于，生成各个功率变换单元的运行电压以及H桥整流器的总电压的过程为：
    为实现各个级联单元直流侧母线电压稳定平衡，将各级联单元直流侧实际输出电压滤波后与电压参考值比较，各个差值均经PI调节得到对应的相应运行控制电压幅值，在系统运行过程中，实时将各单元运行控制电压相加得到H桥整流器的总电压。
  6. 如权利要求1所述的一种高功率密度单相级联H桥整流器的控制方法，其特征在于，生成平均调制信号因子的过程为：
    H桥整流器的总电压除以网侧输入电压幅值标幺化，再与网侧电压锁相环输出相位相乘，作为网侧电流参考值；
    将实际检测的电网电流瞬时值与网侧电流参考值比较，再经网侧滤波电感电流闭环无静差跟踪，输出平均调制信号因子。
  7. 一种高功率密度单相级联H桥整流器的控制系统，所述的一种高功率密度单相级联H桥整流器包括：交流网侧滤波电感和至少两个级联的功率变换单元；每个功率变换单元包括并联连接的H桥功率单元、解耦单元和直流侧等效 负载；每个解耦单元均为一个独立式降压型有源功率解耦回路，所述解耦单元用于缓冲二次纹波功率，以减小直流母线电容的容量；其特征在于，
    直流母线电压控制模块，其用于闭环控制各个功率变换单元直流母线电压，生成各个功率变换单元的运行电压以及H桥整流器的总电压；
    单位功率因数整流模块，其用于将H桥整流器的总电压作为外环给定值，闭环控制网侧电流，生成平均调制信号因子；
    电压均衡模块，其用于将系统总电压平均值分别与各功率单元电压比较，差值标幺化后与网侧电压相位相乘后的结果作为偏差调制信号因子，同时实现电压均衡与直流母线电压二次脉动的抑制；
    调制信号生成模块，其用于将各个功率变换单元的偏差调制信号因子与平均调制信号因子叠加，生成各个功率变换单元的最终调制信号；
    驱动信号生成模块，其用于将各个功率变换单元的最终调制信号与载波比较，采用单极倍频载波移相调制算法产生各个功率变换单元的开关管的驱动信号；
    有源功率解耦控制模块，其用于实时各个功率变换单元的提取直流母线上的二次纹波电流，在线计算解耦单元中功率开关管占空比，以实现直流母线上二次纹波功率向解耦单元的转移。
  8. 如权利要求7所述的一种高功率密度单相级联H桥整流器的控制系统，其特征在于，在所述直流母线电压控制模块中，生成各个功率变换单元的运行电压以及H桥整流器的总电压的过程为：
    为实现各个级联单元直流侧母线电压稳定平衡，将各级联单元直流侧实际输出电压滤波后与电压参考值比较，各个差值均经PI调节得到对应的相应运行 控制电压幅值，在系统运行过程中，实时将各单元运行控制电压相加得到H桥整流器的总电压。
  9. 如权利要求7所述的一种高功率密度单相级联H桥整流器的控制系统，其特征在于，在所述单位功率因数整流模块中，生成平均调制信号因子的过程为：
    H桥整流器的总电压除以网侧输入电压幅值标幺化，再与网侧电压锁相环输出相位相乘，作为网侧电流参考值；
    将实际检测的电网电流瞬时值与网侧电流参考值比较，再经网侧滤波电感电流闭环无静差跟踪，输出平均调制信号因子。
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