WO2020194637A1 - Procédé de commande et dispositif de commande pour véhicule électrique - Google Patents

Procédé de commande et dispositif de commande pour véhicule électrique Download PDF

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Publication number
WO2020194637A1
WO2020194637A1 PCT/JP2019/013468 JP2019013468W WO2020194637A1 WO 2020194637 A1 WO2020194637 A1 WO 2020194637A1 JP 2019013468 W JP2019013468 W JP 2019013468W WO 2020194637 A1 WO2020194637 A1 WO 2020194637A1
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Prior art keywords
command value
axis current
axis
value
calculated
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PCT/JP2019/013468
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English (en)
Japanese (ja)
Inventor
翔 大野
弘征 小松
藤原　健吾
中島　孝
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日産自動車株式会社
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Priority to PCT/JP2019/013468 priority Critical patent/WO2020194637A1/fr
Publication of WO2020194637A1 publication Critical patent/WO2020194637A1/fr

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    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L15/00—Methods, circuits, or devices for controlling the traction-motor speed of electrically-propelled vehicles
    • B60L15/20—Methods, circuits, or devices for controlling the traction-motor speed of electrically-propelled vehicles for control of the vehicle or its driving motor to achieve a desired performance, e.g. speed, torque, programmed variation of speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/64—Electric machine technologies in electromobility
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

Definitions

  • the present invention relates to a control method for an electric vehicle and a control device.
  • vibration suppression control that reduces torsional vibration of a drive shaft that connects the motor and drive wheels has been used.
  • the above-mentioned vibration suppression control is performed because the rotor magnetic flux generated in the motor fluctuates. It is difficult to apply it as it is.
  • JP5939316B discloses a method of applying the above-mentioned vibration damping control to an induction motor in which the rotor magnetic flux fluctuates.
  • JP5939316B is a control method for applying the above vibration damping control to an induction motor by correcting the torque current based on the exciting current ( ⁇ -axis current). Therefore, even if the control method is applied to a field winding type synchronous motor in which it is necessary to consider the current flowing through the field winding of the rotor (f-axis current) and the d-axis current that generates reluctance torque. It is difficult to obtain a vibration damping effect.
  • An object of the present invention is to provide a technique for applying vibration damping control for reducing torsional vibration of a drive shaft connecting a motor and a drive wheel to a field winding type synchronous motor.
  • the method for controlling an electric vehicle is to use an electric vehicle using a winding field synchronous motor as a drive source, which includes a rotor having a rotor winding and a stator having a stator winding.
  • This is a control method for an electric vehicle that controls a stator current flowing through a stator winding and a rotor current flowing through a rotor winding.
  • a basic torque command value is set based on vehicle information, and based on the basic torque command value and vehicle information, a d-axis current command value for a stator current, a first q-axis current command value, and The f-axis current command value for the rotor current is calculated, and the magnetic flux estimated value, which is the estimated value of the torque generated in the rotor, is calculated based on the d-axis current command value and the f-axis current command value, and the first q
  • the first torque command value is calculated based on the shaft current command value and the estimated magnetic flux value, and the vibration of the drive shaft torque transmission system of the electric vehicle is suppressed with respect to the first torque command value based on the vehicle information.
  • the vibration damping torque command value is calculated, and the second q-axis current command value is calculated based on the magnetic flux estimated value and the vibration damping torque command value. Then, the stator current and the rotor current are controlled based on the second q-axis current command value, the d-axis current command value, and the f-axis current command value.
  • FIG. 1 is a schematic configuration diagram of a vehicle system to which the electric vehicle control method of the first embodiment is applied.
  • FIG. 2 is a flowchart showing a flow of processing performed by the electric motor controller.
  • FIG. 3 is a diagram showing an example of an accelerator opening degree-torque table.
  • FIG. 4 is a block diagram of the motor control system of the first embodiment.
  • FIG. 5 is a block diagram of the q-axis current control unit.
  • FIG. 6 is a block diagram of the d-axis current control unit.
  • FIG. 7 is a block diagram of the f-axis current control unit.
  • FIG. 8 is a block diagram of the vibration damping control calculation unit.
  • FIG. 9 is a block diagram of the magnetic flux estimator.
  • FIG. 1 is a schematic configuration diagram of a vehicle system to which the electric vehicle control method of the first embodiment is applied.
  • FIG. 2 is a flowchart showing a flow of processing performed by the electric motor controller.
  • FIG. 3 is
  • FIG. 10 is a block diagram of a reluctance torque equivalent magnetic flux estimator.
  • FIG. 11 is a block diagram of the field magnetic flux estimator.
  • FIG. 12 is a block diagram of a vibration damping torque command value calculator.
  • FIG. 13 is a diagram for explaining the equation of motion of the electric vehicle.
  • FIG. 14 is a diagram showing the characteristics of the bandpass filter H (s).
  • FIG. 15 is a time chart showing the control result by the control method of the electric vehicle.
  • FIG. 16 is a block diagram of the motor control system of the second embodiment.
  • FIG. 17 is a block diagram of the f-axis current control unit.
  • FIG. 18 is a block diagram of the f-axis F / F compensator.
  • FIG. 19 is a block diagram of an f-axis current model.
  • FIG. 20 is a block diagram of an f-axis current F / B model.
  • FIG. 21 is a block diagram of the f-axis limit processing unit.
  • FIG. 22 is another example of the block diagram of the f-axis limit processing unit.
  • FIG. 23 is a block diagram of the f-axis F / B compensator.
  • FIG. 24 is a block diagram of an f-axis robust compensator.
  • FIG. 25 is a flowchart showing the control process of the motor.
  • FIG. 26 is a block diagram of a reluctance torque equivalent magnetic flux estimator.
  • FIG. 27 is a block diagram of the field magnetic flux estimator.
  • FIG. 28 is a block diagram of the vibration damping control calculation unit.
  • FIG. 29 is a time chart showing the control result by the control method of the electric vehicle.
  • FIG. 1 is a block diagram showing a configuration example of a motor control system 100 to which the electric vehicle control method according to the embodiment of the present invention is applied.
  • An electric vehicle is a vehicle that is provided with at least one winding field type synchronous motor (hereinafter, also simply referred to as a motor) as a part or all of the drive source of the vehicle and can run by the driving force of the motor. It includes electric vehicles and hybrid vehicles.
  • the battery 1 discharges the drive power of the winding field type synchronous motor 4 and charges the regenerative power of the motor 4.
  • the electric motor controller 2 (hereinafter, also simply referred to as a controller) is composed of, for example, a central processing unit (CPU), a read-only memory (ROM), a random access memory (RAM), and an input / output interface (I / O interface). To.
  • the controller 2 has a vehicle speed V, an accelerator opening degree ⁇ , an electric angle ⁇ re of the motor 4, a stator current of the motor 4 (iu, iv, iw in the case of three-phase alternating current), and a rotor current (if) of the motor 4.
  • Signals of various vehicle variables indicating the vehicle state such as are input as digital signals.
  • the controller 2 generates a PWM signal for controlling the motor 4 based on the input signal. Further, the controller 2 generates a drive signal of the inverter 3 according to the generated PWM signal.
  • the inverter 3 is supplied from the battery 1 by turning on / off two switching elements (for example, power semiconductor elements such as IGBTs and MOS-FETs) provided for each phase in order to control the stator current.
  • the direct current is converted to alternating current or inversely converted, and a desired current is passed through the motor 4.
  • two pairs (four in total) of switching elements for example, power semiconductor elements such as IGBTs and MOS-FETs
  • a desired current is passed through the rotor winding.
  • two of the two pairs of switching elements located diagonally may be replaced with diodes.
  • the winding field type synchronous motor 4 (hereinafter, simply referred to as “motor 4”) has a rotor having a rotor winding (field winding) and a stator having a stator winding (armature winding). It is a winding field type synchronous motor equipped with.
  • the motor 4 serves as a drive source for the vehicle.
  • the motor 4 is controlled by controlling the rotor current flowing through the rotor winding and the stator current flowing through the stator winding.
  • the motor 4 generates a drive torque by the current supplied from the inverter 3, and transmits the drive force to the left and right drive wheels 9 via the speed reducer 5 and the drive shaft 8.
  • the motor 4 recovers the kinetic energy of the vehicle as electric energy by generating a regenerative driving force when the motor 4 is rotated by the drive wheels 9 while the vehicle is traveling.
  • the inverter 3 converts the alternating current generated during the regenerative operation of the motor 4 into a direct current and supplies it to the battery 1.
  • the current sensor 7 detects the three-phase currents iu, iv, and iwa (stator current) flowing in the stator winding of the motor 4, and also detects the current if (rotor current) flowing in the rotor winding of the motor 4. To do. However, since the sum of the three-phase AC currents iu, iv, and iw is 0 for the stator current, the current of any two phases may be detected and the current of the remaining one phase may be obtained by calculation.
  • the rotation sensor 6 is, for example, a resolver or an encoder, and detects the rotor phase ⁇ of the motor 4.
  • FIG. 2 is a flowchart showing the flow of processing performed by the controller 2.
  • the processes according to steps S201 to S204 are programmed in the controller 2 so as to be constantly executed at regular intervals while the vehicle system is running.
  • step S201 a signal indicating the vehicle state is input to the controller 2.
  • the DC voltage value V dc (V) of the battery 1 is input.
  • the vehicle speed V (km / h) is acquired by communication from a meter (not shown), a vehicle speed sensor, or another controller such as a brake controller.
  • the controller 2 obtains the vehicle speed v (m / s) by multiplying the rotor mechanical angular velocity ⁇ m by the tire driving radius r and dividing by the gear ratio of the final gear, and changes from m / s to km / s.
  • the vehicle speed V (km / h) is obtained by multiplying by the unit conversion coefficient (3600/1000).
  • the accelerator opening ⁇ (%) is obtained from an accelerator opening sensor (not shown).
  • the accelerator opening degree ⁇ (%) may be obtained from another controller such as a vehicle controller (not shown).
  • the electric angle ⁇ re (rad) of the motor 4 is acquired from the rotation sensor 6.
  • the electric angular velocity ⁇ re is divided by the pole pair number p of the electric motor to obtain the motor rotation speed detection value ⁇ m (rad / s) which is the mechanical angular velocity of the motor 4. It is obtained by multiplying the obtained motor rotation speed detection value ⁇ m by the unit conversion coefficient (60 / (2 ⁇ )) from rad / s to rpm.
  • the currents iu, iv, if, and if (A) flowing through the motor 4 are acquired from the current sensor 7.
  • the DC current value V dc (V) is detected by a voltage sensor (not shown) provided in the DC power supply line between the battery 1 and the inverter 3.
  • the DC voltage value V dc (V) may be detected by a signal transmitted from the battery controller (not shown).
  • step S202 the motor torque command value calculation process is executed.
  • the motor torque command value (motor torque command value () is obtained by referring to the accelerator opening-torque table shown in FIG. 3 based on the operation information such as the accelerator opening ⁇ and the vehicle speed V input in step S201.
  • Basic torque command value) T m * is set.
  • step S203 the vibration damping control calculation process is executed. Specifically, the controller 2 is based on the motor torque command value T m * set in step S202, without wasting the response of the drive shaft torque, and the driving force transmission system vibration (torsion vibration of the drive shaft 8 or the like). ) Is suppressed, the q-axis current command value I q2 * , the d-axis current command value i d1 * , and the f-axis current command value i f1 * are calculated. The details of the vibration damping control calculation process will be described later.
  • step S204 the current control calculation process is executed.
  • the d-axis current i d , the q-axis current i q, and the f-axis current i f are obtained by the q-axis current command value i q2 * , d-axis current command value i d1 *, and f-axis obtained in step S203.
  • Current control is performed to match each with the current command value I f1 * .
  • FIG. 4 is a diagram showing a configuration example of the motor control system 100, and is a control block diagram of the current control calculation processing unit 2a provided by the controller 2 as one functional unit.
  • the controller 2 uses the current control calculation processing unit 2a to execute the current control calculation processing according to step S204.
  • the current control calculation processing unit 2a includes a stator PWM conversion unit 401, a rotor PWM conversion unit 402, a look-ahead compensation unit 403, coordinate conversion units 404 and 410, a non-interference control unit 405, and a q-axis current control unit. It includes a 406, a d-axis current control unit 407, an f-axis current control unit 408, a voltage command value calculation unit 409, and an A / D conversion unit 411.
  • the stator PWM conversion unit 401 performs PWM_Duty to the stator switching element included in the inverter 3 based on the three-phase voltage command values v u * , v v * , and v w * output from the coordinate conversion unit 410 described later.
  • Drive signals (high voltage element drive signals) D uu * , D ul * , D vu * , D vl * , D wu * , D wl * are generated and output to the inverter 3.
  • the rotor PWM conversion unit 402 generates PWM_Duty drive signals D fu * and D fl * for the rotor switching element included in the inverter 3 based on the f-axis voltage command value v f * described later, and causes the inverter 3 to generate PWM_Duty drive signals D fu * and D fl *. Output.
  • the inverter 3 has an AC voltage v u , v v , for controlling the dq axis currents i d and i q flowing in the stator winding of the motor 4 based on the PWM_Duty drive signal generated by the stator PWM conversion unit 401.
  • V w is generated and supplied to the motor 4.
  • the inverter 3 generates an f-axis voltage v f for controlling the f-axis current if flowing in the rotor winding of the motor 4 based on the PWM_Duty drive signal generated by the rotor PWM conversion unit 402. It is supplied to the motor 4.
  • the current sensor 7 detects at least two phase currents, for example, u-phase current i u and v-phase current i v , among the three-phase AC currents supplied from the inverter 3 to the motor 4.
  • the detected two-phase currents i u and iv are converted into digital signals by the A / D (analog / digital) conversion unit 411 and input to the coordinate conversion unit 404.
  • the current sensor 7 detects the f axis current i f fed from the inverter 3 to the motor 4.
  • the detected f-axis current if is converted into a digital signal by the A / D conversion unit 411 and output to the f-axis current control unit 408.
  • the look-ahead compensation unit 403 inputs the electric angle ⁇ re and the electric angular velocity ⁇ re, and adds the product of the electric angular velocity ⁇ re and the dead time of the control system to the electric angle ⁇ re to perform the look-ahead compensation. Calculate the electrical angle ⁇ re '. After pre-reading compensation, the electric angle ⁇ re'is output to the coordinate conversion unit 410.
  • the coordinate conversion unit 404 converts the three-phase AC coordinate system (uvw axis) to the orthogonal two-axis DC coordinate system (dq axis). Specifically, the coordinate conversion unit 404 performs coordinate conversion processing from the u-phase current i u , the v-phase current i v , the w-phase current i w , and the electric angle ⁇ re using the following equation (1). By doing so, the d-axis current i d and the q-axis current i q are calculated.
  • the non-interference control unit 405 has an electric angular velocity ⁇ re , a d-axis current reference response id_ref , a q-axis current reference response i q_ref , and f output from the q-axis, d-axis, and f-axis current control units 406 , 407 , and 408 .
  • axis current nominal response i F_REF enter the differential value s ⁇ i d_ref the d-axis current nominal response, and a partial value s ⁇ i F_REF of f-axis current nominal response, d-axis, between the q-axis, and the f-axis
  • the non-interference voltages v d_dcpl , v q_dcpl , and v f_dcpl required to cancel the interference voltage are calculated using the voltage equation shown by the following equation (2).
  • the following equation (2) is a voltage equation of the winding field type synchronous motor 4 which is the controlled object of the present invention.
  • each parameter of the above equation (2) is as follows. Note that s in the equation is a Laplace operator. id : d-axis current i q : q-axis current i f : f-axis current v d : d-axis voltage v q : q-axis voltage v f : f-axis voltage L d : d-axis inductance L q : q-axis inductance L f : F-axis inductance M: Mutual inductance between stator / rotor L d ': d-axis dynamic inductance L q ': q-axis dynamic inductance L f ': f-axis dynamic inductance M': Controller / rotor Dynamic mutual inductance between R a : Controller winding resistance R f : Rotor winding resistance ⁇ re : Electric angular velocity
  • the voltage equation of the above equation (2) can be diagonalized as shown in the following equation (3).
  • the characteristics from the voltage to the current of the d-axis, the q-axis, and the f-axis have a first-order delay as shown in the following equations (4), (5), and (6), respectively. ..
  • the q-axis current control unit 406 desires the q-axis current i q , which is a measured value of the actual current (actual current), to the q-axis current command value (second q-axis current command value) i q2 * without steady deviation.
  • the first q-axis voltage command value v q_dsh for tracking with responsiveness is calculated and output to the voltage command value calculation unit 409. Details of the q-axis current control unit 406 will be described later with reference to FIG.
  • d-axis current control unit 407 the actual current (actual current) measurements at a d-axis current i d a d-axis current command value i d1 * in steady-state error without first for follow a desired response
  • the d-axis voltage command value v d_dsh is calculated and output to the voltage command value calculation unit 409. Details of the d-axis current control unit 407 will be described later with reference to FIG.
  • the f-axis current control section 408 the actual current (actual current) a measure of the f-axis current i f the f-axis current command value i f1 * the steady-state error without first for follow a desired response of
  • the f-axis voltage command value v f_dsh is calculated and output to the voltage command value calculation unit 409. Details of the f-axis current control unit 408 will be described later with reference to FIG. 7.
  • FIG. 5 is a diagram illustrating details of the q-axis current control unit 406 of the present embodiment.
  • the q-axis current control unit 406 includes a control block 501, gains 502 and 503, an integrator 504, a subtractor 505, and an adder 506.
  • Control block 501 q-axis current command value i q2 * transfer characteristics of the first order lag of the response delay to simulate the actual current i q for 1 / ( ⁇ q s + 1 ) is.
  • the control block 501 inputs the q-axis current command value i q2 * and outputs the q-axis current norm response i q_ref .
  • the transfer characteristic 1 / ( ⁇ q s + 1 ) in such a tau q is a q-axis current nominal response time constant.
  • the gain 502 is a proportional gain K pq and is represented by the following equation (7).
  • the gain 502 takes the deviation between the q-axis current command value i q2 * and the q-axis current i q as an input, and outputs the value obtained by multiplying the input value by the proportional gain K pq to the adder 506.
  • the gain 503 is a proportional gain K iq and is represented by the following equation (8).
  • the gain 503 takes the deviation between the q-axis current command value i q2 * and the q-axis current i q as an input, and outputs the value obtained by multiplying the input value by the proportional gain K iq to the integrator 504.
  • the output of the integrator 504 is input to the adder 506.
  • the adder 506 calculates the first q-axis voltage command value V q_dsh by adding the output of the gain 502 and the output of the integrator 504.
  • the q-axis current control unit 406 sets the gains of the gains 502 and 503 as shown in the above equations (7) and (8), whereby the q-axis current command value i q2 * to the q-axis current i
  • the transmission characteristics up to q can be matched with the normative response represented by the following equation (9).
  • FIG. 6 is a diagram illustrating details of the d-axis current control unit 407 of the present embodiment.
  • the d-axis current control unit 407 includes control blocks 601 and 602, gains 603 and 604, an integrator 605, a subtractor 606, and an adder 607.
  • the control block 601 has a first-order delay transmission characteristic (d-axis current transmission characteristic) 1 / ( ⁇ d s + 1) that simulates the response delay of the actual current (d-axis current id ) with respect to the d-axis current command value id1 * . ..
  • Control block 601 receives as input the d-axis current command value i d1 *, and outputs the d-axis current nominal response i d_ref.
  • the transfer characteristic 1 / ( ⁇ d s + 1 ) in such a tau d is the d-axis current nominal response time constant.
  • Control block 602 the transfer characteristic s / ( ⁇ d s + 1 ) for calculating a differential value of d-axis current nominal response i d_ref against d-axis current command value i d1 * is.
  • Control block 602 receives as input the d-axis current command value i d1 *, and outputs the d-axis current nominal response differential value s ⁇ i d_ref.
  • the gain 603 is a proportional gain K pd and is expressed by the following equation (10).
  • Gain 603 inputs the deviation between the d-axis current command value i d1 * and d-axis current i d, and outputs a value obtained by multiplying a proportional gain K pd input values to the adder 607.
  • the gain 604 is a proportional gain K id and is represented by the following equation (11).
  • Gain 604 inputs the deviation between the d-axis current command value i d1 * and d-axis current i d, and outputs a value obtained by multiplying a proportional gain K id to the input value to the integrator 605.
  • the output of the integrator 605 is input to the adder 607.
  • the adder 607 calculates the first d-axis voltage command value v d_dsh by adding the output of the gain 603 and the output of the integrator 605.
  • the d-axis current control unit 407 sets the gains of the gains 603 and 604 as shown in the above equations (10) and (11), whereby the d-axis current command value i d1 * to the d-axis current i
  • the transmission characteristics up to d can be matched with the normative response represented by the following equation (12).
  • FIG. 7 is a diagram illustrating details of the f-axis current control unit 408 of the present embodiment.
  • the f-axis current control unit 408 includes control blocks 701 and 702, gains 703 and 704, an integrator 705, a subtractor 706, and an adder 707.
  • the control block 701 has a first-order delay transmission characteristic 1 / ( ⁇ f s + 1) that simulates a response delay of the actual current if with respect to the f-axis current command value i f1 * .
  • the control block 701 receives the f-axis current command value if f1 * as an input, and outputs the f-axis current reference response if_ref .
  • the transfer characteristic 1 / ( ⁇ f s + 1 ) in such a tau f is the f-axis current nominal response time constant.
  • the control block 702 is a transmission characteristic s / ( ⁇ f s + 1) for calculating the differential value of the f-axis current normative response if_ref with respect to the f-axis current command value if f1 * .
  • the control block 702 receives the f-axis current command value if f1 * as an input, and outputs the f-axis current norm response differential value s ⁇ if_ref .
  • the gain 703 is a proportional gain K pf and is represented by the following equation (13).
  • Gain 703 inputs the deviation between the f-axis current command value i f1 * and f-axis current i f, and outputs a value obtained by multiplying a proportional gain K pf the input value to the adder 707.
  • the gain 704 is a proportional gain K if and is represented by the following equation (14).
  • Gain 704 inputs the deviation between the f-axis current command value i f1 * and f-axis current i f, and outputs a value obtained by multiplying a proportional gain K an if the input value to the integrator 705.
  • the output of the integrator 705 is input to the adder 707.
  • the adder 707 calculates the first f-axis voltage command value v f_dsh by adding the output of the gain 703 and the output of the integrator 705.
  • the f-axis current control unit 408 sets the gains of the gains 703 and 704 as in the above equations (13) and (14), whereby the f-axis current command value i f1 * to the f-axis current i
  • the transmission characteristics up to f can be matched with the normative response represented by the following equation (15).
  • the voltage command value calculation unit 409 has a first q-axis voltage command value v q_dsh and a first d, which are outputs of the q-axis current control unit 406, the d-axis current control unit 407, and the f-axis current control unit 408.
  • the shaft voltage command value v d_dsh and the first f-axis voltage command value v f_dsh are corrected by using the non-interference voltages v q_dcpl , v d_dcpl , and v f_dcpl which are the outputs of the non-interference control unit 405 (in this embodiment). to add.
  • the voltage command value calculation unit 409 outputs the second q-axis voltage command value v q * and the second d-axis voltage command value v d * obtained by the correction to the coordinate conversion unit 410.
  • the second f-axis voltage command value v f * is output to the rotor PWM converter 402.
  • the coordinate conversion unit 410 converts the orthogonal 2-axis DC coordinate system (dq-axis) rotating at the electric angular velocity ⁇ re into the 3-phase AC coordinate system (uvw phase). Specifically, the coordinate conversion unit 410 uses the input second d-axis voltage command value v d * , the second q-axis voltage command value v q * , and the pre-reading compensation post-electric angle ⁇ re '. By performing the coordinate conversion process using the following equation (16), the voltage command values v u * , v v * , and v w * of each uvw phase are calculated.
  • step S203 the details of the vibration damping control process executed in step S203 (see FIG. 2) will be described.
  • FIG. 8 is a control block diagram of the vibration damping control calculation unit 2b provided in front of the current control calculation processing unit 2a shown in FIG. 1 as a functional unit of the controller 2. That is, the output from the vibration damping control calculation unit 2b is input to the current control calculation processing unit 2a.
  • the controller 2 uses the vibration damping control calculation unit 2b to execute the vibration damping control process according to step S203.
  • the vibration damping control calculation unit 2b includes a first current command value calculator 801, a magnetic flux estimator 802, a first torque command value calculator 803, a vibration damping torque command value calculator 804, and a second q. It is configured to include an axial current command value calculator 805.
  • the first current command value calculator 801 inputs the motor torque command value T m * , the motor rotation speed (mechanical angular speed) ⁇ rm, and the DC voltage V dc, and the q-axis current command values i q1 * , d. Calculate the shaft current command value i d1 * and the f-axis current command value i f1 * .
  • the first current command value calculator 801 includes each of the q-axis current command value i q1 * , the d-axis current command value i d1 * , and the f-axis current command value if f1 * , and the motor torque command value (basic torque command).
  • T m * motor rotation speed (mechanical angular speed) ⁇ rm , and map data that defines the relationship with the DC voltage V dc are stored in advance, and each value is calculated by referring to the map data.
  • the calculated q-axis current command value i q1 * is output to the first torque command value calculator 803, and the d-axis current command value i d1 * and the f-axis current command value i f1 * are the magnetic flux estimator 802. Is output to.
  • FIG. 9 is a control block diagram of the magnetic flux estimator 802.
  • the magnetic flux estimator 802 includes a reluctance torque equivalent magnetic flux estimator 901, a field magnetic flux estimator 902, and an adder 903.
  • Reluctance torque equivalent flux estimator 901 inputs the d-axis current command value i d1 *, calculates the reluctance equivalent flux estimation value [phi] r ⁇ .
  • Field flux estimator 902 inputs the f-axis current command value i f1 *, calculates the magnetic field flux estimate .phi.f ⁇ .
  • the adder 903 calculates the magnetic flux estimated value ⁇ ⁇ by adding the reluctance equivalent magnetic flux estimated value ⁇ r ⁇ and the field magnetic flux estimated value ⁇ f ⁇ .
  • FIG. 10 is a control block diagram of the reluctance torque equivalent magnetic flux estimator 901.
  • the relaxation torque equivalent magnetic flux estimator 901 includes a phase lead compensator 1001 and a multiplier 1002.
  • the phase lead compensator 1001 has a transmission characteristic (d-axis current transmission characteristic (see control block 601)) in which the q-axis current response is phase-advance compensated with respect to the transmission characteristic of the first-order delay simulating the d-axis current response delay. ⁇ q s + 1) / ( ⁇ d s + 1). The phase lead compensator 1001 outputs a value obtained by performing phase lead compensation using the transfer characteristic ( ⁇ q s + 1) / ( ⁇ d s + 1) on the d-axis current command value id1 * to the multiplier 1002.
  • the multiplier 1002 calculates the output of the phase lead compensator 1001 multiplies the difference L d -L q and d-axis inductance L d and q-axis inductance L q, the reluctance torque equivalent flux estimation value [phi] r ⁇ To do.
  • the d-axis inductance L d and the q-axis inductance L q the values at arbitrary operating points (representative operating points) of the motor 4 may be used, or may be obtained by referring to the map data stored in advance.
  • the reluctance torque generated in the rotor by the dq axis currents i d and i q is expressed by the following equation (17). Therefore, it is possible to define the following formula (17) in the section (L d -L q) i d and reluctance torque equivalent flux.
  • p n is the logarithm of the motor 4.
  • the vibration suppression control calculation unit 2b uses the reluctance torque equivalent magnetic flux estimated value ⁇ r ⁇ in which the phase advance compensation of the q-axis current response is performed by the phase advance compensator 1001, so that the q-axis current in consideration of the q-axis current response delay is taken into consideration.
  • the command value (second q-axis current command value) i q2 * can be calculated.
  • FIG. 11 is a control block diagram of the field magnetic flux estimator 902.
  • the field magnetic flux estimator 902 includes a phase lead compensator 1101 and a multiplier 1102.
  • the phase lead compensator 1101 advances and compensates for the q-axis current response with respect to the transmission characteristic of the first-order delay simulating the f-axis current response delay (see control block 701) ( ⁇ q s + 1) / ( ⁇ f ). s + 1).
  • Phase lead compensator 1101 outputs a value obtained by performing phase-lead compensation using the f-axis current command value i f1 * the transfer characteristic ( ⁇ q s + 1) / ( ⁇ f s + 1) to the multiplier 1102.
  • the multiplier 1102 calculates the field magnetic flux estimated value ⁇ f ⁇ by multiplying the output of the phase lead compensator 1101 by the mutual inductance M f between the stator and the rotor.
  • the mutual inductance M f may be a value at an arbitrary operating point (representative operating point) of the motor 4, or may be obtained by referring to map data stored in advance.
  • the vibration suppression control calculation unit 2b uses the field magnetic flux estimated value ⁇ f ⁇ in which the phase advance compensator 1101 compensates for the advance of the q-axis current response, so that the q-axis current command value in consideration of the q-axis current response delay. i q2 * can be calculated.
  • the description will be continued by returning to FIG.
  • the first torque command value calculator 803 multiplies the q-axis current command value i q1 * , the estimated magnetic flux value ⁇ ⁇ , and the number of pole pairs p n of the motor 4 to obtain the first torque command value (vibration suppression). Pre-control torque command value) Calculate T m1 * . The calculated first torque command value T m1 * is output to the vibration damping torque command value calculator 804.
  • the vibration damping torque command value calculator 804 performs feed forward control, which is a filter process for removing the natural vibration frequency component of the drive shaft torque transmission system of the vehicle, and motor rotation with respect to the first torque command value T m1 * .
  • the vibration damping torque command value T mfin * is calculated by performing vibration damping control called feedback control based on the number ⁇ rm .
  • FIG. 12 is a control block diagram of the vibration damping torque command value calculator 804.
  • the vibration damping torque command value calculator 804 includes a feed forward compensator 1201, a control block 1202, a feedback compensator 1203, an adder 1204, and a subtractor 1205.
  • the control block 1202, the feedback compensator 1203, and the subtractor 1205 may be collectively referred to as the feedback controller 1207.
  • the feed forward compensator 1201 When the feed forward compensator 1201 receives the input of the first torque command value T m1 * , the feed forward compensator 1201 performs a filter process consisting of Gr (s) / Gp (s) on the first torque command value T m1 * . Calculate the second torque command value T m2 * .
  • Gr (s) and Gp (s) will be described with reference to FIG. 13 described later.
  • the control block 1202 performs a filter process consisting of the transmission characteristic Gp (s) on the vibration damping torque command value T mfin * output from the adder 1204, and calculates the motor angular velocity estimated value ⁇ m ⁇ .
  • the second torque command value T m2 * output from the feed forward compensator 1201 and the third torque command value T m3 * output from the feedback compensator 1203 are added.
  • the vibration damping torque command value T mfin * is calculated.
  • the feedback controller 1207 first, in the subtractor 1205, the actual motor angular velocity ⁇ m is subtracted from the motor angular velocity estimated value ⁇ m ⁇ . Then, the feedback compensator 1203 performs a filter process consisting of H (s) / Gp (s) on the difference between the motor angular velocity estimated value ⁇ m ⁇ output from the subtractor 1205 and the actual motor angular velocity ⁇ m. , Calculate the third torque command value T m3 * .
  • H (s) is configured such that the difference between the denominator order and the numerator order is larger than the difference between the denominator order and the numerator order of Gp (s).
  • the control system can be stabilized by H (s) / Gp (s).
  • FIG. 13 is a diagram in which the driving force transmission system of the vehicle is converted into a control block 601. Each parameter in the figure is as shown below.
  • J m Motor inertia
  • J w Drive wheel inertia (for one axis)
  • M Vehicle mass
  • K d Torsional rigidity of drive shaft (drive shaft)
  • K t Factor related to friction between tire and road surface
  • N al Overall gear ratio
  • r Tire load radius
  • ⁇ m Motor angular velocity
  • ⁇ w Drive wheel angular velocity
  • T m Motor torque
  • T d Drive shaft torque
  • V Body speed
  • Gp (s) can be expressed as the following equation (26).
  • ⁇ p and ⁇ p in Eq. (26) are the damping coefficient and the natural vibration frequency of the drive shaft torsional vibration system, respectively.
  • the ideal normative model Gr (s) indicating the response target of the motor rotation speed to the torque input to the vehicle is the following equation (27)
  • the inverse characteristic of the linearly approximated vehicle transfer characteristic Gp (s) and The transfer function Ginv (s) having the characteristics of Gr (s) / Gp (s), which is composed of the normative model Gr (s) can be expressed by the following equation (28).
  • ⁇ m and ⁇ m in the equations (27) and (28) are the damping coefficient and the natural vibration frequency of the drive shaft torsional vibration system, respectively.
  • H (s) is a feedback element that reduces only vibration when a bandpass filter is used. At this time, if the characteristics of the filter are set as shown in FIG. 14, the greatest effect can be obtained. That is, the transfer function H (s) has substantially the same damping characteristics on both the low-pass side with a low frequency and the high-pass side with a high frequency, and the torsional resonance frequency of the drive system is on the logarithmic axis (log scale). It is set to be near the center of the pass band.
  • the second q-axis current command value calculator 805 shown in FIG. 8 has a vibration damping torque command value T mfin * output from the vibration damping torque command value calculator 804 and a magnetic flux output from the magnetic flux estimator 802.
  • the q-axis current command value (second q-axis current command value) i q2 * is calculated using the following equation (31).
  • the calculated q-axis current command value i q2 * is input to the q-axis current control unit 406 of the current control calculation processing unit 2a shown in FIG.
  • the magnetic flux estimated value ⁇ ⁇ is the d-axis current command value i d1 * and the f-axis current command value i f1 set according to the motor torque command value T m * set based on the vehicle information.
  • the q-axis current command value i q2 * of the present embodiment is set according to the motor torque command value T m * in consideration of the d-axis current command value i d1 * and the f-axis current command value i f1 *. It is calculated by correcting the q-axis current command value i q1 * .
  • the vibration damping control calculation unit 2b considers the influence of the reluctance torque generated by the d-axis current command value i d1 * and the field magnetic flux generated by the f-axis current command value i f1 * , and drives the drive shaft. It is possible to suppress the occurrence of torsional vibration of the torque transmission system.
  • FIG. 15 is a time chart showing the control results of the present embodiment and the comparative example.
  • the horizontal axis represents time
  • the vertical axis represents the motor torque command value [Nm], the vehicle front-rear acceleration [m / s2], and the f-axis voltage [V] in order from top to bottom on the left side, and from the top on the right side.
  • the q-axis current command value [A] the d-axis current command value [A]
  • the f-axis current command value [A] are shown.
  • the solid line in the figure shows the present embodiment
  • the dotted line shows a comparative example.
  • the f-axis current norm response time constant ⁇ f is set to a value at which f-axis voltage saturation does not occur.
  • the motor torque command value is changed (started up) in steps at the timing of time t1 from the state where the vehicle is stopped to accelerate.
  • the f-current command value is increased stepwise as shown in the lower right part of the figure in order to change the rotor magnetic flux at time t1
  • the f-axis voltage is increased as shown in the lower left part of the figure. After rising, converges to a predetermined value.
  • the d current command value for controlling the magnetic field component falls.
  • the f-axis current is not taken into consideration when calculating the q-axis current command value, and the influence of the field magnetic flux generated by the f-axis current and the reluctance torque generated by the d-axis current is further affected. Not considered. Therefore, as shown in the upper right part of the figure, the q-axis current command value for controlling the torque component increases in a stepwise manner, and as shown in the middle left part of the figure, the vehicle front-rear acceleration vibrates.
  • the q-axis current i q2 * is calculated in consideration of the f-axis current i f1 * and the d-axis current i d1 * as shown in FIG. Therefore, the q-axis current is controlled so as to decrease after the rise at time t1 and then converge to a predetermined value.
  • the motor torque that suppresses the drive shaft torsional vibration is realized by the q-axis current command value i q2 * calculated in consideration of the d-axis current id and the f-axis current if, which is shown in the middle left of the figure. As shown, the vibration of the vehicle front-rear acceleration is suppressed.
  • the method for controlling an electric vehicle according to the first embodiment is in an electric vehicle using a winding field type synchronous motor 4 as a drive source, which includes a rotor having a rotor winding and a stator having a stator winding.
  • This is a control method for an electric vehicle that controls a stator current flowing through a stator winding and a rotor current flowing through a rotor winding.
  • the motor torque command value T m * is set based on the operation information
  • the d-axis current command for the stator current is based on the motor torque command value T m * and the motor angular speed ⁇ m which is the vehicle information.
  • the values i d1 *, the first q-axis current command value i q1 *, and the f-axis current command value i f1 * for the rotor current are calculated, and the d-axis current command value i d1 * and the f-axis current command value i are calculated.
  • the magnetic flux estimated value ⁇ ⁇ which is an estimated value of the magnetic flux generated in the rotor, is calculated based on f1 *, and the first torque command is calculated based on the first q-axis current command value i q1 * and the magnetic flux estimated value ⁇ ⁇ .
  • the torque command value T mfin * is calculated, and the second q-axis current command value i q2 * is calculated based on the magnetic flux estimated value ⁇ ⁇ and the vibration damping torque command value T mfin * .
  • stator current and the rotor current are controlled based on the second q-axis current command value i q2 * , the d-axis current command value i d1 *, and the f-axis current command value i f1 * .
  • the second q-axis current command value i q2 * can be calculated in consideration of the d-axis current command value i d1 * and the f-axis current command value i f1 *. Therefore, the d-axis current command value i d1 and the reluctance torque generated by the *, taking into account the influence of the field magnetic flux generated by the f-axis current command value i f1 *, the electric vehicle as a driving source for winding field synchronous motor 4 driving shaft torque transmission Vibration suppression control that suppresses the torsional vibration of the system can be applied.
  • the feed forward compensator 1201 is used for the first calculation.
  • the second torque command value T m2 * is calculated by performing feed forward control with respect to the torque command value T m1 * of.
  • the vibration damping torque command value T mfin * is calculated by adding the second torque command value T m2 * and the third torque command value T m3 * .
  • the vibration damping torque command value calculator 804 calculates the vibration damping torque command value T m2 * calculated by the feed forward control and the calculation by the feedback control.
  • the vibration damping torque command value T mfin * is calculated, so that the vibration that may occur in the vehicle can be suppressed accurately.
  • the transmission characteristic Gp (s) of the linear approximation model in the electric vehicle and the transmission characteristic Gp (s) of the linear approximation model with respect to the first torque command value T m1 * is calculated by performing Gr (s) / Gp (s) filtering based on the transmission characteristic Gr (s) of the normative model.
  • the control block 1202 filters the vibration damping torque command value T mfin * including the second torque command value T m2 * based on the transmission characteristic Gp (s) of the linear approximation model.
  • the standard motor angular velocity ⁇ m ⁇ is obtained.
  • the linear approximation model Gp (s) and the bandpass filter H for the difference between the reference motor angular velocity ⁇ m ⁇ calculated by the subtractor 1205 and the measured motor angular velocity ⁇ m
  • the third torque command value T m3 * is calculated.
  • the second torque command value T m2 * calculated by feedforward control and the third torque command value T m3 * calculated by feedback control are added to each other.
  • the vibration damping torque command value T mfin * is calculated. Therefore, even when a disturbance or a model error occurs, the feedforward control and the feedback control composed of a plurality of transfer functions are performed, so that the vibration of the drive shaft torque transmission system of the vehicle can be suppressed.
  • the second q-axis current command value i q2 * is calculated by dividing the vibration damping torque command value T mfin * by the magnetic flux estimated value ⁇ ⁇ . To. As a result, it is possible to calculate the q-axis current command value i q2 * that realizes the vibration control torque command value T mfin * with vibration control control.
  • to calculate the estimated value is the field flux estimate of the magnetic field flux of the rotor .phi.f ⁇ based on the f-axis current command value i f1 *, d based on the axis current value i d1 *, and calculates the equivalent flux estimation value of the reluctance torque generated in the rotor [phi] r ⁇
  • the magnetic flux estimation value phi ⁇ is the field flux estimate .phi.f ⁇ equivalent flux estimation value [phi] r ⁇ and Is calculated by adding.
  • damping control calculates the q-axis current command value i q2 * for realizing the damping torque command value T MFIN * subjected be able to.
  • the field magnetic flux estimated value ⁇ ⁇ simulates the response delay of the f-axis current if constituting the rotor current with respect to the f-axis current command value i f1 *. It is calculated using a transmission characteristic (phase advance compensator 1101) configured to compensate the q-axis current response with respect to the f-axis current transfer characteristic. Thus, it is possible to calculate the d-axis current i q-axis current command value q-axis current response delay is considered for d i q2 *.
  • the f-axis current transfer characteristic is a transfer function of the first-order lag.
  • the equivalent magnetic flux estimated value ⁇ r ⁇ simulates the response delay of the d-axis current id constituting the stator current to the d-axis current command value i d1 * . It is calculated using a transmission characteristic (phase advance compensator 1001) configured to compensate the q-axis current response with respect to the shaft current transmission characteristic. As a result, the q-axis current command value i q2 * can be calculated in consideration of the q-axis current response delay with respect to the f-axis current if .
  • the control method of the electric vehicle of the present embodiment is a control method applied on the premise that the f-axis current if is controlled in consideration of the f-axis voltage saturation, and is particularly provided by the vibration damping control calculation unit 2b.
  • the configuration related to feedforward and the configuration related to feedback of the magnetic flux estimator 802 are different from those of the first embodiment.
  • FIG. 16 is a diagram showing a configuration example of the motor control system 200 of the second embodiment.
  • the power supply voltage V dc of the battery 1 and the non-interference voltage v f_dcpl which is the output of the non-interference control unit 405 are input to the f-axis current control unit 408. 1 Different from the embodiment.
  • FIG. 17 is a control block diagram of the f-axis current control unit 408.
  • the first f is such that the f-axis current if input from the A / D conversion unit 411 follows the f-axis current command value if f1 * with a desired response without steady deviation.
  • the shaft voltage command value v f_dsh Calculate the shaft voltage command value v f_dsh .
  • the f-axis current control unit 408 calculates the f-axis current normative response if_ref and the differential values s ⁇ if_ref of the f-axis current normative response to be used in the subsequent processing.
  • the f-axis current control unit 408 is composed of an f-axis F / F (feed forward) compensator 1701, an f-axis F / B compensator 1702, an f-axis robust compensator 1703, and an f-axis limit processing unit 1704. , The details of each will be described below.
  • the f-axis F / F compensator 1701 takes the f-axis current command value i f1 * as an input, and in addition to the f-axis F / F compensation voltage v f_ff , the f-axis current normative response if_ref and its differential value. Calculate the differential value s ⁇ if_ref of the f-axis current normative response.
  • the f-axis F / F compensator 1701 outputs the f-axis current normative response if_ref and the differential value s ⁇ if_ref of the f-axis current normative response, which is the differential value thereof, to the non-interference control unit 405, and f.
  • the shaft current norm response if_ref is output to the f-axis F / B compensator 1702. Details of the f-axis F / F compensator 1701 will be described later with reference to FIG. Although not shown, the power supply voltage V dc output from the battery 1 and the non-interference voltage v f_dcpl output from the non-interference control unit 405 are input to the f-axis F / F compensator 1701. ..
  • the f-axis F / B compensator 1702 is a compensator that performs general feedback compensation.
  • f-axis F / B compensator 1702 with respect to the f-axis current nominal response i F_REF calculated in the f-axis F / F compensator 1701, is negatively fed back to the f-axis current i f that is measured by the current sensor 7 F
  • the f-axis F / B compensation voltage v f_fb is calculated so that the f-axis current if follows the f-axis current normative response if_ref .
  • the f-axis F / B compensator 1702 outputs the f-axis F / B compensation voltage v f_fb to the adder 1705. Details of the f-axis F / B compensator 1702 will be described later with reference to FIG.
  • the f-axis F / B compensator 1702 is an example of a block that executes the F / B compensation step.
  • the f-axis robust compensator 1703 includes a first f-axis voltage command value v f_dsh calculated by the f-axis limit processing unit 1704 described later and finally output from the f-axis current control unit 408, and an f-axis current if . Based on, the f-axis robust compensation voltage v f_rbst for ensuring the robustness of the system is calculated. The f-axis robust compensator 1703 outputs the f-axis robust compensation voltage v f_rbst to the adder 1706. Details of the f-axis robust compensator 1703 will be described later with reference to FIG. 24.
  • Two adders 1705 and 1706 are provided in front of the f-axis limit processing unit 1704.
  • the f-axis F / B compensation voltage v f_fb is added by the adder 1705 to the f-axis F / F compensation voltage v f_ff calculated by the f-axis F / F compensator 1701, and further, the f-axis by the adder 1706.
  • the robust compensation voltage v f_rbst is added. Then, the final addition value is input to the f-axis limit processing unit 1704.
  • the f-axis limiting processor 1704 with respect to the f-axis F / F compensation voltage v F_ff a F / F command value, F / B is a compensation value f-axis F / B compensation voltage v F_fb and, The sum of the f-axis robust compensation voltage v f_rbst, which is the f-axis robust compensation value, is input.
  • the f-axis limit processing unit 1704 limits the input voltage command value and calculates the first f-axis voltage command value v f_dsh .
  • the f-axis limit processing unit 1704 outputs the f-axis voltage command value v f_dsh to the voltage command value calculation unit 409 and the f-axis robust compensator 1703.
  • the f-axis limit processing unit 1704 performs the same processing as the f-axis limit processing unit 303 described later with reference to FIGS. 22 and 23.
  • FIG. 18 is a detailed block diagram of the f-axis F / F compensator 1701.
  • the f-axis F / F compensator 1701 has an f-axis current model 1801, an f-axis current pseudo F / B model 1802, and an f-axis limit processing unit 1803.
  • the f-axis current model 1801 is a filter that models the normative response characteristics from the f-axis voltage to the f-axis current.
  • the f-axis current model 1801 performs filtering processing on the f-axis F / F compensation voltage v f_ff output from the f-axis limit processing unit 1803 described later using a normative response model from voltage to current on the f-axis.
  • the f-axis current normative response if_ref which is the normative response, is calculated and output to the non-interference control unit 405 and the f-axis F / B compensator 202.
  • the f-axis current model 1801 outputs the differential value s ⁇ if_ref of the f-axis current normative response, which is the differential value of the f-axis current normative response if_ref, to the non-interference control unit 405 for use in the subsequent processing. To do. Details of the f-axis current model 1801 will be described later with reference to FIG.
  • the f-axis current pseudo F / B model 1802 In the f-axis current pseudo F / B model 1802, the f-axis current normative response i output from the f-axis current model 1801 to the f-axis current command value i f1 * calculated by the vibration damping control calculation unit 2b. f_ref is negatively fed back.
  • the f-axis current pseudo F / B model 1802 has a pseudo FB voltage command value v f_pse_fb in order to make the f-axis current normative response if_ref follow the f-axis current command value if f1 * with a desired response without steady deviation. Is calculated and output to the f-axis limit processing unit 1803. Details of the f-axis current pseudo F / B model 1802 will be described later with reference to FIG.
  • the f-axis limit processing unit 1803 limits the pseudo FB voltage command value v f_fb_psu output from the f-axis current pseudo F / B model 1802, calculates the f-axis F / F compensation voltage v f_ff, and addser . Output to 205 and f-axis current model 1801. Details of the f-axis limit processing unit 1803 will be described later with reference to FIGS. 21 and 22.
  • the power supply voltage V dc output from the battery 1 and the non-interference voltage v f_dcpl output from the non-interference control unit 405 are input to the f-axis limit processing unit 1803.
  • the f-axis F / F compensation voltage v f_ff output from the f-axis F / F compensator 1701 passes through the adder 1705, the adder 1706, and the f-axis limit processing unit 1704.
  • the first f-axis voltage command value v f_dsh is calculated.
  • the f-axis current model is not the F / B system in which the measured f-axis current if is negatively fed back to the f-axis current pseudo F / B model 1802.
  • a pseudo F / B system is constructed in which the f-axis current normative response if_ref calculated in 1801 is negatively fed back.
  • the f-axis voltage v f is generated by the battery 1, the upper limit of the f-axis voltage v f is limited by the power supply voltage V dc of the battery 1 and saturated. Therefore, in the f-axis F / F compensator 1701 shown in FIG. 18, an f-axis limit processing unit 1803 that models saturation at the power supply voltage V dc is provided to limit the first f-axis voltage command value v f_dsh . Then, the f-axis F / F compensation voltage v f_ff is calculated. By feeding back the f-axis F / F compensation voltage v f_ff in consideration of voltage saturation to the f-axis current pseudo F / B model 1802, the accuracy of rotation control can be improved.
  • FIG. 197 is a detailed block diagram of the f-axis current model 1801.
  • the f-axis current model 1801 has a multiplier 1901, a subtractor 1902, a divider 1903, and an integrator 1904.
  • the multiplier 1901 is one of the final outputs of the f-axis current model 1801, and the rotor winding resistance R f is multiplied by the f-axis current normative response if_ref output from the integrator 1904 described later. , The multiplication result is output to the subtractor 1902. The result of this multiplication corresponds to the voltage value of the normative response.
  • the subtractor 1902 subtracts the voltage value of the normative response output from the multiplier 1901 from the f-axis F / F compensation voltage v f_ff output from the f-axis limit processing unit 1803, and outputs the subtracted value to the divider 1903. To do.
  • the divider 1903 divides the difference calculated by the subtractor 1902 with the f-axis dynamic inductance L f ' , and outputs the division result to the non-interference control unit 405 and the integrator 1904. In this way, the differential value s ⁇ if_ref of the f-axis current normative response is calculated.
  • Integrator 1904 the differential value s ⁇ i F_REF of f-axis current norms response outputted from the divider 1903 integration processing to calculate the f-axis current nominal response i F_REF, non-interference of the f-axis current nominal response i F_REF It outputs to the control unit 405, the f-axis F / B compensator 1702, and the multiplier 1901.
  • the f-axis current normative response if_ref which is one of the final outputs, is multiplied by the rotor winding resistance R f by the multiplier 1901, and the f-axis is the input. Negative feedback is given to the F / F compensation voltage v f_ff .
  • the negative feedback divider 1903 the result value of is divided by the f-axis dynamic inductance L f ', f axis current nominal response i F_REF based on the f-axis F / F compensation voltage v F_ff, and its differential value s ⁇ If_ref can be obtained.
  • FIG. 20 is a detailed block diagram of the f-axis current pseudo F / B model 1802.
  • the f-axis current pseudo-F / B model 1802 has a filter 2001, a filter 2002, and a subtractor 2003.
  • the filter 2002 multiplies the f-axis current norm response if_ref output from the f-axis current model 301 by the gain G bf , and outputs the filtered value to the subtractor 2003.
  • the subtractor 2003 calculates the pseudo F / B voltage command value v f_fb_psu by subtracting the output value of the filter 2002 from the output value of the filter 2001, and sends the pseudo FB voltage command value v f_fb_psu to the f-axis limit processing unit 1803. Is output. That is, a pseudo F / B control is configured by negatively feeding back the f-axis current norm response if_ref, which is not a measured value.
  • ⁇ f is an f-axis current control norm response time constant (f-axis current norm response time constant).
  • FIG. 21 is a detailed block diagram of the f-axis limit processing unit 1803.
  • the f-axis limit processing unit 1803 includes a comparator 2101, a reversing device 2102, a comparator 2103, and a subtractor 2104, 2105.
  • the subtractor 2104 provided in front of the comparator 2101, a subtraction value obtained by subtracting the f-axis non-interference voltage v f_dcpl output from the non-interference control unit 405 from the power supply voltage V dc of the battery 1 is obtained. Then, the comparator 2101 compares the pseudo FB voltage command value v f_pse_fb , which is the output value from the f-axis current pseudo F / B model 1802, with the subtracted value in the subtractor 2104, and transfers a smaller value to the comparator 2103. Is output.
  • the inverting device 2102 inverts the sign of the power supply voltage V dc .
  • a subtractor 2105 is provided in front of the comparator 2103.
  • the subtractor 2105 a subtraction value obtained by subtracting the f-axis non-interference voltage v f_dcpl output from the non-interference control unit 405 from the output of the reversing device 2102. Is required.
  • the comparator 2103 compares the output value of the comparator 2101 with the subtracted value of the subtractor 2105, and outputs a larger value to the f-axis current model 1801 and the adder 1705.
  • the f-axis limit processing unit 1803 adds the f-axis non-interference voltage v f_dcpl to the pseudo FB voltage command value v f_pse_fb , which is the output value of the f-axis current pseudo F / B model 1802.
  • Limit processing based on the power supply voltage V dc offset to the minus by the f-axis non-interference voltage v f_dcpl , specifically, the upper limit is "V dc -v f_dcpl " and the lower limit is " -V dc -v f_dcpl "is performed.
  • the f-axis limit processing unit 1803 may be configured as shown in FIG.
  • FIG. 22 is another example of a detailed block diagram of the f-axis limit processing unit 1803.
  • the f-axis limit processing unit 1803 has a comparator 2201, a reversing device 2202, a comparator 2203, a subtractor 2204, and an adder 2205.
  • An adder 2205 is provided in front of the comparer 2201.
  • the adder 2205 the f-axis non-interference voltage v f_dcpl output from the non-interference control unit 405 and the f-axis current pseudo F / B model 1802 output.
  • the pseudo FB voltage command value vf_pse_fb to be added is added.
  • the comparator 2201 compares the power supply voltage V dc of the battery 1 with the addition result in the adder 2205, and outputs a smaller value to the comparator 2203.
  • the inverting device 2202 inverts the sign of the power supply voltage V dc .
  • the comparator 2203 compares the output from the comparator 2201 with the output from the inverting device 2202 and outputs a large value to the subtractor 2204.
  • the subtractor 2204 calculates the f-axis F / F compensation voltage v f_ff by subtracting the f-axis non-interference voltage v f_dcpl output from the non-interference control unit 405 from the output value of the comparator 2203.
  • the subtractor 2204 outputs the f-axis F / F compensation voltage v f_ff to the f-axis current model 1801 and the adder 1706 constituting the f-axis current control unit 408.
  • the f-axis non-interference voltage v f_dcpl is added to the pseudo FB voltage command value v f_pse_fb which is the output value of the f-axis current pseudo F / B model 1802.
  • Limit processing based on the power supply voltage V dc offset negatively by the f-axis non-interference voltage v f_dcpl in order to obtain a margin specifically, the upper limit is "V dc -v f_dcpl " and the lower limit is "-”. Restriction processing such as "V dc -v f_dcpl " is performed.
  • FIG. 23 is a detailed block diagram of the f-axis F / B compensator 1702.
  • the f-axis F / B compensator 1702 has a block 2301, a multiplier 2302, and a subtractor 2303.
  • the block 2301 is a delay filter, and performs delay processing for the dead time L of the control system.
  • the block 2301 delays the f-axis current normative response if_ref with respect to the input of the f-axis current normative response if_ref output from the f-axis F / F compensator 1701, and the f-axis current normative response if_ref and the f-axis current.
  • calculating a dead time after treatment f axis current nominal response i F_REF 'in order to match the phase of the i f and outputs to the subtractor 2303 provided upstream of the multiplier 2302.
  • Block 2301 is an example of a block that executes a delay step.
  • a dead time after treatment f axis current nominal response i F_REF 'output from block 2301 subtracts the f axis current i f which is output from the A / D conversion unit 411 calculates the subtraction result.
  • the multiplier 2302 is input with the subtraction result in the subtracter 2303 calculates the f-axis F / B compensation voltage v F_fb by multiplying the f-axis F / B gain K f, the f-axis F / B compensation voltage v F_fb Is output to the adder 1705.
  • the value of the f-axis F / B gain K f is determined by adjusting it experimentally so that the stability such as the gain margin and the phase margin satisfies a predetermined standard.
  • the f-axis F / B compensator 1702 calculates the f-axis F / B compensation voltage v f_fb based on the f-axis current if .
  • FIG. 24 is a detailed block diagram of the f-axis robust compensator 1703.
  • the f-axis robust compensator 1703 is composed of a block 2401, a block 2402, a block 2403, and a subtractor 2404.
  • Block 2401 calculates the first f-axis voltage estimated value v F_est1 filtering process on the input of the f-axis current i f which is output from the A / D conversion unit 411, the f-axis voltage estimated value v F_est1 Output to the subtractor 2404.
  • the block 2401 is a delay filter having the characteristics of (L f ' ⁇ s + R f ) / ( ⁇ h_f ⁇ s + 1) including the low-pass filter 1 / ( ⁇ h_f ⁇ s + 1) of the block 2403 described later.
  • Block 2402 is the same delay filter as block 1801.
  • the block 2402 delays the first f-axis voltage command value v f_dsh output from the f-axis limit processing unit 204 by the dead time L of the control system, and delays the second f-axis voltage estimated value v f_est2. Is calculated. Then, the block 2402 outputs the second f-axis voltage estimated value v f_est2 to the block 2403.
  • Block 2403 is a low-pass filter having a characteristic of 1 / ( ⁇ h_f ⁇ s + 1).
  • the block 2403 performs a low-pass filter process on the second f-axis voltage estimated value v f_est2 output from the block 2402, and calculates the third f-axis voltage estimated value v f_est3 . Then, the block 2403 outputs the third f-axis voltage estimated value v f_est3 to the subtractor 2404.
  • the subtractor 2404 calculates the f-axis robust compensation voltage v f_rbst into the adder 1706 by subtracting the first f-axis voltage estimate v f_est1 from the third f-axis voltage estimate v f_est3 .
  • the first f-axis voltage command value v f_dsh is processed to process the delay filter block 2401 and the low-pass filter block 2403, and the first f-axis based on the measured value is processed.
  • the voltage estimate v f_est1 the f-axis low- pass compensation voltage v f_rbst for further improving stability is calculated.
  • FIG. 25 is a flowchart showing the control process of the motor 4 described with reference to FIGS. 16 to 24 described above. These controls are performed by the controller 2 executing a predetermined program.
  • step S1 the A / D conversion unit 411 acquires the current values (u-phase current i us , v-phase current i vs , and f-axis current if ) and the electric angle ⁇ re of the motor 4.
  • step S2 the motor rotation speed ⁇ rm , which is the mechanical angular velocity, and the electric angular velocity ⁇ re are calculated based on the electric angle ⁇ re acquired in step S1.
  • step S3 the prefetch compensator 403 based on the electric angle theta re calculated at step S2, and calculates the post-prefetch compensation electrical angle theta re '.
  • step S4 the coordinate conversion unit 404 calculates the d-axis current i d and the q-axis current i q based on the u-phase current i u and v-phase current i v calculated in step S1.
  • step S5 the d-axis current command value id * , the q-axis current command value i q * , and the f-axis current command are based on the motor rotation speed ⁇ rm , the torque command value T * , and the power supply voltage V dc. The value if * is calculated.
  • step S6 the q-axis current control unit 406, the d-axis current control unit 407, and the f-axis current control unit 408 perform the first d-axis voltage command value v d_dsh , the d-axis current normative response id_ref , and the d-axis current.
  • step S7 the non-interference control section 405, and the electrical angular velocity omega re calculated in step S2, the d-axis current nominal response i d_ref, the differential value s ⁇ i d_ref the d-axis current nominal response calculated in step S6,
  • the non-interference voltages v d_dcpl , v q_dcpl , and v f_dcpl are calculated according to the q-axis current normative response i q_ref , the f-axis current normative response if_ref , and the differential values s ⁇ if_ref of the f-axis current normative response.
  • step S8 the voltage command value calculation unit 409 uses the first d-axis voltage command value v d_dsh , the first q-axis voltage command value v q_dsh , and the first f-axis voltage command calculated in step S6. for each value v F_dsh, step S7 incoherent voltage v D_dcpl calculated, v Q_dcpl, and, v F_dcpl by adding the second d-axis voltage command value v d *, a second q-axis The voltage command value v q * and the second f-axis voltage command value v f * are calculated.
  • step S9 the coordinate conversion unit 410 sets the second d-axis voltage command value v d * , the second q-axis voltage command value v q * , and the second f-axis voltage command calculated in step S8.
  • the voltage command values v u * , v v * , and v w * of each phase of uvw are calculated.
  • the controller 2 executes the processes of steps S1 to S9 to generate a command value for controlling the motor 4.
  • the voltage command values v u * , v v * , and v w * calculated in step S9 are fixed to the motor 4 via the stator PWM converter 401 and the inverter 3. It is applied to the winding on the child side.
  • the second f-axis voltage command value v f * calculated in step S8 is applied to the winding on the rotor side of the motor 4 via the rotor PWM conversion unit 402. In this way, the rotation control of the motor 4 is performed.
  • a motor control method for controlling the f-axis current i f in consideration of the f-axis voltage saturation When vibration damping control processing is applied to such motor control, the reluctance torque equivalent magnetic flux estimator 901 and the field reluctance estimator 902 constituting the magnetic flux estimator 802 included in the vibration damping control calculation unit 2b are f. It is necessary to perform processing in consideration of shaft voltage saturation.
  • FIG. 26 shows a control block diagram of the reluctance torque equivalent magnetic flux estimator 901 in the magnetic flux estimator 802 of the present embodiment.
  • the relaxation torque equivalent magnetic flux estimator 901 of the present embodiment is composed of a multiplier 2601.
  • the multiplier 2601 has a difference L between the d-axis inductance L d and the q-axis inductance L q with respect to the d-axis current command value id1 * output from the first current command value calculator 801 of the second embodiment. Multiply d ⁇ L q to calculate the reluctance torque equivalent magnetic flux estimated value ⁇ r ⁇ .
  • the values at arbitrary operating points (representative operating points) of the motor 4 may be used, or may be obtained by referring to the map data stored in advance.
  • the configuration of the reluctance torque equivalent magnetic flux estimator 901 shown in the present embodiment can be simplified as compared with the configuration of FIG. 10 shown in the first embodiment.
  • FIG. 27 is a control block diagram of the field magnetic flux estimator 902 of the second embodiment.
  • the field magnetic flux estimator 902 of the present embodiment includes control blocks 2701 and 2704, a multiplier 2702, a control block 2703, a limiter 2705, and an adder 2706.
  • the control block 2701 is an f-axis model that models the transmission characteristics from the f-axis voltage v f to the f-axis current if .
  • the f-axis model has a characteristic of ( ⁇ q s + 1) / (L f s + R f ).
  • Control block 2701 enter the f axis current nominal response v Fc_lim considering f-axis voltage saturation characteristic outputted from the limiter 2705, in consideration of the transfer characteristic from the f-axis voltage v f to f axis current i f f
  • the shaft current reference response if_ref is calculated and output to the multiplier 2702 and the control block 2704.
  • the multiplier 2702 calculates the field magnetic flux estimated value ⁇ f ⁇ by multiplying the f-axis current norm response if_ref by the mutual inductance M f between the stator and the rotor.
  • the mutual inductance Mf may be a value at an arbitrary operating point (representative operating point) of the motor 4, or may be obtained by referring to map data stored in advance.
  • the control block 2703 is composed of a gain G af .
  • the gain G af is represented by the above equation (32).
  • Control block 2703 outputs obtained by multiplying the gain G af the f-axis current command value i f1 * input values to the adder 2706.
  • the control block 2704 is a filter composed of a gain G bf and 1 / ( ⁇ qs +1).
  • the gain G bf is shown in the above equation (32).
  • the control block 2704 outputs the value obtained by filtering the f-axis current norm response if_ref to the adder 2706.
  • the adder 2706 calculates the f-axis voltage command value v fc by adding the output values of the control blocks 2703 and 2704.
  • the calculated f-axis voltage command value v fc is output to the f-axis limiter 2705.
  • the field magnetic flux estimator 902 of the present embodiment includes the control block 2703 and the control block 2704, and the gain G af is multiplied by the f-axis current command value i f1 * and the f-axis current normative response i.
  • the current F / B system (f-axis current F / B model) is constructed by multiplying f_ref by the gain G bf .
  • the f-axis limiter 2705 simulates the f-axis voltage saturation characteristic by limiting the f-axis current command value v fc according to the power supply voltage V dc .
  • the field magnetic flux estimator 902 can calculate the f-axis current norm response if_ref in consideration of the f-axis voltage saturation characteristic in the control block 2701 arranged at the subsequent stage .
  • the q-axis current is provided by the phase lead compensation ( ⁇ qs +1) of the q-axis current response on the control blocks 2701 and 2704 included in the magnetic flux estimator 802.
  • the q-axis current command value i q2 * can be calculated in consideration of the response delay. That is, the field magnetic flux estimated value ⁇ f ⁇ of the present embodiment is an f-axis model that models the characteristics from the f-axis voltage vf to the f-axis current if that constitutes the rotor current, and the f-axis current command value if f1 *.
  • a pseudo F / B system composed of an f-axis current F / B model in which and the output of the f-axis model are input, and an f-axis limiter 2705 that limits the output of the f-axis current F / B model.
  • the q-axis current response is phase-advanced and compensated for the f-axis model and the f-axis current F / B model.
  • the vibration damping control calculation unit 2b can appropriately simulate the f-axis current response when there is f-axis voltage saturation.
  • FIG. 28 is a control block diagram of the vibration damping control calculation unit 2b of the second embodiment.
  • the vibration damping control calculation process is composed of a feedforward compensator 2801 and a feedback compensator 2802.
  • the feed forward compensator 2801 subtracts the value obtained by integrating the F / B gain from the torque command value and the pseudo torsion angular velocity with the vehicle model 2804 composed of the vehicle parameter and the dead zone model simulating the gear back crash. It is composed of a drive shaft torsional velocity F / B model 2805, and calculates a second torque command value T m2 * and a first motor angular velocity estimated value ⁇ m1 ⁇ .
  • a vehicle model composed of a dead zone model simulating a gear back crash and modeling a torque transmission system is used, and the first motor angular velocity estimated value ⁇ m1 ⁇ is set according to the motor torque T m .
  • the pseudo drive shaft torsional velocity ⁇ d ⁇ is obtained.
  • the pseudo drive shaft torsional velocity ⁇ d ⁇ is obtained by simulating the torsional angular velocity generated in the drive shaft 8 to which the torque of the motor 4 is transmitted.
  • the drive shaft torsion angle speed F / B model 2805 feedback control is performed in which the pseudo drive shaft torsion angle velocity ⁇ d ⁇ multiplied by the gain is subtracted from the first torque command value T m1 * before the vibration suppression control. , The second torque command value T m2 * is obtained.
  • the torsional angular velocity FB model is used, and this torsional angular velocity FB model is obtained by using a part of the vehicle model.
  • the feedback compensator 2802 In the feedback compensator 2802, the first motor angular velocity estimated value ⁇ m1 ⁇ and the motor angular velocity ⁇ m are input. Then, the feedback compensator 2802 outputs a third torque command value T m3 * from these inputs.
  • the feedback compensator 2802 uses the transmission characteristic Gp (s) of the vehicle model shown in the block 2806 with respect to the third torque command value T m3 * output from itself in order to perform feedback control.
  • the motor angular velocity estimated value (second motor angular velocity estimated value) ⁇ m2 ⁇ corresponding to the second torque command value T m2 * is calculated.
  • the adder 2807 the first motor angular velocity estimated value ⁇ m1 ⁇ and the second motor angular velocity estimated value ⁇ m2 ⁇ are added to calculate the third motor angular velocity estimated value ⁇ m3 ⁇ .
  • the third torque command value T m3 * is calculated by processing the filter H (s) / Gp (s) shown in the block 2809 with respect to this deviation.
  • the filter H (s) / Gp (s) is composed of an inverse characteristic of the transmission characteristic Gp (s) of the vehicle model and a bandpass filter H (s).
  • the second torque command value T m2 * output from the feedforward compensator 2801 and the third torque command value T m3 * output from the feedback compensator 2802 are added. Then, the vibration damping torque command value T mfin * is calculated.
  • FIG. 29 is a time chart showing the control results of the present embodiment and the comparative example.
  • the horizontal axis represents time
  • the vertical axis represents the motor torque command value [Nm], the vehicle front-rear acceleration [m / s2], and the f-axis voltage [V] in order from top to bottom on the left side, and from the top on the right side.
  • the q-axis current command value [A] the d-axis current command value [A]
  • the f-axis current command value [A] are shown.
  • the solid line in the figure shows the present embodiment, and the dotted line shows a comparative example.
  • the f-axis current norm response time constant ⁇ f is set to a value at which f-axis voltage saturation occurs.
  • FIG. 29 shows a scene in which the motor torque command value is changed (started up) in steps at the timing of time t1 while the vehicle is decelerating due to the regenerative torque of the motor 4. ..
  • the motor torque command value (vibration damping torque command value) that suppresses the drive shaft torsional vibration is calculated in consideration of the influence of the backlash of the gear. are doing.
  • the motor torque command value is changed (started up) in steps at the timing of time t1 to accelerate.
  • the f current command value is increased stepwise as shown in the lower right part of the figure, the f-axis voltage rises as shown in the lower left part of the figure.
  • the f-axis current normative response time constant ⁇ f is set to a value at which f-axis voltage saturation occurs. Therefore, the f-axis voltage is saturated as shown in the lower left part of the figure. Then, as shown in the middle right of the figure, the d current command value changes stepwise from a positive value to a negative value.
  • the f-axis current is not taken into consideration when calculating the q-axis current command value, and the influence of the field magnetic flux generated by the f-axis current and the reluctance torque generated by the d-axis current is further affected. Not considered. Therefore, as shown in the upper right part of the figure, the q-axis current command value also increases stepwise. Then, as shown in the middle left of the figure, the vehicle front-rear acceleration vibrates after increasing due to the influence of gear back crash.
  • the q-axis current i q2 * is calculated in consideration of the f-axis current i f1 * and the d-axis current i d1 * as shown in FIG. .. Therefore, the q-axis current is controlled so as to decrease after the rise at time t1 and then converge to a predetermined value.
  • the motor torque that suppresses the drive shaft torsional vibration is realized by the q-axis current command value i q2 * calculated in consideration of the d-axis current id and the f-axis current if, which is shown in the middle left of the figure. As you can see, the vehicle front-rear acceleration vibration is suppressed.
  • the feed forward compensator 2801 is configured by a dead zone model simulating a gear back crash with respect to the output second torque command value T m2 * , and torque is transmitted.
  • the vehicle model 2804 that models the system, the first motor angular velocity estimated value ⁇ m1 ⁇ and the pseudo drive shaft torsional velocity ⁇ d ⁇ can be obtained according to the motor torque T m .
  • the drive shaft torsional velocity F / B model 2805 feedback control is performed in which the pseudo drive shaft torsional velocity ⁇ d ⁇ multiplied by the gain is subtracted from the first torque command value T m1 * before the vibration damping control. Therefore, the second torque command value T m2 * is obtained.
  • the output third torque command value T m3 * is filtered by Gp (s) based on the linear approximation model in the electric vehicle.
  • the adder 2807 the first motor angular velocity estimated value ⁇ m1 ⁇ calculated in the feed forward control and the second motor angular velocity estimated value ⁇ m2 ⁇ are added to obtain a third motor angular velocity estimated value. Calculate ⁇ m3 ⁇ .
  • the difference between the third motor angular velocity estimated value ⁇ m3 ⁇ and the measured motor angular velocity ⁇ m is filtered based on a linear approximation model and a bandpass filter to perform a third filter process. Calculate the torque command value T m3 * .
  • vibration damping torque command value T mfin * is calculated by adding the second torque command value T m2 * and the third torque command value T m3 * .
  • the second torque command value T m2 * calculated by feedforward control using the dead zone model and the third torque command value T m3 * calculated by feedback control are added.
  • the vibration damping torque command value T mfin * is calculated. Therefore, in the case of re-acceleration during deceleration due to regenerative torque, vibration in the drive shaft torque transmission system of the vehicle can be suppressed even if gear backlash or the like occurs.
  • the vibration control calculation unit 2b provides phase advance compensation ( ⁇ qs +1) for the q-axis current response to the control blocks 2701 and 2704 provided in the magnetic flux estimator 902.
  • the q-axis current command value i q2 * can be calculated in consideration of the q-axis current response delay.
  • the magnetic field flux estimate ⁇ f of the present embodiment ⁇ includes a f-axis model 2701 models the characteristics of to f axis current i f that constitute the rotor current from the f-axis voltage vf, f-axis current command value i
  • a pseudo F composed of an f-axis current F / B model 2704 in which f1 * and the output of the f-axis model are input, and an f-axis limiter 2705 that limits the output of the f-axis current F / B model.
  • the vibration damping control calculation unit 2b can appropriately simulate the f-axis current response when the f-axis voltage is saturated.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Transportation (AREA)
  • Mechanical Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

Dans ce procédé de commande pour un véhicule électrique : une valeur de commande de couple de base est établie sur la base d'un état de fonctionnement ; une valeur de commande de courant d'axe d et une première valeur de commande de courant d'axe q par rapport au courant de stator, et une valeur de commande de courant d'axe f par rapport au courant de rotor sont calculées sur la base de la valeur de commande de couple de base et des informations de véhicule ; une valeur estimée de flux magnétique, qui est une valeur estimée de flux magnétique généré dans le rotor, est calculée sur la base de la valeur de commande de courant d'axe d et de la valeur de commande de courant d'axe f ; une première valeur de commande de couple est calculée sur la base de la première valeur de commande de courant d'axe q et de la valeur estimée de flux magnétique ; le calcul est effectué pour commander la vibration du système de transmission de couple d'arbre d'entraînement du véhicule électrique, sur la base des informations de véhicule, par rapport à la première valeur de commande de couple, une valeur de commande de couple d'amortissement de vibration étant par là-même calculée ; et une seconde valeur de commande de courant d'axe q est calculée sur la base de la valeur estimée de flux magnétique et de la valeur de commande de couple d'amortissement de vibration. En outre, le courant de stator et le courant de rotor sont commandés sur la base de la seconde valeur de commande de courant d'axe q, de la valeur de commande de courant d'axe d et de la valeur de commande de courant d'axe f.
PCT/JP2019/013468 2019-03-27 2019-03-27 Procédé de commande et dispositif de commande pour véhicule électrique WO2020194637A1 (fr)

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Publication number Priority date Publication date Assignee Title
CN112977094A (zh) * 2021-04-26 2021-06-18 比亚迪股份有限公司 电驱动系统控制方法、电驱动系统及车辆
CN116039398A (zh) * 2023-03-08 2023-05-02 阿维塔科技(重庆)有限公司 一种电动汽车充电控制方法、装置及计算机可读存储介质

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WO2014104164A1 (fr) * 2012-12-28 2014-07-03 日産自動車株式会社 Dispositif de commande de moteur et procédé de commande de moteur
WO2014103586A1 (fr) * 2012-12-28 2014-07-03 日産自動車株式会社 Dispositif de commande de moteur et procédé de commande de moteur
JP2015035885A (ja) * 2013-08-08 2015-02-19 日産自動車株式会社 モータ制御装置

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Publication number Priority date Publication date Assignee Title
WO2014104164A1 (fr) * 2012-12-28 2014-07-03 日産自動車株式会社 Dispositif de commande de moteur et procédé de commande de moteur
WO2014103586A1 (fr) * 2012-12-28 2014-07-03 日産自動車株式会社 Dispositif de commande de moteur et procédé de commande de moteur
JP2015035885A (ja) * 2013-08-08 2015-02-19 日産自動車株式会社 モータ制御装置

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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN112977094A (zh) * 2021-04-26 2021-06-18 比亚迪股份有限公司 电驱动系统控制方法、电驱动系统及车辆
CN116039398A (zh) * 2023-03-08 2023-05-02 阿维塔科技(重庆)有限公司 一种电动汽车充电控制方法、装置及计算机可读存储介质
CN116039398B (zh) * 2023-03-08 2023-09-26 阿维塔科技(重庆)有限公司 一种电动汽车充电控制方法、装置及计算机可读存储介质

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