WO2017033304A1 - 半導体スイッチング素子の制御回路および半導体装置 - Google Patents

半導体スイッチング素子の制御回路および半導体装置 Download PDF

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Publication number
WO2017033304A1
WO2017033304A1 PCT/JP2015/073981 JP2015073981W WO2017033304A1 WO 2017033304 A1 WO2017033304 A1 WO 2017033304A1 JP 2015073981 W JP2015073981 W JP 2015073981W WO 2017033304 A1 WO2017033304 A1 WO 2017033304A1
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Prior art keywords
current
circuit
switching element
temperature
semiconductor switching
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PCT/JP2015/073981
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English (en)
French (fr)
Japanese (ja)
Inventor
山本　剛司
河本　厚信
伸介 神戸
Original Assignee
三菱電機株式会社
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Application filed by 三菱電機株式会社 filed Critical 三菱電機株式会社
Priority to CN201580082680.9A priority Critical patent/CN108026888B/zh
Priority to DE112015006836.8T priority patent/DE112015006836T5/de
Priority to JP2017536129A priority patent/JP6465213B2/ja
Priority to PCT/JP2015/073981 priority patent/WO2017033304A1/ja
Priority to US15/561,077 priority patent/US10128735B2/en
Publication of WO2017033304A1 publication Critical patent/WO2017033304A1/ja

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F02—COMBUSTION ENGINES; HOT-GAS OR COMBUSTION-PRODUCT ENGINE PLANTS
    • F02P—IGNITION, OTHER THAN COMPRESSION IGNITION, FOR INTERNAL-COMBUSTION ENGINES; TESTING OF IGNITION TIMING IN COMPRESSION-IGNITION ENGINES
    • F02P3/00—Other installations
    • F02P3/02—Other installations having inductive energy storage, e.g. arrangements of induction coils
    • F02P3/04—Layout of circuits
    • F02P3/0407—Opening or closing the primary coil circuit with electronic switching means
    • F02P3/0435—Opening or closing the primary coil circuit with electronic switching means with semiconductor devices
    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F02—COMBUSTION ENGINES; HOT-GAS OR COMBUSTION-PRODUCT ENGINE PLANTS
    • F02P—IGNITION, OTHER THAN COMPRESSION IGNITION, FOR INTERNAL-COMBUSTION ENGINES; TESTING OF IGNITION TIMING IN COMPRESSION-IGNITION ENGINES
    • F02P3/00—Other installations
    • F02P3/02—Other installations having inductive energy storage, e.g. arrangements of induction coils
    • F02P3/04—Layout of circuits
    • F02P3/05—Layout of circuits for control of the magnitude of the current in the ignition coil
    • F02P3/051—Opening or closing the primary coil circuit with semiconductor devices
    • F02P3/053—Opening or closing the primary coil circuit with semiconductor devices using digital techniques
    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F02—COMBUSTION ENGINES; HOT-GAS OR COMBUSTION-PRODUCT ENGINE PLANTS
    • F02P—IGNITION, OTHER THAN COMPRESSION IGNITION, FOR INTERNAL-COMBUSTION ENGINES; TESTING OF IGNITION TIMING IN COMPRESSION-IGNITION ENGINES
    • F02P3/00—Other installations
    • F02P3/02—Other installations having inductive energy storage, e.g. arrangements of induction coils
    • F02P3/04—Layout of circuits
    • F02P3/055—Layout of circuits with protective means to prevent damage to the circuit, e.g. semiconductor devices or the ignition coil
    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
    • G05F1/56—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices
    • G05F1/561—Voltage to current converters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01T—SPARK GAPS; OVERVOLTAGE ARRESTERS USING SPARK GAPS; SPARKING PLUGS; CORONA DEVICES; GENERATING IONS TO BE INTRODUCED INTO NON-ENCLOSED GASES
    • H01T15/00—Circuits specially adapted for spark gaps, e.g. ignition circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/08—Modifications for protecting switching circuit against overcurrent or overvoltage
    • H03K17/082—Modifications for protecting switching circuit against overcurrent or overvoltage by feedback from the output to the control circuit
    • H03K17/0828—Modifications for protecting switching circuit against overcurrent or overvoltage by feedback from the output to the control circuit in composite switches
    • F—MECHANICAL ENGINEERING; LIGHTING; HEATING; WEAPONS; BLASTING
    • F02—COMBUSTION ENGINES; HOT-GAS OR COMBUSTION-PRODUCT ENGINE PLANTS
    • F02P—IGNITION, OTHER THAN COMPRESSION IGNITION, FOR INTERNAL-COMBUSTION ENGINES; TESTING OF IGNITION TIMING IN COMPRESSION-IGNITION ENGINES
    • F02P3/00—Other installations
    • F02P3/02—Other installations having inductive energy storage, e.g. arrangements of induction coils
    • F02P3/04—Layout of circuits
    • F02P3/055—Layout of circuits with protective means to prevent damage to the circuit, e.g. semiconductor devices or the ignition coil
    • F02P3/0552—Opening or closing the primary coil circuit with semiconductor devices
    • F02P3/0554—Opening or closing the primary coil circuit with semiconductor devices using digital techniques
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/08—Modifications for protecting switching circuit against overcurrent or overvoltage
    • H03K2017/0806—Modifications for protecting switching circuit against overcurrent or overvoltage against excessive temperature
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K2217/00—Indexing scheme related to electronic switching or gating, i.e. not by contact-making or -breaking covered by H03K17/00
    • H03K2217/0027—Measuring means of, e.g. currents through or voltages across the switch

Definitions

  • the present invention relates to a semiconductor switching element control circuit and a semiconductor device.
  • an igniter semiconductor device that performs on / off control of a semiconductor switching element for use in addition to a spark plug of an internal combustion engine is known.
  • the temperature characteristic of the diode is used to detect the temperature of the semiconductor switching element using a current signal obtained through the diode, and this is used for overheat protection.
  • An ignition system for an internal combustion engine such as an automobile engine includes an ignition coil (inductive load, hereinafter also referred to as “L load”) for generating a high voltage applied to a spark plug.
  • L load an ignition coil
  • L load noise a displacement current
  • the L load noise may cause erroneous temperature detection, and there is a problem that unnecessary overheat protection is performed in accordance with the erroneous temperature detection.
  • the present invention has been made to solve the above-described problems, and provides a semiconductor switching element control circuit and a semiconductor device that suppress unnecessary unnecessary overheat protection due to erroneous detection of temperature. With the goal.
  • the control circuit of the semiconductor switching element is: A control circuit comprising a control terminal, a main electrode terminal and a current sense terminal, and controlling a semiconductor switching element having a diode connected to the main electrode terminal or the current sense terminal, An overheat detection circuit that emits an overheat detection signal when the temperature detected based on the output of the diode is equal to or higher than a preset temperature; and A current detection circuit that generates a current detection signal when an output value of the current sense terminal is equal to or greater than a predetermined set current value; the overheat detection signal from the overheat detection circuit; and the current detection signal from the current detection circuit.
  • a cutoff circuit that turns off the semiconductor switching element when both are input; and Is provided.
  • the semiconductor device is A semiconductor device for turning on and off a current flowing through an inductive load for energizing an ignition plug of an internal combustion engine,
  • a semiconductor switching element comprising a main electrode terminal and a current sense terminal, wherein the main electrode terminal is connected to the inductive load;
  • a diode connected to the main electrode terminal or the current sense terminal;
  • An overheat detection circuit that emits an overheat detection signal when the temperature detected based on the output of the diode is equal to or higher than a preset temperature; and
  • a current detection circuit that generates a current detection signal when an output value of the current sense terminal is equal to or greater than a predetermined set current value; the overheat detection signal from the overheat detection circuit; and the current detection signal from the current detection circuit.
  • a cutoff circuit that turns off the semiconductor switching element when both are input; and Is provided.
  • the semiconductor switching element since it is determined whether or not overheat protection is necessary based on both the overheat detection signal and the current detection signal, it is determined that the load current is flowing to the semiconductor switching element to some extent and that the temperature is abnormal. Only in this case, the semiconductor switching element can be turned off. As a result, even if the overheat detection signal is erroneously output, unnecessary overheat protection can be prevented from being performed.
  • FIG. 1 is a diagram showing an igniter semiconductor device according to a first embodiment of the present invention
  • FIG. 3 is a time chart for explaining the operation of the igniter semiconductor device according to the first exemplary embodiment of the present invention
  • FIG. 6 is a diagram showing a semiconductor switching element according to a modification applicable to the first to third embodiments of the present invention. It is a figure which shows the control circuit of the semiconductor switching element concerning the related technology relevant to embodiment of this invention. It is a figure which shows the relationship between the reverse direction saturation current of the diode relevant to embodiment of this invention, and diode temperature. It is a figure which shows the control circuit of the semiconductor switching element concerning Embodiment 4 of this invention. It is a time chart for demonstrating operation
  • switching element prevents the switching element from deteriorating when the semiconductor switching element that drives the inductive load (hereinafter also simply referred to as “switching element”) becomes an abnormal temperature. And a circuit method.
  • Japanese Unexamined Patent Publication No. 2011-124269 discloses an ignition system for an internal combustion engine such as an automobile engine.
  • This publication proposes an overheat protection circuit that shuts off a switching element in order to prevent the switching element from deteriorating when the switching element abnormally generates heat.
  • overheating protection of the ignition system according to the above publication has the following problems to be solved.
  • Soft interruption is an interruption method in which the load current gradually approaches zero, that is, an interruption method in which the current is gradually reduced.
  • hard cut-off is a method of forcibly cutting off the current of the switching element, that is, a method of cutting off the current sharply.
  • soft shutoff it is preferable to perform soft shutoff in order to shut off a switching element when overheating is detected. This is because when a hard shutdown is performed, the load current is cut off at a timing unrelated to the ignition signal timing that is normally scheduled by the engine control computer, so that the spark plug is energized at a timing different from the correct engine ignition timing. This is because there is a high possibility that it will end up.
  • the igniter power semiconductor device has a “current limiting function” for limiting the load current to a preset current limiting value, and prevents erroneous ignition at an unnecessary timing.
  • This “current limiting function” is used for the purpose of self-protection of switching elements.
  • the load current cut-off timing coincides with the ignition timing normally scheduled by the engine control computer, and the joule loss is suppressed by lowering the current limit value depending on the temperature during abnormal heat generation. The calorific value is suppressed.
  • the current limit value gradually decreases depending on the temperature.
  • an igniter semiconductor device including an overheat protection device that can quickly shut off when an abnormal temperature is reached, using the temperature characteristics of the reverse saturation current of the diode.
  • FIG. 1 is a diagram showing an igniter semiconductor device 10 according to a first embodiment of the present invention.
  • FIG. 2 is a time chart for explaining the operation of the igniter semiconductor device 10 according to the first embodiment of the present invention.
  • the ignition system induction ignition system for an internal combustion engine such as an automobile engine shown in FIG. 1 includes each component for generating a high voltage to be applied to the spark plug 7.
  • this ignition system includes a transformer 6 as an ignition coil (inductive load, hereinafter also referred to as “L load”), a semiconductor device 10 that is a so-called igniter that controls a current flowing through the transformer 6, and a computer.
  • L load inductive load
  • the semiconductor device 10 includes a semiconductor switching element 5 (hereinafter also simply referred to as “switching element 5”) connected to the transformer 6, and a control circuit 1 that controls on / off (that is, conduction and interruption) of the switching element 5. Yes.
  • the semiconductor device 10 includes a switching element 5 including a gate terminal G that is a control terminal, a collector terminal C that is a main electrode terminal, and an emitter terminal E, a diode Ds2 having an anode connected to the emitter terminal E of the switching element 5, and a drive circuit. 4, and a control circuit 1 that controls the switching element 5 in response to the signal from 4.
  • the diode Ds2 is a temperature sensitive diode that shows an output corresponding to the temperature.
  • the control circuit 1 receives a control signal from the ECU 3, which is a control computer, via the drive circuit 4.
  • a drive voltage signal Von is input from the drive circuit 4 to the control circuit 1.
  • the IGBT (insulated gate bipolar transistor) 5a which is the main component of the switching element 5, has a small sense proportional to the collector current Ic in addition to the general collector terminal C, emitter terminal E, and gate terminal G as electrode terminals.
  • a device having a current sense terminal SE through which a current Isense flows is employed.
  • the ratio between the collector current Ic and the sense current Isense is, for example, about 1/1000, and the sense current Isense is very small.
  • the switching element 5 uses a clamp Zener diode Ds0 and a diode Ds2 in addition to the IGBT 5a.
  • the switching element 5 is connected to the primary winding 6 a of the transformer 6, and the secondary winding 6 b of the transformer 6 is connected to the spark plug 7.
  • the diode Ds2 is built in the switching element 5.
  • the diode Ds2 may be externally attached to the switching element 5, for example, may be built in the control circuit 1.
  • the switching element 5 is formed by forming an IGBT 5a and a diode Ds2 as a temperature-sensitive element for detecting the temperature of the IGBT 5a on the same substrate.
  • the diode Ds2 may be a Zener diode as shown in the circuit diagram, for example, a Schottky barrier diode.
  • the switching element 5 includes a temperature sense terminal TSD connected to the cathode of the diode Ds2.
  • a reverse saturation current Is3 of the diode Ds2 flows through the temperature sense terminal TSD. Based on the reverse direction saturation current Is3, the temperature of the switching element 5 (that is, the element temperature Tigbt which is the temperature of the IGBT 5a) is detected.
  • the main component of the switching element 5 is an IGBT.
  • a MOSFET having a current sense terminal may be used instead of the IGBT.
  • a wide band gap semiconductor may be used as the semiconductor material.
  • Silicon carbide (SiC), gallium nitride-based material (GaN), or diamond may be used as the wide band gap semiconductor.
  • the transformer 6 is an ignition coil.
  • a power source Vp such as a battery is connected to one end of the primary winding 6a, and the semiconductor device 10 (specifically, the collector terminal C of the switching element 5) is connected to the other end.
  • the semiconductor device 10 specifically, the collector terminal C of the switching element 5
  • one end of the secondary winding 6b is connected to the power supply Vp, and the other end is connected to a spark plug 7 whose one end is grounded.
  • the switching element 5 can energize / cut off the primary current of the transformer 6 (ignition coil).
  • the ECU 3 and the drive circuit 4 are connected to a GND_ECU that is a control system ground.
  • the control circuit 1 and the switching element 5 are connected to GND_PW which is a power system ground.
  • an ON signal of the ECU 3 is transmitted to the control circuit 1 via the drive circuit 4.
  • the ON signal is transmitted to the gate terminal G of the IGBT 5 a that is a power semiconductor switching element.
  • the IGBT 5a is driven, a current flows through the transformer 6, that is, the inductive load (L load).
  • an off signal is input to the gate terminal G, and the IGBT 5a is turned off.
  • collector-emitter voltage Vce rises to about 500V, and high voltage V2 is excited in secondary winding 6b of transformer 6.
  • V2 is minus 30 kV or more.
  • the time chart shown in FIG. 2 will be described as follows. (1) By applying an ON signal to the gate terminal G, the IGBT 5a is turned ON. (2) A load current Ic as shown in FIG. 2 flows according to a time constant determined by the inductance and wiring resistance of the L load. (3) The signal at the gate terminal G is turned off at the timing Tig at which the fuel is to be ignited. (4) The gate drive signal of the IGBT 5a is also turned off, and the load current Ic is cut off. (5) This current interruption induces a change in the chain magnetic flux in the transformer 6 and induces a high voltage depending on the turn ratio on the secondary side. As a result, discharge occurs in the spark plug 7 in the engine cylinder.
  • the area surrounded by the broken line X1 in FIG. 2 is an area where the “current limiting function” is operated by the “current limiting circuit 19” described later. In this region X1, a large Joule loss occurs in the switching element 5.
  • a clamp Zener diode Ds0 is provided between C and G so that Vce is clamped at about 500V.
  • the value of the load current Ic varies depending on the ON time of the control signal and the voltage Vp.
  • the semiconductor device 10 is configured such that when the load current Ic becomes a certain value or more, a “current limiting function” that limits the current so as to reach the current level works.
  • the current limiting function is a protection function that prevents a load current Ic exceeding a certain level from flowing. This is to avoid the risk of melting of the winding of the transformer 6 or magnetic saturation of the transformer 6. That is, according to the current limiting function of the semiconductor device 10, winding fusing due to overcurrent, suppression of magnet demagnetization for adjusting the reluctance (magnetic resistance) of the transformer 6, and suppression of magnetic saturation of the core material are performed.
  • the setting value that determines the maximum allowable current value in the current limiting function is also referred to as “current limiting value Ilm”.
  • the current limit value Ilm can be set to a value such as 10 A or 14 A, for example.
  • FIG. 3 is a diagram illustrating the control circuit 1 for the switching element according to the first embodiment of the present invention.
  • the control circuit 1 includes a Schmitt trigger circuit B1 that shapes the applied input signal on the ECU 3 side, and a turn-on delay circuit 13 that outputs a control signal EST obtained by adding a predetermined delay to the output of the Schmitt trigger circuit B1.
  • Each circuit element in the control circuit 1 operates by receiving a control power supply Vreg.
  • the control circuit 1 includes a first current mirror circuit composed of a PMOSFET 44 and a PMOSFET 45, a resistor Rg1, a detection resistor Rs1, and an NMOSFET 71.
  • One end of the resistor Rg1 is connected to a connection point between the gate terminal G of the IGBT 5a and the PMOSFET 45, and the other end of the resistor Rg1 is connected to the ground.
  • One end of the detection resistor Rs1 is connected to the current sense terminal SE, and the other end of the detection resistor Rs1 is connected to the emitter terminal E.
  • the output signal of the inverter INV1 is applied to the gate of the NMOSFET 71, and when the NMOSFET 71 becomes conductive, the emitter terminal E and the gate terminal G are connected via the resistor Rg2.
  • the first current mirror circuit generates a first mirror current Ig2 that is a duplicate of the first input current Ig1, and uses the first mirror current Ig2 as an input signal to the gate terminal G of the switching element 5.
  • the control circuit 1 further includes a current limiting circuit 19, an overheat detection circuit 16, a current detection circuit 18, and a cutoff circuit 11.
  • the control circuit 1 includes an overheat detection circuit 16 for performing temperature detection based on the diode Ds2.
  • the overheat detection circuit 16 detects the temperature of the switching element 5 and outputs an overheat detection signal OT depending on the temperature.
  • the overheat detection circuit 16 generates an overheat detection signal OT when the temperature detected based on the output of the diode Ds2 is equal to or higher than a preset temperature.
  • the control circuit 1 includes a current detection circuit 18.
  • the current detection circuit 18 monitors the load current Ic of the switching element 5 and generates an output corresponding to the load current Ic.
  • Isense is input to the current detection circuit 18.
  • the current detection circuit 18 detects a current proportional to the load current Ic based on this Isense, and outputs a current detection signal Enable depending on the load current Ic flowing through the switching element 5.
  • the current detection circuit 18 generates a current detection signal Enable when the output value of the current sense terminal SE is equal to or greater than a predetermined “set current value”.
  • the anode of the diode Ds2 is connected to the emitter terminal E of the switching element 5, and the cathode of the diode Ds2 is connected to the overheat detection circuit 16 described above.
  • One end of the current detection circuit 18 is connected to a connection point between the detection resistor Rs1 and the current sense terminal SE of the switching element 5.
  • the cutoff circuit 11 turns off the switching element 5 when both the overheat detection signal OT from the overheat detection circuit 16 and the current detection signal Enable from the current detection circuit 18 are input. As a result, the cutoff circuit 11 causes the load that flows through the switching element 5 when the load current Ic that flows through the switching element 5 is equal to or higher than a predetermined set current and the temperature detected by the diode Ds2 is equal to or higher than the predetermined set temperature. Cut off current. Specifically, the cutoff circuit 11 turns off the switching element 5 by reducing the first input current Ig1.
  • the cutoff circuit 11 includes an AND circuit 12, a latch circuit 14, and a PMOSFET 48.
  • the AND circuit 12 calculates a logical product of the overheat detection signal OT and the current detection signal Enable.
  • the latch circuit 14 latches the output value (Qbar) in response to the output change of the AND circuit 12.
  • the PMOSFET 48 is a switch element that operates in response to the output value of the latch circuit 14 to reduce the first input current Ig1. More specifically, the outputs of the current detection circuit 18 and the overheat detection circuit 16 are input to the AND circuit 12.
  • the output signal OUTA of the AND circuit 12 is input to the set terminal S of the latch circuit 14.
  • the reset terminal R of the latch circuit 14 is connected to the output of the inverter INV1.
  • the Q bar terminal Qbar of the latch circuit 14 is connected to the gate of the PMOSFET 48.
  • the latch circuit 14 keeps operating the switch element, thereby maintaining the stop of the switching element 5. Once the overheat is detected by the latch circuit 14, overheat protection can be continued until the control signal EST is turned off.
  • the current limiting circuit 19 includes an amplifier 22, a voltage / current conversion circuit 20, a constant current source I_base, and a second current mirror circuit composed of PMOSFETs 42 and 43.
  • the second current mirror circuit replicates the second input current If1 flowing through the PMOSFET 43 and generates a second mirror current If2 flowing through the PMOSFET 42.
  • the constant current source I_base is connected to the junction of the second mirror current If2 and the first input current Ig1.
  • the current limit circuit 19 adjusts the second input current If1 based on the voltage of the current sense terminal SE so as to limit the load current Ic of the switching element 5 to the set current limit value Ilm. A specific operation will be described later using the related technique of FIG.
  • the anode of the diode Ds2 is connected to the emitter terminal E, and the cathode of the diode Ds2 is connected to the overheat detection circuit 16.
  • the interruption circuit 11 performs temperature detection based on the value of the reverse saturation current Is3 flowing through the cathode of the diode Ds2.
  • FIG. 14 is a diagram showing a control circuit 401 of the switching element 5 according to related technology related to the embodiment of the present invention.
  • the operation of the current limiting circuit 19 will be described with reference to FIG. 14, but the current limiting circuit 19 shown in FIG. 3 also performs the same circuit operation. Therefore, the following description can be applied to the control circuit 1 with respect to the configuration in which the control circuit 1 and the control circuit 401 have the same reference numerals.
  • the limitation of the collector current Ic by the current limiting circuit 19 is realized by the following mechanism.
  • the sense current Isense of the IGBT 5a is supplied to the detection resistor Rs1 in the control circuit 401, and a sense voltage Vsense corresponding to the collector current Ic of the IGBT 5a is generated in the detection resistor Rs1.
  • the sense voltage Vsense is compared with the voltage value of the reference voltage source Vref by the amplifier 22, and a voltage corresponding to the difference is input to the voltage-current conversion circuit 20.
  • the voltage-current conversion circuit 20 outputs a current If1 corresponding to the difference between the sense voltage Vsense and the voltage value of the reference voltage source Vref.
  • This current If1 becomes the second input current If1 input to the second current mirror circuit constituted by the PMOSFET 42 and the PMOSFET 43.
  • the second current mirror circuit outputs a second mirror current If2 corresponding to the mirror ratio in response to the input of the second input current If1.
  • the second mirror current If2 is also referred to as “current limiting signal If2”. Since the current limit signal If2 works to reduce the current Ig2 that generates the gate drive voltage of the IGBT 5a, when the current limit signal If2 increases, the gate voltage decreases and the collector current Ic is prevented from increasing. That is, the collector current Ic is limited to a predetermined constant value because the entire system operates so as to perform a negative feedback operation.
  • the current limit signal If2 is generated by the second input current If1, and flows into the base reference current source (constant current source) I_base. Since the input current Ig1 of the first current mirror circuit composed of the PMOSFET 44 and the PMOSFET 45 varies with the inflow of the current limiting signal If2, the current Ig2 of the first current mirror circuit varies, and the gate voltage for driving the switching element 5 Fluctuates. As the load current Ic increases, the current Ig2 decreases, and the voltage generated by the resistor Rg1 decreases. In this way, negative feedback control that suppresses the load current Ic is realized.
  • the resistor Rg1 has a resistance value on the order of several tens of k ⁇ .
  • the control circuit 401 further includes an overheat protection circuit 410.
  • the overheat protection circuit 410 includes a diode Ds1 whose anode is connected to the emitter terminal E, and a third current mirror circuit composed of a PMOSFET 46 and a PMOSFET 47.
  • the diode Ds1 is also a temperature sensitive diode that exhibits an output corresponding to the temperature.
  • the cathode of the diode Ds1 is connected to the reference side of the third current mirror circuit.
  • the output current Is2 of the third current mirror circuit acts to reduce the current Ig2 that generates the gate drive voltage of the IGBT 5a, as in the above-described current limiting function.
  • FIG. 15 is a diagram showing an example of the relationship between the reverse saturation current Is3 of the diode Ds2 and the temperature of the diode Ds2 related to the embodiment of the present invention.
  • the reverse saturation current of the Schottky barrier diode rises rapidly near 170 ° C.
  • a Schottky barrier diode having the same characteristics as in FIG. 15 can be employed.
  • the overheat detection in the first embodiment is performed using the reverse saturation current Is3 of the diode Ds2.
  • the reverse saturation current Is3 increases with the same temperature characteristics as in FIG.
  • a modification may be adopted in which temperature detection is performed using a forward voltage of a diode connected in the forward direction, for example, by inverting the connection direction of the diode Ds2.
  • the current dependency is high, so that highly accurate current control is required.
  • the temperature detection is performed using the reverse saturation current of the diode Ds2 as in the first embodiment, the voltage dependency is small when the diode Ds2 is used below the reverse breakdown voltage. There is an advantage that the accuracy is not necessarily high.
  • FIG. 4 is a time chart for explaining the operation of the switching element control circuit 1 according to the first embodiment of the present invention.
  • the reverse saturation current Is3 reaches a predetermined threshold value.
  • This threshold value is the overheat cutoff judgment current Ithot in FIG.
  • the overheat detection circuit 16 generates an overheat detection signal OT by inverting the output. Thereby, an abnormally high temperature can be detected. That is, the overheat cutoff determination current Ithot is the value of the reverse saturation current Is3 when the element temperature Tigbt reaches the set temperature Tm1 (for example, 210 ° C.).
  • a displacement current 100 (also referred to as L load noise 100) is generated by the L load when the switching element 5 is turned on.
  • a displacement current 100 is generated at times t1 to t2 and t6 to t7.
  • the overheat detection signal OT rises due to the displacement current 100 (reference numeral 101).
  • the load current Ic of the switching element 5 is monitored by the detection resistor Rs1, and the logical product of the output dependent on the load current Ic (that is, the Enable signal in FIG. 4) and the output of the overheat detection circuit 16 ( OUTA) in FIG. 4 is the overheat cutoff judgment output.
  • the Enable signal is not generated (that is, the current detection circuit 18 When the output is low), the logical product OUTA does not go high, and as a result, the blocking circuit 11 does not operate.
  • the load current Ic is equal to or higher than a predetermined set current value, and the protection (that is, the protection that completely cuts off the load current of the switching element 5) is applied only when an abnormal temperature is detected.
  • the “set current value” here is, for example, a predetermined value of 1 A or less, and is a value for detecting that a certain amount of load current Ic flows through the switching element 5.
  • the current limiting function described above is used to cut off the switching element 5. If it is determined that the temperature is abnormal and protection is required, the PMOSFET 48 is turned on, and If3 flows into the reference current source (constant current source) I_base. However, the relationship of If3 ⁇ I_base holds for the magnitude of the current. As a result, the current Ig2 in the first current mirror circuit composed of the PMOSFET 44 and the PMOSFET 45 is reduced to 0A. As a result, the gate drive signal (that is, the gate voltage) of the switching element 5 can be attenuated to 0V, and the switching element 5 can be turned off.
  • FIG. 5 is a diagram showing a relationship between the current limit value Ilm and the element temperature Tigbt in the switching element control circuit 1 according to the first embodiment of the present invention.
  • a characteristic Cv0 in FIG. 5 indicates the temperature dependence of the current limit value Ilm in the above-described comparative example in FIG.
  • the characteristic Cv1 in FIG. 5 shows the temperature dependence of the current limit value Ilm in the first embodiment. Focusing on the position indicated by the arrow X3, when the temperature detected by the diode Ds2 reaches a predetermined set temperature Tm1 (210 ° C. as an example in the first embodiment), the current limit value Ilm is quickly attenuated to 0A. Can do.
  • the latch circuit 14 detects an abnormal temperature, it achieves a function of maintaining protection until the control signal EST is turned off. As a result, it is prevented that load current Ic oscillates due to repeated overheat interruption and recovery.
  • the oscillation of the load current Ic is an abnormal temperature detection ⁇ gate signal OFF (that is, load current Ic cutoff) ⁇ temperature decrease ⁇ gate signal ON (that is, recovery) ⁇ temperature increase ⁇ abnormal temperature detection ⁇ load current Ic cutoff.
  • the interruption and the return are repeated.
  • L load noise (displacement current) 100 associated with the operation of the switching element 5 may cause erroneous temperature detection by the diode Ds2, but as described above, according to the first embodiment, the L load noise is generated. Even if 100 occurs, unless the Enable signal is generated, overheat protection (that is, the switching element 5 is forcibly turned off) is not performed. That is, in the first embodiment, since it is determined whether or not overheat protection is necessary based on the result of logical operation by the AND circuit 12, the load current Ic is flowing to some extent (that is, the load current Ic is set in advance). Only when it is determined that the temperature is equal to or higher than the current value and the temperature is abnormal, it can be interrupted. As a result, unnecessary overheat protection associated with the L load noise 100 can be suppressed.
  • FIG. 6 is a diagram illustrating an example of an overheat detection circuit provided in the switching element control circuit 1 according to the first embodiment of the present invention.
  • the overheat detection circuit 116 shown in FIG. 6 is a circuit applicable as the overheat detection circuit 16 of FIG. 3 and includes a current comparator.
  • the current comparator compares the detection-side current Is4 corresponding to the reverse saturation current of the diode Ds2 with the current value of the reference current source (constant current source) Iref1. Since the reverse saturation current characteristic of the diode Ds2 is used for temperature detection, it is preferable to use a current comparator instead of a voltage comparator for the overheat detection circuit 116. This eliminates the need for an IV conversion circuit as compared with the case of using a voltage comparator, thereby enabling a reduction in the size of the circuit.
  • a current obtained by subtracting the reverse saturation current Is3 from the constant current source I_base2 (ie, I_base2-Is3) is passed through the series circuit of the resistor Rd1, the Schottky barrier diode Dz1, and the NMOSFET 74.
  • This current generates a voltage Vd applied to the diode Ds2.
  • the voltage Vd is, for example, 3V.
  • the NMOSFET 74 and the NMOSFET 73 constitute a fourth current mirror circuit.
  • the NMOSFET 73 is connected to a PMOSFET 51 of a fifth current mirror circuit to be described later, and a common detection-side current Is4 flows through them.
  • the relationship between the current values of the two constant current sources included in the overheat detection circuit 116 is I_ref1 ⁇ I_base2.
  • a difference (I_base2-I_ref1) between I_base2 and I_ref1 corresponds to the overheat determination threshold value Ihot.
  • the detection-side current Is4 flows to the PMOFET 51, and a mirror current that duplicates the detection-side current Is4 flows to the PMOSFET 50.
  • a connection point between the PMOSFET 50 and the constant current source I_ref1 is an output Vout1.
  • the relationship of I_ref1 ⁇ Is4 is maintained, so that the output Vout1 is at a high level.
  • the relationship I_ref1> Is4 is established, and the output Vout1 becomes a low level.
  • Vout1 is input to a CMOS circuit composed of a PMOSFET 49 and an NMOSFET 72, and the output of the CMOS circuit becomes an overheat detection signal OT. Thereby, the overheat determination at the time of abnormal temperature can be performed.
  • FIG. 7 is a block diagram showing an example of a current detection circuit provided in the switching element control circuit 1 according to the first exemplary embodiment of the present invention.
  • the current detection circuit 118 shown in FIG. 7 is converted by a voltage-current conversion circuit 181 that converts a voltage corresponding to the difference between the voltage generated in the detection resistor Rs1 and the reference voltage source Vref2 into a current, and the voltage-current conversion circuit 181.
  • a current comparator 182 that compares the current with a current value of a reference current source (constant current source) I_ref2.
  • the current detection in the current detection circuit 118 in FIG. 7 is realized by the following mechanism.
  • the sense current Isense of the switching element 5 is energized to the detection resistor Rs1 in the control circuit 1.
  • a voltage proportional to the load current Ic of the switching element 5 is generated in the detection resistor Rs1.
  • the voltage generated in the detection resistor Rs1 is compared with the reference voltage source Vref2 by the amplifier 183.
  • the output of the amplifier 183 is input to the voltage / current conversion circuit 181.
  • the voltage-current conversion circuit 181 outputs a current Iout2 corresponding to the difference between the voltage generated in the detection resistor Rs1 and the reference voltage source Vref2.
  • This output current Iout2 is compared with a reference current source (constant current source) I_ref2 by a current comparator 182, and current is detected by outputting a determination voltage Enable.
  • FIG. 8 is a diagram illustrating an example of a current detection circuit included in the control circuit 1 for the switching element according to the first exemplary embodiment of the present invention, and an example in which the block diagram illustrated in FIG. 7 is configured as a specific circuit. It is.
  • the current detection circuit 118 includes a sixth current mirror circuit composed of a PMOSFET 52 and a PMOSFET 53 having one end connected to the voltage Vreg, and a seventh current mirror circuit composed of an NMOSFET 55 and an NMOSFET 56 that are vertically stacked on the sixth current mirror circuit. It has.
  • a resistor R1a and a detection resistor Rs1 are connected in series to the NMOSFET 55.
  • a resistor R1b and a detection resistor Rs2 are connected in series to the NMOSFET 56.
  • the PMOSFET 54 and the PMOSFET 52 constitute an eighth current mirror circuit, and the connection point between the PMOSFET 54 and the constant current source Iref2 is the output point of the Enable signal.
  • the drain-source current of the PMOSFET 54 flows toward the constant current source Iref2.
  • a sense voltage Vsense is generated by a sense current Isense proportional to the load current Ic.
  • the detection resistor Rs1 is, for example, 30 ⁇ .
  • the sense current Isense is, for example, 1/1000 of the load current Ic, and is, for example, several mA to several tens mA.
  • the voltage generated in the detection resistor Rs1 by the sense current Isense is, for example, several tens to several hundreds mV.
  • the detection-side current Is5 is determined by the sense voltage Vsense generated in the detection resistor Rs1 and the detection resistor Rs2 used for VI conversion.
  • the detection resistor Rs2 has, for example, 5 k ⁇ . As the load current Ic increases, the voltage generated at the detection resistor Rs1 also increases, and the detection-side current Is5 also increases.
  • the sense current Isense is detected to be 1 mA or more is referred to as “current detection”.
  • the detection threshold voltage for example, 30 mV
  • the voltage-current conversion circuit converts the current into, for example, about 5 ⁇ A, and performs comparison with the reference current source (constant current source) I_ref2 as a current comparator.
  • the risk of misjudgment is reduced and accuracy can be improved.
  • Iref_2 When the relationship between the converted detection-side current Is5 and the reference-side current I_ref2 is Iref_2> Is5, the output Enable is at a low level. When I_ref2 ⁇ Is5, the output Enable is at a high level. That is, Iref_2 corresponds to the comparator threshold value.
  • the temperature of the switching element 5 is detected using the reverse saturation current Is3 of the diode Ds2. For this reason, when the current comparator is used, an IV conversion circuit is not required as compared with the voltage comparator, and the circuit can be reduced in size. Further, in the current detection circuit 18, the voltage generated in the detection resistor Rs1 is as small as several tens mV. For this reason, it is possible to improve the determination accuracy as compared with the voltage comparator by performing the current conversion by the voltage-current conversion circuit and determining the output by the current comparator.
  • FIG. The igniter semiconductor device according to the second embodiment has a circuit configuration similar to that of the semiconductor device 10 according to the first embodiment except that the control circuit 1 is replaced with the control circuit 201.
  • the control circuit 201 according to the second embodiment has a circuit configuration similar to that of the control circuit 1 according to the first embodiment except that an attenuation circuit 210 is added. Therefore, in the following description, the same or corresponding components as those in the first embodiment will be described with the same reference numerals, and differences from the first embodiment will be mainly described, and common items will be briefly described. Or omitted.
  • FIG. 9 is a diagram showing a control circuit 201 of the switching element according to the second embodiment of the present invention.
  • the control circuit 201 according to the second embodiment includes an attenuation circuit 210 for giving a temperature characteristic to the current limit value Ilm.
  • the attenuation circuit 210 includes a diode Ds1 that is a temperature sensing element, and a second current mirror circuit that includes a MOSFET 46 and a MOSFET 47.
  • the attenuation circuit 210 is similar to the related art overheat protection circuit 410 shown in FIG.
  • the attenuation circuit 210 attenuates the gate drive signal (that is, the gate voltage) so as to reduce the load current Ic when the sensed temperature of the Zener diode Ds1 reaches a predetermined reference temperature (Tm2) or higher.
  • the reference temperature Tm2 is a value lower than the set temperature Tm1.
  • the load current Ic is reduced by the characteristic shown in FIG. 10 by providing the current limit value Ilm with a temperature characteristic and lowering it at a point where the reference temperature Tm2 or more (for example, 170 ° C.) is exceeded. That is, in particular, according to the second embodiment, the load current Ic is reduced as the temperature increases, and the rate of decrease in the load current Ic is increased as the temperature is further increased. The curve is decreasing.
  • a voltage lower than the control power supply Vreg by a voltage corresponding to the threshold value of the PMOSFET 47 is applied between the anode and the cathode of the diode Ds1.
  • the reverse saturation current Is3 shows an exponential increase with temperature.
  • a diode Ds2 having a size and a specification that selects a current in the order of affecting the constant current source I_base from around the temperature Tm2 (170 ° C.) is selected. Thereby, in the operation below the temperature Tm2 (170 ° C.), the drive of the switching element 5 is not affected.
  • the reverse saturation current Is1 of the diode Ds1 flows through the PMOSFET 47.
  • the PMOSFET 47 and the PMOSFET 46 constitute a third current mirror circuit. Therefore, the reverse saturation current Is1 flowing through the PMOSFET 47 generates the drain current Is2 of the PMOSFET 46.
  • the drain current Is2 has the same effect as the current limit signal If2 of the current limit circuit 19, and works to reduce the current Ig2 that generates the gate drive voltage of the IGBT 5a.
  • FIG. 10 is a diagram showing the relationship between the current limit value Ilm and the temperature in the control circuit 1 of the switching element according to the second embodiment of the present invention.
  • Tm2 for example, 170 ° C.
  • a temperature characteristic is given to the current limit value Ilm on the high temperature side. Due to this temperature characteristic, the current limit value Ilm gradually decreases as the temperature rises.
  • Tm1 170 ° C. to 210 ° C.
  • the current limit value Ilm is gradually decreased to suppress the Joule loss generated in the switching element 5, thereby suppressing heat generation.
  • the switching element responds to the output of the AND circuit 12 indicating the occurrence of overheating. 5 is forcibly shut off. That is, when an abnormally high temperature exceeding the set temperature Tm1 (210 ° C.) is reached, the switching element 5 is completely shut off.
  • Tm1 eg, 210 ° C.
  • the burden on the switching element 5 is also small, and the effect of preventing the life reduction is further enhanced.
  • Embodiment 3 FIG.
  • the igniter semiconductor device according to the third embodiment has a circuit configuration similar to that of the igniter semiconductor device 10 according to the first embodiment except that the control circuit 1 is replaced with a control circuit 301.
  • the control circuit 301 according to the third embodiment has a circuit configuration similar to that of the control circuit 1 according to the first embodiment except that an attenuation circuit 310 is added. Therefore, in the following description, the same or corresponding components as those in the first embodiment will be described with the same reference numerals, and differences from the first embodiment will be mainly described, and common items will be briefly described. Or omitted.
  • FIG. 11 is a diagram showing a control circuit 301 of the switching element according to the third embodiment of the present invention.
  • the attenuation circuit 310 has a circuit configuration similar to that of the attenuation circuit 210, and further includes a PMOSFET 57, an AND circuit 312, and a current limit attenuation start temperature detection circuit 316.
  • the PMOSFET 57 is interposed between the PMOSFET 46 and the constant current source I_base and switches the electrical connection therebetween.
  • the gate of the PMOSFET 57 is controlled by the output signal of the AND circuit 312.
  • the AND circuit 312 receives a signal from the current limit decay start temperature detection circuit 316 and a current detection signal Enable from the current detection circuit 18.
  • the current limit decay start temperature detection circuit 316 can detect the temperature of the IGBT 5a based on the reverse saturation current Is3 of the diode Ds2.
  • another current limiting attenuation start temperature detection circuit 316 and an AND circuit 312 are provided in the control circuit 301. This is a device for starting the attenuation of the current limit value Ilm from a desired start temperature different from the temperature detected by the overheat detection circuit 16.
  • the overheat detection method is performed using the reverse saturation current Is3 of the diode Ds2.
  • the reverse saturation current Is3 of the diode Ds2 rises to a preset overheat cutoff judgment current Ithot, in other words, when the temperature of the IGBT 5a rises to the set temperature Tm1, it is judged as an abnormal temperature.
  • the AND circuit 312 calculates the logical product of the current detection circuit 18 that detects the load current Ic and the current limit attenuation start temperature detection circuit 316.
  • the AND circuit 312 issues an overheat determination output when the overheat detection signal OT and the current detection signal Enable are input (details are the same as OUTA in FIG. 4).
  • the PMOSFET 57 is turned on.
  • the PMOSFET 57 is always off until reaching the current limit decay start temperatures Tm2 to Tm4 lower than the set temperature Tm1. For this reason, it can prevent that current limiting value Ilm falls in the temperature range below setting temperature Tm1, and can suppress that ignition energy falls in a low temperature range.
  • FIG. 12 is a diagram showing a relationship between the current limit value Ilm and the temperature in the control circuit 1 of the switching element according to the third embodiment of the present invention.
  • the temperature at which the current limit is started can be freely set by the current limit attenuation start temperature detection circuit 316 independently of the overheat detection circuit 16.
  • the temperature characteristics Cv2, Cv3, and Cv4 of the current limit value Ilm as shown in FIG. 12 can be freely designed.
  • the current limit decay start temperature Tm2 (170 degrees)
  • the current limit decay start temperature Tm3 160 ° C. as an example
  • the current limit decay start temperature Tm4 180 ° C. as an example
  • the characteristics Cv2 to Cv4 when the threshold temperature of the detection circuit 316 is set are illustrated. According to the third embodiment, it is not necessary to adjust the temperature characteristics of the diode Ds1 (specifically, size adjustment, impurity concentration adjustment, etc.), and the customizability is greatly improved.
  • FIG. 13 is a diagram showing a switching element 5 according to a modification applicable to the first to third embodiments of the present invention.
  • the anode of the temperature detection diode Ds ⁇ b> 2 built in the switching element 5 is connected to the current sense terminal SE instead of the emitter terminal E of the switching element 5.
  • Parasitic capacitances C Q1 and C Q2 exist between the gate (G) and the emitter (E), and between the gate (G) and the sense emitter (SE). The distance between GE is larger.
  • the capacitance between GE increases.
  • the displacement current 100 (L load noise 100) generated in the diode Ds2 increases when the load current Ic at the time of starting the switching element 5 flows. In this case, the risk of erroneously detecting the temperature increases.
  • the gate (G) -sense emitter (SE) capacitance is smaller than the gate (G) -emitter (E) capacitance. Therefore, by connecting the anode of the diode Ds2 to the sense emitter (SE) side and reducing the capacitance between the gates, it is possible to suppress the occurrence of the L load noise 100 at the time of startup.
  • Embodiment 4 FIG.
  • a logic circuit that performs a logical operation other than the logical product may be provided instead of the AND circuit 12.
  • an OR circuit that calculates a logical sum, a NOR circuit that calculates a negative logical sum, or a NAND circuit that calculates a negative logical product may be provided instead of the AND circuit 12.
  • an OR circuit that calculates a logical sum, a NOR circuit that calculates a negative logical sum, or a NAND circuit that calculates a negative logical product may be provided.
  • NOR circuit that calculates a negative logical product
  • FIG. 16 is a diagram showing a control circuit 501 for a semiconductor switching element according to the fourth embodiment of the present invention.
  • the igniter semiconductor device according to the fourth embodiment has a circuit configuration similar to that of the igniter semiconductor device 10 according to the first embodiment except that the control circuit 1 is replaced with a control circuit 501.
  • the control circuit 501 according to the fourth embodiment is implemented except that the AND circuit 12, the overheat detection circuit 16, and the current detection circuit 18 are replaced with the NOR circuit 512, the overheat detection circuit 516, and the current detection circuit 518, respectively.
  • the same circuit configuration as that of the control circuit 1 according to the first embodiment is provided. Therefore, in the following description, the same or corresponding components as those in the first embodiment will be described with the same reference numerals, and differences from the first embodiment will be mainly described, and common items will be briefly described. Or omitted.
  • FIG. 17 is a time chart for explaining the operation of the control circuit 501.
  • FIG. 18 is a diagram illustrating an example of the overheat detection circuit 516 included in the control circuit 501.
  • the overheat detection circuit 516 shown in FIG. 18 is obtained by deleting the inverter composed of the PMOSFET 49 and the NMOSFET 72 included in the overheat detection circuit 116 shown in FIG.
  • the overheat detection circuit 516 has an output that is high in the normal temperature range, and outputs when an abnormal temperature is detected (that is, when the detected temperature based on the output of the diode Ds2 is equal to or higher than a predetermined set temperature Tm1). Let be low.
  • FIG. 19 is a diagram illustrating an example of the current detection circuit 518 provided in the control circuit 501.
  • a current detection circuit 518 shown in FIG. 19 is obtained by adding an inverter composed of a PMOSFET 49 and an NMOSFET 72 to the output stage of the current detection circuit 118 shown in FIG.
  • the current detection circuit 518 is similar to the current detection circuit 18 in that the current detection signal Enable is generated when the output value of the current sense terminal SE is equal to or greater than the “set current value” described in the first embodiment. However, the current detection circuit 518 outputs a high output if the output value of the current sense terminal SE is less than the set current value, and low as the current detection signal Enable when the output value of the current sense terminal SE is equal to or greater than the set current value. Emit output.
  • the overheat detection circuit 516 emits a low output as the overheat detection signal OT
  • the current detection circuit 518 emits a low output as the current detection signal Enable at the timing indicated by reference numeral X5 in the time chart of FIG.
  • the output signal OUTN of the NOR circuit 512 becomes high. Since the output signal OUTN is input to the set terminal S of the latch circuit 14, the circuit operation similar to that of the first embodiment is realized thereafter.
  • the NOR circuit 512 In the fourth embodiment, the case where the NOR circuit 512 is applied in the circuit configuration of the first embodiment has been described. However, the NOR circuit 512 may be used in the second and third embodiments as well. Various modifications described in the first to third embodiments may be applied to the fourth embodiment. Further, instead of the negative logic circuit 512, a logical circuit that calculates a logical sum (OR logic) or a negative logical product (NAND logic) may be applied. The logic circuit switches the output signal when both the overheat detection signal from the overheat detection circuit and the current detection signal from the current detection circuit are input, so that the circuit operation described in the first to fourth embodiments is realized. In addition, the logic circuit, the subsequent circuit (latch circuit side), and the preceding circuit (overheat detection circuit, current detection circuit) only need to match the logic.
  • the set current value used for the comparison determination of the load current Ic (the set current value Ithen in the embodiment) does not need to be a large current value as in the overcurrent determination.
  • the set current value (set current value Ithen in the embodiment) used for the comparison determination of the load current Ic is a value for determining that a certain amount of current flows through the IGBT 5a as described above, for example, 1A or less. It can be set to a predetermined current value.
  • Control circuit 4 drive circuit, 5 switching element, 5a IGBT, 6 transformer, 6a primary winding, 6b secondary winding, 7 spark plug, 10 igniter semiconductor device, 11 cutoff circuit 12, 312 AND circuit, 13 turn-on delay circuit, 14 latch circuit, 16, 116, 516 overheat detection circuit, 18, 118, 518 current detection circuit, 19 current limit circuit, 20, 181 current voltage conversion circuit (VI Conversion circuit), 22, 183 amplifier, 100 displacement current (L load noise), 101 output fluctuation (temperature erroneous detection), 182 current comparator, 210, 310 attenuation circuit, 316 current limit attenuation start temperature detection circuit, 410 overheat protection circuit 512 NOR circuit, B1 Schmitt trigger times , Ds0 clamp Zener diode, INV1 inverter, Ds1, Ds2 diode, Dz1 Schottky barrier diode, Enable current detection signal, OT overheat detection signal, SE current sense terminal, EST control

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