Classifications

• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B7/00—Generation of oscillations using active element having a negative resistance between two of its electrodes
  • H03B7/12—Generation of oscillations using active element having a negative resistance between two of its electrodes with frequency-determining element comprising distributed inductance and capacitance
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B7/00—Generation of oscillations using active element having a negative resistance between two of its electrodes
  • H03B7/02—Generation of oscillations using active element having a negative resistance between two of its electrodes with frequency-determining element comprising lumped inductance and capacitance
  • H03B7/06—Generation of oscillations using active element having a negative resistance between two of its electrodes with frequency-determining element comprising lumped inductance and capacitance active element being semiconductor device
  • H03B7/08—Generation of oscillations using active element having a negative resistance between two of its electrodes with frequency-determining element comprising lumped inductance and capacitance active element being semiconductor device being a tunnel diode
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
  • H03B9/00—Generation of oscillations using transit-time effects
  • H03B9/12—Generation of oscillations using transit-time effects using solid state devices, e.g. Gunn-effect devices
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H11/00—Networks using active elements
  • H03H11/46—One-port networks
  • H03H11/52—One-port networks simulating negative resistances
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
  • H03H7/01—Frequency selective two-port networks
  • H03H7/0123—Frequency selective two-port networks comprising distributed impedance elements together with lumped impedance elements
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
  • H03H7/01—Frequency selective two-port networks
  • H03H7/17—Structural details of sub-circuits of frequency selective networks
  • H03H7/1741—Comprising typical LC combinations, irrespective of presence and location of additional resistors
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
  • H03H7/38—Impedance-matching networks
  • H03H7/383—Impedance-matching networks comprising distributed impedance elements together with lumped impedance elements

Definitions

• the present invention relates to an oscillation element including a negative resistance element and an oscillator using this oscillation element.
• a negative resistance element involves a resonator and is useful in an application field of an electromagnetic wave oscillation element.
• an electromagnetic wave including at least a part of a frequency band from a millimeter waveband to a terahertz band (which is higher than or equal to 30 GHz and lower than or equal to 30 THz) (hereinafter, will be simply referred to as "terahertz wave”) can be generated by using the negative resistance element.
• terahertz wave an electromagnetic wave including at least a part of a frequency band from a millimeter waveband to a terahertz band (which is higher than or equal to 30 GHz and lower than or equal to 30 THz) (hereinafter, will be simply referred to as "terahertz wave”) can be generated by using the negative resistance element.
• terahertz wave the configurations, NPL 1 discloses a monolithic oscillator obtained by integrating the negative resistance element and the resonator to each other on a substrate. Slot antennas are integrated to each other on
• Fig. 9 illustrates the oscillator according to NPL 1.
• the negative resistance element uses a resonant tunneling diode S-RTD 91 provided with a Schottky barrier on a collector side.
• the resonator is a slot antenna.
• the slot antenna is constituted by a metallic pattern 92 on the semiconductor substrate and provided with capacitances 93 and 94 in end parts.
• NPL 1 is further provided with a rectifier diode 95.
• the rectifier diode 95 functions as a stabilization circuit configured to suppress a parasitic oscillation which becomes a problem in the oscillator using the negative resistance element.
• the parasitic oscillation refers to a parasitic oscillation in a frequency region on a lower frequency side than a desired oscillation frequency f osc . Since the above-described parasitic oscillation decreases an oscillation output in the oscillation frequency fosc the presence of the stabilization circuit may be considered to be important for the oscillator using the negative resistance element.
• the stabilization circuit having a low impedance from the viewpoint of the negative
• resistance element is arranged in a location within 1/4 of a wavelength X osc corresponding to the oscillation frequency fosc towards a power supply side as viewed from the S-RTD 91.
• the shunt rectifier diodes 95 are integrated as the stabilization circuit in a location within X osc /
• a resistance 97 corresponds to a total of an internal resistance of the power supply 96 and a resistance of an electric wire.
• the above-described stabilization circuit in the related art for the suppression of the parasitic oscillation is a circuit configured to attenuate an amplitude of the parasitic oscillation in the low-frequency region where the oscillation is not desired by using a diode or a resistor.
• the oscillation element or the oscillator that oscillates the electromagnetic wave in the millimeter waveband or the terahertz band since the frequency range of the above- described low-frequency region is relatively widened, the amplitude of the parasitic oscillation is to be attenuated across the extremely wide band. For that reason, the design may be complicated, or the oscillation at the desired frequency may also be attenuated.
• an oscillation element that oscillates an electromagnetic wave
• the oscillation element including: a negative resistance element; and a resonator including a first conductor and a second conductor, in which the
• the negative resistance element and the resonator are arranged on a substrate, the negative resistance element is
• the first conductor and the second conductor are capacitively coupled to each other, and when a capacitance between the first conductor and the second conductor is set as C, an inductance of the first conductor and the second conductor is set as Li, a speed of the oscillated
• Fig. 1 is a cross sectional view of an oscillation element according to a first exemplary embodiment.
• Fig. 2A is a schematic diagram of a configuration of a one-conductor resonator in a related art and an
• Fig. 2B is a schematic diagram of a configuration of a two-conductor resonator in the related art and an impedance plot thereof.
• Fig. 2C is a schematic diagram of a configuration of a resonator according to the first exemplary embodiment and an impedance plot thereof.
• Fig. 2D is a Smith chart of the one-conductor resonator in the related art.
• Fig. 2E is a Smith chart of the two-conductor resonator in the related art.
• Fig. 2F is a Smith chart of the resonator according to the first exemplary embodiment.
• Fig. 3 is a schematic diagram of an oscillator using the oscillation element according to the first
• Fig. 4 is a cross sectional view of a part of an oscillation element according to a second exemplary embodiment .
• Fig. 5 is a schematic diagram of an oscillator using the oscillation element according to the second exemplary embodiment.
• Fig. 6A is a schematic diagram of an oscillator according to Example 1.
• Fig. 6B is a cross sectional view of an oscillation element according to Example 1.
• Fig. 7A is a Smith chart as a simulation result of the oscillation element according to Example 1.
• Fig. 7B is an expanded view in the vicinity of 1 GHz in the Smith chart of the oscillation element according to Example 1.
• Fig. 8A is a schematic diagram of an oscillator according to Example 2.
• Fig. 8B is a cross sectional view of an oscillation element according to Example 2.
• Fig. 9 is a schematic diagram of a configuration of an oscillator according to a related art technology.
• An oscillator that oscillates a terahertz wave includes an oscillation element provided with a negative resistance element and a resonator of a distributed constant type, and a power supply that supplies a bias voltage to the negative resistance element.
• the bias voltage from the power supply is supplied to the negative resistance element via a bias supply unit including an electric wire and a conductor.
• a parasitic low-frequency oscillation (parasitic oscillation) in the terahertz wave oscillator is generated in many cases because of a structure accompanied by this bias supply unit. Therefore, the bias supply unit is caused to have a characteristic in an oscillation element 100 according to the present exemplary embodiment. That is, a structure is adopted where a part of the bias supply unit is included in a part of a resonator 110 of a distributed constant type and is also capacitively coupled thereto.
• the bias supply unit is caused to have a characteristic in the oscillation element 100 according to the present exemplary embodiment. That is, the structure is adopted where the part of the bias supply unit is included in the part of the resonator 110 of the distributed constant type and also is capacitively coupled thereto.
• the resonator 110 is a resonator of a two-conductor type in which two conductors 104 and 106 are capacitively coupled to each other. These conductors constitute an integrated
• Fig. 2A is a schematic diagram illustrating a configuration of a resonator 2 constituted by a single conductor and an impedance plot of the resonator 2
• Fig. 2D is a Smith chart of the resonator 2.
• the impedance of the resonator 2 of the distributed constant type which is constituted by the single conductor becomes zero in a DC limit as illustrated in Fig. 2A.
• the Smith chart of Fig. 2D since the impedance of the resonator 2 turns to be clockwise as the frequency is increased, the next resonance point is a lowest order parallel resonance point 11 of the resonator 2.
• the parallel resonance point 11 of the resonator 2 is the oscillation frequency f 0S c itself of the
• Fig. 2B is a schematic diagram illustrating a resonator 3 of the distributed constant type which includes two conductors and an impedance plot of the resonator 3, and Fig. 2E is a Smith chart of the resonator 3.
• an undesired loop 20 of a resonance circuit is generated in the wide band that is higher than or equal to DC and lower than f OSc as illustrated in the Smith chart of Fig. 2E.
• This undesired loop is typically generated by a parasitic capacitance between the two conductors, a junction capacitance specific to the negative resistance element 1, an inductance of a wiring or an electric wire of the bias supply. nit which is not illustrated in the drawing, or the like. Since the loop 20 generally contains a parallel resonance point 21, a
• the bias voltage can be applied at both the end parts of the negative resistance element 1 because of the two conductors having different potentials.
• Fig. 2C illustrates a resonator 4 of the
• Fig. 2C is a schematic diagram illustrating the resonator 4 according to the present exemplary embodiment and an impedance plot of the resonator 4, and Fig. 2F is a Smith chart of the resonator.
• the impedance of the resonator 4 becomes oo in the DC limit but is put into a situation close to the resonator 2 by setting the capacitance C between the two capacitively- coupled conductors —> ⁇ .
• frequency fi a series resonant frequency fi (hereinafter, will be referred to as "frequency fi") formed by the inductance Li of the two conductors and the capacitance C to be sufficiently low.
• the frequency fi is represented by Expression (1) .
• the capacitance C - ⁇ is difficult to establish, but it is possible to further restrict the frequency region where the problematic parasitic oscillation is to be suppressed than the band that is higher than or equal to DC and lower than f OS c- That is, since the phase matching does not occur in the band that is higher than or equal to fi and lower than f OS cr it is possible to narrow the frequency region where the parasitic oscillation is to be suppressed down to the band that is higher than or equal to DC and lower than fi . In the case of the narrow frequency region, it is facilitated to control a loop 22 on the Smith chart of Fig. 2F arid it is simple to delete the resonance point in the band that is higher than or equal to DC and lower than f x . Of course, it is possible to apply the bias voltage at both the end parts of the negative resistance element 1 since the two conductors having the different potentials are arranged.
• the frequency fi is set to be lower than a predetermined frequency that has been set in advance. This can be achieved by adjusting the size of the
• Fig. 1 is a cross sectional view of the oscillation element (semiconductor die) 100.
• the oscillation element 100 includes a negative resistance element 101 (hereinafter, will be referred to as "element 101”) and the resonator 110.
• the element 101 and the resonator 110 are arranged on a substrate 105.
• the resonator 110 includes a first conductor 106 (hereinafter, will be referred to as "conductor 106") including a first conductive layer 102 (hereinafter, will be referred to as “conductive layer 102”) and a second conductive layer 103 (hereinafter, will be referred to as “conductive layer 103"), and also a second conductor 104 (hereinafter, will be
• capacitor 104 provided with a third conductive layer.
• the element 101 are electrically
• the conductors 104 and 106 may include a plurality of
• the conductor 104 and the substrate 105 may be formed of a single conductor each.
• the conductor 104 and the substrate 105 may be
• conductor 104 according to the present exemplary embodiment is constituted by the third conductive layer alone, the conductor 104 may be referred to as the conductive layer 104 in some cases in the following explanation.
• the resonator 110 corresponds to a part where a resonance region is formed by the conductor 104 and the conductor 106 and a part where the conductor 104 and the conductor 106 are arranged opposite each other to be capacitively coupled to each other among the conductor 104 and the conductor 106 arranged on the substrate 105.
• the resonator 110 includes a part where a resonance region 108 in which the electromagnetic wave is resonated is formed and a part where the conductive layer 103 and the conductive layer 104 are arranged opposite each other. For that reason, the conductor 104 also exists on an outer side with respect to the resonator 110. It is noted that a part between the conductor 106 and the substrate 105 and a part between the conductor 104 and the substrate 105 may be hollow or may include a dielectric.
• the substrate 105 is a conductive substrate and is a semiconductor die cut into a necessary and sufficient size for creating the oscillation element 100.
• the substrate 105 is in contact with the conductive layer 104 on the substrate 105 and can apply the bias voltage to the element 101.
• the conductive layer 102 and the conductive layer 103 are electrically short-circuited and connected to each other, and the bias voltage is applied to the other pole of the element 101.
• the conductive layer 102 and the conductive layer 103 can be treated as a continuous conductor having a common potential, and the bias voltage can be supplied to the other pole of the element 101.
• the resonance region 108 is hollow, and the terahertz wave oscillated from the element 101 is resonated.
• the configuration of the resonance region 108 is not limited to the hollow structure, and the oscillator of the
• the resonator 110 has the inductance Li between the conductor 106 and the conductor 104.
• the inductance Li is mainly an inductance derived from a
• the inductance Li may be collectively referred to as "inductance of the
• the conductor 106 and the conductor 104 are identical to each other.
• the resonator 110 according to the present exemplary
• the embodiment includes a relatively large capacitance C between the conductor 106 and the conductor 104. Therefore, the inductance Li and the capacitance C are provided in series between poles of the element 101.
• the typical element 101 oscillates at a parallel resonant frequency (oscillation frequency) f OS c (hereinafter, will be referred to as "frequency f 0S c" ) ⁇
• frequency f 0S c parallel resonant frequency
• the element 101 does not oscillate at the frequency fi defined by the inductance Li of the resonator 110 which is derived from the conductive layer 102 and the capacitance C between the conductive layer 103 and the conductive layer 104.
• the oscillation element 100 does not attenuate the oscillation at the desired frequency f OS c and can suppress the parasitic oscillation in the frequency region that is higher than or equal to fi and lower than f 0S c. This is the same as above in the above- described explanation.
• the frequency fi is set to be lower than the frequency of the parasitic oscillation derived from the wiring.
• the structure accompanied by the bias supply unit corresponding to the cause of the parasitic low-frequency oscillation is roughly divided into two components including an electric wire 132 from a power supply 131 to the oscillation element 100 and a wiring of the oscillation element 100.
• the "wiring" mentioned in the present specification is a conductor passing through a part from the supply of the bias voltage from the power supply 131 via the electric wire 132 to the oscillation element 100 up to the supply to the element 101.
• the wiring refers to the conductors 104 and 106.
• the bias voltage from the power supply 131 is electrically connected to the oscillation element 100 via the electric wire 132 according to the present exemplary embodiment, but the configuration is not limited to this.
• a conductor that electrically connects the power supply 131 to the oscillation element 100 may be provided. That is, the conductor is not limited to the electric wire, and the conductor may have a plate-like shape or the like.
• the electric wire 132 causes the parasitic
• the electric wire 132 is several mm to several m, and the electric wire having a length of several m functions as the oscillator of the distributed constant type at approximately 10 MHz.
• the wiring causes the parasitic oscillation because of the inductance in proportion to the length of the wiring of the oscillation element 100 or the distributed constant circuit.
• the frequency of the parasitic oscillation is set to be higher than or equal to a value obtained by diving the speed C of the electromagnetic wave in the wiring by the length d of the diagonal line of the semiconductor die 105 (C/d) .
• the speed C of the electromagnetic wave in the wiring is slower than the speed C 0 of the electromagnetic wave in vacuum.
• the semiconductor die 105 is set as ⁇ ⁇ , the speed C of the electromagnetic wave in a case where the wiring is located on the surface of the semiconductor die 105 is represented by Expression (2) .
• the frequency fi is preferably appropriately determined to be lower than the resonant frequency of the parasitic distributed constant type that may be generated in the semiconductor die 105.
• the oscillation element 100 has the semiconductor die smaller than or equal to 20 mm x 20 mm
• the distributed constant oscillator at approximately 3 GHz or higher is obtained.
• the oscillation element that oscillates the terahertz wave the oscillation element typically includes the semiconductor die smaller than or equal to 20 mm x 20 mm at a maximum, the frequency fi is preferably set to be lower than 3 GHz.
• the frequency fi is set to be lower than a frequency f 2 , the parasitic oscillation of the latter can be avoided. For that reason, the frequency fi may be
• the capacitance that can be integrated on the same substrate 105 is approximately 100 nF at a maximum.
• the frequency f i is preferably selected to be 100 MHz or higher. According to the present exemplary embodiment, the case in which the frequency f i is set at approximately 1 GHz will be described as an example.
• a method of decreasing the frequency f i includes a method of increasing the capacitance C of the resonator 110. Although it depends on the configuration of the resonator 110 and the shape of the resonance region 108, typically, the frequency f i can be achieved at approximately 1 GHz when the capacitance C is 0.1 nF or higher with respect to the oscillator having the lowest order frequency f osc at 0.1 THz or lower.
• the frequency f i can be achieved at approximately 1 GHz when the capacitance C is 1 nF or higher with respect to the oscillator having the frequency f OS c at 1 THz or lower, and the frequency f i can be achieved at approximately 1 GHz when the capacitance C is 10 nF or higher with respect to the oscillator having the frequency fosc at 10 THz or lower.
• the inductances of the two conductors 104 and 106 that constitute the resonator 110 have an order of an inverse number of the frequency f osc in The International System of Units. That is, the resonator 110 having the frequency f osc at 0.1 THz has the order of 10 pH, and the resonator 110 having the frequency f osc at 1 THz has the order of 1 pH.
• Fig. 3 is an
• the oscillator 150 includes the oscillation element 100 and a power circuit 130 (hereinafter, will be referred to as "circuit 130").
• the circuit 130 includes the power supply 131 that supplies the bias voltage to the oscillation element 100 and the electric wire 132, and the parasitic oscillation may be induced in the frequency region that is higher than or equal to DC and lower than fi , in particular, the frequency region that is higher than or equal to 10 MHz and lower than fi .
• a shunt rectifier diode (shunt element) 133 (hereinafter, will be referred to as "diode 133”) is
• the diode 133 and the electric wire 132 are connected to each other between the power supply 131 and the oscillation element 100.
• the length of the electric wire 132 in the location where the electric wire 132 and the oscillation element 100 are connected to each other and the diode 133 is 7.5 cm or shorter.
• the amplitude of the parasitic oscillation in the frequency region that is higher than or equal to DC and lower than fi can be attenuated.
• the insertion of the diode 133 is equivalent to the decrease in a Q value of a resonance at a parallel resonance point of a loop 120 on the Smith chart in Fig. 2F.
• the loop 120 edges closer to the left side on the same chart.
• the oscillator 150 according to the present exemplary embodiment only the parasitic low- frequency oscillation can be suppressed without attenuating the amplitude of the terahertz wave at the desired frequency f osc .
• the configuration is not limited to the rectifier diode, and a shunt element including a resistor or the like can attain the same effect.
• an oscillator obtained by putting the oscillation element 100 using a short electric wire (not illustrated) having 7.5 cm or shorter which is sufficiently smaller than 1/4 of the
• parasitic oscillation in the frequency region that is higher than or equal to DC and lower than fi is not limited to this, and various methods can be applied to the method.
• oscillation element 100 have been described above. With the oscillation element 100, it is possible to provide the oscillation element and the oscillator in which the
• the oscillation element 100 can suppress the parasitic oscillation in the frequency region that is higher than or equal to the frequency fi and lower than the frequency f osc - That is, when the oscillator is configured by using the oscillation element according to the present exemplary embodiment, since the parasitic
• the band where the amplitude of the parasitic oscillation is to be attenuated is narrowed down by the configuration of the power supply circuit or the like. For that reason, the suppression of the parasitic
• FIG. 4 is a cross sectional view of the oscillation element
• the oscillation element 200 includes a negative resistance element 201 (hereinafter, will be referred to as "element 201”) and a resonator 210.
• the element 201 and the resonator 210 are arranged on a substrate 205.
• resonator 210 includes a first conductor 206 (hereinafter, will be referred to as “conductor 206") which includes a first conductive layer 202 (hereinafter, will be referred to as “conductive layer 202"), and also a second conductive layer 203 (hereinafter, will be referred to as “conductive layer 203") and a second conductor 204 (hereinafter, will be referred to as “conductor 204”) provided with a third conductive layer.
• the element 201 is electrically connected to each of the conductors 204 and 206. It is noted that the conductor 204 may be referred to as the conductive layer 204 in some cases in the following explanation.
• the substrate 205 is in contact with the conductive layer 204.
• One pole of the element 201 is supplied with the bias voltage via the conductor 204.
• the conductive layer 202 and the conductive layer 203 are electrically short- circuited and connected to each other and form the other pole of the element 201. That is, the conductive layers 202 and 203 can be treated as a single conductor having a common potential, and the other pole of the element 201 is supplied with the bias voltage via the conductor 206.
• the resonator 210 is the oscillator of the
• the region 208 may be hollow or may be filled with the dielectric. According to the present exemplary embodiment, among the oscillators of the distributed
• the element 201 sandwiched between the conductive layer 202 and the conductive layer 204 forms an oscillator of a waveguide type which extends in a direction perpendicular to the paper surface of Fig. 4.
• the inductance Li of the oscillator which is derived from the conductive layer 202 and a series resistance R s can be set to be smaller as compared with the first exemplary embodiment without changing the product of a junction capacitance Cd of the element 201 and a negative differential resistance -R d (R d > 0) of the element 201. It is however noted that, in the low-frequency region where the entire structure of the oscillation element can be
• Expression (4) can be obtained by solving a differential equation.
• the typical series resistance R s in the case of the resonator 210 of the waveguide type is typically from approximately 0.1 ⁇ to approximately 1 ⁇ .
• the inductance Li of the resonator 210 is demanded to be 10 "12 H or higher and 10 "13 H or lower, and in a case where the element 201 is the Esaki diode, the inductance Li is demanded to be 10 -11 H or higher and 10 ⁇ 12 H or lower.
• resonator 210 on an inner side as seen from the element 201 takes the following configuration.
• the respective impedances of the conductive layers 202, 203, and 204 alternately repeat the series resonance and the parallel resonance on the frequency axis. According to this, the phase matching does not occur in principle in the band that is higher than or equal to the frequency fi and lower than the frequency f osc of the resonator 210. For that reason, the parasitic oscillation is suppressed in the frequency region that is higher than or equal to fi and lower than f OSc - This is the same as the description made with reference to Figs. 2A to 2F according to the first exemplary embodiment.
• the frequency fi is set to be lower than or equal to 200 MHz. This is because an upper limit of the frequency at which the power supply circuit can be regarded as the lumped constant element instead of the distributed constant circuit is 200 MHz.
• the resonator 210 Since the value of the frequency fi is low, the demanded capacitance C is relatively large. Since the resonator 210 according to the present exemplary embodiment has the structure of the waveguide type which extends in a direction perpendicular to the paper surface of Fig. 4, it is simple to form the relatively large capacitance C between the conductive layer 203 and the conductive layer 204 which are arranged to face each other and are capacitively coupled to each other, and this configuration is preferably adopted according to the present exemplary embodiment. Similarly as in the first exemplary embodiment, since the capacitance that can be integrated on the same substrate 205 is
• Fig. 5 illustrates a configuration of the oscillator 250.
• the oscillator 250 includes the oscillation element 200 and a power supply circuit 230 (hereinafter, will be referred to as "circuit 230") .
• the circuit 230 includes a power supply 231 and an electric wire 232, and the parasitic oscillation may be induced particularly in the frequency region that is higher than or equal to 10 MHz among the frequency region that is higher than or equal to DC and lower than fi.
• the shunt element is connected to the electric wire 232 that connects the oscillation element 200 to the power supply 231 to suppress the parasitic
• the resonator 210 In the wavelength region where the resonator 210 can be considered to be sufficiently small, that is, the low-frequency region where the entire structure can be represented by the lumped constant element, when the
• the series resistance R s of the resonator 210 which is derived from the conductive layer 202 and the inductance Li are respectively replaced by the series resistance R s2 and the inductance L 2 of the electric wire 232.
• the capacitance C can be adjusted by the area of the part where the conductive layer 203 and the conductive layer 204 are arranged opposite each other.
• the negative resistance -R d (R d > 0) can be
• the loop 22 on the Smith chart in Fig. 2F becomes smaller and also edges closer to the left side or the bottom side on the same chart. That is, when the oscillator 250 is configured by using the oscillation element 200, the parasitic oscillation in the frequency region that is higher than or equal to fi and lower than f osc can be suppressed by the oscillation element 200. For that reason, it is
• the oscillation element 100 can suppress the parasitic oscillation in the frequency region that is higher than or equal to the frequency fi and lower than the frequency f osc - Thus, when the oscillator is
• the band where the amplitude of the parasitic oscillation is to be attenuated is narrowed down by the configuration of the power supply circuit or the like. For that reason, the suppression of the parasitic
• FIG. 6A is a schematic diagram of the oscillator 350
• Fig. 6B is a cross sectional view of the
• the oscillator 350 uses resonant tunneling diodes as negative resistance elements 301a and 301b. According to the present example, since two resonators 310 of the
• the resonators 310 each include a first conductor 306 (hereinafter, will be referred to as
• conductor 306 and a second conductor 304 (hereinafter, will be referred to as "conductor 304").
• a first conductive layer 302, a second conductive layer 303, and the third conductive layer (second conductor) 304 are each a metallic layer using Ti/Pd/Au metal having a thickness of 200 nm.
• these conductive layers are respectively referred to as metallic layer (first metallic layer) 302, metallic layer (second metallic layer) 303, and metallic layer (third metallic layer) 304.
• a substrate 305 is a conductive n-InP substrate.
• conductor 304 is arranged on the substrate 305 that is in contact with the metallic layer 304 and supplies the bias voltage to one pole of each of the RTD 301a and 301b.
• a dielectric benzocyclobutene (BCB) 32 is arranged between the metallic layer 302 and the metallic layer 303.
• the metallic layer 302 and the metallic layer 303 are electrically short-circuited and connected to each other in a BCB channel part 315 and form the other pole of the negative resistance element 301.
• the two resonators 310 include regions surrounded by the metallic layers 302, 303, and 304 and are separated from each other by a metal filled in the BCB channel part 315.
• the wavelength ⁇ herein is not a wavelength ⁇ 0 in vacuum but is an effective wavelength on which the
• the metallic layers 302 and 303 that surround the resonant region 308 and the metallic layer 304 have an inverted-F antenna structure.
• the two resonators 310 have symmetrical shapes to each other while the BCB channel part 315 is set as the center. This is because a coupled oscillation is mutually performed at a certain oscillation frequency.
• the oscillators having different shapes may be used as the resonator 310, and the oscillator on one side may be omitted.
• a silicon nitride (SiN) film 33 having a thickness of 100 nm is provided between the metallic layer 303 and the metallic layer 304 to form the capacitance C .
• the frequency fi ( fi 1/ ⁇ 2 ( LiC ) ⁇ ) based on the inductance L x of the resonator 310 which is derived from the metallic layer 302 and the capacitance C between the metallic layer 303 and the metallic layer 304 exists between both poles of the RTD 301a and 301b.
• the frequency fi can be adjusted to be lower than a
• predetermined frequency by adjusting the area where the metallic layer 303 and the metallic layer 304 are overlapped with each other.
• Each of the RTD 301a and 301b is constituted by including a multiquantum well structure based on
• a triple-barrier structure is used herein as the multiquantum well structure. More specifically, the multiquantum well structure is configured by a semiconductor multilayer film structure of AlAs (1.3 nm) /InGaAs (7.6 nm) /InAlAs (2.6
• InGaAs is a well layer, and lattice matching InAlAs and non-matching
• AlAs are barrier layers. These layers are set as undoped layers where carrier doping is not intentionally performed.
• the above-described multiquantum well structure is sandwiched by electric contact layers based on n-InGaAs where an electron concentration is 2 x 10 18 cm -3 .
• a peak current density is approximately 280 kA/cm 2 .
• negative differential resistance region corresponds to a range from approximately 0.7 V to approximately 0.9 V.
• a mesa structure having a diameter of 2 ⁇ is used as the RTD 301a.
• a peak current 10 mA and the negative differential resistance -20 ⁇ are obtained for each of the RTDs 301a and 301b.
• the metallic layer 302 is a resonance circuit for the terahertz wave which uses an inverted-F antenna, and according to the present example, the length 1 ⁇ of the resonator 310 in which the resonant frequency is designed to be at approximately 0.6 THz is 75 ⁇ , and a length in a direction perpendicular to this (hereinafter, will be referred to as vertical direction) is 150 ⁇ .
• a thickness of the BCB 32 is approximately 3 um according to the present example.
• the oscillation frequency fosc is approximately 0.5 THz by being shifted from the resonant frequency in each of the resonance regions 308 by the amount corresponding to the reactances of the RTD 301a and 301b where the influence is increased in the terahertz waveband.
• Each of the above-described dimensions is a design element and can be appropriately changed in
• the area where the metallic layer 303 and the metallic layer 304 are overlapped with each other is set as approximately 0.75 mm x approximately 0.75 mm according to the present example.
• the capacitance C approximately 0.33 nF between the metallic layer 303 and the metallic layer 304 is secured by using the above-described units, the above-described frequency fi becomes approximately 3 GHz.
• the oscillator 350 using an oscillation element 300 can narrow the frequency region where the parasitic oscillation is to be suppressed down to the frequency region that is higher than or equal to DC and lower than 3 GHz.
• the oscillation element 300 can be manufactured by the following manufacturing method. First, a semiconductor multilayer film is subjected to epitaxial growth on the n- InP substrate 305 by a molecular beam epitaxy ( BE) method, a metal organic vapor phase epitaxy (MOVPE) method, or the like. That is, the RTD structure based on n-InP/n-InGaAs, InGaAs/InAlAs and n-InGaAs are sequentially subjected to epitaxial growth. Next, the RTD 301 is etched into a circular mesa shape. Dry etching based on electron beam (EB) lithography and inductive coupled plasma (ICP) is used for the etching. Photolithography may be used instead.
• EB electron beam
• ICP inductive coupled plasma
• the Ti/Pd/Au metallic layer (third metallic layer) 304 is formed on the etched surface by a lift-off method, and a film of the silicon nitride layer 100 nm is formed by using a spattering method.
• the Ti/Pd/Au metallic layer (second metallic layer) 303 is formed by the lift-off method, and the capacitance C is completed.
• the unnecessary silicon nitride layer that does not contribute to the capacitance is removed by etching by using a pattern similar to the second metallic layer 303.
• the first exemplary embodiment may be used for the oscillator 350 in which a power supply circuit 330
• circuit 330 (hereinafter, will be referred to as "circuit 330") is connected to the oscillation element 300.
• a voltage supply at approximately 0.8 V is prepared as a power supply 331 included in the circuit 330 such that the negative
• differential resistance region of the RTD 301a and 301b can be biased at a voltage from approximately 0.7 V to
• An electric wire 332 may be
• the electric wire 332 may be connected to the metallic layer 303 and the metallic layer 304 in the vicinity of the end part of the oscillation element (semiconductor die) .
• a shunt resistor (shunt element) 333 is arranged within approximately 2.5 cm towards the power supply 331 side from the location where the electric wire 332 is connected to the metallic layer 303 or the metallic layer 304. To more simply realize this configuration, an
• the integrated resistor may be provided between the metallic layer 303 and the metallic layer 304 in the connecting location.
• a resistant value of the shunt resistor 333 is lower than or equal to approximately 10 ⁇ that is equal to the absolute value of a combined resistance of the two negative resistance elements 301a and 301b at approximately -10 ⁇ , the parasitic oscillation in the frequency region that is higher than or equal to DC and lower than 3 GHz can be reliably suppressed.
• Fig. 7A and Fig. 7B illustrate simulation
• Fig. 7A is a Smith chart in which the impedance of the oscillator that also serves as a bias supply structure on an outer side with respect to both end parts of the RTD 301a and 301b is plotted from approximately 1 GHz to approximately 1200 GHz.
• Fig. 7B is an expanded view in the vicinity of 1 GHz in the Smith chart of Fig. 7A.
• Fig. 7A and Fig. 7B it may be understood from Fig. 7A and Fig. 7B that an intersecting point of a straight line where the reactance of the Smith chart (imaginary part of the impedance) is zero and a curved line indicating a frequency dependency of the impedance of the resonator 310 is 3 GHz. That is, it may be understood that the series resonance point of the resonator 310 has a frequency of approximately 3 GHz, and the next resonance point is approximately 0.6 THz corresponding to the lowest order parallel resonance point of the resonator 310.
• a high-frequency electromagnetic field simulator HFSS ver. 13 based on a three-dimensional finite element method which is manufactured by Ansys Inc. is used for the
• the frequency range where the parasitic oscillation is to be attenuated can be further narrowed down than the related art technology.
• FIG. 8A is a schematic diagram of the oscillator 450
• Fig. 8B is a cross sectional view of the oscillation element 400.
• a negative resistance element 401 according to the present example is a resonant tunneling diode (hereinafter, will be referred to as "RTD 401") .
• RTD 401 resonant tunneling diode
• Example 1 resides in that a shape of the RTD 401 is not the mesa shape but is a stripe appearance.
• a resonator 410 of a distributed constant type is an oscillator of a gain waveguide type along a longitudinal direction of the stripe shaped element 401
• the electromagnetic wave is distributed as a standing wave as a loop, a node, a loop, and the like in the
• both ends are open ends, when a length of a first metallic layer
• first conductive layer 402 (hereinafter, will be referred to as "metallic layer 402") in the vertical direction is set as 1 3
• a metallic layer 402, a second metallic layer (second conductive layer) 403 (hereinafter, will be referred to as “metallic layer 403”), a third metallic layer (third conductive layer) 404 (hereinafter, will be referred to as “metallic layer 404"), and an n-InP substrate 405 which are the other constituting components are similar to those according to Example 1.
• the RTD 401 sandwiched by the two metallic layers as illustrated in the second exemplary embodiment may be used, and the substrate 405 at that time may not be a lattice matching system with the RTD 401.
• the metallic layer 402 and the metallic layer 403 sandwich a BCB 42 serving as a dielectric and are also electrically short-circuited and connected to each other in the BCB channel parts 415 and 416.
• the wavelength ⁇ is not a wavelength ⁇ 0 in vacuum but is an effective wavelength on which the wavelength shortening effects are received by the effects of a dielectric constant of the BCB 42 and the shapes of the metallic layer 402 and the metallic layer 403.
• the resonator 410 Since the resonator 410 is of the gain waveguide type, the resonator 410 also depends on the semiconductor multilayer film of the RTD 401 in addition to the shapes of the metallic layer 402 and the metallic layer 403 and has a relatively strong wavelength shortening effect. This is because a TM0 mode in the RTD 401, that is, a semi-TEM mode is selected as a waveguide mode.
• the resonator 410 is filled with the BCB 42 and is also closed by the BCB channel parts 415 and 416. Therefore, a TE mode in which a magnetic line has a component H z in the vertical direction is established in the region filled with the BCB 42.
• a distance 1 2 between the RTD 401 and the BCB channel parts 415 and 416 is designed as follows.
• Expression (7) in the side wall region where the BCB 42 is filled.
• a longitudinal direction of the negative resistance element 401 (vertical direction) is set as a Z direction, and a direction intersecting with this is set as a y direction (horizontal direction).
• the component H z in the vertical direction and the component E x are expressed by way of general
• a wave number in the y direction is set as y
• a wave number in the z direction is set as ⁇ ⁇ .
• H z Aexp (jcot + j p z z + j y y) + Bexp ( rot + j p z z - j P y y) ⁇ ⁇ ⁇ (6)
• E x ⁇ ⁇ /(1 ⁇ 2 - ⁇ ⁇ 2 ) x ⁇ Aexp (j t + j p z z + j P y y) + Bexp (j t +
• the frequency fi can be adjusted by adjusting the area where the metallic layer 403 and the metallic layer 404 are overlapped with each other.
• the oscillation element of the gain waveguide type at the oscillation frequency of 0.3 THz is designed such that the length 1 3 of the first metallic layer 402 in the vertical direction is set as approximately 100 um, and the distance 1 2 between the RTD 401 and the BCB channel parts 415 and 416 is set as approximately 35 um. Since the power is supplied from the two locations including the BCB channel parts 415 and 416, the counts of the inductance Li of the resonator 410 which is derived from the metallic layer 402 and the series resistance R s are reduced by half. However, the inductance Li in the order of 10 "11 H remains.
• the design is made such that the area where the second metallic layer 403 and the third metallic layer 404 constituting the resonator 410 are overlapped with each other is adjusted, and the capacitance C can secure, for example, 10 nF.
• a hafnium film 43 having an overlapped area of approximately 0.1 mm x approximately 5 mm and a thickness of approximately 20 nm is used.
• the frequency fi becomes 500 MHz.
• the second exemplary embodiment may be used for the oscillator 450 in which the power supply circuit is
• a power supply 431 prepares a voltage source at approximately 0.8 V such that the negative resistance region of the RTD 401 can be biased to approximately 0.7 V to approximately 0.9 V.
• An electric wire 432 uses a relatively thick electric wire in which the series resistance R s2 is approximately 0.1 ⁇ to set the demand for the inductance L 2 to be 10 ⁇ 8 H or lower. Therefore, the electric wire 432 and the oscillation element 400 may be connected to each other by using wire bonding at approximately 10 ⁇ 9 H or the like, and a plurality of electric wires 432 may of course be used in parallel.
• the frequency range where the parasitic oscillation is to be attenuated can be further narrowed down than the related art technology.
• an imaging system may be constructed by including the oscillation element according to the above- described exemplary embodiments and examples and the
• the oscillator using this oscillation element, and also an image forming apparatus configured to detect the millimeter wave and the terahertz wave as an image. Since the energy of background black-body radiation is small in the frequency region from the millimeter waveband to the terahertz band as being different from an infrared region, active illumination based on the above-described oscillation element and the oscillator is normally used.
• An electronic device including a rectifier element such as a Schottky barrier diode or an FET and a thermal conversion device such as a micro bolometer, a pyrodetector, or a Golay cell may be used as the image forming apparatus using the electromagnetic wave.
• the terahertz wave having information of a subject which is generated from the subject irradiated and transmitted or reflected is obtained by the image forming apparatus.
• the present imaging system corresponds to a focus plane array type, and image pickup by a single shot can be carried out.
• oscillator using this oscillation element is expected to be applied as an illumination unit that performs the active illumination based on the millimeter wave and the terahertz wave which can be used for a manufacturing management, a medical image diagnosis, a safety management, or the like, or a transmission unit of an ultra-fast communication device.

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• 2015
  • 2015-02-09 JP JP2015023663A patent/JP6562645B2/ja active Active
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