WO2011147205A1 - 正交频分复用系统频偏补偿和均衡的方法和装置 - Google Patents
正交频分复用系统频偏补偿和均衡的方法和装置 Download PDFInfo
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- 238000004891 communication Methods 0.000 description 7
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Classifications
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2657—Carrier synchronisation
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/0202—Channel estimation
- H04L25/022—Channel estimation of frequency response
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/0202—Channel estimation
- H04L25/0224—Channel estimation using sounding signals
- H04L25/0228—Channel estimation using sounding signals with direct estimation from sounding signals
- H04L25/023—Channel estimation using sounding signals with direct estimation from sounding signals with extension to other symbols
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2689—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation
- H04L27/2695—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation with channel estimation, e.g. determination of delay spread, derivative or peak tracking
Definitions
- the present invention relates to a mobile communication system, and in particular to a method and apparatus for frequency offset compensation and equalization of an Orthogonal Frequency Division Multiplexing (OFDM) communication system.
- OFDM Orthogonal Frequency Division Multiplexing
- the received signal is affected by the frequency offset.
- the Doppler effect also causes a frequency offset.
- the frequency is too large, the receiver performance is degraded.
- frequency offset compensation is required in the receiver.
- a typical method for frequency offset compensation in OFDM communication systems is to compensate the phase difference of channel estimates corresponding to different single carrier data symbols after channel estimation is completed, and then perform equalization and multi-antenna combining processing.
- this method cannot suppress large inter-carrier interference due to frequency offset, resulting in poor reception performance.
- the technical problem to be solved by the present invention is to provide a frequency offset compensation and equalization method and apparatus, which suppress interference between subcarriers when a frequency offset is large, and improve reception performance.
- the present invention provides a method for frequency offset compensation and equalization of an orthogonal frequency division multiplexing system, including:
- Phase compensation is performed on the channel estimate /2 to obtain a channel estimate at the data symbol / ; and an estimated value of the transmitted symbol X is obtained from M(/) and /.
- N k is a subcarrier identifier, which ranges from 0 to w - 1 , m is the number of subcarriers used for frequency offset estimation, ⁇ ⁇ , F is the subcarrier spacing, and N is an orthogonal frequency division multiplexing (OFDM) symbolic
- FFT Fast Fourier Transform
- the step of obtaining an estimated value of the transmitted signal ⁇ according to M(/) and / comprises:
- the steps of obtaining an estimated value of the transmitted symbol X based on ⁇ (/) and / include:
- the received data symbol, h is the channel estimation value corresponding to the received data symbol on the ith antenna, diag(h;) is a diagonal matrix, and the diagonal element is the channel estimation value h; , ⁇ ⁇ i ⁇ N, N is the number of antennas, is noise, ⁇ 2 is the noise power spectral density, is a constant related to digital modulation, / is an identity matrix, represents the conjugate transpose of the matrix, represents the conjugate rotation of the matrix S Set.
- the step of obtaining an estimated value of the transmitted signal X according to the M(/) and / comprises:
- the channel estimate, ifcg ⁇ .) is a diagonal matrix whose diagonal elements are channel estimates.
- N is the number of antennas
- A/ ⁇ N is the number of antennas
- ⁇ is the noise power Density
- a constant associated with digital modulation a constant associated with digital modulation
- / is an identity matrix, representing the conjugate transpose of the matrix, representing the conjugate transpose of the matrix S.
- the present invention also provides an apparatus for frequency offset compensation and equalization of an orthogonal frequency division multiplexing system, comprising: a channel estimation unit configured to acquire a channel estimation value at a reference symbol/; and a frequency offset value/; frequency offset acquisition a unit, configured to obtain a Toeplitz matrix M(/) according to the frequency offset value/acquisition; a channel compensation unit configured to perform phase compensation on the channel estimation value/2 to obtain a corresponding channel estimation value on each data symbol/;
- An equalization estimating unit is arranged to obtain an estimated value of the transmitted signal X based on M(/) and /.
- k / is the subcarrier identification, and its value ranges from 0 to m - 1
- m is the number of subcarriers used for the frequency offset estimation
- ⁇ is the subcarrier spacing
- N is the FFT point of one OFDM symbol.
- h' h, (2 fAt")
- At is a directional time difference between the data symbol and the reference symbol.
- the equalization estimating unit is further configured to be a single-user single antenna.
- y Get the estimated value of the transmitted symbol ⁇ :
- the equalization estimating unit is further configured to be in the case of a single-user multi-antenna, M(f)diag(h;
- the received data symbol, h is the channel estimation value corresponding to the received data symbol on the ith antenna, diag(h;) is a diagonal matrix, and the diagonal element is the channel estimation value h; , ⁇ ⁇ i ⁇ N, N is the number of antennas, is noise, ⁇ 2 is the noise power spectral density, is a constant related to digital modulation, / is an identity matrix, represents the conjugate transpose of the matrix, represents the conjugate rotation of the matrix S Set.
- the equalization estimating unit is further configured to be in the case of a multi-user multi-antenna, ⁇ ⁇ y 2 + w 2 according to
- the channel estimate, ifcg ⁇ .) is a diagonal matrix whose diagonal elements are channel estimates.
- N is the number of antennas
- A/ ⁇ N is the number of antennas
- ⁇ is the noise power Density
- a constant associated with digital modulation a constant associated with digital modulation
- / is an identity matrix, representing the conjugate transpose of the matrix, representing the conjugate transpose of the matrix S.
- the invention can make the receiver of the communication system more accurately compensate the interference of the frequency offset suppression subcarriers when the frequency deviation of the signal is large, and improve the performance of the receiver when the frequency offset is large.
- 1 is a flow chart of a frequency offset compensation and equalization method of the present invention
- FIG. 2 is a block diagram of a frequency offset compensation and equalization apparatus of the present invention.
- the invention is a method for frequency offset compensation and equalization of data symbols suitable for an orthogonal frequency division multiplexing communication system.
- the invention combines frequency offset compensation, equalization and multi-antenna combining.
- the present invention will be described in detail by taking a 3GPP LTE uplink receiver as an example.
- A is the identifier of the subcarrier allocated to a mobile terminal, which is the transmission signal on the Ath subcarrier, ⁇ is the received signal on the kth subcarrier, and H is the channel estimation value on the first subcarrier.
- ⁇ is the number of fast Fourier transform (FFT) points of an OFDM symbol
- N 2048.
- the interference between the A subcarriers is a noise term, where: m- ⁇ sin [ ⁇ ) ( . N-l) f . l-k
- w is the number of subcarriers used for the frequency offset estimation.
- Equation (1) can be expressed in matrix form as
- the frequency offset compensation and equalization method includes:
- Step 110 Obtain a channel estimation value / 2 and a frequency offset value at the reference symbol /;
- Step U0 according to the frequency offset value / acquisition Toeplitz matrix M (/); M (f) definition see equation (4) and its simplified value method;
- the directional time difference between the reference symbol and the reference symbol, M is less than 0 when the data symbol precedes the reference symbol, and At is greater than 0 when the data symbol is after the reference symbol.
- Step 140 Obtain an estimated value of the transmitted signal X according to M(/) and /;
- the methods for obtaining estimates include:
- ⁇ 2 is the noise power spectral density of digital modulation
- / is an identity matrix.
- ⁇ ⁇ .
- simplification can be based on ⁇ (/)- ⁇ (-/) to obtain an estimate of X:
- B dia g , h ' , , represents the conjugate transpose of matrix S
- ⁇ 2 is the noise power spectral density
- ⁇ is a constant associated with digital modulation
- / is an identity matrix.
- MX 1 is the inverse matrix of ⁇ (/; ⁇ . 2) Single-user multi-antenna case
- X ( ⁇ 2 ⁇ + ⁇ ⁇ ⁇ ) * A H y ( 11 )
- y is converted to represent the conjugate transpose of the matrix, ⁇ 2
- noise power spectral density a constant associated with digital modulation.
- / is an identity matrix.
- ⁇ ⁇ .
- the data symbol received by the first antenna is the data symbol received by the second antenna, and is the data symbol received by the Nth antenna (the typical value of N is 2, 4, 8);
- A is the data symbol transmitted by the first user
- x 2 is the data symbol transmitted by the second user
- x M is the data symbol transmitted by the first user
- / 2 is the frequency offset of the second user
- / M is the frequency offset of the Mth user
- lr NM is the channel estimate corresponding to the data symbol transmitted by the first user received by the Nth antenna
- the minimum mean square error estimate for X is:
- the invention also provides an apparatus for frequency offset compensation and equalization of an orthogonal frequency division multiplexing system, comprising: a channel estimation unit, configured to acquire a channel estimation value/; and a frequency offset value/ at a reference symbol;
- a frequency offset obtaining unit configured to obtain a Toeplitz matrix M(/) according to the frequency offset value/channel
- a channel compensation unit configured to perform phase compensation on the channel estimation value /2 to obtain a channel estimation value corresponding to each data symbol/;
- An equalization estimating unit configured to obtain an estimated value of the transmitted data symbol X according to the M(/) and the /;
- N k is the subcarrier identification, and its value ranges from 0 to w, which is the number of subcarriers used for the frequency offset estimation, ⁇ , F is the subcarrier spacing, and N is the FFT point of one OFDM symbol.
- the diagonal matrix the diagonal element is a channel estimation value at each data symbol /
- the ⁇ is a received data symbol
- noise Represents the conjugate transpose of the matrix ⁇
- ⁇ 2 is the noise power spectral density, which is a constant associated with digital modulation.
- Number, / is an identity matrix
- the equalization estimation unit is in the case of a single user multiple antenna
- M(f)di g ⁇ h ⁇ obtains the estimated value of the transmitted data symbol x:
- the first antenna is connected
- the received data symbol, h is the channel estimation value corresponding to the data symbol received by the ith antenna, ⁇ (/ ⁇ is a diagonal matrix, the diagonal element is the channel estimation value, is noise, and ⁇ 2 is noise Power spectral density, a constant associated with digital modulation, / is an identity matrix, representing the conjugate transpose of the matrix, and ⁇ represents the conjugate transpose of the matrix S.
- the equalization estimation unit is in the case of a multi-user multi-antenna, according to
- y is the data symbol received on the ith antenna
- ⁇ is the channel estimation value corresponding to the data symbol received on the ith antenna
- ifcg ⁇ .) is a diagonal matrix
- the diagonal element is the channel estimation value.
- N is the number of antennas
- A/ ⁇ N is the number of antennas
- ⁇ is the noise power density
- / is an identity matrix, representing the conjugate of the matrix Transpose, indicating the conjugate transpose of matrix S.
- a matrix M(/) is introduced, which suppresses interference between large subcarriers due to frequency offset, and improves reception performance.
- the invention can make the receiver of the communication system compensate the frequency offset more accurately when the frequency deviation of the signal is large, suppress the interference between the subcarriers, and improve the performance of the receiver when the frequency offset is large.
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Description
正交频分复用系统频偏补偿和均衡的方法和装置
技术领域
本发明涉及移动通信系统, 具体涉及一种正交频分复用 ( OFDM )通讯 系统的频偏补偿和均衡的方法和装置。
背景技术
通讯系统的发射机和接收机的晶振没有对准时, 接收信号会受到频偏的 影响。 当无线通讯系统中的移动终端快速移动时, 多普勒效应也会造成频偏。 当频偏较大时, 会降低接收机性能。 为消除频偏的影响需要在接收机中进行 频偏补偿。
目前在 OFDM通讯系统中频偏补偿的一种典型的方法是在信道估计完成 后, 补偿各个不同单载波数据符号对应的信道估计的相位差, 然后再做均衡 和多天线合并处理。 这种方法在频偏较大时, 例如高速列车场景下, 不能抑 制因为频偏导致的较大子载波间干扰, 导致接收性能不佳。
发明内容
本发明要解决的技术问题是提供一种频偏补偿和均衡方法和装置, 抑制 频偏较大时子载波间干扰, 提高接收性能。
为了解决上述问题, 本发明提供了一种正交频分复用系统频偏补偿和均 衡的方法, 包括:
获取参考符号处的信道估计值/;和频偏值 / ;
根据频偏值 /获取 Toeplitz矩阵 M(/);
对信道估计值 /2进行相位补偿得到数据符号处的信道估计值/ ; 以及 根据 M(/)和/ 获取发射符号 X的估计值。
其中, Μ(/)=( ) ,
ak,i 当 ≠射
/|> (:时, akt =0 , (:为预先给定的常数, j,当 = /时
>exp| -jn I ex πγ ~ 7— |,白 A:≠/时
N N
Nsin
N k , /为子载波标识, 其取值范围为 0至 w - 1 , m为频偏估计使用的子载 波个数, γ丄, F为子载波间隔, N为一个正交频分复用 (OFDM)符号的
F
快速傅立叶变换(FFT)点数。
优选地, 上述方法中, h' = h-exp( 2 fAt), At为数据符号和参考符号之间 的有向时间差。 优选地, 上述方法中, 在单用户单天线情况下, 根据 M(/)和/ 获取发射 信号 Λ·的估计值的步骤包括:
χ = {σ2βΙ + ΑΗΑ) λ*ΑΗγ·,
或者 X = (AHA)~1 *AHy;
或者 χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y,
或者 = [BHB)~X *BH *M (-/) * y;
其中, 为对角矩阵, 其对角元素为信道估计值 A', _,为接收到的
数据符号, w为噪声,
表示矩阵 的共轭转置, σ2为噪 声功率谱密度, 为和数字调制相关的一个常数, /为恒等矩阵; Β = άύ '、, 表示矩阵 S的共轭转置。
优选地, 在单用户多天线情况下, 根据 Μ(/)和/ 获取发射符号 X的估计 值的步骤包括:
M{f)diag(h;
M f)diag\h2
根据 x + W = 取发射符号 X的估计值为: yN
M{f)diag(hN χ = (σ2βΙ + ΑΗΑ) *AHy, 或者 x = A Aj * A y; 或者 χ = (σ2βΙ + ΒΗΒ *BH *M(- 或者 = (BHB)~ *BH *M(-f
的接收到的数据符号, h;是第 i根天线上的接收到的数据符号对应的信道估计 值, diag(h; )为对角矩阵, 其对角元素为信道估计值 h; , \ <i<N, N为天线 数, 为噪声, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为 恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。 优选地, 在多用户多天线情况下, 根据所述 M(/)和/ 获取发射信号 X的 估计值的步骤包括:
χ = (σ2βΙ + ΑΗΑ)~1 *AHy,
或者 X = (AHA)~1 *AHy,
或者 Λ Λ:. χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y
或者 = ΒΗΒ *Bh *M (-/) * y;
'Ν,ΐ) M(fM)diag(hNM) diag(h ) diag(] 2) diag iXM)
M(f2) diag(h2 , ) diag(h22 ) di g i2M)
M(f) = B =
M(fM) diag(hNl) diag(hN2) ■■■ diag(hNM) y,为第 i根天线上接收到的数据符号, 】是第 i根天线上接收到的数据符 号对应的信道估计值, ifcg^.)为对角矩阵, 其对角元素为信道估计值 ., 为噪声, 为用户数, N为天线数, A/<N, Κί<Ν, σ2为噪声功率语密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。 本发明还提供了一种正交频分复用系统频偏补偿和均衡的装置, 包括: 信道估计单元, 其设置为获取参考符号处的信道估计值/;和频偏值 /; 频偏获取单元, 其设置为根据所述频偏值 /获取 Toeplitz矩阵 M(/); 信道补偿单元, 其设置为对信道估计值 /2进行相位补偿得到各数据符号 上对应的信道估计值 / ;
均衡估计单元, 其设置为根据 M(/)和/ 获取发射信号 X的估计值。
或者,
k , /为子载波标识, 其取值范围为 0至 m- 1, m为频偏估计使用的子载 波个数, γ丄, F为子载波间隔, N为一个 OFDM符号的 FFT点数。
F
优选地, 上述装置中, h' = h,( 2 fAt"), At为数据符号和参考符号之间 的有向时间差。 优选地, 上述装置中, 均衡估计单元还设置为在单用户单天线情况下,
y获取发射符号 χ的估计值为:
χ = (σ2βΙ + ΑΗΑ)~1 *AHy,
或者 X = (AHA)~1 *AHy
或者 χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y,
或者 = [BHB)~X *Bh *M (-/) * y;
其中, 所述 为对角矩阵, 其对角元素为信道估计值/ , 所述;为 接收到的数据符号, 《为噪声, 所述 =Μ(/μ^(/ ;), 表示矩阵 的共轭 转置, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为恒等矩
阵; B = diag(i') , 表示矩阵 S的共轭转置。
优选地, 上述装置中, 均衡估计单元还设置为在单用户多天线情况下, M(f)diag(h;
M(f)di g t2
根据 x + W = 获取发射符号 X的估计值为: yN
M{f)diag(hN χ = (σ2βΙ + ΑΗΑ) *AHy, 或者 χ = (ΑΗΑ)~ * AHy;
或者 X = (σ2βΙ + ΒΗΒγ *ΒΗ *M{-f)*y, 或者 χ = {ΒΗΒ)~λ *Β Η*Μ(- - )* ';
的接收到的数据符号, h;是第 i根天线上的接收到的数据符号对应的信道估计 值, diag(h; )为对角矩阵, 其对角元素为信道估计值 h; , \ <i<N, N为天线 数, 为噪声, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为 恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。
优选地, 上述装置中, 均衡估计单元还设置为在多用户多天线情况下, ηλ y2 +w2 根据
χ = (σ2βΙ + ΑΗΑ)~1 *AHy,
或者 X = (AHA)~1 *AHy,
或者 χ = (σ2βΙ + ΒΗΒ *BH *M(-f)*y,
或者 Χ = (ΒΗΒ ~ *ΒΗ *M{-f)*y
Λ Λ:. 'Ν,ΐ) ■Ν,Ι ) M(fM)diag(hNM) diag(h ) diag(] 2) di g(^M)
M(f2) diag(h2 , ) diag(h22 ) di g(h2'M)
M(f) = Β =
M(fM) diag(hNl) diag(hN2) ■■■ diag(hNM) y,为第 i根天线上接收到的数据符号, 】是第 i根天线上接收到的数据符 号对应的信道估计值, ifcg^.)为对角矩阵, 其对角元素为信道估计值 ., 为噪声, 为用户数, N为天线数, A/<N, Κί<Ν, σ2为噪声功率语密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。
本发明可以使通讯系统接收机在信号频偏较大时较为准确地补偿频偏 抑制子载波间干扰, 提高接收机在频偏较大时的性能。
附图概述
图 1是本发明频偏补偿和均衡方法流程图;
图 2是本发明频偏补偿和均衡装置框图。
本发明的较佳实施方式
本发明是一种适用于正交频分复用通讯系统的数据符号的频偏补偿和均 衡的方法, 本发明将频偏补偿、 均衡和多天线合并联合起来完成。
以 3GPP LTE上行接收机为例对本发明作详细叙述如下。
A为分配给某个移动终端的子载波的标识, 为第 A个子载波上的发射 信号 , γη为第 k个子载波上的接收信号, H 为第 个子载波上信道估计值。 为归一化 是频偏值,比如, F=15000 赫兹时 ,
Ν为一个 OFDM符号的快速傅立叶变换 (FFT)点数,
15000
在 LTE系统中, 即一个符号的釆样点数, 系统带宽为 20M时, N为 2048,
为其它子载波对第 A个子载波间的干扰, 为噪声项, 其中: m-\ sin [πγ) ( . N-l) f . l-k
: ∑ H(l exp I ]τνγ exp ―] π- (2) l=0,l≠k Ν Ν
Nsin
Ν 其中, w为频偏估计使用的子载波个数。
式( 1 )可以用矩阵形式表示为
y=M(f)diag(X)h + n (3 ) 在参考信号所在符号处
+ «, Ρ为移动终端发射的参考信 号,矩阵 为对角矩阵,其对角元素为参考信号。 Μ(/)为 Toeplitz矩阵, (/;) = ( )的各
为简化计算, 考虑到子载波间干扰主要来自邻近的子载波, 可以按如下 方式取值:如果 μ- /|>C,则 =0,其中常数 (:的三种典型取值分别为 1、 2、 3, 其他情况下取值同式(4) 。
本发明提供的频偏补偿和均衡方法包括:
步骤 110, 获取参考符号处的信道估计值/ 2和频偏值 /;
步骤 U0,根据频偏值 /获取 Toeplitz矩阵 M(/); M(f)的定义参见式( 4 ) 及其简化取值方式;
步骤 130,对信道估计值 进行相位补偿得到各数据符号处的信道估计值 即根据公式 / = /^Χρ( ·2;τ/Δ 得到各个数据符号处的信道估计值 / , At为 数据符号和参考符号之间的有向时间差, 数据符号在参考符号之前时 M小于 0, 数据符号在参考符号之后时 At大于 0。
步骤 140, 根据 M(/)和/ 获取发射信号 X的估计值;
获取 的估计值的方法包括:
1 )单用户单天线情况下,
对应数据符号, 有:
M{f)diag{h')x + n = y (5 ) 其中, ;为接收到的数据符号, 《为噪声。
令 = M ( f)diag(h') , 得到 JC的估计值:
χ = (σ2βΙ + ΑΗΑ) λ*ΑΗγ (6) 或者 = ( )— ^ ( 7 ) 其中, 表示矩阵 的共轭转置, σ2为噪声功率谱密度, 为和数字调 制相关的一个常数。 /为恒等矩阵。 在 QPSK调制时, β = \。
进一步地, 可以基于 Μ(/)— Μ(-/)进行简化, 得到 X的估计值:
χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y (8) 或者 = {ΒΗβγ *Bh *M (-/) * y (9) 其中, B = diag、h'、, 表示矩阵 S的共轭转置, σ2为噪声功率谱密度, β 为和数字调制相关的一个常数, /为恒等矩阵。 M X1是 Μ(/;ΐ的逆矩阵。
2)单用户多天线情况下
对应数据符号, 有:
M{f)diag(h M{f)diag(h2
x + W = ( 10) yN
M{f)diag(hN 其中, ,.为第 根天线上的接收到的数据符号, 是第 根天线上的接收 到的数据符号对应的信道估计, ^为噪声。
M {†)diag{h M{f)diag(h^
X = (σ2βΙ + ΑΗΑ) * AHy ( 11 ) 或者, 也可以使用迫零估计: = μ^4)— ^^ ( 12) 其中, 表示矩阵 的共轭转置, σ2为噪声功率谱密度, 为和数字调 制相关的一个常数。 /为恒等矩阵。 在 QPSK调制时, β = \。
进一步地, 可以基于 Μ —^Λ^-/)进行简化, 由此得到
χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y (14) 或者 JC的迫零估计值: χ = {ΒΗβγ *BH *M(-f)*y (15) 3) 多用户多天线情况下
对应数据符号, 有如下系统方程:
=M(f1)diag(^)x1 +M(/2)i¾?g(/¾2)x2 +〜 + Μ(/Μ)ί¾^(/ζ;Μ
y2
其中, 为第 1根天线接收到的数据符号, 为第 2根天线接收到的数 据符号, 为第 N根天线接收到的数据符号 (N的典型数值为 2、 4、 8) ;
A为第 1个用户发射的数据符号, x2为第 2个用户发射的数据符号, xM 为第 个用户发射的数据符号;
为第 1个用户的频偏 ,/2为第 2个用户的频偏 , /M为第 M个用户的频偏; 为第 1根天线接收到的第 1个用户发射的数据符号对应的信道估计, / ^为第 1根天线接收到的第 2个用户发射的数据符号对应的信道估计, lrNM为 第 N根天线接收到的第 个用户发射的数据符号对应的信道估计,其中 M<N
«,(1 N)为噪声项。
记:
或者, 也可以使用迫零估计: = ( ^4)— ^ (18) 其中, 表示矩阵 的共轭转置, σ2为噪声功率谱密度, 为和数字调 制相关的一个常数。 /为恒等矩阵。 在 QPSK调制时, β = \。
进一步地, 以基于 Μ(/)— ^Λ^-/)进行简化, 由此得到
其中 M
diag{} , ) diag(hx 2 )
diag(h2 ) diag(h 2) diag(hlM)
令 S = diag(hNl) diag(hN2) ■■■ diag(hNM) 由此得到 x的最小均方误差估计值:
χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y (20) 或者 x的迫零估计值: J^^^^—^S^^M (21 ) 本发明还提供一种正交频分复用系统频偏补偿和均衡的装置, 包括: 信道估计单元, 用于获取参考符号处的信道估计值 /;和频偏值 /;
频偏获取单元, 用于根据所述频偏值 /获取 Toeplitz矩阵 M(/); 信道补偿单元, 用于对信道估计值 /2进行相位补偿得到各数据符号上对 应的信道估计值/ ;
均衡估计单元, 用于根据所述 M(/)和所述 / 获取发射的数据符号 X的估 计值;
ak,i =
sin(^ ) l-k\ ( N- exp jTT ~ 当 ≠射
{l-k + \ N N
Nsin
N 或者,
ak,i =
sin(^ ) ( . l-k\ ( . Ν-Ιλ _
!>exp -jn exp jTT ~ 7— |,3A:≠/0T n{l-k + N N
Nsin
N k , /为子载波标识, 其取值范围为 0至 w为频偏估计使用的子载 波个数, γ丄, F为子载波间隔, N为一个 OFDM符号的 FFT点数。
F
其中, 所述 / = /;· exp ( ·2τ/Δ^, 所述 At为所述数据符号和所述参考符号之 间的有向时间差。
其中 , 所述均衡估计单元在单用户单天线情况下 , 根据 M(f)diag(h,)x + n = y, 获取发射的数据符号 x的估计值为:
χ = (σ2βΙ + ΑΗΑ)~1 *AHy,
或者 x = {AHA)~X*AHy,
或者 χ = (σ2βΙ + ΒΗΒ)~1 *BH *M(-f)*y,
或者 = [BHB)~X *Bh *M (-/) * y;
其中, 所述 为对角矩阵, 其对角元素为各数据符号处的信道估计 值/ , 所述 <为接收到的数据符号, 《为噪声,
表 示矩阵 ^的共轭转置, σ2为噪声功率谱密度, 为和数字调制相关的一个常
数, /为恒等矩阵; B = diO 表示矩阵 S的共轭转置,
其中, 所述均衡估计单元在单用户多天线情况下,
M(f)di g^h} 获取发射的数据符号 x的估计值为:
χ = (σ2βΙ + ΑΗΑ 或者 χ = (ΑΗΑ)~ *AHy 或者 X = (σ2βΙ + ΒΗΒγ *ΒΗ *M{-f)*y, 或者 = {ΒΗΒ)~λ *Β
收到的数据符号, h;是第 i根天线接收到的数据符号对应的信道估计值, ·^(/^为对角矩阵, 其对角元素为信道估计值 , 为噪声, σ2为噪声功率 谱密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的 共轭转置, ^表示矩阵 S的共轭转置。
其中, 所述均衡估计单元在多用户多天线情况下, 根据
=M(fl)diag( !ll)xl + M ( ^diagiJ^ +… + M ( ^diagiJ^ Μ、χΜ +ηλ y2 +w2 yN
+ nN 获取发射的数据符号 X的估计值为: χ = {σ2βΙ + ΑΗΑ) λ*ΑΗγ ; 或者
或者 χ = (σ2βΙ + ΒΗΒ)~ *BH *M{-f)*y 或者 x = [BHB)~X *Bh *M (-/) * y;
Mif^diagih' M(f2)diag(h 2) ■■■ M(fM)diag(hM) M(f,)diag(h2' ) M(f2)diag(h2'2) ■■■ M(fM)diag(h2'M)
diag(h ) diag(] 2) … diag(] M )
diag(h2 , ) diag(h22) … diag(h2M )
M(f) = diag(hNl) diag(hN2) ■■■ diag(hNM)
y,为第 i根天线上接收到的数据符号, 】是第 i根天线上接收到的数据符 号对应的信道估计值, ifcg^.)为对角矩阵, 其对角元素为信道估计值 ., 为噪声, 为用户数, N为天线数, A/<N, Κί<Ν, σ2为噪声功率语密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。
本发明所述频偏补偿和均衡方法, 在进行估计时, 引入了矩阵 M(/), 抑 制了因为频偏导致的较大子载波间干扰, 提高了接收性能。
工业实用性
本发明可以使通讯系统接收机在信号频偏较大时较为准确地补偿频偏, 抑制子载波间干扰, 提高接收机在频偏较大时的性能。
Claims
1、 一种正交频分复用系统频偏补偿和均衡的方法, 包括:
获取参考符号处的信道估计值/;和频偏值 /;
根据所述频偏值 /获取 Toeplitz矩阵 M(/);
对所述信道估计值 进行相位补偿得到数据符号处的信道估计值/ ;以及 根据所述 M(/)和/ 获取发射符号 X的估计值。
2、 如权利要求 1所述的方法, 其中,
ak,i =
sin(^ ) l-k N_\
exp ίτνγ- 当 ≠射
N N
Nsin
k, /为子载波标识, 其取值范围为 0至 w为频偏估计使用的子载 波个数, γ丄, F为子载波间隔, N为一个正交频分复用 OFDM符号的快速
F
傅立叶变换 FFT点数。
3、 如权利要求 1所述的方法, 其中, 所述 / = /;· exp( '2;r/At), At为所述数据符号和所述参考符号之间的有向时 间差。
4、 如权利要求 1、 2或 3所述的方法, 其中,
χ = (σ2βΙ + ΑΗΑ)~1 *AHy,
或者
或者
其中, 所述 为对角矩阵, 其对角元素为所述信道估计值/ , 所述 ;为接收到的数据符号, 《为噪声, 所述 =Μ(/μ^(/ ;), 表示矩阵 的 共轭转置, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为恒 等矩阵; B = diag、h' , 表示矩阵 S的共轭转置。
5、 如权利要求 1、 2或 3所述的方法, 其中,
在单用户多天线情况下, 根据所述 M(/)和/ 获取发射信号 X的估计值的 步骤包括:
根据
获取所述发射符号 X的估计值为:
χ = (σ2βΙ + ΑΗΑ)~ *AHy
或者
的接收到的数据符号, h:是第 i根天线接收到的数据符号对应的信道估计值, diag(h; )为对角矩阵, 其对角元素为所述信道估计值 h; , \ <i<N, N为天线 数, 为噪声, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为 恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。
6、 如权利要求 1、 2或 3所述的方法, 其中,
在多用户多天线情况下, 根据所述 M(/)和/ 获取发射信号 X的估计值的 步骤包括:
χ = {σ2βΙ + ΑΗΑ) λ*ΑΗγ; 或者 Χ = (ΑΗΑ)~' *AHy, 或者 x = (σ2βΙ + ΒΗΒ *ΒΗ *M(-f)*y, 或者 Χ = (ΒΗΒ)~ *ΒΗ *Μ{
diag(h ) diag(] 2) diag iXM)
diag(h2 , ) diag(h22 ) di g i2M)
M(f) Β
M(fM) diag(hN ) diag(hN ) di g(hNM) y,为第 根天线上接收到的数据符号, .是第 根天线上接收到的数据符 号对应的信道估计值, ifcg^.)为对角矩阵, 其对角元素为信道估计值 ., 为噪声, 为用户数, N为天线数, A/<N, \ <i<N, σ2为噪声功率语密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。
7、 一种正交频分复用系统频偏补偿和均衡的装置, 包括:
信道估计单元, 其设置为获取参考符号处的信道估计值/;和频偏值 /; 频偏获取单元, 其设置为根据所述频偏值 /获取 Toeplitz矩阵 M(/); 信道补偿单元, 其设置为对所述信道估计值 /2进行相位补偿得到数据符 号处的信道估计值/ ; 以及
均衡估计单元, 其设置为根据所述 M(/)和所述 / 获取发射符号 X的估计 值。
8、 如权利要求 7所述的装置, 其中,
所 = ( ;)
当 ≠射
k, /为子载波标识, 其取值范围为 0至 w为频偏估计使用的子载 波个数, γ丄, F为子载波间隔, N为一个 OFDM符号的 FFT点数。
F
9、 如权利要求 7所述的装置, 其中,
所述 / = /;.eXp( '2;r/At),所述 At为所述数据符号和所述参考符号之间的有 向时间差。
10、 如权利要求 7-9中任一项所述的装置, 其中,
所述均衡估计单元还设置为在单用户单天线情况下根据 M ( f)diag {h') x + n = y获取发射符号 JC的估计值:
或者
或者
其中, 所述 为对角矩阵, 其对角元素为所述信道估计值/ , 所述 ;为接收到的数据符号, 《为噪声, 所述 =Μ(/μ^(/ ;), 表示矩阵 的 共轭转置, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为恒 等矩阵; B = diag、h'、, 表示矩阵 S的共轭转置。
11、 如权利要求 7-9中任一项所述的装置, 其中,
所述均衡估计单元还设置为在单用户多天线情况下, 根据 yN
获取发射符号 Λ·的估计值:
χ = (σ2βΙ + ΑΗΑ)~1 *AHy,
或者 x = (AH Α *AHy 或者 χ = (σ2βΙ + ΒΗΒ)~ *ΒΗ*Μ (- f)*y
或者 χ = --{ΒΗΒ)~] *ΒΗ *Μ
的接收到的数据符号, h:是第 i根天线上的接收到的数据符号对应的信道估计 值, 为对角矩阵, 其对角元素为所述信道估计值/ Ki<N, N为 天线数, 为噪声, σ2为噪声功率谱密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的共轭转置, ^表示矩阵 S的共轭转置。
12、 如权利要求 7-9中任一项所述的装置, 其中,
χ = (σ2βΙ + ΑΗΑ) *ΑΗγ;
或者 χ = (ΑΗλ *AHy, 或者 χ = (σ2βΙ + ΒΗΒ) *BH *M{-f)*y 或者 χ = ΒΗΒ *Βη *Μ (-/) * y;
Λ Λ:. A
'Ν,ΐ) ■Ν,Ι ) M(fM)diag(hNM) diag(h ) diag(] 2) di g(^M)
M(f2) diag(h2 , ) diag(h22 ) di g(h2'M)
M(f) = Β =
M(fM) diag(hNl) diag(hN2) ■■■ diag(hNM) y,为第 i根天线上接收到的数据符号, 】是第 i根天线上接收到的数据符 号对应的信道估计值, ifcg^.)为对角矩阵, 其对角元素为信道估计值 ., 为噪声, 为用户数, N为天线数, A/<N, Κί<Ν, σ2为噪声功率语密度, 为和数字调制相关的一个常数, /为恒等矩阵, 表示矩阵 的共轭转置, 表示矩阵 S的共轭转置。
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- 2011-01-27 US US13/576,814 patent/US8897121B2/en active Active
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CN102263719A (zh) | 2011-11-30 |
EP2515459A1 (en) | 2012-10-24 |
US8897121B2 (en) | 2014-11-25 |
CN102263719B (zh) | 2014-04-09 |
US20120300610A1 (en) | 2012-11-29 |
EP2515459A4 (en) | 2017-05-17 |
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