Classifications

• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M1/00—Details of apparatus for conversion
  • H02M1/40—Means for preventing magnetic saturation
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M3/00—Conversion of dc power input into dc power output
  • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
  • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
  • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
  • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
  • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
  • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M1/00—Details of apparatus for conversion
  • H02M1/0003—Details of control, feedback or regulation circuits
  • H02M1/0006—Arrangements for supplying an adequate voltage to the control circuit of converters

Definitions

• the present invention relates to a switching power supply and, more specifically, relates to the pulse width modulation (PWM) of the switching power supply.
• PWM pulse width modulation
• the PWM controller is an integrated circuit used in the switching power supply to control and regulate the switching duty cycle. Being subject to environmental regulations, the power system design of computers and other electrical products have been required to meet the power management and energy conservation standards.
• the power management is to manage the system to consume power during operation and only very little power is consumed during the non-operation mode. For the power management application in the power supply, how to save the power in light load and no load condition is a main concern.
• One object of the PWM modulator is to optimize the saving of the power consumption and to reduce acoustic noise when the oscillation frequency falls into the audio band.
• FIG. 1 shows a typical flyback power supply circuit, in which a PWM controller 100 controls and regulates the output power.
• the PWM-control starts when the power supply is turned on.
• a capacitor 220 is charged via a resistor 210 until the supply voltage N cc of the PWM controller 100 reaches the start-threshold. Then the PWM controller 100 starts to output PWM signal and drive the entire power supply.
• the supply voltage Ncc is provided from the auxiliary bias winding of a transformer 400 through a rectifier 230.
• a resistor 240 converts the switching current information of the transformer 400 into voltage signal for PWM control and over-power protection.
• the feedback voltage N FB is derived from the output of an optical-coupler 250.
• the input of the optical-coupler 250 is connected to the output of the power supply No through a resistor 290 and a zener diode 280 to form the feedback loop.
• the voltage V FB controls the on-time T O duration of the PWM signal through the PWM controller 100 and dominantly decides the output power.
• the power loss of the power supply is an important concern. Those major losses including the transformer core loss, the transistor switching loss, and the snubber power loss are proportional to the switching frequency F.
• T ON the on-time T ON is required. The saturation of magnetic components such as inductors and transformers may result without the limitation of the T ON duration and will cause over-stress damage to the switching devices such as the transistors and rectifiers.
• an audio noise will be created when the switching frequency falls into the audio band (such as 200Hz - 8KHz) in light load and no load conditions.
• Another object is to reduce the acoustic noise when the switching frequency falls into the audio band in light load and no load conditions.
• REGULATOR EFFICIENCY UNDER LIGHT LOAD (incorporated herein by reference) disclosed an oscillation frequency control circuit capable of varying an oscillation frequency of the oscillator circuit in response to load conditions. Another method is disclosed in U.S. Pat. No. 6,366,070 Bl, "SWITCHING VOLTAGE
• REGULATOR WITH DUAL MODULATION CONTROL SCHEME which disclosed the regulator employs three operation modes, which operate at constant switching frequency for heavy load condition, use dual modulation control scheme for moderate load condition, and entering "pulse-skipping" for light load condition.
• the disadvantage of foregoing prior arts are: (1) Narying the switching frequency without the limitation of maximum on-time may result in saturation of magnetic components and cause over-stress damage to switching devices such as transistors and rectifiers; (2) The modulation of switching frequency is only controlled by the load condition and not correlated with the supply voltage. As the switching frequency is reduced too low for saving more power in light load and no load conditions, the auxiliary bias winding of the transformer or inductor might be unable to provide sufficient power for the supply voltage of the PWM controller.
• the PWM control may work improperly under such condition.
• the switching frequency might be decreased to the audio band. If the magnetic components are not well impregnated, the audio band switching frequency might generate an undesirable acoustic noise.
• the present invention provides an adaptive off-time modulation for saving power and reducing acoustic noise.
• the off-time modulation is achieved by moderating a bias current of an oscillator in the PWM controller.
• the maximum on-time of the PWM signal is kept as a constant. Reducing the bias current increases the off-time of switching period and thus the switching period is extended.
• the feedback voltage derived from the voltage feedback loop and the supply voltage are taken as variables to correlate with the off-time modulation.
• the bias current is modulated as a function of the feedback voltage and the supply voltage.
• a threshold voltage is a constant that defines the level of light load.
• a limit voltage defines the low-level of the supply voltage.
• a first differential signal is generated by subtracting the threshold voltage from the feedback voltage.
• a second differential signal is generated by subtracting the attenuated supply voltage from the limit voltage.
• the sum of the first differential signal and the second differential signal is converted into the bias current.
• the bias current is clamped by a limiter to set up the minimum switching period in normal load and full load conditions. Once the feedback voltage is decreased lower than the threshold voltage, the bias current is reduced and the off-time of the switching period is extended continuously. When the supply voltage is lower than the limit voltage, the bias current is increased and a maximum off-time of the switching period is determined.
• a control circuit provides two entrance voltages.
• a reference resistor transfers the reference current derived from the bias current into a voltage signal to the input of the control circuit.
• a first entrance voltage determines a level of audio switching frequency. As the switching frequency falls into the audio band in light load and no load conditions, the control circuit will output an OFF signal to turn off the oscillator of the PWM controller.
• a second entrance voltage determines a level of starting up the oscillator of the PWM controller. Once the supply voltage is decreased or the feedback voltage is increased and the input voltage of the control circuit is greater than the second entrance voltage, the oscillator will restart working again.
• the adaptive off-time modulation improves the efficiency and saves the power consumption of the power supply in light load and no load conditions.
• the control circuit applied in the present invention turns off the oscillator as the switching frequency falls into the audio band, which greatly reduced the acoustic noise.
• FIG. 1 shows a typical flyback circuit of a power supply
• FIG 2 illustrates a block diagram of a preferred embodiment of an adaptive off-time modulator of the present invention
• FIG. 3 displays a circuit diagram illustrating a preferred embodiment of the oscillator shown in FIG. 2 according to the present invention
• FIG. 4 shows a circuit diagram of a preferred embodiment of the bias current synthesizer shown in FIG. 2 according to the present invention.
• FIG. 2 shows a block diagram of an adaptive off-time modulator according to one embodiment of the present invention.
• a bias current synthesizer 10 generates a bias current I for an oscillator 30 to determine an off-time of the PWM signal.
• a threshold voltage VA is subtracted from a feedback voltage V FB via an adder 11 and generates a first differential signal 11a.
• the output of the adder 11 is connected to the input of a limiter 22.
• a voltage attenuated from the supply voltage Vcc by the attenuator ⁇ is subtracted from a limit voltage Nx via an adder 12 and generates a second differential signal 12a.
• the output of the adder 12 is connected to the input of a limiter 24.
• the outputs of the limiter 22 and the limiter 24 are connected to two inputs of an adder 13 respectively.
• a N-to-I converter 15 transfers the voltage 13a derived from the output of the adder 13 into a current signal 15 a.
• This current signal 15a is clamped by a limiter 26 to create a modulated bias current I M - Reducing the bias current I M extends an oscillation period of the oscillator 30.
• the oscillator 30 outputs a pulse signal Np to drive a S-R register 41 and initiates a PWM cycle.
• the S-R register 41 is reset through a comparator 45 when the current sense input Vs is higher than the feedback voltage V FB -
• the logic low of the pulse signal Vp represents the off-time of the oscillator 30.
• An AND gate 43 ensures the PWM signal output is turned off during the off-time of the oscillator 30.
• the bias current I is a function of the supply voltage Vcc and the feedback voltage V FB -
• the limiter 22 scales the first differential signal 11a by K A and clamps its output into a range of zero to a first-maximum N A .
• the limiter 24 scales the second-differential signal 12a by K B and clamps its output into a range of zero to a second-maximum N ⁇ .
• the limiter 26 clamps the output of the V-to-I converter 15 into a range of zero to a maximum-current I MAX to set up the minimum switching period in normal load and fully load conditions.
• FIG. 3 illustrates one embodiment of the oscillator 30 according to the present invention shown in FIG. 2.
• the pulse signal Np is derived from the output of a ⁇ A ⁇ D gate 35.
• the output of a comparator 31 outputs a logic high signal to the input of a ⁇ A ⁇ D gate 34.
• the output of a comparator 32 outputs a logic low signal to the input of the ⁇ A ⁇ D gate 35.
• the output of the ⁇ A ⁇ D gate 35 remains in logic high.
• the ⁇ A ⁇ D gate 34 outputs a logic low signal and drives a NOT gate 33 to turn on a switch 36.
• the constant current source Ic starts to charge the capacitor C TB -
• the NAND gate 34 outputs a high signal to turn off the switch 36 and enable a switch 37 to discharge the capacitor C TB -
• a MOSFET 39 mirrors a discharge current from the bias current I M flowing through a MOSFET 38. This discharge current determines the off-time of the V P pulse signal.
• adjusting the bias current IM can achieve the off-time modulation of the oscillator 30.
• the on-time of the switching period is controlled through the feedback loop to regulate the output power of the power supply.
• a MOSFET 40 is used to cut off the bias current I M the oscillator 30. As the switching frequency falls into the audio band, the MOSFET 40 will be turned on and stop the bias current from flowing into the oscillator 30.
• the constant current source I c determines the maximum on-time of the switching period. Increasing the switching period by only increasing the off-time duration prevents the magnetic components, such as inductors and transformers, from being saturated. The maximum on-time
• FIG. 4 shows an embodiment of the bias current synthesizer 10 of the PWM controller which includes a constant current source I T , a first current mirror composed of a MOSFET 71 and a MOSFET 73, a second current mirror composed of a MOSFET 81 and a MOSFET 83, a third current mirror composed of a MOSFET 88 and a MOSFET 89, a first buffer amplifier 77, a second buffer amplifier 87, a first operation amplifier (op amp) 76, a first N-to-I MOSFET 75, a second operation amplifier (op amp) 86, a second N-to-I MOSFET 85, a first resistor 72 (R72), a second resistor 82 (R82), an attenuator composed of a third resistor 91 (R91) and a fourth resistor 92 (R92), a first switch 95, a comparator 93, a second switch 96, a reference resistor RF and a NOT gate 94.
• the feedback voltage V FB is connected to the positive input terminal of the first op amp 76.
• the threshold voltage N A is connected to the positive input terminal of the first buffer amplifier 77.
• the output of the first buffer amplifier 77 is connected to the negative input terminal of the first op amp 76 via the first resistor 72.
• the source of the first N-to-I MOSFET 75 is connected to the negative input terminal of the first op amp 76.
• the output of the first op amp 76 is connected to the gate of the first N-to-I MOSFET 75 for developing a first source- follow circuit to drive the first resistor 72.
• the drain of the first N-to-I MOSFET 75 is connected to the drain of the min-or MOSFET 71.
• the drain and the gate of the mirror MOSFET 71 and the gate of the mirror MOSFET 73 are connected together.
• the source of the mirror MOSFET 71 and the source of the mirror MOSFET 73 are connected to the constant current source
• the limit voltage Nx is connected to the positive input terminal of the second op amplifier 86.
• the supply voltage Ncc is connected to the positive input terminal of the second buffer amplifier 87 via the resistor 91.
• the resistor 92 is connected between the positive input terminal of the second buffer amplifier 87 and the ground.
• the output of the second buffer amplifier 87 is connected to the negative input terminal of the second op amplifier 86 via the second resistor 82.
• the source of the second N-to-I MOSFET 85 is connected to the negative input terminal of the second op amp 86.
• the output of the second op amp 86 is connected to the gate of the second N-to-I MOSFET 85 for developing a second source- follow circuit to drive the second resistor 82.
• the drain of the second N-to-I MOSFET 85 is connected to the drain of the mirror MOSFET 81.
• the drain and the gate of the mirror MOSFET 81 and the gate of the MOSFET 83 are connected together.
• the source of the mirror MOSFET 81 and the source of the mirror MOSFET 83 are connected to the constant current source I T .
• the drain of the mirror MOSFET 73 and the mirror MOSFET 83 are connected together and output the bias current I M -
• the threshold voltage N A is subtracted from the feedback voltage V FB in the first op amplifier 76 to generate a first output, which is input to the gate of the first N-to-I MOSFET 75 and converted to a first current l ⁇ .
• the first current mirror mirrors the first current ⁇ ⁇ and outputs a current
• the limit voltage Nx is subtracted from the supply voltage N C c attenuated by the attenuator to generate a second output, which is connected to the gate of the second N-to-I MOSFET 85 and converted to a second current I 2 .
• the second current mirror mirrors the second current I 2 and outputs a current Ivcc through the drain of the mirror MOSFET 83.
• the current I FB is applied to the current Ivcc to form the bias current I M -
• the bias current I M reduces linearly and extends the off-time of the oscillation period of the oscillator 30 shown in FIG. 2.
• the feedback voltage V FB is high and/or the supply voltage Vcc is low, the I M increases gradually.
• I FB W FB - V A )IR 12 - ⁇ x M A (1)
• the range of the I M is clamped as (O ⁇ T M ⁇ T T ) .
• M A is the transfer ratio of the first current mirror;
• M B is the transfer rate of the second current mirror; is equal to [R92 / ( R91 + R92)].
• the minimum off-time of the PWM signal is determined by the constant current source I T as shown in equation (5) and (7)-(9).
• a third current mirror is composed of a mirror MOSFET 88 and a mirror MOSFET 89.
• the gate of the mirror MOSFET 88 is connected to the gate of the mirror MOSFET 71.
• the gate of the mirror MOSFET 89 is connected to the gate of the mirror MOSFET 81.
• the source of the mirror MOSFET 88 and the source of the mirror MOSFET 89 are connected to the constant current source I T .
• the drain of the mirror MOSFET 88 and the drain of the mirror MOSFET 89 are connected together to output a reference current Ip in proportion to the variation of the bias current I M -
• the 4 includes a comparator 93, a first switch 95, a second switch 96, a reference resistor R F and a NOT gate 94.
• the reference resistor Rp is connected to the negative input terminal of the comparator 93.
• the output of the comparator 93 is connected to the control terminal of the second switch 96 and the input terminal of the NOT gate 94.
• the output terminal of the NOT gate 94 is connected to the control terminal of the first switch 95.
• a first entrance voltage V ⁇ is connected to the input terminal of the first switch 95.
• a second entrance voltage NT2 is connected to the input terminal of the second switch 96.
• the outputs of the first switch 95 and the second switch 96 are connected together to the positive input terminal of the comparator 93.
• the output of the comparator 93 is a logic low voltage.
• the NOT gate 94 inverts this logic low voltage and generates a logic high voltage to turn on the first switch 95 that makes the connection of the first entrance voltage V ⁇ to the positive input terminal of the comparator 93.
• the first entrance voltage N ⁇ defines the threshold that the switching frequency falls into the audio band. Once the switching frequency reduces and falls into the audio band due to the light load and no load conditions, the reference current Ip will reduces in proportion to the decrease of the bias current I M -
• the reference resistor R F converts the reference current I F into a reference voltage V F in the negative input terminal of the comparator 93.
• the comparator will output a logic high OFF signal, which turns off the first switch 95 and turns on the second switch 96.
• This OFF signal is connected to the gate of a MOSFET 40 shown in FIG. 3.
• the source of the MOSFET 40 is connected to the ground.
• the drain of the MOSFET 40 is connected to the drain of the mirror MOSFET 38.
• the logic high OFF signal turns on the MOSFET 40 and cuts off the bias current of the oscillator.
• the reference current I F will increase in proportion to the bias current I M -
• the second entrance voltage VT2 defines the threshold to restart the oscillation of the oscillator 30.
• the comparator 93 will output a logic low voltage to turn off the MOSFET 40 and the second switch 96.
• the NOT gate 94 inverts the low logic voltage to turn on the first switch 95 and make the comiection of the first entrance voltage N ⁇ to the positive input terminal of the comparator 93 again.
• the logic low OFF signal makes the bias current I M flow into the oscillator 30 again.
• the oscillator 30 will restart the oscillation according to the variation of the load conditions.
• the PWM controller including an adaptive off-time modulation of the present invention can reduce the power consumption for the power supply in light load and no load conditions. Furthermore, the acoustic noise will be greatly reduced when the PWM frequency falls into the audio band in light load and no load conditions by the control of the control circuit.

Landscapes

• Engineering & Computer Science (AREA)
• Power Engineering (AREA)
• Dc-Dc Converters (AREA)

Priority Applications (2)

Applications Claiming Priority (1)

Publications (1)

Family

ID=35150297

Family Applications (1)

Country Status (2)

Cited By (3)

Families Citing this family (11)

Citations (4)

• 2004
  • 2004-04-13 WO PCT/CN2004/000341 patent/WO2005101624A1/en active Application Filing
  • 2004-04-13 CN CN2004800427247A patent/CN1938924B/zh not_active Expired - Fee Related

Patent Citations (4)

Also Published As

Similar Documents

Legal Events