Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/10—Frequency-modulated carrier systems, i.e. using frequency-shift keying
  • H04L27/14—Demodulator circuits; Receiver circuits
  • H04L27/144—Demodulator circuits; Receiver circuits with demodulation using spectral properties of the received signal, e.g. by using frequency selective- or frequency sensitive elements
  • H04L27/152—Demodulator circuits; Receiver circuits with demodulation using spectral properties of the received signal, e.g. by using frequency selective- or frequency sensitive elements using controlled oscillators, e.g. PLL arrangements

Definitions

• the invention is directed to a method and a system for carrier frequency recovery, especially for TFM signal receiver structures.
• Continuous phase modulation is a class of modulation techniques, which are efficient in power and bandwith at the same time.
• the RF signal envelope is constant and phase varies in a continuous manner.
• the constant envelope allows the use of nonlinear amplifiers, simplifying receiver and transmitter design.
• Tamed frequency modulation is one continuous phase modulation scheme.
• One of the main benefits of TFM is the very narrow bandwith needed in comparison with most of the other modulation schemes.
• CPM signals can be described by
• ⁇ 1, ⁇ 3, ...., +(M-1), /*,. is the modulation index, which may vary from interval to interval
• q(t) is the phase response function
• g(t) is the frequency response
• E b is the bit energy
• bit interval is the modulation index, which may vary from interval to interval
• q(t) is the phase response function
• g(t) is the frequency response
• E b is the bit energy
• /o is the carrier frequency
• ⁇ P is an arbitrary initial phase.
• M 2
• h t 0.5
• the bit period has the same length as a symbol period.
• Characteristic for TFM modulation is, that the phase shift of the modulated carrier over one bit interval is determined not only by the current bit but by three consecutive input binary signals in accordance with the encoding rule:
• phase changes of ⁇ /2 are obtained, if three successive bits have the same polarity, and the phase remains constant for three bits of alternating polarity.
• Phase changes of ⁇ /4 are connected with the bit configurations ++ -, + — , -++ and — +.
• the signal space diagram of TFM is shown in Figure 1.
• TMF modulation is described further in the article entitled “Tamed Frequency Modulation, A Novel Method to Achieve Spectrum Economy in Digital Transmission”, Fank de Jager and Cornelis B. Dekker, IEEE Trans.on Comm. Vol. COM-26, NO. 5, May 1978, pp. 534-542.
• CPM modulation is described further in the book “Digital Phase Modulation” by John B. Anderson, Tor Aulin and Carl-Erik Sundberg, Plenum Publishing Corporation, 233 Spring Street, New York, N.Y. 10013, on pages 15 to 53.
• Direct conversion receivers are receivers, which do not use intermediate frequencies for filtering and detection of received signals.
• the received RF signal is mixed with a local oscillator signal, whose frequency corresponds to the carrier frequency of the RF signal.
• Direct conversion receivers have many advantages. For example, the bandwith filtering can be performed at low i.e. audio frequencies, allowing realisation of very narrow bandwiths with sharp edges. Also, intermediate frequency filters are not needed.
• direct conversion receivers have not been used for receiving TFM signals due to an inherent problem of direct conversion receiver structures, namely the existence of a DC offset at the mixer output as a result of imperfections of the mixer structure.
• the DC offset results from leaking of the local oscillator signal to the RF port of the mixer and subsequent mixing of the leaked signal with the local oscillator signal itself.
• the random fluctuations in phase of the leakage signal result in a relatively slowly and randomly changing DC offset signal.
• Low frequency phase noise resulting from transmitter phase noise or local oscillator phase noise can be presented as an example.
• the DC offset problem can be thought of as a very low frequency phase noise.
• Figure 2 illustrates the problem created by the DC offset in the detection of TFM modulated data. Without any DC offset, the received signal has the constellation shown with white circles.
• the vectors Si and s 2 drawn with a thick dashed line, represent signals detected at two consecutive sampling times. Without any DC offset, the detection of vectors Si and s 2 is straightforward. However, DC offset changes the situation considerably. DC offset has the effect of moving the signal constellation in the IQ-diagram, as depicted by the black circles.
• the vectors s' ⁇ and s' 2 show the corresponding sampled signal vectors in the presence of DC offset.
• any detector optimized for detecting the vectors Si and s 2 has difficulties in recognizing vectors s and s' 2 . Therefore, the presence of DC offset results easily in a high detection error rate.
• W094/28662 describes a double differential detector structure, which is described in the following. If r n is defined as the signal vector at time nTb, the difference vector between two received vectors over one bit interval is defined as
• ⁇ n Z(A n ) - Z(A n _ 1 ) (5)
• the quantity ⁇ n is not a sufficient measure to make a detection, since the difference vector A n can be zero.
• the phase does not change with certain information bit patterns. Therefore, the quantity ⁇ n can also be zero for some bit patterns, which makes it impossible to make decisions about the bits based on the value of the quantity ⁇ n .
• Carrier frequency offset is the difference between the transmitter frequency and the receiver local oscillator frequency.
• the carrier error information can be extracted from the phase difference between two adjacent received signal vectors. If there is no frequency offset and the received signal vector at a time instant nT b is denoted as r roast , the phase difference over one bit interval can be expressed as
• ⁇ n r ⁇ + ⁇ - r n (6)
• a ⁇ n A ⁇ n + 2 ⁇ AfT b (7)
• the frequency offset information can be extracted by
• An object of the invention is to realize a method for carrier frequency recovery, which allows carrier frequency recovery during reception of a tamed frequency modulated signal in the presence of DC offset or other forms of low frequency noise. It is also an object of the invention to realize a system for carrier frequency recovery, which is able to perform carrier frequency recovery during reception of a tamed frequency modulated signal in the presence of DC offset or other forms of low frequency noise.
• the objects are reached by comparing the length of a sample difference vector against the length of a difference vector resulting from the decision of data and by observing the rotation direction of the sample difference vector, and determining the carrier frequency error substantially on the basis of said comparison and said observation.
• the carrier frequency recovery method according to the invention is characterized by that, which is specified in the characterizing part of the independent method claim directed to a carrier frequency recovery method.
• the data transmission method according to the invention is characterized by that, which is specified in the characterizing part of the independent method claim directed to a data transmission method.
• the carrier frequency recovery system according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a carrier frequency recovery system.
• the mobile communication means according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a mobile communication means.
• the base station according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a base station.
• the radio link system according to the invention is characterized by that, which is specified in the characterizing part of the independent claim directed to a radio link system.
• the length of a sample difference vector is compared against the length of a difference vector resulting from the decision of data.
• the result of the comparison is combined with an observation of the direction of the sample difference vector, a result corresponding to the carrier frequency error is obtained.
• the difference vector over one bit interval is free of the effect of DC offset. Any further information extracted from the difference vector will also free of the effect of DC offset.
• carrier error information can be extracted from
• a constant error in frequency is equivalent to a continuously increasing phase error.
• direction factor which has the value of 1 if the received signal vector changes from r M _, to r n clockwise i.e. if the received signal vector corresponds to clockwise rotation, and a value of -1 if the received signal vector changes from r n _ to r n counter-clockwise i.e. if the received signal vector corresponds to counter-clockwise rotation.
• the carrier error function be expressed as
• ⁇ n Z(A ) - Z(A l hole- ⁇ ) (13) where ⁇ n is the phase difference between two differential vectors ⁇ 1 * and ⁇ i .
• the phase change over one bit interval is determined by three consecutive transmitted bits. Further, the quantity ⁇ n is determined by four consecutive information bits.
• the direction factor ⁇ can be determined on the basis of the detected data and the quantity ⁇ n , which will be shown later in this specification.
• Figure 3 illustrates the possible state changes resulting from two consecutive symbols. Since according to equation (3) the phase change of one symbol can be 0, ⁇ /4, or ⁇ /2, the concatenation of two symbols result in a phase change of either one of ⁇ 0, ⁇ /4, ⁇ /2, 3 ⁇ /4, ⁇ or one of ⁇ 0, - ⁇ /4, - ⁇ /2, -3 ⁇ /4, - ⁇ . Therefore, the difference of two consecutive signal vectors may be one of the vectors shown in Figure 3.
• Figure 3 illustrates the case of ⁇ 0, ⁇ /4, ⁇ /2, 3 ⁇ /4, ⁇ .
• the length of the difference vector may be one of z, s, lj, l 2 , or l 3 .
• Length z refers to a zero differential vector
• length 5 to a differential vector from one to a neighbouring constellation point
• length lj refers to the length of a differential vector bypassing one constellation point on the constellation point circle
• length l 2 refers to the length of a differential vector bypassing two constellation points on the constellation point circle
• length l 3 refers to the length of a differential vector bypassing three constellation points on the constellation point circle.
• bit sequence 011 results in a change of phase of ⁇ /4. This change of phase corresponds to the difference vector with length of s in Figure 3, resulting in a value of s for the quantity Similarly, the bit sequence 110 results again in a change of phase of ⁇ /4, resulting in a value of s for the quantity ⁇ ! caravan .
• ⁇ ! M is then a vector A 2 n having a length of lj, as equation (7) shows.
• the difference of the angles of two consecutive difference vectors is then ⁇ /4.
• the differential vectors corresponding to bit pattern 0010 and have the same values as the corresponding differential vectors corresponding to bit pattern 1101, as can be seen from the previous table. Therefore, based on the differential vector lengths alone, the complement bit patterns cannot be distinguished from each other at the receiver. This problem can be alleviated using a differential encoding.
• differential encoding two consecutive bits having the same value correspond to a "0" bit, and two consecutive bits having differing values correspond to a "1" bit.
• the differentially encoded bit patterns 0010 and 1101 correspond to the same decoded bit pattern, Oil.
• any bit patterns which are complements of each other produce the same differentially decoded bit patterns.
• the length of a differential vector provides enough data for detecting the received symbols.
• the data can be detected on the basis of for example the quantity ⁇ 2 consult , using a Viterbi algorithm and the the 4-state trellis of
• the value of the weight of the correct path should be close to zero, and ideally would be zero. Therefore, the correct path can be found in the Viterbi algorithm by finding the path with the smallest accumulated weight.
• these weights can be formulated in many other ways as well. For example, one can formulate the weights in such a way, that the weight of a particular path has a maximum, when the path is the correct path. In that case, a maximum of the cumulative sum of the path weights would be sought in the Viterbi algorithm.
• the exponent k may be adjusted for obtaining an optimum performance.
• the value of k may be for example 1, in which case the weights are the absolute value of the difference between the length of the differential vector between received vectors over two bit intervals and the ideal length of the vector corresponding to the path in question.
• the value of k may further be for example 2, in which case the weights are the square of the absolute value of the difference between the length of the differential vector between received vectors over two bit intervals and the ideal length of the vector corresponding to the path in question.
• Other values of k are also possible, and the optimum value can be found out, for example, by simulation or experiment.
• ⁇ n is used in determination of ⁇ .
• the difference signal vectors are directed in the clockwise direction and ⁇ n ⁇ 0.
• bit patterns 0011 and 1100 the difference vectors have opposing directions, and are advantageously not used in the determination of the direction factor.
• the value of ⁇ n is undefined, since one or both of »n-l and ⁇ ! are zero vectors.
• the following table is obtained from the previously shown table by reorganization of the rows, by adding columns listing the data bits dn corresponding to the transmitted differentially encoded bit patterns bn, and by adding a column for the values of direction factor we wish to obtain.
• ⁇ d n _ x denotes the complement of d n _
• bit d n _ 2 can be used to obtain the decision of ⁇ . In those cases, where the direction factor is nonzero and the length of ⁇ naut is /;, the value of bit d n _ 2 is zero. In those cases, where the direction factor is nonzero and the length of Aont is s, the value of bit d n _ 2 is one. Therefore, using n-1 instead of n in the carrier error function (12), the value of 'n-1
• This embodiment is advantageous, since the hardware implementation of the computation of the cross product is simpler than the computation of ⁇ increment .
• Figure 1 illustrates the symbol constellation of tamed frequency modulation
• FIG. 3 illustrates the difference vectors calculated over two bit intervals
• Figure 4 illustrates the 4-state trellis used in an advantageous embodiment of the invention
• FIG. 5 illustrates a block diagram of an advantageous embodiment of the invention
• Figure 6 illustrates a block diagram of a further advantageous embodiment of the invention
• Figure 7 illustrates a block diagram of a mobile communication means according to an advantageous embodiment of the invention.
• FIG. 8 illustrates further advantageous embodiments of the invention.
• FIG. 5 illustrates a block diagram of a receiver structure according to an advantageous embodiment of the invention.
• a receiver structure can be used, for example, in mobile communication means and in base stations of cellular mobile communication networks.
• the receiver structure comprises a sampling part 200 and a carrier frequency recovery part 500.
• the sampling part receives the RF signal, and converts the received signal to baseband I and Q signals with mixers 210a, 210b.
• a local oscillator 205 provides the signal to be mixed with the received RF signal, and a phase shifter 206 produces a 90° phase shift to the local oscillator signal taken to the mixer 210b.
• the local oscillator 205 receives a control signal from the carrier frequency recovery part 500.
• the output signals of the mixers 210a and 210b are taken to matched filters 215a, 215b, whose filtering properties are optimized for filtering TFM signals.
• the output signals from the filters 215a, 215b is taken to switch elements 225, which produce samples of the quadrature I and Q signals. Each corresponding pair of I and Q samples define a signal sample vector.
• the switch elements 225 are controlled by a sampling oscillator 220.
• the sampling part 200 is an example of a structure, which can be used in a receiver structure according to the invention. As a man skilled in the art knows, many other structures can be used to produce a sampled, downconverted signal. The invention is not limited to using the sampling part structure 200 shown in Figure 5.
• the switch elements 225 performing the sampling comprise typically A/D converters or other types of sampling means.
• the resulting I and Q samples are taken to the carrier frequency recovery part 500 and a demodulator part 300.
• the signals are first taken to a delay block 515a, 515b and an adding block 520a, 520b.
• the delay blocks 515a, 515b delay the I and Q samples for the period of one T b , and the delayed samples are subtracted from the not delayed samples in the adding blocks 520a, 520b, whereby the difference of consecutive I and Q samples is obtained.
• the differences are taken to a difference vector length calculating block 522 and a difference vector angle calculating block 523.
• the calculating blocks 522 and 523 may, for example, utilize look-up tables for obtaining the length and angle values corresponding to each pair of I and Q difference samples.
• the angle values are taken to a delay block 525 and an adding block 530, which calculate the difference ⁇ n between successive angle values.
• the resulting angle difference values ⁇ n are subsequently delayed in a delay block 535b for a time, which corresponds to the delay in the demodulator part 300, in order to synchronize the arrival times of corresponding angle difference values ⁇ n and demodulated data bits from the demodulator part.
• the length values are delayed in a delay blocks 535a for the same reason.
• the length values and the angle difference values are taken to a carrier error calculation block 540, which receives the demodulated data bits as well from the demodulator part 300.
• the carrier error calculation block 540 calculates a carrier error function such as, for example, the carrier error function defined by equation 17.
• the carrier error function values are taken to smoothing and oscillator control block 545, which smoothes the carrier error values and controls the oscillator of the receiver according to the smoothed result.
• the smoothing is performed by averaging within an averaging window, i.e. by averaging a certain predefined number of the latest values, such as for example the latest 512 or 1024 values.
• the output of the block 545 is taken to the control input of the local oscillator 205 of the receiver structure.
• the demodulating part 300 may in other embodiments of the invention take its input values also from the output of the adding blocks 520a, 520b, in which case the demodulating part 300 needs to be able to demodulate on the basis of difference samples. This can be performed for example on the basis of the length of the difference vectors as described previously.
• Figure 6 shows another advantageous embodiment of the invention.
• the structure of Figure 6 corresponds to the embodiment of equation 19, in which a cross product of ⁇ dad_ ! and ⁇ flower is used in obtaining the direction factor ⁇ .
• a similar sampling part 200 as in the embodiment of Figure 5 is used.
• a cross product calculation block 570 is used instead of angle calculation block 523 as in Figure 6.
• the cross product calculation block 570 may comprise two delay blocks
• the delay blocks 550a, 550b produce the delayed I and Q difference samples ⁇ ' «- ⁇ and , while the two multiplication blocks 555a,555b and the adding block 560 perform the multiplications and the subtraction requred for forming the cross product ⁇ Vi ⁇ q n - ⁇ 9 »- ⁇ - Acute .
• the result values of the cross product calculation are subsequently delayed in a delay block 535b for a time, which corresponds to the delay in the demodulator part 300, in order to synchronize the arrival times of the cross product values and demodulated data bits from the demodulator part.
• Fig. 7 shows a block diagram of a digital mobile communication means according to an advantageous embodiment of the invention.
• the mobile communication means comprises a microphone 301, keyboard 307, display 306, earpiece 314, antenna duplexer or switch 308, antenna 309 and a control unit 305, which all are typical components of conventional mobile communication means.
• the mobile communication means contains typical transmission and receiver blocks 304, 311.
• Transmission block 304 comprises functionality necessary for speech and channel coding, encryption, and modulation, and the necessary RF circuitry for amplification of the signal for transmission.
• Receiver block 311 comprises the necessary amplifier circuits and functionality necessary for demodulating and decryption of the signal, and removing channel and speech coding.
• the signal produced by the microphone 301 is amplified in the amplifier stage 302 and converted to digital form in the A/D converter 303, whereafter the the signal is taken to the transmitter block 304.
• the transmitter block encodes the digital signal and produces the modulated and amplified RF-signal, whereafter the RF signal is taken to the antenna 309 via the duplexer or switch 308.
• the receiver block 311 demodulates the received signal and removes the encryption and channel coding.
• the resulting speech signal is converted to analog form in the D/A converter 312, the output signal of which is amplified in the amplifier stage 313, whereafter the amplified signal is taken to the earpiece 314.
• the control unit 305 controls the functions of the mobile communication means, reads the commands given by the user via the keypad 307 and displays messages to the user via the display 307.
• the mobile communication means comprises a carrier frequency recovery part 500 performing carrier frequency recovery according to the invention.
• the carrier frequency recovery part 500 advantageously has the structure shown in Figure 5. However, other structures according to the invention for carrier frequency recovery can also be used.
• Figure 8 shows an example of an embodiment of the invention.
• a carrier frequency recovery part 500 according to the invention is used in at least some base stations 360 of a mobile communication network for demodulating TFM signals received from the mobile communication means 350.
• Figure 8 further illustrates a base station controller 370 controlling the base stations 360 and two radio link units 371 for connecting the base station controller 370 to the rest of the mobile communication network 380.
• Figure 8 also illustrates a further advantageous embodiment of the invention, namely the use of carrier frequency recovery parts 500 according to the invention in radio links.
• the demodulation method according to the invention is very suitable to be used in continuous high speed communications, where a continuous demodulation is needed. High speed radio links are one example of such an advantageous application of the invention.
• the inventive carrier error recovery method can also be used in conventional TFM receiver structures, in which the amount of DC offset and low frequency noise is not disturbingly large.
• the difference of consecutive received sample vectors is used for carrier error recovery, also the difference over two consecutive bit intervals may be used for obtaining carrier error recovery.
• sample vector refers to a pair of corresponding I and Q signal samples.

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• Physics & Mathematics (AREA)
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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
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• Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

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Citations (3)

• 1998
  • 1998-05-29 FI FI981227A patent/FI106501B/sv active
• 1999
  • 1999-05-27 AU AU45169/99A patent/AU4516999A/en not_active Abandoned
  • 1999-05-27 WO PCT/FI1999/000455 patent/WO1999063720A1/en active Application Filing

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