US4680526A - Method of controlling inverter-driven induction motor - Google Patents

Method of controlling inverter-driven induction motor Download PDF

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US4680526A
US4680526A US06/766,945 US76694585A US4680526A US 4680526 A US4680526 A US 4680526A US 76694585 A US76694585 A US 76694585A US 4680526 A US4680526 A US 4680526A
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Prior art keywords
inverter
induction motor
current
voltage
command
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Toshiaki Okuyama
Takayuki Matsui
Noboru Fujimoto
Yuzuru Kubota
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Hitachi Ltd
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Hitachi Ltd
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Priority claimed from JP59173713A external-priority patent/JPS6152176A/ja
Priority claimed from JP60072875A external-priority patent/JPH0687678B2/ja
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Assigned to HITACHI, LTD., 6, KANDA, SURUGADAI 4-CHOME, CHIYODA, KU, TOKYO, JAPAN, A CORP OF JAPAN reassignment HITACHI, LTD., 6, KANDA, SURUGADAI 4-CHOME, CHIYODA, KU, TOKYO, JAPAN, A CORP OF JAPAN ASSIGNMENT OF ASSIGNORS INTEREST. Assignors: FUJIMOTO, NOBORU, KUBOTA, YUZURU, MATSUI, TAKAYUKI, OKUYAMA, TOSHIAKI
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/04—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for very low speeds

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  • the present invention relates to a control method for an induction motor, wherein a primary current thereof is resolved into a torque current component and an exciting current component which are separately controlled independent of each other for controlling relational speed of the induction motor.
  • a vector control method In connection with the control of an induction motor, a vector control method has heretofore been known in which the primary current of the motor is resolved into a torque component and an excitation component which are separately controlled independent of each other.
  • a speed control of the induction motor can be accomplished with a high speed response as in the case of a direct current motor.
  • the vector control method for the induction motor may be generally classified into a magnetic flux detection type vector control and a slip frequency control type vector control method.
  • the former type control method is disclosed, for example, in U.S. Pat. No. 3,824,437 (corresponding to West-German Laid-Open (DOS) No. 1941312).
  • DOS West-German Laid-Open
  • a typical example of the latter type is disclosed in Japanese Patent Application Laid-Open No. 11125/1976.
  • a magnetic flux detector has to be incorporated in the induction motor. Consequently, this type control method can not be employed in an induction motor designed for general purpose.
  • the slip frequency control type vector control method has attracted attention in the recent years and is actually adopted in practical applications.
  • the slip frequency control type vector control method for the induction motor is based on the control of the output frequency of an inverter unit in dependence on the rotational speed of the motor. Consequently, signal conductors or cables are required for the output of a speed detector (or angular position detector) as well as for interconnection between the speed detector and the inverter unit, thus involving a complicated configuration of the system and troublesome procedure for application of this type of vector control method to the existing induction motor.
  • An object of the present invention is to provide a control method for an induction motor which allows the rotational speed of the induction motor to be controlled with an improved accuracy without using detectors such as speed detectors voltage detectors or the like.
  • Another object of the invention is to provide a control method for an induction motor which can assure highly accurate control of the induction motor by preventing control performance characteristics from being degraded due to internal voltage drop in a voltage-type inverter driving the induction motor and by suppressing occurrences of ripple in torque.
  • the output voltage and frequency of an inverter are controlled in dependence on a component of the primary current of an induction motor driven by the inverter, which component is in phase with a reference phase of the output voltage of the inverter, and that the rotational speed or frequency of the induction motor is controlled in accordance with a frequency command which in turn is determined from the aforementioned current component.
  • a voltage drop in the inverter circuit is arithmetically determined on the basis of an inverter output current detection signal, wherein the output voltage command is correctively modified on the basis of the result of the arithmethic determination.
  • FIG. 1 is a circuit diagram showing a configuration of a control circuit for a PWM (pulse width modulation) inverter system according to an exemplary embodiment of the invention
  • FIG. 2a is a view illustrating a general equivalent circuit of an induction motor
  • FIG. 2b is a view illustrating an equivalent circuit of the induction motor in which primary leakage reactance is eliminated;
  • FIG. 3 is a view showing graphically characteristic relationship between a slip frequency and a current for illustrating the control operation
  • FIG. 4a-4f is a view showing signal waveforms for illustrating operation of the PWM inverter
  • FIG. 5 is a schematic circuit diagram for illustrating the working principle underlying a method of measuring internal voltage drop of the inverter according to the invention
  • FIG. 6 is a view for illustrating graphically a characteristic relation between the internal voltage drop of the inverter and a preset command value of a function generator
  • FIGS. 7, 8, 9, 10 and 11 are circuit diagrams showing, respectively, circuit arrangements of the control circuit according to other embodiments of the invention.
  • FIGS. 12 and 13 are views for illustrating in flow charts processings executed in the control circuit shown in FIG. 11.
  • FIG. 1 shows a PWM (pulse width modulation) type control system for an induction motor according to an embodiment of the present invention.
  • a numeral 1 denotes a PWM inverter for converting a DC voltage to an AC voltage of a variable frequency.
  • the inverter 1 is composed of self-extinction elements interconnected in Graetz connection, each of the self-extinction elements having a feedback diode connected in antiparallel therewith.
  • a solid state switching element such as a transistor or gate-turn-off (GTO) thyristor may be mentioned employed.
  • An induction motor 2 is connected to AC output terminals of phases U, V and W of the inverter 1.
  • the primary currents i u , i v , i w of U-, V- and W-phases of the induction motor 2 are detected by current detectors 3U, 3V and 3W, respectively.
  • a reference numeral 6 denotes a speed command circuit for producing a speed command signal ⁇ r * which is applied to an adder 7 and compared with an estimated speed signal ⁇ r supplied from an adder 39, resulting in a deviation between the signals ⁇ r * and ⁇ r which is outputted from the adder 7.
  • a speed deviation amplifier 9 correspondingly outputs a command signal i q * for a torque current component i q of the motor current, as described hereinafter.
  • the command signal i q * is applied to an adder 11 and a coefficient multiplier 29.
  • the adder 11 determines a deviation between the command signal i q * and the torque component signal i q derived through a coordinate transformation circuit 32 (described hereinafter in more detail).
  • the deviation signal called outputted from the adder 11 is supplied to a current deviation amplifier 35 which in turn produces as the output signal a frequency control signal ⁇ , in correspondence with the deviation signal.
  • the signal ⁇ is applied to an adder 36 where the signal ⁇ is added with an output signal ⁇ r ** supplied from a first order delay circuit 37 (a signal having a time lag of first order relative to the speed command signal ⁇ r *), whereby a frequency indicating command signal ⁇ 1 * is produced by the adder 36.
  • An oscillator 12 produces a sinusoidal signal having a frequency proportional to the aforementioned speed command signal ⁇ i * and a constant amplitude (this signal serves as a phase reference signal for the inverter output voltage).
  • the sinusoidal signal output from the oscillator 12 is supplied to coordinate transformation circuits 17, 18 and 32, respectively.
  • the current detection signals i u , i v and i w are supplied to a three-phase/two-phase conversion circuit 31, whereby the three phase signals i u , i v and i w are converted to two phase signals i.sub. ⁇ and i.sub. ⁇ .
  • These signals i.sub. ⁇ and i.sub. ⁇ are supplied to the coordinate transformation circuit 32 which transforms the signals i.sub. ⁇ and i.sub. ⁇ to a component signal i q which is in phase with the latter, respectively, on the basis of the sinusoidal signal supplied from the oscillator 12.
  • the signal i q is applied to the aforementioned adder 11 and a function generator 38 which generates a slip frequency signal ⁇ s (of an estimated value) on the basis of the input signal i q .
  • the estimated slip frequency signal ⁇ s is supplied to an adder 39 to be subtracted from the aforementioned frequency command signal ⁇ 1 *, whereby the estimated value ⁇ r of rotational speed of the induction motor 2 is outputted from the adder 39 to be subsequently applied to the adder 7 mentioned above.
  • a reference numeral 13 denotes an exciting current command circuit which produces as the output thereof a motor exciting current command i d * which is then supplied to an arithmetic unit 14 for determining the induced electromotive force (also referred to as EMF arithmetic unit) and to a coefficient multiplier 40 which functions to multiply the input signal i d * with a factor K to output a signal K.i d * which in turn is supplied to an adder 28.
  • a coefficient circuit 29 serves to multiply the torque current command signal i q * with the factor K to produce an output signal K.i q * which is then applied to an adder 30 to be added with the output signal V q * of the EMF arithmetic unit 14, resulting in an updated voltage command signal V q ** to be subsequently applied to an adder 15.
  • Signals ⁇ V d * and ⁇ V q * applied to the address 15 and 28, respectively, will be described hereinafter. It should be mentioned here that the output signal V d *** from the adder 28 and the output signal V q *** of the adder 15 are applied to the input of the coordinate transformation circuit 18 which serves to transform quantities in a rotating magnetic field coordinate system (or i.e.
  • stator coordinate system i.e. stator reference frame
  • sinusoidal signal produced by the oscillator 12 as the result of which two-phase AC signals V.sub. ⁇ * and V.sub. ⁇ * in the stator reference frame are derived from the output of the coordinate transformation circuit 18.
  • the signals V.sub. ⁇ * and V.sub. ⁇ * are then supplied to a two-phase/three-phase converter 19, whereby voltage command signals V u *, V v * and V w * are drived from the phase number converter 19, which signal have a frequency, proportional to the frequency command signal ⁇ 1 * and being mutually phase-shifted by 120°.
  • the signals V u *, V v * and V w * are then applied to a switch circuit 20.
  • the output signals of the switch circuit 20 are applied to comparators 24U, 24V and 24W, respectively, to be compared with a carrier signal for pulse width modulation as supplied from an oscillator 25, whereby pulse width modulating or PWM pulse signals for turning on and off the associated switching elements of the PWM inverter 1 are produced, respectively.
  • a gate circuit 26 served to supply gate signals to the switching elements of the PWM inverter 1 in response to the output pulse signals produced by the comparators 24U, 24V and 24W, respectively.
  • circuit components which play roles in compensating for the voltage drop occurring in the voltage type inverter 1.
  • Function generators 4U, 4V and 4W produce signals ⁇ V u *, ⁇ V v * and ⁇ V w * which are proportional to voltage drops in the inverter and motor windings as well as wirlng conductors, in dependence on magnitude and polarities of the output signals from the current detectors 3U, 3V and 3W.
  • These signals ⁇ V u *, ⁇ V v * and ⁇ V w * are applied to a three-phase/two-phase converter 16 to be converted to two phase signals ⁇ V.sub. ⁇ * and ⁇ V.sub. ⁇ * which are then supplied to the coordinate transformation circuit 17 where the signals ⁇ V.sub. ⁇ * and ⁇ V.sub. ⁇ * are transformed on the basis of the sinusoidal signal produced by the oscillator 12 to component signals ⁇ V d * and ⁇ V q * which are, respectively, delayed by 90° and in phase with the induced eletromotive force.
  • the signal ⁇ V q * outputted from the circuit 17 is applied to the adder 15 to be added with the signal V q ** outputted from the adder 30, whereby the updated voltage command signal V q *** is produced by the adder 15.
  • the other output signal ⁇ V d * of the coordinate transformation circuit 17 is applied to the adder 28 to be added with the signal K.i d * to create a voltage command signal V d ***.
  • the circuit configuration described above serves to compensate for the voltage drop in the inverter 1.
  • FIG. 2a shows a general equivalent circuit of an induction motor. Since ⁇ can be selected arbitrarily, the primary leakage reactance can be made zero by correspondingly selecting the value of ⁇ , whereby an equivalent circuit shown in FIG. 2b is resulted. In connection with the equivalent circuits.
  • R 2 ' secondary resistance (after conversion to equivalence),
  • R 1 primary resistance
  • the primary current I 1 and torque T are functions only of the slip angular frequency ⁇ s . Accordingly, the slip angular frequency ⁇ s and the primary current I 1 can be definitely determined as a function of the torque T, which in turn means that the torque T and the primary current I 1 can be controlled in dependence on the slip angular frequency ⁇ s .
  • the condition that the exciting current I 0 ' be constant can be realized by controlling the voltage magnitude
  • the oscillator 12 generates a sinusoidal signal (providing phase reference for the induced electromotive force) having a frequency proportional to the frequency command signal ⁇ 1 *.
  • a voltage command calculator 42 of the circuit configuration shown as enclosed in a broken line block in FIG. 1 the frequency command ⁇ 1 * is multiplied with the aforementioned sinusoidal signal to create an electromotive force command e 1 * having magnitude and frequency proportional to the frequency command ⁇ 1 *.
  • the voltage drops R i i 1 due to the primary resistance are derived from the detected current signal i 1 (i u , i v i w ) and added to the electromotive force command e 1 * to create the primary voltage command V 1 * (V u *, V v *, V w *) in the form of three-phase sinusoidal signal.
  • d-axis with the axis orthogonal to the d-axis termed q-axis
  • d- and q-axis components of the primary current and the primary voltage are represented by i d , i q and V d , V q , respectively.
  • the oscillator 12 generates two phase sinusoidal signals of a frequency proportional to the frequency command signal outputted from the adder 36.
  • the two phase sinusoidal signals are out of phase by 90° with each other and thus given by cos ⁇ 1 t and sin ⁇ 1 t.
  • the former serves as the phase reference signal for the excitation component of U-phase, while the latter serves as the phase reference signal for the induced electromotive force of U-phase.
  • the voltage command signals of three phases V u *, V v * and v w * are derived in accordance with the following expressions: ##EQU4##
  • V u *, V v * and V w * as derived can be expressed as follows:
  • V u terminal voltage of U-phase
  • the primary voltage command V 1 * is compared with the carrier signal for performing the pulse width control well known in the art, whereby the output voltage of the PWM inverter 1 is so controlled that the instantaneous value of the output voltage is in proportion to the primary voltage command V 1 *.
  • the motor voltage is controlled in accordance with the expression (6) to maintain the exciting current
  • torque T bears a proportional relation to the current component i q , thus allowing the current component i q to be considered equivalent to torque.
  • the current component i q is detected by the current component detecting circuit designated by a broken line block 43 in FIG. 1 in accordance with
  • i U , i V , i W instantaneous values of U-, V- and W-phase currents
  • the current component i q thus detected is compared with the torque command T* ( ⁇ i q *) at the adder 11, wherein the resultant deviation is utilized for adjusting the slip angular frequency (output frequency of the inverter) ⁇ s in accordance with the frequency control signal ⁇ which depends on the deviation, whereby torque is proportionally controlled in dependence on the torque command T*.
  • torque can be so controlled as to follow the torque command T* with high response speed.
  • the slip angular frequency ⁇ s bears relation given by the expression (18) to the current component i q . Accordingly, the slip angular frequency ⁇ s can be determined on the basis of the current component i q . This determination is accomplished by the function generator 38.
  • a typical example of the ⁇ s -i q characteristic is illustrating FIG. 3. So far as the rated torque is not exceeded, the current component i q is approximately proportional to the slip angular frequency ⁇ s . In that case, the function generator 38 may be spared.
  • FIG. 3 further illustrates the characteristic of the primary current
  • the current component i q approximately coincides with the primary current
  • can be determined from the magnitude of the primary current detection signal i 1 .
  • the estimated rotational speed (calculated value) ⁇ r can be arithmetically determined in accordance with
  • the speed deviation amplifier 9 produces the torque command signal T* ( ⁇ i q *) which depends on the deviation between the speed command signal ⁇ r * and the estimated value ⁇ r thereof, wherein torque is controlled in proportion to the torque command T*, to perform the speed control.
  • T* torque command signal
  • the estimated value ⁇ r of rotational speed is arithmetically determined by subtracting the slip angular frequency ⁇ s from the primary frequency ⁇ 1 , decreasing of the rotational speed due to slip which is peculiar to the induction motor can be correctively compensated for. Accordingly, the speed control can be accomplished with higher accuracy when compared with the hitherto known V/F control technique.
  • the first order delay circuit 37 serves to vary the frequency command ⁇ 1 * in accompaniment to change of the speed command ⁇ r * so that the frequency control signal ⁇ mentioned hereinbefore is meaningful as the command for the slip angular frequency ⁇ s in the steady state.
  • Each of the switching elements which constitute essential parts of the inverter 1 can be turned on or off only with a time lag or delay. Accordingly, in order to prevent a short-circuit from occurring between the bridge arms of positive and negative polarities, the gate signal associated with the PWM pulse is imparted with on-delay time t d as indicated by broken lines in FIG. 4 at (b) and (c).
  • the on-delay time t d is usually set at a value which is twice or thrice as great as the turn-off time of the switching element.
  • the U-phase current will then flow through the diodes connected in anti-parallel with the switching elements of the negative arm during the on-delay period. Consequently, the U-phase potential is minum (-) during the on-delay period.
  • the V-phase current of negative polarity will flow through the anti-parallel diodes of the positive arm during the on-delay period. Consequently, the V-phase potential is plus (+) during the on-delay period.
  • the line voltage (V U-N-V V-N ) between U- and V-phases is of a waveform indicated by broken lines and hatched areas at (d) in FIG. 4 and is apparently smaller than the solid-line waveform depicted on the assumption that the on-delay time is neglected.
  • the turn-off time of the switching element varies in dependence on magnitude of the current flowing through the switching element (i.e. the output current of the inverter 1). Accordingly, time taken for the switching elements of the positive and negative arms to be turned off simultaneously during the on-delay period varies as a function of magnitude of the output current of the inverter 1. In other words, magnitude of the hatched portion of the waveform indicated by broken line in FIG. 4 at (d) for which the on-delay time is taken into consideration varies in dependence on magnitude of the output current of the inverter 1.
  • the voltage drop due to the on-delay phenomenon occurs every on/off period of the switching element. So far as the PWM switching frequency is sufficiently high when compared with the output frequency of the inverter, the voltage drop (fundamental wave component) is in phase with the output current of the inverter. In this respect, the voltage drop under consideration behaves like a voltage drop produced across an ohmic resistor.
  • the magnitude of the voltage drop bears non-linear relation to the inverter output current. This characteristic can be measured in the manner mentioned below.
  • FIG. 5 illustrates an equivalent circuit in which the motor is supplied with DC currents from the inverter.
  • E represents the aforementioned voltage drop in the inverter
  • R represent resistances of the motor windings and interconnecting or wiring conductors between the inverter and the motor.
  • the voltage drops E and R.i dc can be determined on the basis of the voltage command V dc *.
  • the resistance R is previously known, the internal voltage drop E of the inverter can be handled separately from the voltage drop R.i dc due to the resistances of the motor windings and the interconnecting conductors.
  • An example of the result of the measurement is graphically illustrated in FIG. 6. As will be seen in the figure, magnitude of the voltage drop is in a non-linear relation to the output current. Consequently, in case the inverter output current is an alternating current, the voltage drop can contain harmonic components.
  • the internal voltage drop E of the inverter and the voltage drop R.i due to resistances of the motor windings and the wiring conductors are previously measured, wherein the characteristic data determined through the measurement are stored in memories incorporated in the function generators 4U, 4V and 4W, respectively, so that the data can be read out in accordance with the inverter output currents for thereby compensating the voltage drops E and R.i associated with the output currents, respectively.
  • This voltage-drop compensation can be accomplished by the circuitry composed of the function generators 4U, 4V and 4W, the three-phase/two-phase converter 16, the coordinate transformation circuit 17 and the adders 15 and 28.
  • the voltage drop signals ⁇ V U *, ⁇ V V * and ⁇ V W * are converted to two phase signals ⁇ V.sub. ⁇ * and ⁇ V.sub. ⁇ * which are then transformed to the quantities ⁇ V d * and ⁇ V q * pertaining to rotating field coordinate system to be subsequently applied to the adders 15 and 28 for modifying correctively the voltage command signals V d * and V q * to V d *** and V q *** which are the voltage commands modified to compensate for the voltage drops.
  • the induced electromotive force of the induction motor can be controlled so as to coincide with the voltage command V d * and V q * by compensating for the actual voltage drops with the aid of the corresponding estimated quantities stored as mentioned above.
  • K.i represents the quantity relating to the output current as mentioned above. It should however be noted that the quantity under consideration need not necessarily be proportional to the current i but may bear any other relation to the latter, to a similar effect.
  • the aforementioned voltage drop E is inherent to the inverter, the voltage drop R.i may vary in dependence on the motor connected actually to the inverter as well as the resistance of the wiring conductors as employed.
  • the contents of the function generators 4U, 4V, 4W are to be set in consideration of both the constant voltage drop E and the variable voltage drop Ri. However, this procedure may be troublesome. In view of this, description will be made of a method of automatically measuring these voltage drops and automatically setting the contents or data to be loaded in the function generators.
  • the above metnod can be carried out with the aid of the circuitry including the DC voltage command circuit 21, the coefficient multipliers 22U, 22V and 22W, the switch circuit 20 and a meter 5.
  • the switch circuit 20 is changed over to the position designated by a. In this state, DC current i dc will flow from the inverter.
  • a characteristic like the one shown in FIG. 6 can be obtained. This characteristic is set at the function generators 4U, 4V and 4W.
  • the switch circuit 20 Upon actual operation of the motor, the switch circuit 20 is changed over to the position denoted by b to thereby allow the output voltage to be controlled in dependence on the output signals V U *, V V * and V W * of the two-phase/three-phase converter 19, whereby the voltage drops in concern are compensated for by the signals ⁇ V U *, ⁇ V V * and ⁇ V W * outputted from the function generators 4U, 4V and 4W, as described hereinbefore. Additionally, in order to suppress the DC component of the inverter output current, the quantity relating to the latter may be subtracted from the result of the aforementioned measurement so that the more appropriate characteristic data can be placed in the function generators.
  • FIG. 7 shows another embodiment of the invention which differs from the system shown in FIG. 1 in that the exciting current i is detected to control the inverter output voltage in accordance with deviation between the detected exciting current i d and the associated command value i d *.
  • the exciting current i d can be maintained constant by controlling the ratio
  • the exciting current may undergo variations transiently upon abrupt acceleration or deceleration. It is therefore proposed according to the instant embodiment of the invention that the exciting current i d is detected and that the amplitude of the primary voltage command V 1 * is correctively controlled so that the exciting current i d assumes constantly a predetermined value.
  • a reference numeral 43 denotes an exciting current component detector for detecting a component i d of the exciting current which is out of phase by 90° relative to the phase reference of the inverter output voltage (induced electromotive force), 33 denotes an adder or adding point for deriving the deviation between the detected exciting current component i d and the relevant command value i d *, 34 denotes an exciting current deviation amplifier for amplifying the deviation, and 42 denotes a multiplier for multiplying the output signal of the amplifier 34 with the frequency command ⁇ 1 * for outputting a signal for setting the amplitude of the primary, voltage command V 1 *.
  • Other circuit components the same as or equivalent to those shown in FIG. 1 are denoted by like reference numerals and further description of these components is omitted.
  • the current component detector 43 In operation, the current component detector 43, indicated as enclosed by a broken line in FIG. 7 detects the exciting current component i d in accordance with
  • the deviation or difference between the exciting current command value i d * and the exciting current i d as detected is amplified through the exciting current deviation amplifier 34 to be subsequently multiplied with the frequency calculator indicated generally broken line block 42 in FIG. 7 to prepare an amplitude setting signal for the primary voltage command V 1 *.
  • the voltage command calculator 42 multiplies the amplitude setting signal with a phase reference signal for the induced electromotive force supplied from the oscillator 12 to produce voltage commands V 1 * for U-, V- and W-phases, respectively.
  • Other operation of the system shown in FIG. 7 is same as described hereinbefore in conjunction with FIG. 1. Because the exciting current i d is controlled to coincide constantly with the exciting current command value i d * in this way, it is possible to suppress excursion or fluctuation of the exciting current i d .
  • FIG. 8 shows still another embodiment of the invention which differs from the one shown in FIG. 7 in that the d-axis component of the inverter output voltage is controlled in dependence on difference or deviation between the exciting current i d as detected and the associated command values i d *.
  • the exciting current i d can be certainly maintained constant by correctively controlling the amplitude of the primary voltage command V 1 * so that the exciting current i d as detected assumes constantly a predetermined value.
  • the exciting current i d and the torque current i q undergo fluxuation in response to change in the load, bringing about mutual interference which tends to degrade the control characteristic.
  • the embodiment shown in FIG. 8 is so configured that the exciting current i d as detected is controlled to assume a predetermined value constantly, as described below.
  • the exciting current command circuit 13 outputs the exciting current command i d * for the motor.
  • This signal i d * is applied to the induced EMF arithmetic unit 14, the adder 33 and the coefficient multiplier 40.
  • the adder 33 outputs the deviation between the exciting current command i d * and the detected exciting current i d , the deviation signal being applied to the current deviation amplifier 34 which in turn outputs a voltage command signal V d * which depends on the current deviation mentioned above.
  • This signal V d * is then applied to the adder 41.
  • the coefficient multiplier 40 multiplies the command signal i d * with a coefficient k.
  • the output signal k.i d * of the coefficient multiplier 40 is applied to the adder 41 where the former is added with the output signal V d * of the current deviation amplifier 34, resulting in an updated voltage command signal V d ** being outputted, which is then applied to the adder 28.
  • the deviation between the exciting current command i d * issued by the command circuit 13 and the detected exciting current i d is amplified through the amplifier 34 to derive the voltage command V d *.
  • the d-axis voltage component V d i.e. voltage component orthogonal to the induced electromotive force
  • V d * is controlled as a function of the voltage command V d * so that the detected exciting current i d coincides with the exciting current command i d *.
  • the slip frequency ⁇ s is estimated on the basis of the signal i q by the slip calculator 38.
  • the estimated slip signal ⁇ s is subtracted from the frequency command signal ⁇ * by the adder 39 to detect the estimated speed signal ⁇ r .
  • Deviation between the speed command signal ⁇ r * and the estimated speed signal ⁇ r is amplified by the amplifier 9 to derive the torque current command i q *.
  • deviation between the torque current command i q * and the detected torque current i q is amplified by the amplifier 35, whereby the frequency command ⁇ 1 * is determined on the basis of the output deviation signal ⁇ of the amplifier 35.
  • the frequency command ⁇ 1 is increased, which causes the slip frequency to increase, which in turn causes the torque current i q to be increased.
  • the above description also holds true in the reverse operation of what has been described above.
  • the torque current i q is controlled so as to coincide with the torque current command i q *.
  • the signals ⁇ 1 * and i d * are multiplied with each other to derive the voltage command V q * which provides a basis for controlling the q-axis voltage component V q (i.e. voltage component in phase with the induced electromotive force).
  • the ratio between the induced electromotive force and the frequency is controlled to be constant (i.e. flux is constant).
  • control of the induction motor can be accomplished in the same manner as the case of a DC motor by controlling the motor flux so as to assume a predetermined value depending on the exciting current command i d *, while the slip frequency and torque are controlled in dependence on the torque current command i q *.
  • FIG. 9 shows a further embodiment of the invention.
  • like parts as those shown in FIG. 8 are denoted by like reference numerals and repeated description thereof is omitted.
  • the embodiment shown in FIG. 9 differs from the one shown in FIG. 8 in that the outputs of the function generators 4U, 4V and 4W are added to AC voltage command signal U u *, V v * and V w *, respectively.
  • the two-phase/three-phase converter 19 outputs the voltage command signals V u *, V v * and V w * which have a frequency proportional to the frequency command signal ⁇ 1 * and are out of phase with one another by 120°.
  • These voltage command signals V u *, V v * and V w * are applied to the adders 40U, 40V and 40W, respectively, through which the voltage command signal V u *, V v * and V w * are added in terms of vector with the output signals ⁇ V u *, ⁇ V v * and ⁇ V w * of the function generators 4U, 4V and 4W, respectively, with the illustrated polarity, whereby the voltage command signal V u **, V v ** and V w ** are produced and supplied to the switch circuit 20.
  • the embodiment shown in FIG. 9 can assure the similar effect as that of the embodiment shown in FIG. 8 and enjoy additional advantage in that the converters 16 and 17 required in the case of the system shown in FIG. 8 can be spared, to simplify the system structure.
  • FIG. 10 shows a further embodiment of the invention.
  • like parts as those shown in FIG. 9 are denoted by like reference numerals and description thereof is omitted.
  • the system shown in FIG. 10 differs from the one shown in FIG. 9 in that current command signals i u *, i v * and i w * are employed as the input signals to the function generators 4U, 4V and 4W, respectively.
  • the coordinate transformation circuit 44 functions to transform the output signal i d * of the exciting current command circuit 13 and the output signal i q * of the speed deviation amplifier 9 (both of which are quantities in the rotating field orientated frame) to the quantities in the stator reference frame (coordinate system) on the basis of the sinusoidal signal produced by the oscillator 12, whereby two-phase AC signal i.sub. ⁇ * and i.sub. ⁇ * are produced.
  • the system shown in FIG. 10 can assure the similar effect to that of the embodiment shown in FIG. 9 and provides additional advantage that the current commands i u *, i v * and i w * are insusceptible to influence of harmonic noise contained in the motor current because the current command contains only the fundamental wave component.
  • FIG. 11 shows another exemplary embodiment of the invention applied to a digital control system.
  • the primary currents i u , i v and i w or U-, V- and W-phases of an induction motor 2 are detected by current detectors 3U, 3V and 3W, respectively, and applied to analoque-to-digital (A/D) converters 46, respectively.
  • a digital arithmetic circuit 56 is composed of a processor 55 for executing arithmetic processings, a program memory 54 storing a control program, a data memory 53 for storing data, input/output ports 49 and 51, a counter 50 and address/data bus 52 interconnecting these elements.
  • the processor 55 Prior to the operation of the induction motor 2, the processor 55 causes the inverter 1 to supply a DC current to the induction motor 2 for measuring the characteristic relation between the DC voltage command and the DC current (i.e. the characteristic illustrated graphically in FIG. 6) in accordance with the processing procedure of the control program stored in the memory 54. The result of the measurement is loaded in the data memory 53.
  • the processor 55 fetches the value of the counter 50 which is adpated to count the output signal of a voltage-to-frequency (V/F) converter 47 by way of the bus 52 in accordance with the processing procedure given by the control program stored in the memory 54 to thereby arithmetically determine the voltage commands V u *, V v * and V w * for the induced EMF of the induction motor 2. Additionally, the processor 55 fetches the output current of the inverter 1 through the input/output port 51 and reads out from the data memory 53 by way of the bus 52 the voltage command values ⁇ V u *, ⁇ V v * and ⁇ V w * having such magnitude and polarity as required for compensating for the voltage drop.
  • V/F voltage-to-frequency
  • the induction motor 2 is supplied with a DC current i dc from the inverter 1 prior to the actual operation of the motor on the same principle as that described hereinbefore in conjunction with the system shown in FIG. 1.
  • an output frequency command f* for the inverter is set to zero (i.e. DC current) while the DC voltage command V dc * is initialized to zero at a block or step 12a.
  • incremental values ⁇ V dc * of the DC voltage command V dc * for varying the DC current i dc of the inverter 1 from zero to a rated current value and the aforementioned data are stored in the data memory 53 at a leading address M.
  • the content of the data memory 51 is cleared to zero.
  • the voltage command V u ** for U-phase is set to V dc *, the while the voltage command V v ** for V-phase is set to -V dc * with the voltage command V w ** for W-phase being set to zero.
  • the voltage commands are then outputted through the input/output port 49.
  • the output current i u of the inverter 1 corresponding to the DC voltage command V dc * is detected by the current detector 3U and supplied to the A/D converter 46.
  • the digital signal I outputted from the A/D converter 46 is fetched through the input/output port 51 by way of the bus 52.
  • K represents the same constant as the one appearing in the expression (23).
  • the DC voltage command V dc * is incremented by the incremental value ⁇ V dc * to prepare the updated DC voltage command V dc *.
  • the procedure then returns to the step 12d to repeat the execution of the steps 12d to 12h.
  • the content at the address of the memory 53 where no data has been written is determined through interpolation technique and written in the data memory by way of the bus 52 at a step 12i. Further, in order to prepare the content of the data memory for the current of negative polarity, the data for the current of positive polarity are read out by way of the bus 52, inverted in respect to the sign and again written in the data memory 53 by way of the bus 52 at a step 12j.
  • the output pulse signal of the V/F converter 47 which converts the output signal of the speed command circuit 6 to a train of pulses is counted by the counter 50 and fetched by way of the bus 52 to arithmetically determine the amplitude values of the voltage command pattern signals V u *, V v * and V w * for the induced electromotive force of the induction motor 2.
  • the output current i u , i v and i w of the inverter 1 are detected by the current detectors 3U, 3V and 3W, respectively, and applied to the A/D converter 46, the digital outputs IU, IV and IW thereof being fetched through the input/output port 51 by way of the bus 52.
  • the contents of the data memory 53 are read out in dependence on the magnitude and polarity of IU, IV and IW. The contents thus read out are represented by ⁇ V u *, ⁇ V v * and ⁇ V w *, respectively.
  • the voltage command pattern signals V u *, V v * and V w * determined at the step 13a are, respectively, added with ⁇ V u *, ⁇ V v * and ⁇ V w * read out at the step 13c.
  • the results of the addition at the step 13d are outputted through the input/output port 49 by way of the bus 52.
  • the instant embodiment can assure the advantageous effect mentioned hereinbefore in conjunction with FIG. 9. Besides, the processing can be performed continuously from the preparation processing preceeding to the operation of the motor to the processing in the course of operation of the induction motor. By virtue of this feature, the advantageous effect mentioned in conjunction with FIG. 9 can be retained even when combination of the inverter and the induction motor is altered.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
US06/766,945 1984-08-21 1985-08-19 Method of controlling inverter-driven induction motor Expired - Fee Related US4680526A (en)

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JP59-173713 1984-08-21
JP59173713A JPS6152176A (ja) 1984-08-21 1984-08-21 誘導電動機のベクトル制御方法
JP60072875A JPH0687678B2 (ja) 1985-04-05 1985-04-05 電圧形インバ−タの制御方法
JP60-72875 1985-04-05

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Cited By (52)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4767976A (en) * 1986-04-14 1988-08-30 Hitachi, Ltd. Control system for PWM inverter
US4808903A (en) * 1987-04-13 1989-02-28 Hitachi, Ltd. Vector control system for induction motors
US4862054A (en) * 1988-10-31 1989-08-29 Westinghouse Electric Corp. Tacho-less vector control adaptive system for motor drive
US4958117A (en) * 1989-09-29 1990-09-18 Allen-Bradley Company, Inc. Frequency control based on sensing voltage fed to an induction motor
US4965504A (en) * 1988-09-20 1990-10-23 Hitachi, Ltd. Control apparatus for inverter for driving AC motor
US5021724A (en) * 1989-11-25 1991-06-04 Hitachi, Ltd. Controlling apparatus for induction motor
US5066899A (en) * 1989-03-24 1991-11-19 Kabushiki Kaisha Okuma Tekkosho Control method of induction motor and device therefor
US5123080A (en) * 1987-03-20 1992-06-16 Ranco Incorporated Of Delaware Compressor drive system
US5126642A (en) * 1991-01-31 1992-06-30 Ranco Incorporated Of Delaware Variable speed motor control
US5140248A (en) * 1987-12-23 1992-08-18 Allen-Bradley Company, Inc. Open loop motor control with both voltage and current regulation
US5155305A (en) * 1989-10-16 1992-10-13 Otis Elevator Company Delayed start of elevator car deceleration and creep using VVVF technology
US5264773A (en) * 1991-02-22 1993-11-23 Mitsubishi Denki K.K. Controller for induction motor
US5270498A (en) * 1990-05-24 1993-12-14 Mitsubishi Denki Kabushiki Kaisha Elevator speed control apparatus with drive motor residual secondary magnetic flux control
US5355070A (en) * 1993-03-10 1994-10-11 A. C. Propulsion, Inc. Induction motor drive stability control circuit
DE4413809A1 (de) * 1993-04-28 1994-11-03 Hitachi Ltd Verfahren zum Steuern eines Induktionsmotors und von einem solchen betriebene Vorrichtungen
US5387855A (en) * 1987-09-24 1995-02-07 Kabushiki Kaisha Toshiba Induction motor control system
US5457375A (en) * 1994-05-27 1995-10-10 Emerson Electric Co. Sensorless commutation controller for a poly-phase dynamoelectric machine
US5471126A (en) * 1988-12-24 1995-11-28 Fanuc Ltd. Slip frequency control method for induction motors
US5477114A (en) * 1992-11-30 1995-12-19 Kabushiki Kaisha Meidensha Method and system for estimating inertia of 2-mass system during speed control
US5481173A (en) * 1993-01-11 1996-01-02 Kabushiki Kaisha Meidensha Vector control system for induction motor
US5502360A (en) * 1995-03-10 1996-03-26 Allen-Bradley Company, Inc. Stator resistance detector for use in electric motor controllers
US5598081A (en) * 1993-09-27 1997-01-28 Matsushita Electric Works, Ltd. Vector control method for controlling a rotor speed of an induction motor
US5623190A (en) * 1993-09-17 1997-04-22 Fuji Electric Co., Ltd. Method and apparatus for controlling an AC motor and detecting malfunctions
US5635811A (en) * 1994-06-30 1997-06-03 Siemens Aktiengesellschaft Method and apparatus for standstill recognition when restarting a power converter-fed AC motor without a tachometer generator
US5646499A (en) * 1994-08-25 1997-07-08 Matsushita Electric Industrial Co.,Ltd. Inverter control apparatus
US5689167A (en) * 1995-10-24 1997-11-18 Micro Linear Corporation Two-phase electronic control of a three-phase induction motor
US5714897A (en) * 1996-06-19 1998-02-03 Micro Linear Corporation Phase-shifted triangle wave generator
US5767653A (en) * 1995-10-24 1998-06-16 Micro Linear Corporation Variable speed AC induction motor controller
US5793168A (en) * 1996-08-23 1998-08-11 Micro Linear Corporation Active deceleration circuit for a brushless DC motor
US5844397A (en) * 1994-04-29 1998-12-01 Reda Pump Downhole pumping system with variable speed pulse width modulated inverter coupled to electrical motor via non-gap transformer
US5859518A (en) * 1997-12-22 1999-01-12 Micro Linear Corporation Switched reluctance motor controller with sensorless rotor position detection
US5936856A (en) * 1998-02-13 1999-08-10 Baldor Electric Companh Method and apparatus for active common-mode voltage compensation in induction motor systems
US5962999A (en) * 1997-07-30 1999-10-05 Matsushita Electric Industrial Method of controlling a torque ripple of a motor having interior permanent magnets and a controller using the same method
US6040673A (en) * 1998-05-29 2000-03-21 Matsushita Electric Industrial Co., Ltd. Motor control apparatus
US6088246A (en) * 1997-06-17 2000-07-11 Mitsubishi Denki Kabushiki Kaisha Method of and device for controlling pulse width modulation inverter
US6184638B1 (en) * 1998-05-29 2001-02-06 Mitsubishi Denki Kabushiki Kaisha Control system for an induction motor
US6255798B1 (en) * 1995-10-30 2001-07-03 Hitachi, Ltd. Control apparatus for electric vehicle
US6259226B1 (en) * 1998-09-03 2001-07-10 Mitsubishi Denki Kabushiki Kaisha Controller for AC motor
US6528966B2 (en) * 2000-08-18 2003-03-04 Lg Industrial Systems Co., Ltd. Sensorless vector control apparatus and method thereof
US20030090232A1 (en) * 2001-11-06 2003-05-15 International Rectifier Current ripple reduction by harmonic current regulation
US20030160585A1 (en) * 2002-02-26 2003-08-28 Lg Industrial Systems Co., Ltd. Motor torque control apparatus and method
US20040036439A1 (en) * 2002-08-20 2004-02-26 Samsung Electronics Co., Ltd Apparatus and method of controlling a stepper motor
US20040080968A1 (en) * 2001-03-19 2004-04-29 Antti Tarkiainen Vector control of an induction motor
FR2859583A1 (fr) * 2001-10-12 2005-03-11 Mitsubishi Electric Corp Appareil de conversion de courant electrique
US20050190094A1 (en) * 2002-04-17 2005-09-01 Danfoss Drives A/S Method for measuring currents in a motor controller and motor controller using such method
US7161324B1 (en) * 2003-07-16 2007-01-09 Mitsubishi Denki Kabushiki Kaisha Device for estimating pole position of synchronous motor
GB2421090B (en) * 2003-08-18 2007-03-21 Yaskawa Denki Seisakusho Kk Control method of voltage source inverter
US20070251236A1 (en) * 2004-02-05 2007-11-01 Pierre Barthelet Motor Control and Driver for Electric Boosting Application
US20080277209A1 (en) * 2005-12-20 2008-11-13 Edward Piedra Elevator Drive Control Strategy
US9421611B2 (en) 2014-03-07 2016-08-23 Kennametal Inc. Composite cutting insert and method of making same
JP2019507566A (ja) * 2015-12-15 2019-03-14 サフラン エレクトロニクス アンド ディフェンス 電気機械アクチュエータシステムの監視方法
US10333446B2 (en) * 2017-09-22 2019-06-25 Fuji Electric Co., Ltd. Controller for induction motor

Families Citing this family (12)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2708408B2 (ja) * 1986-05-09 1998-02-04 株式会社日立製作所 電圧制御形ベクトル制御インバータの制御装置
DE3820125C2 (de) * 1987-06-12 1994-04-07 Hitachi Ltd Verfahren zum Steuern eines wechselrichtergespeisten Asynchronmotors
US5155797A (en) * 1987-09-08 1992-10-13 Kabushiki Kaisha Meidensha Control system for controlling revolution speed of electric motor
EP0306922B1 (de) * 1987-09-08 1995-01-25 Kabushiki Kaisha Meidensha Regelsystem zur Regelung der Umdrehungsgeschwindigkeit eines elektrischen Motors
JP2625969B2 (ja) * 1988-03-11 1997-07-02 株式会社明電舎 誘導電動機のベクトル制御装置
JP2646633B2 (ja) * 1988-03-26 1997-08-27 株式会社安川電機 誘導電動機の磁束演算方法
JPH01270793A (ja) * 1988-04-21 1989-10-30 Fanuc Ltd Ｐｗｍ制御における電流制御装置
DE59203113D1 (de) * 1992-01-27 1995-09-07 Siemens Ag Einrichtung zum Regeln der Drehzahl einer Drehfeldmaschine.
DE4310778C2 (de) * 1993-03-26 1995-02-09 Licentia Gmbh Verfahren zur zeitdiskreten Regelung des Stromes eines über einen Wechselrichter gespeisten Asynchronmotors
DE59305367D1 (de) * 1993-07-26 1997-03-13 Siemens Ag Verfahren und vorrichtung zur kompensation einer fehlerspannung bei einem pulswechselrichter
FI96371C (fi) * 1994-05-13 1996-06-10 Abb Industry Oy Menetelmä verkkovaihtosuuntaajan kautta siirrettävän tehon säätämiseksi
DE10161927C2 (de) * 2001-12-17 2003-11-13 Siemens Ag Verfahren zum Bestimmen von Parametern eines Wechselrichters

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3989991A (en) * 1974-10-03 1976-11-02 Westinghouse Electric Corporation Method and circuit for the derivation of an analog slip frequency signal of an induction motor in a tachometerless motor drive
US4451771A (en) * 1982-01-18 1984-05-29 Hitachi, Ltd. Apparatus for controlling AC motor
US4459534A (en) * 1981-05-29 1984-07-10 Hitachi, Ltd. Method for controlling induction motor and apparatus therefor
US4503376A (en) * 1982-10-04 1985-03-05 Hitachi, Ltd. Control method for induction motors

Family Cites Families (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4317074A (en) * 1979-09-12 1982-02-23 Borg-Warner Corporation Inverter-motor system with different control characteristics for inverter voltage and frequency
JPS5949797B2 (ja) * 1980-09-10 1984-12-05 富士電機株式会社 交流機の電流制御方式
JPS59169369A (ja) * 1983-03-16 1984-09-25 Toshiba Corp 交流電流制御装置

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3989991A (en) * 1974-10-03 1976-11-02 Westinghouse Electric Corporation Method and circuit for the derivation of an analog slip frequency signal of an induction motor in a tachometerless motor drive
US4459534A (en) * 1981-05-29 1984-07-10 Hitachi, Ltd. Method for controlling induction motor and apparatus therefor
US4451771A (en) * 1982-01-18 1984-05-29 Hitachi, Ltd. Apparatus for controlling AC motor
US4503376A (en) * 1982-10-04 1985-03-05 Hitachi, Ltd. Control method for induction motors

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US4767976A (en) * 1986-04-14 1988-08-30 Hitachi, Ltd. Control system for PWM inverter
US5123080A (en) * 1987-03-20 1992-06-16 Ranco Incorporated Of Delaware Compressor drive system
US4808903A (en) * 1987-04-13 1989-02-28 Hitachi, Ltd. Vector control system for induction motors
US5387855A (en) * 1987-09-24 1995-02-07 Kabushiki Kaisha Toshiba Induction motor control system
US5140248A (en) * 1987-12-23 1992-08-18 Allen-Bradley Company, Inc. Open loop motor control with both voltage and current regulation
US4965504A (en) * 1988-09-20 1990-10-23 Hitachi, Ltd. Control apparatus for inverter for driving AC motor
US4862054A (en) * 1988-10-31 1989-08-29 Westinghouse Electric Corp. Tacho-less vector control adaptive system for motor drive
US5471126A (en) * 1988-12-24 1995-11-28 Fanuc Ltd. Slip frequency control method for induction motors
US5066899A (en) * 1989-03-24 1991-11-19 Kabushiki Kaisha Okuma Tekkosho Control method of induction motor and device therefor
US4958117A (en) * 1989-09-29 1990-09-18 Allen-Bradley Company, Inc. Frequency control based on sensing voltage fed to an induction motor
US5155305A (en) * 1989-10-16 1992-10-13 Otis Elevator Company Delayed start of elevator car deceleration and creep using VVVF technology
US5021724A (en) * 1989-11-25 1991-06-04 Hitachi, Ltd. Controlling apparatus for induction motor
US5270498A (en) * 1990-05-24 1993-12-14 Mitsubishi Denki Kabushiki Kaisha Elevator speed control apparatus with drive motor residual secondary magnetic flux control
US5126642A (en) * 1991-01-31 1992-06-30 Ranco Incorporated Of Delaware Variable speed motor control
US5264773A (en) * 1991-02-22 1993-11-23 Mitsubishi Denki K.K. Controller for induction motor
US5477114A (en) * 1992-11-30 1995-12-19 Kabushiki Kaisha Meidensha Method and system for estimating inertia of 2-mass system during speed control
US5481173A (en) * 1993-01-11 1996-01-02 Kabushiki Kaisha Meidensha Vector control system for induction motor
US5355070A (en) * 1993-03-10 1994-10-11 A. C. Propulsion, Inc. Induction motor drive stability control circuit
DE4413809A1 (de) * 1993-04-28 1994-11-03 Hitachi Ltd Verfahren zum Steuern eines Induktionsmotors und von einem solchen betriebene Vorrichtungen
CN1042184C (zh) * 1993-04-28 1999-02-17 株式会社日立制作所 感应电动机的磁通确定方法及其驱动和控制方法和系统
US5623190A (en) * 1993-09-17 1997-04-22 Fuji Electric Co., Ltd. Method and apparatus for controlling an AC motor and detecting malfunctions
US5598081A (en) * 1993-09-27 1997-01-28 Matsushita Electric Works, Ltd. Vector control method for controlling a rotor speed of an induction motor
US5844397A (en) * 1994-04-29 1998-12-01 Reda Pump Downhole pumping system with variable speed pulse width modulated inverter coupled to electrical motor via non-gap transformer
US5457375A (en) * 1994-05-27 1995-10-10 Emerson Electric Co. Sensorless commutation controller for a poly-phase dynamoelectric machine
US5635811A (en) * 1994-06-30 1997-06-03 Siemens Aktiengesellschaft Method and apparatus for standstill recognition when restarting a power converter-fed AC motor without a tachometer generator
US5646499A (en) * 1994-08-25 1997-07-08 Matsushita Electric Industrial Co.,Ltd. Inverter control apparatus
US5502360A (en) * 1995-03-10 1996-03-26 Allen-Bradley Company, Inc. Stator resistance detector for use in electric motor controllers
US5689167A (en) * 1995-10-24 1997-11-18 Micro Linear Corporation Two-phase electronic control of a three-phase induction motor
US5767653A (en) * 1995-10-24 1998-06-16 Micro Linear Corporation Variable speed AC induction motor controller
US6255798B1 (en) * 1995-10-30 2001-07-03 Hitachi, Ltd. Control apparatus for electric vehicle
US5714897A (en) * 1996-06-19 1998-02-03 Micro Linear Corporation Phase-shifted triangle wave generator
US5793168A (en) * 1996-08-23 1998-08-11 Micro Linear Corporation Active deceleration circuit for a brushless DC motor
US6088246A (en) * 1997-06-17 2000-07-11 Mitsubishi Denki Kabushiki Kaisha Method of and device for controlling pulse width modulation inverter
US5962999A (en) * 1997-07-30 1999-10-05 Matsushita Electric Industrial Method of controlling a torque ripple of a motor having interior permanent magnets and a controller using the same method
US5859518A (en) * 1997-12-22 1999-01-12 Micro Linear Corporation Switched reluctance motor controller with sensorless rotor position detection
US5936856A (en) * 1998-02-13 1999-08-10 Baldor Electric Companh Method and apparatus for active common-mode voltage compensation in induction motor systems
GB2355870B (en) * 1998-05-29 2001-09-12 Mitsubishi Electric Corp Control system for an induction motor
US6040673A (en) * 1998-05-29 2000-03-21 Matsushita Electric Industrial Co., Ltd. Motor control apparatus
US6184638B1 (en) * 1998-05-29 2001-02-06 Mitsubishi Denki Kabushiki Kaisha Control system for an induction motor
GB2355870A (en) * 1998-05-29 2001-05-02 Mitsubishi Electric Corp A control system for controlling the variable speed of an induction motor without an angular velocity detector
US6259226B1 (en) * 1998-09-03 2001-07-10 Mitsubishi Denki Kabushiki Kaisha Controller for AC motor
US6528966B2 (en) * 2000-08-18 2003-03-04 Lg Industrial Systems Co., Ltd. Sensorless vector control apparatus and method thereof
US6956352B2 (en) 2001-03-19 2005-10-18 Abb Oy Vector control of an induction motor
US20040080968A1 (en) * 2001-03-19 2004-04-29 Antti Tarkiainen Vector control of an induction motor
FR2859583A1 (fr) * 2001-10-12 2005-03-11 Mitsubishi Electric Corp Appareil de conversion de courant electrique
US20030090232A1 (en) * 2001-11-06 2003-05-15 International Rectifier Current ripple reduction by harmonic current regulation
US6777907B2 (en) * 2001-11-06 2004-08-17 International Rectifier Corporation Current ripple reduction by harmonic current regulation
US20030160585A1 (en) * 2002-02-26 2003-08-28 Lg Industrial Systems Co., Ltd. Motor torque control apparatus and method
US6828744B2 (en) * 2002-02-26 2004-12-07 Lg Industrial Systems Co., Ltd. Motor torque control apparatus and method
US20050190094A1 (en) * 2002-04-17 2005-09-01 Danfoss Drives A/S Method for measuring currents in a motor controller and motor controller using such method
US20040036439A1 (en) * 2002-08-20 2004-02-26 Samsung Electronics Co., Ltd Apparatus and method of controlling a stepper motor
US7161324B1 (en) * 2003-07-16 2007-01-09 Mitsubishi Denki Kabushiki Kaisha Device for estimating pole position of synchronous motor
US20070018605A1 (en) * 2003-07-16 2007-01-25 Mitsubishi Denki Kabushiki Kaisha Device for estimating pole position of synchronous motor
US7362069B2 (en) 2003-08-18 2008-04-22 Kabushiki Kaisha Yaskawa Denki Voltage source inverter control method
GB2421090B (en) * 2003-08-18 2007-03-21 Yaskawa Denki Seisakusho Kk Control method of voltage source inverter
US20070251236A1 (en) * 2004-02-05 2007-11-01 Pierre Barthelet Motor Control and Driver for Electric Boosting Application
US7545115B2 (en) * 2004-02-05 2009-06-09 Honeywell International Inc. Motor control and driver for electric boosting application
US20080277209A1 (en) * 2005-12-20 2008-11-13 Edward Piedra Elevator Drive Control Strategy
US8104584B2 (en) * 2005-12-20 2012-01-31 Otis Elevator Company Elevator drive control strategy
US9421611B2 (en) 2014-03-07 2016-08-23 Kennametal Inc. Composite cutting insert and method of making same
JP2019507566A (ja) * 2015-12-15 2019-03-14 サフラン エレクトロニクス アンド ディフェンス 電気機械アクチュエータシステムの監視方法
US10451502B2 (en) * 2015-12-15 2019-10-22 Safran Electronics & Defense Method for monitoring an electromechanical actuator system
US10333446B2 (en) * 2017-09-22 2019-06-25 Fuji Electric Co., Ltd. Controller for induction motor

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EP0175154A3 (en) 1986-12-30
DE3584603D1 (de) 1991-12-12
EP0175154B1 (de) 1991-11-06
EP0175154A2 (de) 1986-03-26

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