US4584635A - Flux centering and power control for high frequency switching power - Google Patents

Flux centering and power control for high frequency switching power Download PDF

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Publication number
US4584635A
US4584635A US06/584,195 US58419584A US4584635A US 4584635 A US4584635 A US 4584635A US 58419584 A US58419584 A US 58419584A US 4584635 A US4584635 A US 4584635A
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transformer
signal
operable
power supply
current
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US06/584,195
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Alexander G. MacInnis
William B. Nunnery
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International Business Machines Corp
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International Business Machines Corp
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Priority to US06/584,195 priority Critical patent/US4584635A/en
Assigned to INTERNATIONAL BUSINESS MACHINES CORPORATION A CORP. OF NY reassignment INTERNATIONAL BUSINESS MACHINES CORPORATION A CORP. OF NY ASSIGNMENT OF ASSIGNORS INTEREST. Assignors: MACINNIS, ALEXANDER G., NUNNERY, WILLIAM B.
Priority to JP59219416A priority patent/JPS60183970A/ja
Priority to EP84114673A priority patent/EP0155369B1/en
Priority to DE8484114673T priority patent/DE3472514D1/de
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/538—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration
    • H02M7/53803—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration with automatic control of output voltage or current
    • H02M7/53806—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration with automatic control of output voltage or current in a push-pull configuration of the parallel type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3376—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current
    • H02M3/3378—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current in a push-pull configuration of the parallel type

Definitions

  • the invention relates to power supplies in general and more particularly to switch mode power supplies in which a power transformer is directly coupled to the bulk voltage.
  • switch mode power supplies for providing power to different types of loads.
  • the prior art switch mode power supplies consist of a power transformer with an input coil or winding and an output winding.
  • the input winding is connected to a supply voltage which provides electrical energy to the transformer.
  • a switching circuit which may be a bridge or a two-switch push pull design, is connected to the input winding.
  • the circuit switches the direction of magnetic flux within the input winding as a result of voltage developed across the input winding.
  • the voltage is induced onto the output winding.
  • the voltage is then rectified and supplied to the attached load.
  • a feedback error voltage is developed and is used to generate pulse width modulated signals which drive the switching circuits.
  • One of the prior art techniques involves sensing the current in each of the two switches directly and using the signal to control the switch turnoff signals.
  • This technique is referred to as a current balancing or current mode technique.
  • the current has a ramp or saw tooth waveform and it is compared to a reference voltage.
  • the intention is to keep the switch currents equal and indirectly keep the DC component of the magnetizing current (I m ) at 0.
  • This technique presents several problems. Essentially, bipolar switches usually have unpredictable turnoff times which may be a substantial portion of on-time, so the true peak switch current, and the current expected and measured by the control system are quite different from one another. The result is that a DC component is added to I m , tending to cause saturation.
  • an inductor is placed in series with the DC input terminal of the transformer primary (center tapped only). This tends to hold the primary current constant, which tends to maintain transformer stability.
  • Non-dissipative coupling of the inductor's stored energy is a problem.
  • leakage in coupled inductors causes spikes which appear on the switches. Leakage also causes a problem with regulation and efficiency, especially in systems with power over about 200 watts. Snubbers interfere with efficiency, ripple and regulation.
  • a practical coupled inductor has to meet isolation specifications which cause increased leakage inductance, and the inductor is expensive. Reflected load currents can cause an I m offset.
  • a circuit is added to detect the difference in peak switch currents, and this result is applied to the integrator in such a way as to modify switch times to equalize the peak currents.
  • This technique is effective except for the aforementioned problem of reflected synchronous or transient load current at the primary. As previously stated, the reflection saturates the transformer. It is therefore necessary to ensure that the magnetizing current is much greater than the maximum reflected AC load current at the switch frequency, necessitating a gapped-core transformer or special output filter.
  • the switches driving the primary are essentially controlled in a standard pulsewidth modulated (PWM) fashion.
  • PWM pulsewidth modulated
  • the method senses saturation at the core directly, via an evenly-gapped E-E core and a special winding.
  • the information, representative of core saturation, is used to modify switching times to correct the saturation problem.
  • the gapped transformer implies problems with size, weight, cost and efficiency. In particular, the specially made transformers are expensive and of limited application.
  • the introduction of an air gap implies large magnetizing current. A more detailed description of this technique is given by Patel, Raoji, in an article entitled, "Detecting Impending Core Saturation in Switched-Mode Power Converters," Unitrode Power Supply Design Seminar, 1980, Unitrode Corporation, Lexington, Mass.
  • the improved power supply includes a power transformer having a core with secondary and primary windings.
  • the primary windings include a bulk power supply and a plurality of switching transistors.
  • the secondary winding is coupled by an appropriate circuit to an output terminal which supplies power to a load.
  • a sense winding and serially connected integrating circuitry are used to generate a signal representative of the total magnetic flux within the power transformer.
  • a sensing device and appropriate circuitry are used to generate an offsetting signal representative of the magnetizing current (I m ) within the power transformer.
  • the magnetizing current is equivalent to the magnetizing flux.
  • a composite signal is formed from the offsetting signal and the total magnetic flux signal.
  • a multiphase clocking signal is gated with the composite signal to generate a pulse-width modulated signal which is used to switch the power transistor so that the I m and hence the offsetting magnetic flux are maintained at an acceptable level.
  • the offsetting signal is combined with the total magnetic flux signal, the DC component of the flux is eliminated and the composite signal represents an accurate value of the flux within the core of the power transformer.
  • the sensing device includes a coil and a resistor coupled to the secondary windings. Current is allowed to flow in the resistor only when the power switches are in an off state. This ensures that the value of the composite signal is a true measure of I m .
  • FIG. 1 shows a block diagram of an improved power supply according to the teachings of the present invention.
  • FIG. 2 shows a detailed schematic of the novel circuitry which controls the power supply of FIG. 1.
  • FIG. 3 shows a graphical representation of the signals which are combined to form a pulse-width modulated (PWM) signal.
  • the (PWM) signal is used to switch the power transistors which control current flow through the power transformer.
  • FIG. 1 shows a block diagram of the improved power supply according to the teachings of the present invention.
  • the power supply comprises of a power transformer with a primary coil identified by numeral 10 and a secondary coil identified by numeral 12. A center-tap point on the primary coil is connected by conductor 14 to a voltage supply source identified as V bulk .
  • a switching means identified by numeral 16 is connected to the primary coil.
  • the function of the switching means is to switch current so that it flows bi-directionally within the primary winding, and as a result a voltage is generated across the winding.
  • the switching means comprises of a pair of power transistors identified as switch A (SWA) and switch B (SWB).
  • SWA switch A
  • SWB switch B
  • the power transistors are connected in a push-pull fashion with their emitters connected to a common ground level and their respective bases are connected over conductors 18 and 20, respectively, to a base drive circuit means identified by numeral 22.
  • Base drive circuit means 22 is primarily an amplification type circuit which accepts the control signals which are supplied from control circuit means 24 via conductors 26 and 28, respectively, amplify these signals and feed them over conductors 18 and 20 so that the switching of the power transistors is out of phase. Since the base drive circuit means are conventional circuitry, details will not be given. Suffice it to say that it is within the skill of one skilled in the art to develop an appropriate circuitry for amplifying the signals supplied over conductors 26 and 28, respectively.
  • the output section of the power supply includes the secondary winding 12.
  • a conductor identified by numeral 30 interconnects a center-tap point on the primary winding with a negative potential output terminal.
  • Current sense means 32 is connected to the output of the secondary winding. As will be explained hereinafter, the function of current sense means 32 is to generate a signal which represents the magnetizing current (I m ) in the transformer and feed that current over conductor 34 into control circuit means 24.
  • a signal representative of the flux in the power transformer is generated by sense winding 36.
  • the signal is fed over conductor 38 into control circuit means 24.
  • the details of current sensing means 32 and control circuit means 24 which form the crux of this invention will be given hereinafter. Suffice it to say, at this point, that the sense winding 36 in conjunction with current sense means 32 and control circuit means 24 generates a first electrical signal which represents the total flux in the transformer.
  • a second electrical signal representative of the magnetizing current is also generated.
  • An integrator (to be described later) integrates both signals and generates therefrom a composite signal.
  • the composite signal is gated with a two-phase clock which is 180° out of phase and generates a pulse width modulated signal which is fed over conductors 26 and 28, respectively, into the base drive circuit means 22.
  • the base drive circuit means 22 amplifies the signal and feeds it to the base of switch A and switch B over conductors 18 and 20, respectively.
  • the output from current sense means 32 is fed into diodes D1 and D2, respectively.
  • the function of D1 and D2 is to modulate or rectify the signal outputted from current sense means 32.
  • the rectified signal is then fed through the LC circuit and into the attached load (not shown).
  • Conductor 40 interconnects the positive output terminal to feedback error circuit means 42.
  • the function of feedback error circuit means 42 is to correlate the feedback signal on conductor 40 with a reference signal and generates an error signal on conductor 44.
  • the error signal on conductor 44 is fed into control circuit means 24 from whence it is used in formulating the pulsewidth modulated pulses which are used to control switch A and switch B, respectively.
  • FIG. 2 shows a detailed circuitry for the control means which generates the pulsewidth modulated signals for controlling switch A and switch B (FIG. 1).
  • sense winding 36 which senses the flux in the transformer is connected to an integrator formed by resistors R1, C2 and M1.
  • M1 is a conventional operational amplifier.
  • C1 is connected to R1 and the negative input of M1.
  • C1 serves as an AC coupler and couples the output of the sense winding into the integrator.
  • R2 is connected across M1 and serves as a DC stabilizer to M1. Furthermore, R2 moves the integrator pole away from the origin.
  • V ios is the error voltage (V error ) due to an offset in the magnetizing current (I mag );
  • the signal on conductor 48 is fed into the negative terminal of M1 while the signal on conductor 46 is fed into the positive terminal of M1.
  • a current transformer (I xfmr ) 50 having a core and a winding identified by numeral 52, is connected to the secondary winding 12 of the transformer.
  • the purpose of the current transformer is to generate a current (I m ) which is the equivalent of the magnetizing current in the transformer core.
  • V Im represents the magnetizing voltage and is reconverted into the current via R 10 and fed into the negative terminal of the integrator.
  • the magnetizing current I m is only measured when the power switches SWA and SWB are in the off state. When either of the switches is conducting the output from the secondary winding is fed through D1, D2, L and C to V out .
  • the elements D1, D2, L and C have already been described in the discussion of FIG. 1. They perform the same functions in FIG. 2 and as such further discussion relative to these elements will not be given.
  • the coil 52 is connected to resistor R7.
  • the function of R7 is to generate the measurement voltage V Im when the power transistor switches SWA and SWB (FIG. 1) are in the off state.
  • a current switching circuit means or current circuit control means identified by numeral 54 is connected through R6 to one terminal of the secondary winding.
  • Another current circuit control means 56 is coupled through a resistor R5 to the other terminal of the secondary winding.
  • the function of current circuit control means 54 and 56 is to monitor the voltage on the secondary winding and when there is a voltage the switching means acts as a short and as a result no current flows through R7 and therefore there is no output V Im .
  • the power transistors SWA and SWB (FIG. 1) is off, current flows through R7 and a voltage is presented at point V Im .
  • By using two current circuit control means both the positive and negative swing of the voltage across the secondary winding is monitored.
  • the circuit control means 54 and 56 are conventional opto-isolators. These isolators are off the shelf packages and include a diode and a transistor. When the diode conducts, the transistor is turned on and acts as a short and diverts current away from the resistor R7. It should be noted that other devices other than opto-isolators can be used to control the current across R7. In this embodiment opto-isolators were used because of their isolation properties and the fact that the isolators stay on until the secondary voltage on the power transformer collapses. This property provides a simple way of treating unpredictable storage times (T stg ) in the power switches. However, in a supply using components such as power MOSFETs one could probably use bipolar transistors instead of opto-isolators if T stg is not a problem.
  • T stg unpredictable storage times
  • D3, D4, C3, C4 form peak detectors; this function could be performed by other circuitry, such as integrated peak detectors.
  • resistor R7 is connected to diodes D3 and D4, respectively.
  • Diodes D3 and D4 are poled (that is, connected to conduct) in opposite directions.
  • Capacitor C3 connects the cathode of diode D3 to reference V g while capacitor C4 connects reference V g to the anode of diode D4.
  • V Im is the voltage representation of the magnetizing current signal
  • V g is a reference voltage referred to hereinafter as the artificial ground potential.
  • a pair of resistors R8 and R9 is connected in series and form a voltage divider. The series connected resistors are connected to the cathode and anode of diodes D3 and D4, respectively and generate a second reference voltage identified as V IOS .
  • Voltage divider R3 and R4 combine a signal identified as V ref , a signal identified as V error control on conductor 44 and V g and output two signals on conductors 58 and 60, respectively.
  • the signal on conductor 58 is fed into the positive terminal of comparator means M2 while the signal on conductor 60 is fed into the negative terminal of comparator M3.
  • the positive terminal of comparator M3 is connected to the output of M1 and the negative terminal of M2 is connected to the output of M1.
  • the outputs of M2 and M3 are gated with clock pulses A and B by means of logical "AND" circuit means 62 and 64, respectively.
  • the signals from the AND circuit means are fed over conductors 26 and 28 to drive the power transistors.
  • FIG. 3 shows a series of curves representing the signals which are developed by the circuit of FIG. 2. These signals are used for driving the power transistors of FIG. 1 so that the power transformer operates at a level below its saturation point.
  • the curves are exaggerated to make clear the improvement which the present invention adds to the prior art power supply of the direct coupled topography.
  • the first curve in the figure identified as V flux is a composite signal which represents the estimated flux in the transformer and the effect of V ios (voltage representing the magnetizing current offset).
  • the dotted line shows the effect of V ios on the estimate of the flux.
  • the solid line in FIG. 3 represents the estimate of the flux without applicants' invention.
  • V ref in curve 1 represents V ref in FIG. 2 while the horizontal line identified as V control represents the error feedback control signal.
  • the overshoot of the signal above V control is due to the storage time in the switches.
  • the second and third curves identified as clock A and clock B are two out of phase clock pulses which are used to gate the composite flux signal.
  • the fourth and fifth curves identified as switch A-based drive and switch B-based drive represent the modulated pulses which are used to switch the power transistors in FIG. 1. It should be noted that the width of these pulses differs by the difference between the solid and dotted curves of V flux.
  • the sense winding 36 is a separate secondary of the power transformer.
  • M 1 is an op-amp, and in conjunction with C5 and R4, forms an integrator.
  • C 1 is used for AC coupling, the output from the sense winding into the integrator.
  • R2 is necessary for DC stabilization, moving the integrator's pole away from the origin.
  • M2 and M3 are comparators and V g is an artificial ground.
  • Clock signals A and B are 180° out of phase and operate at a fixed frequency, so either switch A or switch B is enabled.
  • the switch stays on until V flux crosses V control or V ref , (FIG. 3), respectively, at which time the proper comparator M2 or M3 changes state and turns off the switch.
  • the integrator In order for the integrator (FIG. 2) to have an accurate measurement of the flux in the transformer it is necessary to know the long term flux. It should be noted that the flux and the magnetizing current (I m ) are related directly by the B-H curves of the transformer. Furthermore, if the average value of I m is 0, then the average value of the flux is 0 for all transformers.
  • the signal V ios (FIG. 2) or the voltage corresponding to the magnetizing current offset, is inputted to the integrator M1 in such a way as to cause the integrator's output to constantly increase or decrease with a flow proportional to the input V ios .
  • the loop has a maximum phase shift of 90°, from the integrator, so it is unconditionally stable.
  • the only cause of I m offset which necessitates this loop is terms which are essentially constant.
  • the terms include imbalanced power switch storage time, imbalancing switch voltage drop, imbalancing transformer primary winding coupling, and offset voltage and bias current in the integrator's OP-AMP.
  • this loop need not contend with such causes as dynamic load current, and the median value of I m will be maintained at or very near 0 regardless of operating conditions, for similar reasons the gain of the (I m --offset loop) is not critical.
  • the magnetizing current (I m ) is measured when the power switches are in an off state. As a result, a true measure of the magnetizing current I m is obtained.
  • a power supply using the above described control method has excellent open loop regulating properties.
  • the volt second integral of the transformer core is directly held constant over each pulse save for errors due to timing delays. This causes a constant average input voltage to the averaging output filter.
  • Regulation deviations are due to effective series resistance on series output devices (secondary windings, rectifiers, inductors). Primary losses do not affect regulation, unlike feed forward circuits added to conventional pulsewidth modulator controls.
  • the need for feedback is greatly reduced or eliminated, allowing improved regulation and stability with less gain and/or the omission of direct output sensing. This simplifies system design and component choices and reduces the count of expensive isolated power and feedback components.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
US06/584,195 1984-02-27 1984-02-27 Flux centering and power control for high frequency switching power Expired - Fee Related US4584635A (en)

Priority Applications (4)

Application Number Priority Date Filing Date Title
US06/584,195 US4584635A (en) 1984-02-27 1984-02-27 Flux centering and power control for high frequency switching power
JP59219416A JPS60183970A (ja) 1984-02-27 1984-10-20 スイツチング電源
EP84114673A EP0155369B1 (en) 1984-02-27 1984-12-04 Flux centering and power control for high frequency switching power supplies
DE8484114673T DE3472514D1 (en) 1984-02-27 1984-12-04 Flux centering and power control for high frequency switching power supplies

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US06/584,195 US4584635A (en) 1984-02-27 1984-02-27 Flux centering and power control for high frequency switching power

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EP (1) EP0155369B1 (ja)
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US20070070655A1 (en) * 2003-12-11 2007-03-29 Honda Motor Co., Ltd. Dc-dc converter
US20100274319A1 (en) * 2009-04-28 2010-10-28 Cochlear Limited Current leakage detection for a medical implant
US20110050292A1 (en) * 1998-02-05 2011-03-03 City University Of Hong Kong Coreless printed-circuit-board (pcb) transformers and operating techniques therefor
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US20120191160A1 (en) * 2006-08-25 2012-07-26 Cochlear Limited Current leakage detection method and device
US8588911B2 (en) 2011-09-21 2013-11-19 Cochlear Limited Medical implant with current leakage circuitry
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US4969081A (en) * 1989-01-09 1990-11-06 Sundstrand Corporation Inverter switch current sensor with shoot-through current limiting
US5552979A (en) * 1993-11-30 1996-09-03 Philips Electronics North America Corporation Isolated current sensor for DC to high frequency applications
US5736884A (en) * 1995-02-16 1998-04-07 U.S. Philips Corporation Device for generating a control signal dependent on a variable resistance value and apparatus comprising such device
US5629616A (en) * 1995-07-13 1997-05-13 Performance Conrols, Inc. Circuit for measuring current in class-d amplifiers
US8023290B2 (en) 1997-01-24 2011-09-20 Synqor, Inc. High efficiency power converter
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US9143042B2 (en) 1997-01-24 2015-09-22 Synqor, Inc. High efficiency power converter
US20110050292A1 (en) * 1998-02-05 2011-03-03 City University Of Hong Kong Coreless printed-circuit-board (pcb) transformers and operating techniques therefor
US8102235B2 (en) * 1998-02-05 2012-01-24 City University Of Hong Kong Coreless printed-circuit-board (PCB) transformers and operating techniques therefor
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US6534967B1 (en) 2000-09-25 2003-03-18 Mts Systems Corporation Dual totem current sensor for measuring load current in an H-bridge power stage
US6577111B1 (en) * 2001-09-06 2003-06-10 Abb Technology Ag Controlling magnetizing current in a transformer by comparing the difference between first and second positive peak values of the magnetizing current with a threshold
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US20120191160A1 (en) * 2006-08-25 2012-07-26 Cochlear Limited Current leakage detection method and device
US9238140B2 (en) * 2006-08-25 2016-01-19 Cochlear Limited Current leakage detection
US20100274319A1 (en) * 2009-04-28 2010-10-28 Cochlear Limited Current leakage detection for a medical implant
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US10594223B1 (en) 2013-07-02 2020-03-17 Vlt, Inc. Power distribution architecture with series-connected bus converter
US11075583B1 (en) 2013-07-02 2021-07-27 Vicor Corporation Power distribution architecture with series-connected bus converter
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EP0155369A1 (en) 1985-09-25
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EP0155369B1 (en) 1988-06-29
DE3472514D1 (en) 1988-08-04

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