RU2490756C2 - Method of demodulating phase-modulated signals and apparatus for realising said method - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: method of demodulating phase-modulated signals involves transmitting the phase-modulated signal to a demodulator made from a linear four-terminal element, a two-electrode nonlinear element and a selective load; the phase-modulated signal is converted to an amplitude-phase-modulated signal; conversion of the phase-modulated signal to an amplitude-phase-modulated signal is performed by transmitting said signal to the right or left slope of the amplitude-frequency curve; the low-frequency component of the amplitude-phase-modulated signal is transmitted to a differentiating or integrating circuit, respectively; a nonlinear element is used to decompose the spectrum of the amplitude-phase-modulated signal into high-frequency and low-frequency components; a low-pass filter is used to a low-frequency information signal, the amplitude of which varies according to the law of variation of the phase of the phase-modulated input signal, wherein the four-terminal element is resistive; a nonlinear element is connected in a transverse circuit between the output of the source of the phase-modulated signal and the input of the four-terminal element; a high-frequency load is connected between the output of the four-terminal element and the low-pass filter in a transverse circuit; conversion of the phase-modulated signal into an amplitude-phase-modulated signal is performed by forming a quasilinear slope of the amplitude-frequency curve of the high-frequency part of the demodulator in a given frequency band by selecting frequency characteristics of imaginary components of resistance of the high-frequency load x_n and the source of the high-frequency signal x₀ using given mathematical expressions. A device for realising said method is also provided.

EFFECT: demodulating a phase-modulated signal without using a reference oscillator.

2 cl, 4 dwg

Description

The invention relates to the field of radio communications and radar and can be used to demodulate phase-shifted, as well as phase-modulated signals.

A known method of demodulating phase-modulated signals (PMS), consisting in the fact that two nonlinear elements simultaneously are fed in phase out of phase high-frequency PMS and in phase high-frequency reference oscillation with a frequency equal to the carrier frequency of the PMS. As a result, the FMS phase changes in time and the constant phase of the reference oscillation is compared, as a result of which the FMS is converted into an amplitude-modulated and phase-modulated signal (AFMS). In this case, the amplitude changes according to the law of phase change. This signal then undergoes the same transformations as in the amplitude demodulator [S. Baskakov Radio circuits and signals. M.: Higher School, 1988, pp. 286-292]. This means that on nonlinear elements the AFMS spectrum is destroyed (decomposed) into low-frequency and high-frequency components. Then, using the low-pass filter, a low-frequency component is extracted, the amplitude of which changes according to the law of phase change of the input FMS. Then, using the separation capacitance included in the longitudinal chain (sequentially), the constant component that occurs on non-linear elements as a result of interaction with the AFMS is eliminated. After that, low-frequency vibrations containing useful information are allocated to the low-frequency load.

The disadvantage of this method and device for its implementation is that in order to isolate a low-frequency signal, the amplitude of which changes in accordance with the law of phase change of the high-frequency PMS, it is necessary to have a reference oscillator. Another disadvantage is the inability to correct the amplitude modulation coefficient of AFMS, which when passing through resonant circuits leads to a decrease in this characteristic, that is, to the well-known phenomenon of partial demodulation of AFMS or to a decrease in noise immunity. The main disadvantage is the small size of the quasilinear portion of the demodulation characteristic due to the use of an insufficient number of oscillatory circuits and the lack of selection of their parameters according to the criterion of converting FMS to AFMS in a given frequency band or at a given number of frequencies.

The closest in technical essence and the achieved result (prototype) is a method of demodulating phase-modulated signals, which consists in the fact that to demodulate the FMS, a frequency detector is used, consisting of a cascade-connected amplitude limiter, a frequency-modulated signal (HMS) converter in the amplitude-frequency modulated signal (AFMC) in the form of a parallel oscillatory circuit and a conventional amplitude demodulator. Further, the process of isolating the low-frequency component is carried out in the same way as described above. The peculiarity of using a frequency detector for FMS demodulation is that if the frequency of the FMS carrier signal is located on the right slope of the amplitude-frequency characteristic (AFC) of the circuit, then the low-frequency component is fed to the differentiating circuit. If the frequency of the carrier signal of the FMS is located on the left slope of the frequency response of the circuit, then the low-frequency component is fed to the integrating circuit [S. Baskakov Radio circuits and signals. M.: Higher School, 1988, pp. 286-292]. If necessary, between the source of modulated signals and the nonlinear element or between the nonlinear element and the load include a reactive or resistive four-terminal network for matching and additional signal and interference selection. As a result, at the output of the device, we have a low-frequency oscillation, the amplitude of which changes according to the law of variation of the envelope of the input high-frequency phase-modulated oscillation.

The disadvantage of the method and device for its implementation is that after converting the FMS to AFMS, the amplitude modulation coefficient of the AFMS is not controlled and, as a rule, is insignificant in magnitude, which impairs noise immunity [S. Baskakov. Radio circuits and signals. M.: Higher School, 1988, pp. 286-292. Gonorovsky I.S. Radio circuits and signals. M .: Radio and communications, 1986, p. 247-252]. The main disadvantage is the small size of the quasilinear portion of the demodulation characteristic due to the use of an insufficient number of oscillatory circuits and the lack of selection of their parameters according to the criterion of converting FMS to AFMS in a given frequency band or at a given number of frequencies. In addition, the classical theory of radio circuits suggests that a nonlinear element is purely resistive and inertia-free, and therefore it does not react at all to a change in the frequency and phase of the input signal, but only responds to a change in amplitude. Meanwhile, everyday experience shows that nonlinear elements have internal capacitances and inductances, which have a significant impact on the formation of the dependence of their conductivity (resistance or elements of the matrix of conductivities or resistances) on frequency and phase. This is especially significant with increasing frequency, which is currently mainly sought by designers of new systems and means of radio communications.

The technical result of the invention is to provide FMS demodulation without using a reference oscillation generator with FMS to AFMS conversion using the high-frequency part of the demodulator for a given amplitude modulation coefficient of AFMS at high frequency load while increasing the frequency band in which this conversion is possible, which increases the noise immunity of the receiver.

1. The specified result is achieved by the fact that in the method of demodulating phase-modulated signals, which consists in the fact that the phase-modulated signal is supplied to a demodulator made of a linear four-terminal device, a two-electrode nonlinear element and a selective load, the phase-modulated signal is converted into an amplitude-phase-modulated signal, conversion of the phase-modulated signal to the amplitude-phase-modulated signal is carried out by applying this signal to the right or left slope of the frequency response, the low-frequency component The amplitude-phase-modulated signal is fed to a differentiating or integrating circuit, respectively, using a nonlinear element, the spectrum of the amplitude-phase-modulated signal is destroyed into high-frequency and low-frequency components, and a low-frequency information signal is isolated using a low-pass filter, the amplitude of which changes according to the law of phase change of the phase-modulated input signal , additionally, the four-terminal device is resistive, a nonlinear element is included in the transverse circuit between the output a phase-modulated signal source house and a four-terminal input, a high-frequency load is introduced into the transverse circuit between the four-terminal output and the low-pass filter, the phase-modulated signal is converted into an amplitude-phase-modulated signal by forming a quasilinear slope of the frequency response of the high-frequency part of the demodulator in a given frequency band due to the choice of frequency components high load resistances x _n and the source RF signal x ₀ using follows uyuschih mathematical expressions:

, $$\beta =\frac{b}{a}$$

, $$\gamma =\frac{c}{a}$$

- заданные отношения элементов классической матрицы передачи a, b, c, d резистивного четырехполюсника; m, φ - заданные зависимости модуля и фазы передаточной функции от частоты из условия формирования квазилинейных склонов АЧХ и ФЧХ с заданной крутизной и в заданной полосе частот; g, b - заданные зависимости действительной и мнимой составляющих проводимости двухполюсного нелинейного элемента от частоты; r₀, r_n - заданные зависимости действительных составляющих сопротивлений источника высокочастотного сигнала и высокочастотной нагрузки от частоты. $$\alpha =\frac{d}{a}$$

, $$\beta =\frac{b}{a}$$

, $$\gamma =\frac{c}{a}$$

- the given relations of the elements of the classical transmission matrix a , b, c, d of the resistive four-terminal network; m, φ are the given dependences of the module and phase of the transfer function on frequency from the condition for the formation of quasilinear slopes of the frequency response and phase response with a given slope and in a given frequency band; g, b are given frequency dependences of the real and imaginary components of the conductivity of a bipolar nonlinear element; r ₀ , r _n are the given dependences of the actual components of the resistances of the source of the high-frequency signal and the high-frequency load on frequency.

2. The specified result is achieved by the fact that in the device for demodulating phase-modulated signals connected between the source of phase-modulated signals and the low-frequency load and consisting of a converter of phase-modulated signals into an amplitude-phase-modulated signal in the form of a linear four-terminal, two-electrode nonlinear element, low-pass filter, an additional four-terminal in the form of a U-shaped connection of three resistive bipolar, a nonlinear element is included in the transverse circuit between the output house source phase modulated signal and the input of the quadrupole, between the output quadripole and lowpass filter the transverse chain introduced the high-load, and imaginary components of the high frequency load resistances x _n and the source RF signal x ₀ realized reactive two-terminal networks in the form of series-connected two parallel resonant circuits, the values of the parameters of which L _1k , C _1k and L _2k , C _2k are selected from the condition for ensuring the operation of converting a phase-modulated signal into amplitude-phase-modulated signal by forming a quasilinear slope of the frequency response of the high-frequency part of the demodulator in a given frequency band using the following mathematical expressions:

; $$\beta =\frac{r_2{r}_3}{r_2+r_3}$$

; $$\gamma =\frac{r_1+r_2+r_3}{\left(r_2+r_3\right)r_1}$$

- заданные отношения элементов классической матрицы передачи $$a=1+\frac{r_2}{r_3}$$

; $$c=\frac{r_1+r_2+r_3}{r_1{r}_3}$$

; $$d=1+\frac{r_2}{r_1}$$

резистивного четырехполюсника, равные на четырех заданных частотах ω_n=2πf_n; n=1, 2, 3, 4 - номер частоты; r₁, r₂, r₃ - заданные значения сопротивлений резистивных двухполюсников П-образного соединения; m_n, φ_n - заданные значения модуля и фазы передаточной функции на четырех заданных частотах из условия формирования квазилинейных склонов АЧХ и ФЧХ с заданной крутизной и в заданной полосе частот; g_n, b_n - заданные значения действительной и мнимой составляющих проводимости двухполюсного нелинейного элемента на четырех заданных частотах; r_0n, r_нn - заданные значения действительных составляющих сопротивлений источника высокочастотного сигнала и высокочастотной нагрузки на четырех заданных частотах; k=0,н - индекс, характеризующий действительные и мнимые составляющие сопротивлений источника высокочастотного сигнала и высокочастотной нагрузки; x_kn - оптимальные значения мнимых составляющих сопротивлений источника высокочастотного сигнала и высокочастотной нагрузки на четырех заданных частотах. $$\alpha =\frac{r_{\mathrm{one}}+r_2}{r_{\mathrm{one}}+{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3}$$

; $$\beta =\frac{r_2{r}_3}{{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3}$$

; $$\gamma =\frac{r_{\mathrm{one}}+{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_3}{\left({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_3\right)r_{\mathrm{one}}}$$

- given relations of elements of the classical transmission matrix $$a=\mathrm{one}+\frac{r_2}{{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3}$$

; $$c=\frac{r_{\mathrm{one}}+{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_3}{r_{\mathrm{one}}{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3}$$

; $$d=\mathrm{one}+\frac{r_2}{r_{\mathrm{one}}}$$

resistive four-terminal, equal at four given frequencies ω _n = 2πf _n ; n = 1, 2, 3, 4 - frequency number; r ₁ , r ₂ , r ₃ - set resistance values of resistive two-terminal U-shaped connections; m _n , φ _n are the specified values of the module and phase of the transfer function at four given frequencies from the conditions for the formation of quasilinear slopes of the frequency response and phase response with a given slope and in a given frequency band; g _n , b _n - given values of the real and imaginary components of the conductivity of a bipolar nonlinear element at four given frequencies; r _0n , r _nn - set values of the real components of the resistance of the source of the high-frequency signal and high-frequency load at four given frequencies; k = 0, n is the index characterizing the real and imaginary components of the resistances of the source of the high-frequency signal and high-frequency load; x _kn are the optimal values of the imaginary components of the resistances of the high-frequency signal source and the high-frequency load at four given frequencies.

Figure 1 shows a diagram of a device for demodulating phase-modulated RF signals (prototype).

Figure 2 shows the structural diagram of the proposed device according to claim 2.

Figure 3 shows a diagram of a four-terminal device of the proposed device according to claim 2.

Figure 4 shows a diagram of each of the two-terminal circuits, forming the imaginary components of the resistance of the source of high-frequency signal and high-frequency load of the proposed device according to claim 2.

The prototype device (Fig. 1) contains a phase-modulated signal source 1, a four-terminal 2, a nonlinear element 3, a low-pass filter 4 on the elements R, C, a separation capacitance 5 on the element C _p and a low-frequency load 6 on the elements R _n , C _n .

The principle of operation of the device demodulation phase-modulated signals (prototype) is as follows.

Phase-modulated signal from the source 1 is fed to the demodulator (figure 1). The principle of operation of the device that implements this method is that, using a reactive four-terminal 2, which is a parallel oscillatory circuit and connected between the FMS source and the nonlinear element, the FMS is converted into AFMS, using the nonlinear element 3, the AFMS spectrum is destroyed into high-frequency and low-frequency components. The latter are allocated using a low-pass filter 4 and enter the low-frequency load 6. The separation capacitance 5 eliminates the constant component. As a result, at the output of the device we have a low-frequency oscillation, the amplitude of which changes according to the law of the envelope of the high-frequency AFMS, that is, according to the law of the phase change of the input FMS, which changes according to the law of the amplitude of the primary signal.

The disadvantage of the method and device for its implementation is that when passing the FMS through the specified circuit, after converting the FMS to AFMS, the amplitude modulation coefficient of the latter is negligible. This is due to the large width of the FMS spectrum or to the low quality factor of the circuit. On the other hand, the narrower the bandwidth of the circuit, the greater the distortion of the received signal. In the general case, the amplitude modulation coefficient AFMS decreases and becomes, as a rule, unknown. Thus, the main disadvantage is the insolubility within the prototype of the contradiction of the requirements for increasing the steepness and frequency band of the quasilinear slope of the frequency response of the high-frequency part of the demodulator.

The high-frequency part (before the low-pass filter) of the structural diagram of the generalized proposed device according to claim 2 (Fig. 2) consists of an FMS source 1, a resistive four-terminal 2, a two-electrode nonlinear element 3, and a high-frequency load 7. The low-frequency part of the structural circuit contains a low-pass filter 4 , dividing capacitance 5 and low-frequency load 6. Resistive four-terminal 2 is made in the form of a U-shaped connection of three resistive two-terminal (figure 3), the resistances of which can be selected arbitrarily or from Akiho any physical considerations. The frequency dependences of the imaginary components of the resistances of the source of the high-frequency signal and the high-frequency load are selected from the conditions for the formation of a quasilinear slope of the frequency response of the demodulator with the given values of the transfer function modules at four given frequencies of the required frequency band. The implementation of these dependencies is carried out by reactive two-terminal devices in the form of two parallel parallel oscillatory circuits connected in series (Fig. 4), the parameter values of which L _1k , C _1k and L _2k , C _2k are selected from the condition for ensuring the operation of converting a phase-modulated signal into an amplitude-phase-modulated signal by forming a quasilinear the slope of the frequency response of the high-frequency part of the demodulator in a given frequency band using certain mathematical expressions. The real resistances of the source of the high-frequency signal and the high-frequency load can be purely active (this is often found in practice). In this case, the imaginary components of the resistances of the source of the high-frequency signal and the high-frequency load, implemented in this way, are connected in series to the corresponding active resistances. The implementation of the four-pole resistive is an additional opportunity to increase the quasilinear portion of the slope of the frequency response, since the parameters of the resistive elements are independent of the frequency in a very large frequency band.

The principle of operation of this device is that when the FMS is supplied from source 1 with resistance z ₀ , a left or right slope of the frequency response of the demodulator with the given values of the transfer function modules at four given frequencies of the required frequency band will be formed as a result of a special choice of reactive two-terminal elements. This provides a given coefficient of amplitude modulation AFMS in a larger frequency band, which increases the noise immunity of the receiver. At the same time, the AFMS spectrum is destroyed using a nonlinear element 3 connected between the FMS source and the four-terminal network. As a result, the low-frequency oscillation, the amplitude of which varies according to the law of phase change of the input FMS, is allocated to the low-frequency load 6.

Let us prove the feasibility of implementing these properties.

Let the dependences of the real components of the complex load resistances z _n = r _n + jx _n and the FMS source z ₀ = r ₀ + jx ₀ on frequency be known. The dependence of the conductivity of a bipolar controlled element y = g + jb at a selected operating point on frequency is also known. Hereinafter, the argument (frequency) is omitted for simplicity. Thus, a nonlinear element is characterized by a transfer matrix:

$$A_{n\mathrm{uh}}=\left|\begin{array}{cc}\mathrm{one}& 0\\ \mathrm{at}& \mathrm{one}\end{array}\right|.\left(\mathrm{one}\right)$$

Resistive four-terminal (RF) is described by the transfer matrix:

$$A=a\left|\begin{array}{cc}\mathrm{one}& \beta \\ \gamma & \alpha \end{array}\right|,\left(2\right)$$

; $$\beta =\frac{b}{a}$$

; $$\gamma =\frac{c}{a}$$

; a, b, c, d - элементы классической матрицы передачи.Where $$\alpha =\frac{d}{a}$$

; $$\beta =\frac{b}{a}$$

; $$\gamma =\frac{c}{a}$$

; a , b, c, d - elements of the classical transmission matrix.

The general normalized classical generator / modulator transfer matrix is obtained by multiplying matrices (1) and (2) taking into account normalization conditions:

$$A=a\left|\begin{array}{cc}\sqrt{\frac{z_n}{z_0}}& \beta \frac{\mathrm{one}}{\sqrt{z_0{z}_n}}\\ \left(\mathrm{at}+\gamma \right)\sqrt{z_0{z}_n}& \left(\beta y+\alpha \right)\sqrt{\frac{z_0}{z_n}}\end{array}\right|.\left(3\right)$$

Using the well-known connection of the elements of the scattering matrix with the elements of the transmission matrix (3) [Feldstein A.L., Yavich L.R. Microwave four-terminal and eight-terminal synthesis. M .: Communication, 1965. 40 s], we obtain the expression for the transmission coefficient of the high-frequency part (before the low-pass filter) of the demodulator S ₂₁ :

$${\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="00000038.png"\; state="\{\{state\}\}">S</figure-callout>}}_{2\mathrm{one}}=\frac{2\sqrt{z_0{z}_n}}{a\left[\left(z_n+\beta \right)\left({\mathrm{<figure-callout\; id="2"\; label="g"\; filenames="00000038.png"\; state="\{\{state\}\}">g</figure-callout>}}_{220}+\mathrm{<figure-callout\; id="2"\; label="j\; b"\; filenames="00000038.png"\; state="\{\{state\}\}">j\; </figure-callout>}{b}_{}{}_{220}\right)+\left(\gamma z_n+\alpha \right)z_0\right]},\left(\mathrm{four}\right)$$

where g ₂₂₀ = 1 + gr ₀ -bx ₀ ; b ₂₂₀ = gx ₀ + br ₀ .

The root in (4) can be represented as a complex number a + jb, where

; $$b=\pm \sqrt{\frac{\sqrt{x^2+{у}^2-x}}{2}}$$

; x=r₀x_н+r₀x_н; у=r₀x_н+r₀x_н. $$a=\pm \sqrt{\frac{\sqrt{x^2+{\mathrm{at}}^2+x}}{2}}$$

; $$b=\pm \sqrt{\frac{\sqrt{x^2+{\mathrm{at}}^2-\mathrm{<figure-callout\; id="2"\; label="x"\; filenames="00000038.png"\; state="\{\{state\}\}">x</figure-callout>}}}{2}}$$

; x = r ₀ x _n + r ₀ x _n ; y = r ₀ x _n + r ₀ x _n

последнее выражение изменяется a=r_n; b=x_n.After denormalizing the transmission coefficient (4) by multiplying by $$\sqrt{\frac{z_n}{z_0}}$$

the last expression changes a = r _n ; b = x _n .

.The denormalized transmission coefficient is associated with a physically feasible transfer function as follows $$H=\frac{\mathrm{one}}{2}{\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="00000038.png"\; state="\{\{state\}\}">S</figure-callout>}}_{2\mathrm{one}}$$

.

Let it be required to determine the frequency dependences of the imaginary components of the load resistances x _n and the FMS source x ₀ , optimal according to the criterion for ensuring the given dependences of the module m and phase φ of the transfer function on frequency in the interests of generating the frequency response and phase response of the high-frequency part of the demodulator with the required slope and in a given frequency band:

$${\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="00000038.png"\; state="\{\{state\}\}">S</figure-callout>}}_{2\mathrm{one}}=m\left(cos\left(\varphi \right)+jsin\left(\varphi \right)\right).\left(5\right)$$

Substituting (4) into (5) and after simple transformations and dividing the complex equation into real and imaginary parts, we obtain a system of two algebraic equations equivalent to the given dependences of the module m and phase φ of the transfer function on frequency:

$$r_n-ma\left(Rcos\varphi -Isin\varphi \right)=0;x_n-ma\left(Icos\varphi +Rsin\varphi \right)=0,\left(6\right)$$

where R = (r _n + β) g ₂₂₀ + r ₀ (α + γr _n ) -x _n b ₂₂₀ -γx ₀ x _n ; I = (r _n + β) b ₂₂₀ + x ₀ (α + γr _n ) + x _n g ₂₂₀ + yx _n r ₀ .

The solution of system (6) with respect to x ₀ , x _n has the meaning of the dependences of the imaginary components of the resistance of the signal source and the high-frequency load on the frequency, which are optimal according to the criterion of providing the specified frequency response and phase response (approximating functions):

To realize the optimal characteristics (7) by interpolation, it is necessary to form two-terminal networks with resistances x ₀ , x _n of at least N (the number of interpolation frequencies) of the reactive elements, find expressions for their resistances, equate them with the optimal values of the two-terminal resistances at given frequencies determined by formulas (7), and solve the thus formed system of N equations with respect to N selected parameters of the reactive elements. The values of the parameters of the remaining elements can be chosen arbitrarily or on the basis of any other physical considerations, for example, from the condition of physical realizability.

In accordance with this algorithm, mathematical expressions are obtained for determining the values of the parameters L _1k , C _1k and L _2k , C _{2k of a} reactive two-terminal device in the form of two parallel parallel circuits (Fig. 4), optimal by the criterion for ensuring the indicated conditions for the coincidence of real resistances with characteristics ( 7) at four frequencies:

The original system of equations:

Implementation of optimal approximations of the frequency characteristics (7) using (8), (9) provides an increase in the frequency band within which the slope of the frequency response differs from the linear one by no more than a certain small value, since the conditions for the coincidence (9) of the real frequency characteristics (8 ) with optimal (7) at four frequencies of a given frequency band. This allows for a reasonable choice of the positions of the given frequencies relative to each other ω ₁ -ω ₂ , ω ₁ -ω ₃ , ω ₁ -ω ₄ , ω ₂ -ω ₃ , ω ₂ -ω ₄ , ω ₃ -ω ₄ to expand the quasilinear section of the slope Frequency response of the high-frequency part of the demodulator and phase demodulation characteristics. In this case, the index n (frequency number) must be taken into account in the notation of all frequency-dependent quantities.

Any typical circuit with known elements of the classical transmission matrix can be chosen as a resistive four-terminal [Feldstein A.L., Yavich L.R. Microwave four-terminal and eight-terminal synthesis. M .: Communication, 1965. 40 s], for example, a U-shaped connection of three resistive two-terminal devices (Fig. 3), for which:

$$a=\mathrm{one}+\frac{r_2}{{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3};b={\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2;c=\frac{r_{\mathrm{one}}+{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_3}{r_{\mathrm{one}}{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3};d=\mathrm{one}+\frac{r_2}{r_{\mathrm{one}}};\alpha =\frac{r_{\mathrm{one}}+r_2}{{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3};\beta =\frac{r_2{r}_3}{{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+{\mathrm{<figure-callout\; id="3"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_3};\gamma =\frac{r_{\mathrm{one}}+{\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_3}{\left({\mathrm{<figure-callout\; id="2"\; label="r"\; filenames="00000038.png"\; state="\{\{state\}\}">r</figure-callout>}}_2+r_3\right)r_{\mathrm{one}}}.\left(\mathrm{one}0\right)$$

The resistance values r ₁ , r ₂ , r ₃ can be chosen arbitrarily or on the basis of any other physical considerations, for example, from the conditions of physical realizability of the parameters determined using (9), or from the condition of an additional increase in the frequency band within which the specified deviation of the frequency response slope from the linear dependence of the transfer function module on frequency.

The proposed technical solutions are new, because the method that provides the left or right slope of the frequency response of the demodulator with the given dependences of the module and phase of the transfer function of the FMS demodulation device on the frequency in a given frequency band is unknown from publicly available information, which allows the conversion of FMS to AFMS with a given amplitude coefficient AFMS modulation in a larger frequency band, and the demodulation device consists of a nonlinear two-electrode element connected between the output m of the FMS source and the input of the resistive four-terminal, between the four-terminal output and the low-pass filter a high-frequency load is introduced into the transverse circuit, while the four-terminal is made in the form of a U-shaped connection of three resistive two-terminal, and the imaginary components of the high-frequency load resistance x _n and the high-frequency signal source x ₀ implemented by reactive two-terminal devices in the form of two parallel parallel oscillatory circuits connected in series, whose parameter values L _1k , C _1k and L _2k , C _2k are selected as about the corresponding mathematical expressions.

The proposed technical solutions have an inventive step, since it does not explicitly follow from the published scientific data and known technical solutions that the claimed sequence of operations (performing a four-terminal resistive in the form of the above circuit, including a two-pole nonlinear element between the output of the FMS source and the input of the resistive four-terminal into the transverse circuit , the introduction of a high-frequency load between the four-terminal network and the low-pass filter in the transverse circuit, the implementation of imaginary resistance resistances and a high-frequency signal source by reactive two-pole circuits in the form of two parallel parallel oscillatory circuits whose parameter values L _1k , C _1k and L _2k , C _2k are selected according to the corresponding mathematical expressions from the condition of ensuring a given amplitude modulation coefficient AFMS converts the FMS to AFMS without the presence of a reference signal source in a larger frequency band.

The proposed technical solutions are practically applicable, since semiconductor diodes (parametric diodes, p-i-n diodes, LPD, Gunn diodes, etc.), inductances and capacitances formed in the claimed reactive two-terminal circuit can be used for their implementation. The resistance values of reactive bipolar, inductances and capacitances can be determined using mathematical expressions given in the claims.

The technical and economic efficiency of the proposed device is to simultaneously provide the operation of converting the input FMS to AFMS with a given amplitude modulation coefficient due to the formation of a quasilinear frequency response slope with a given slope in a larger frequency band, which helps to increase noise immunity.

Claims (2)

1. The method of demodulating phase-modulated signals, which consists in the fact that the phase-modulated signal is fed to a demodulator made of a linear four-terminal, two-electrode nonlinear element and a selective load, the phase-modulated signal is converted into an amplitude-phase-modulated signal, the phase-modulated signal is converted to the amplitude-phase-modulated signal by applying of this signal to the right or left slope of the frequency response, the low-frequency component of the amplitude-phase modulated signal fed to a differentiating or integrating circuit, respectively, using a non-linear element, the spectrum of the amplitude-phase modulated signal is destroyed into high-frequency and low-frequency components, an information low-frequency signal is isolated using a low-pass filter, the amplitude of which changes according to the law of phase change of the phase-modulated input signal, characterized in that the quadripole is made resistive, the nonlinear element is included in the transverse circuit between the output of the phase-modulated source with drove and the input of the four-terminal, between the output of the four-terminal and the low-pass filter, a high-frequency load is introduced into the transverse circuit, the phase-modulated signal is converted into an amplitude-phase-modulated signal by forming a quasilinear slope of the frequency response of the high-frequency part of the demodulator in a given frequency band by choosing the frequency characteristics of the imaginary component of the load resistance x _n and a high-frequency signal source x ₀ using the following mathematical expressions

$$\alpha =\frac{d}{a}$$

, $$\beta =\frac{b}{a}$$

, $$\gamma =\frac{c}{a}$$

- the given relations of the elements of the classical transmission matrix a , b, c, d of the resistive four-terminal network; m, φ are the given dependences of the module and phase of the transfer function on frequency from the condition for the formation of quasilinear slopes of the frequency response and phase response with a given slope and in a given frequency band; g, b are given frequency dependences of the real and imaginary components of the conductivity of a bipolar nonlinear element; r ₀ , r _n are the given dependences of the actual components of the resistances of the source of the high-frequency signal and the high-frequency load on frequency. 2. A device for demodulating phase-modulated signals connected between a source of phase-modulated signals and a low-frequency load and consisting of a converter of phase-modulated signals into an amplitude-phase-modulated signal in the form of a linear four-terminal, two-electrode nonlinear element, a low-pass filter, characterized in that the four-terminal is made in the form of a U-shaped connection of three resistive bipolar, a nonlinear element is included in the transverse circuit between the output of the phase-modulated source a signal and a four-terminal input, a high-frequency load is introduced into the transverse circuit between the four-terminal output and the low-pass filter, and the imaginary components of the high-frequency load resistances x _n and the high-frequency signal source x _{0 are} realized by reactive two-poles in the form of two parallel oscillatory circuits connected in series, whose parameter values are L _1k , C _1k and L _2k , C _2k are selected from the condition for ensuring the operation of converting the phase-modulated signal into an amplitude-phase-modulated signal Al by forming a quasilinear slope of the frequency response of the high-frequency part of the demodulator in a given frequency band using the following mathematical expressions

- given relations of elements of the classical transmission matrix

resistive four-terminal, equal at four given frequencies ω _n = 2πf _n ; n = 1, 2, 3, 4 - frequency number; r ₁ , r ₂ , r ₃ - set resistance values of resistive two-terminal U-shaped connections; m _n , φ _n are the specified values of the module and phase of the transfer function at four given frequencies from the conditions for the formation of quasilinear slopes of the frequency response and phase response with a given slope and in a given frequency band; g _n , b _n - given values of the real and imaginary components of the conductivity of a bipolar nonlinear element at four given frequencies; r _0n , r _nn - set values of the real components of the resistance of the source of the high-frequency signal and high-frequency load at four given frequencies; k = 0, n is the index characterizing the real and imaginary components of the resistances of the source of the high-frequency signal and high-frequency load; x _kn are the optimal values of the imaginary components of the resistances of the high-frequency signal source and the high-frequency load at four given frequencies.

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