RU2487463C1 - Method of demodulating phase-modulated and frequency-modulated signals and apparatus for realising said method - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: phase-modulated or frequency-modulated signal is transmitted to a demodulator which is made from a four-terminal element, a nonlinear element, an integrating circuit - low-pass filter, a separating capacitor and a low-frequency load; input signals are further amplitude-modulated by transmitting said signals to the left-side slope of the amplitude-frequency characteristic of the demodulator; the nonlinear element is used to decompose the spectrum of said signals; the low-frequency component is transmitted to the integrating circuit - low-pass filter; the low-pass filter is used to select the information low-frequency signal. The four-terminal element is complex and is made from reactive and resistive elements; the nonlinear element is connected in a longitudinal circuit and relationships between elements of the resistance matrix of the complex four-terminal element and frequency are selected in accordance with given mathematical expressions.

EFFECT: wider field of physical implementation.

2 cl, 4 dwg

Description

The invention relates to the field of radio communications and radar and can be used for demodulation of phase-shifted, phase-modulated, frequency-manipulated and frequency-modulated signals.

A known method of demodulating phase-modulated signals (PMS), consisting in the fact that two nonlinear elements simultaneously are fed in phase out of phase high-frequency PMS and in phase high-frequency reference oscillation with a frequency equal to the carrier frequency of the PMS. As a result, the FMS phase changes in time and the constant phase of the reference oscillation is compared, as a result of which the FMS is converted into an amplitude-modulated and phase-modulated signal (AFMS). In this case, the amplitude changes according to the law of phase change. This signal then undergoes the same transformations as in the amplitude demodulator [S. Baskakov Radio circuits and signals. - M.: Higher School, 1988, pp. 286-292]. This means that on nonlinear elements the AFMS spectrum is destroyed (decomposed) into low-frequency and high-frequency components. Then, using the low-pass filter, a low-frequency component is extracted, the amplitude of which changes according to the law of phase change of the input FMS. Then, using the separation capacitance included in the longitudinal chain (sequentially), the constant component that occurs on non-linear elements as a result of interaction with the AFMS is eliminated. After that, low-frequency vibrations containing useful information are allocated to the low-frequency load.

This method and device can also be used to demodulate frequency-modulated signals (HMS), in which the phase changes according to the law of the integral of frequency. For this, low-frequency oscillations containing useful information must be fed to a differentiating circuit.

The disadvantage of this method and device for its implementation is that in order to isolate a low-frequency signal, the amplitude of which changes in accordance with the law of phase change of the high-frequency PMS, it is necessary to have a reference oscillator. Another disadvantage is the inability to correct the amplitude modulation coefficient of AFMS, which when passing through resonant circuits leads to a decrease in this characteristic, that is, to the well-known phenomenon of partial demodulation of AFMS or to a decrease in noise immunity. The main disadvantage is the small size of the quasilinear portion of the phase demodulation characteristic due to the use of only reactive elements and the lack of selection of their parameters according to the criterion of converting PMF to AFMS in a given frequency band or at a given number of frequencies. In the frequency demodulation mode, the main disadvantage is the small size of the quasilinear portion of the frequency demodulation characteristic due to the use of only reactive elements and the lack of selection of their parameters according to the criterion for converting HMS into amplitude-modulated and HMS (AFM) in a given frequency band or for a given number of frequencies.

The closest in technical essence and the achieved result (prototype) is a method of demodulating phase-modulated and frequency-modulated signals, which consists in the fact that for demodulating the FMS and ChMS, a frequency detector is used, consisting of a cascade-connected amplitude limiter, a ChMS frequency response converter in the form of a parallel oscillatory circuit and conventional amplitude demodulator. Further, the process of isolating the low-frequency component is carried out in the same way as described above. The peculiarity of using this frequency detector for FMS demodulation is that if the frequency of the FMS carrier signal is located on the right slope of the amplitude-frequency characteristic (AFC) of the circuit, then the low-frequency component is fed to the differentiating circuit. If the frequency of the carrier signal of the FMS is located on the left slope of the frequency response of the circuit, then the low-frequency component is fed to the integrating circuit. As a result, at the output of the device, we have a low-frequency oscillation, the amplitude of which changes according to the law of the phase change of the input high-frequency phase-modulated oscillation.

The peculiarity of using this frequency detector for demodulating an HMS is that the frequency of the HMS carrier signal is located on the left slope of the frequency response of the circuit. In this case, the amplitude of the frequency response varies according to the law of frequency variation [S. Baskakov. Radio circuits and signals. - M.: Higher School, 1988, pp. 286-292]. If necessary, between the source of modulated signals and the nonlinear element or between the nonlinear element and the load include a reactive or resistive four-terminal network for matching and additional signal and interference selection. As a result, at the output of the device, we have a low-frequency oscillation, the amplitude of which changes according to the law of changing the frequency of the input high-frequency frequency-modulated oscillation.

The disadvantage of the method and device for its implementation is that after converting the FMS to AFMS, the amplitude modulation coefficient of the AFMS is not controlled and, as a rule, is insignificant in magnitude, which impairs noise immunity [S. Baskakov. Radio circuits and signals. - M.: Higher School, 1988, pp. 286-292. Gonorovsky I.S. Radio circuits and signals. - M.: Radio and Communications, 1986, pp. 247-252]. The main disadvantage is the small size of the quasilinear portion of the demodulation characteristic due to the use of only reactive elements and the lack of selection of their parameters according to the criterion of converting PMF to AFMS in a given frequency band or at a given number of frequencies. In the frequency demodulation mode, the main disadvantage is the small size of the quasilinear portion of the frequency demodulation characteristic due to the use of only reactive elements and the lack of selection of their parameters according to the criterion for converting HMS into amplitude-modulated and HMS (AFM) in a given frequency band or for a given number of frequencies.

In addition, the classical theory of radio circuits suggests that a nonlinear element is purely resistive and inertia-free, and therefore it does not react at all to a change in the frequency and phase of the input signal, but only responds to a change in amplitude. Meanwhile, everyday experience shows that nonlinear elements have internal capacitances and inductances, which have a significant impact on the formation of the dependence of their conductivity (resistance or elements of the matrix of conductivities or resistances) on frequency and phase. This is especially significant with increasing frequency, which is currently mainly sought by designers of new systems and means of radio communications.

The next important drawback of all the above methods and devices is that all elements of the four-terminal devices (matching devices) are made reactive, which is associated with the desire of developers not to introduce additional losses by using complex two-terminal devices based on both reactive and resistive elements. When using only reactive or only resistive elements in matching devices, it is not always possible to provide matching conditions for the criterion of providing the required values of the modules and phases of transmission coefficients in two states of a controlled nonlinear element determined by two frequencies of the input FMS or input HMS, in the interests of forming a given slope of the frequency response, since they have certain areas of physical realizability (areas of change in the real and imaginary components of the resistance of the source s drove and loads), within which these coordination conditions are realized (A. Golovkov. Complex electronic devices. - M.: Radio and communications, 1996. - 128 p.).

The technical result of the invention is the expansion of the areas of physical feasibility as the areas of change of the real and imaginary components of the resistance of the signal source and load, within which the required values of the modules and phases of the transmission coefficients in two states, determined by the two frequencies of the input FMS or HMS, are simultaneously provided, in the interests of the formation of a given slope Frequency response for converting PMS to AFMS or ChMS to AFC due to optimization of the circuit and parameter values of the integrated quadrupole and at the same time increasing the frequency band in which this conversion is possible, which increases the noise immunity of the receiver. The possibility of changing the option of including a nonlinear element relative to the matching complex quadrupole further expands the field of physical feasibility.

1. The specified result is achieved by the fact that in the known method of demodulating phase-modulated and frequency-modulated signals, consisting in the fact that the phase-modulated or frequency-modulated signal is fed to a demodulator made of a four-terminal device, a nonlinear element, a low-pass filter, a separation capacitance, and a low-frequency load , the phase-modulated signal is converted into an amplitude-phase-modulated signal, the frequency-modulated signal is converted into an amplitude-frequency-modulated signal, converted The phase-modulated signal is converted into an amplitude-phase-modulated signal and the frequency-modulated signal into an amplitude-frequency-modulated signal by supplying these signals to the left slope of the frequency response of the demodulator, using a non-linear element, the spectrum of the amplitude-phase-modulated signal and the amplitude-frequency-modulated signal are destroyed by high-frequency ones and low-frequency components, the low-frequency component is fed to an integrating low-pass filter circuit, using a low-pass filter, and formation low-frequency signal, the amplitude of which varies according to the law of the phase change of the phase-modulated input signal or according to the law of change of the frequency of the frequency-modulated signal, eliminate the DC component using an isolation capacitor, in addition, the four-terminal is made complex of reactive and resistive elements, the output of the phase-modulated or frequency-modulated signal source connected to the input of a four-terminal, a nonlinear element is included in the longitudinal circuit between the output of four of the pole and the high-frequency load introduced into the transverse circuit in front of the low-pass filter, the specified frequency dependences of the transfer function of the high-frequency part of the demodulator in the interest of forming a given slope of the amplitude-frequency characteristic provide by frequency selection of the element z _{11 of} the complex four-terminal resistance matrix using the following mathematical expressions:

, $$z_{\mathrm{eleven}}=\frac{z_{21}^2+z_{21}{D}_{\mathrm{one}}}{E_{\mathrm{one}}-z_{22}}-z_0$$

,

; Е₁=z_н+z; z₂₁, z₂₂ - заданные зависимости соответствующих элементов матрицы сопротивлений комплексного четырехполюсника от частоты в заданной полосе частот; m₂₁, φ₂₁ - заданные зависимости модуля и фазы передаточной функции от частоты в заданной полосе частот; z - заданная зависимость комплексного сопротивления двухполюсного нелинейного элемента от частоты в заданной полосе частот; z₀, z_n - заданные зависимости комплексных сопротивлений источника фазомодулированного или частотно-модулированного сигнала и высокочастотной нагрузки от частоты в заданной полосе частот.Where $$D_{\mathrm{one}}=\frac{z_n}{m_{21}\left(\cos {\varphi }_{21}+j\sin {\varphi }_{21}\right)}$$

; E ₁ = z _n + z; z ₂₁ , z ₂₂ are the given dependences of the corresponding elements of the resistance matrix of a complex four-terminal network on frequency in a given frequency band; m ₂₁ , φ ₂₁ - the given dependence of the module and phase of the transfer function on frequency in a given frequency band; z is a given dependence of the complex resistance of a bipolar nonlinear element on frequency in a given frequency band; z ₀ , z _n are the given dependences of the complex resistances of the source of the phase-modulated or frequency-modulated signal and the high-frequency load on the frequency in a given frequency band.

2. The specified result is achieved by the fact that in the known device for demodulating phase-modulated and frequency-modulated signals, consisting of a source of phase-modulated or frequency-modulated signal, four-terminal, two-electrode nonlinear element, low-pass filter, separation capacitance and low-frequency load, an additional four-terminal is made complex in in the form of an overlapped T-shaped connection of four complex two-terminal devices, the output of a phase-modulated or frequency-modulated source annogo signal connected to the input of the quadrupole, the nonlinear element is included in a longitudinal path between the output quadripole and introduced into the transverse chain before the lowpass filter high load, the first bipole overlapped T-fitting is formed of serially connected first resistive two-terminal resistance R _1, with the capacity of the capacitor C and in parallel connected to each other of the second resistive two-terminal with resistance R ₂ and coils with inductance L, the parameter in the first bipolar of an overlapped T-shaped connection are defined in accordance with the following mathematical expressions:

- оптимальные значения комплексного сопротивления первого комплексного двухполюсника перекрытого Т-образного соединения на двух частотах; $$D_1=\frac{z_{нn}}{m_{21n}\left(\cos {\varphi }_{21n}+j\sin {\varphi }_{21n}\right)}$$

; Е₁=z_нn+z_n; Z_2n, Z_3n, Z_4n - заданные значения комплексного сопротивления второго, третьего и четвертого комплексных двухполюсников перекрытого Т-образного соединения на двух частотах; m_21n, φ_21n - заданные значения модулей и фаз передаточной функции высокочастотной части демодулятора на двух частотах; z_n - заданные значения комплексного сопротивления двухполюсного нелинейного элемента на двух частотах; z_0n, z_nn - заданные значения комплексных сопротивлений источника фазомодулированного или частотно-модулированного сигнала и высокочастотной нагрузки на двух частотах; ω_1,2=2πf_1,2; n=1, 2 - номера заданных двух частот f_1,2.r ₁ , r ₂ , x ₁ , x ₂ are the optimal values of the real and imaginary components of the resistance of the first complex two-terminal terminal of a closed T-shaped connection at two frequencies; $$Z_{\mathrm{one}n}=r_n+j{x}_n=\frac{\left(Z_{3n}+Z_{\mathrm{four}n}\right)\left[Z_{2n}\left(D_{\mathrm{one}}-E_{\mathrm{one}}-z_{0n}\right)-E_{\mathrm{one}}{z}_{0n}\right]-Z_3{Z}_{\mathrm{four}n}\left(Z_{2n}+z_{0n}\right)}{E_{\mathrm{one}}\left(Z_{2n}+Z_{3n}+Z_{\mathrm{four}n}+z_{0n}\right)+\left(Z_{2n}+Z_{3n}\right)\left(z_{0n}-D_{\mathrm{one}}+Z_{\mathrm{four}}{}_n\right)}$$

- the optimal values of the complex resistance of the first complex two-pole terminal of a blocked T-shaped connection at two frequencies; $$D_{\mathrm{one}}=\frac{z_{nn}}{m_{21n}\left(\cos {\varphi }_{21n}+j\sin {\varphi }_{21n}\right)}$$

; E ₁ = z _nn + z _n ; Z _2n , Z _3n , Z _4n - setpoints of the complex resistance of the second, third and fourth complex two-terminal circuits of the closed T-shaped connection at two frequencies; m _21n , φ _21n - given values of the modules and phases of the transfer function of the high-frequency part of the demodulator at two frequencies; z _n - set values of the complex resistance of a bipolar nonlinear element at two frequencies; z _0n , z _nn are the set values of the complex resistances of the source of the phase-modulated or frequency-modulated signal and the high-frequency load at two frequencies; ω _1,2 = 2πf _1,2 ; n = 1, 2 - numbers of the given two frequencies f _1,2 .

Figure 1 shows a diagram of a device for demodulating phase-modulated and frequency-modulated signals (prototype) that implements the prototype method.

Figure 2 shows the structural diagram of the proposed device according to claim 2, which implements the proposed method according to claim 1.

Figure 3 shows a diagram of a complex quadrupole of the proposed device according to claim 2.

Figure 4 shows a diagram of the first complex two-terminal, which is part of a complex four-terminal device of the proposed device according to claim 2

The prototype device (figure 1) contains a source 1 of phase-modulated and frequency-modulated signals, a four-terminal 2, a nonlinear element 3, a low-pass filter 4 on the elements R, C, a separation capacitance 5 on the element C _p and a low-frequency load 6 on the elements R _n , With _n .

The principle of operation of the device demodulation phase-modulated and frequency-modulated signals (prototype) is as follows.

Phase-modulated or frequency-modulated signal from the source 1 is fed to the demodulator (figure 1). The principle of operation of a device that implements this method is that, using a reactive four-terminal 2, which is a parallel oscillatory circuit and connected between a PMS or PMS source and a non-linear element, the PMS is converted into AFMS or HMS into AFM, using the non-linear element 3, the spectrum is destroyed AFMS or AFMS for high-frequency and low-frequency components. The latter are allocated using a low-pass filter (integrating circuit) 4 and enter the low-frequency load 6. The separation capacitance 5 eliminates the constant component. As a result, at the output of the device, we have a low-frequency oscillation, the amplitude of which changes according to the law of the envelope of the high-frequency AFMS (AFMS), that is, according to the law of the phase change of the input FMS (frequency of the input HMS), which changes according to the law of the amplitude of the primary signal. The disadvantages of the method and device for its implementation are described above.

The high-frequency part (before the low-pass filter) of the structural diagram of the generalized device according to claim 2 (Fig. 2) consists of a cascade-connected source of PMS or ChMS 1, a complex four-terminal 2, a two-electrode nonlinear element 3 (included in the longitudinal circuit) and a high-frequency load 7. The low-frequency part of the structural diagram contains a low-pass filter 4, a separation capacitance 5 and a low-frequency load 6. The complex four-terminal (CFC) 2 is made in the form of an overlapped T-shaped connection of four complex two-pole yusnikov with resistance Z ₁ - 8, Z ₂ - 9, Z ₃ - 10, Z ₄ - 11 (figure 3). The frequency dependences of element z _{11 of} the KH 2 resistance matrix are selected from the condition for the formation of a quasilinear slope of the frequency response of the demodulator with the given values of the transfer function modules at two given frequencies of the required frequency band. These dependencies were realized by selecting an RF circuit in the form of an overlapped T-shaped link, the optimal frequency dependence of the first complex two-terminal - 8 of this link, and realizing this frequency dependence by choosing the first complex two-terminal scheme from the series-connected first resistive two-terminal with resistance R ₁ - 12, a capacitor with capacity C - 13 and connected in parallel to each other of the second resistive bipolar with resistance R ₂ - 14 and coils with inductance L - 15 (figure 4) and the choice of values of the parameters R ₁ , R ₂ , L, C from the conditions for ensuring the operation of converting the FMS to AFMS (HMS to AFMS) by forming a quasilinear slope of the frequency response of the high-frequency part of the demodulator in a given frequency band using certain mathematical expressions.

The principle of operation of this device is that when the FMS is supplied from source 1 with resistance z ₀ as a result of a special choice of the values of the parameters R ₁ , R ₂ , L, C of the elements of the first complex two-terminal device of the blocked T-shaped link, the left slope of the frequency response of the demodulator with preset values of the transfer function modules at two preset frequencies of the required frequency band. This provides a given coefficient of amplitude modulation AFMS (AFMS) in a larger frequency band, which increases the noise immunity of the receiver. At the same time, the AFMS spectrum (AFMS) is destroyed by a nonlinear element 3 connected between the four-terminal network and the high-frequency load. As a result, the low-frequency oscillation, the amplitude of which changes according to the law of the phase change of the input FMS (according to the law of the frequency change of the input HMS), is allocated to the low-frequency load 6, that is, phase (frequency) demodulation is performed. In this case, the resistances of the source of the FMS or ChMS and the load can be chosen arbitrarily. Let us prove the feasibility of implementing these properties.

Let the dependences of the complex load resistances z _n and the source of the high-frequency (PMS or FMS) signal z ₀ on frequency be known. The dependence of the complex resistance of a bipolar nonlinear element z at a selected operating point is also known. Hereinafter, the argument (frequency) is omitted for simplicity. Thus, a nonlinear element is characterized by a transfer matrix:

$$A_{n\mathrm{uh}}=\left|\begin{array}{cc}\mathrm{one}& z\\ 0& \mathrm{one}\end{array}\right|\left(\mathrm{one}\right)$$

The complex four-terminal network (CC) is described by the transfer matrix:

$$A_{\mathrm{from}f\mathrm{at}}=\left[\begin{array}{cc}\frac{z_{\mathrm{eleven}}}{z_{21}}& -\frac{\left|z\right|}{z_{21}}\\ \frac{\mathrm{one}}{z_{21}}& \frac{-z_{22}}{z_{21}}\end{array}\right],\left(2\right)$$

; z₁₁, z₂₁, z₂₂ - определитель и элементы матрицы сопротивлений СФУ с учетом условия взаимности z₁₂=-z_2l [Фельдштейн А.Л., Явич Л.Р. Синтез четырехполюсников и восьмиполюсников на СВЧ. - М.: Связь, 1965. 40 с.].Where $$\left|z\right|=z_{\mathrm{eleven}}{z}_{22}+z_{21}^2$$

; z ₁₁ , z ₂₁ , z ₂₂ - determinant and elements of the matrix of resistances of SFU taking into account the reciprocity condition z ₁₂ = -z _2l [Feldstein A.L., Yavich L.R. Microwave four-terminal and eight-terminal synthesis. - M.: Communication, 1965. 40 p.].

The general normalized classical demodulator transmission matrix is obtained by multiplying matrices (2) and (1) taking into account normalization conditions:

$$A=\left[\begin{array}{cc}\frac{z_{\mathrm{eleven}}}{z_{21}}\sqrt{\frac{z_n}{z_0}}& \left(\frac{z_{\mathrm{eleven}}}{z_{21}}z-\frac{\left|z\right|}{z_{21}}\right)\frac{\mathrm{one}}{\sqrt{z_0{z}_n}}\\ \frac{\mathrm{one}}{z_{21}}\sqrt{z_0{z}_n}& \left(\frac{z}{z_{21}}-\frac{z_{22}}{z_{21}}\right)\sqrt{\frac{z_0}{z_n}}\end{array}\right].\left(3\right)$$

Using the well-known relationship between the elements of the scattering matrix and the elements of the transmission matrix and (3), we obtain the expression for the transmission coefficient of the high-frequency part of the demodulator:

$$S_{21}=\frac{2z_{21}\sqrt{z_0{z}_n}}{\left(z_n+z\right)\left(z_{\mathrm{eleven}}+z_0\right)-z_0{z}_{22}-\left|z\right|}.\left(\mathrm{four}\right)$$

A physically feasible transfer function is related to the transmission coefficient as follows $$H=\frac{\mathrm{one}}{2}{S}_{21}\sqrt{\frac{z_n}{z_0}}$$

Suppose that it is necessary to determine the scheme of a complex four-terminal network and the values of the complex resistances of the two-terminal networks included in it, at which it is possible to ensure the given dependences of the module m ₂₁ and phase φ _{21 of the} transfer function of the high-frequency part of the demodulator on frequency:

$$S_{21}=m_{21}\left(\cos {\varphi }_{21}+j\sin {\varphi }_{21}\right)\left(5\right)$$

After substituting (4) in (5), we obtain a complex equation whose solution has the form of a relationship between the elements of the sought-after impedance matrix, optimal according to the criterion for ensuring the formation of a given quasilinear slope of the frequency response of the high-frequency part of the demodulator (5) in the entire frequency range:

$$z_{\mathrm{eleven}}=\frac{z_{21}^2+z_{21}{D}_{\mathrm{one}}}{E_{\mathrm{one}}-z_{22}}-z_0,\left(6\right)$$

; Е₁=z_н+z.Where $$D_{\mathrm{one}}=\frac{z_n}{m_{21}\left(\cos {\varphi }_{21}+j\sin {\varphi }_{21}\right)}$$

; E ₁ = z _n + z.

The obtained relationship (6) between the elements of the transmission matrix of a complex four-terminal network means that the FMS and ChMS demodulators must contain at least one independent two-terminal network with complex resistance, the value of which must satisfy the equation formed on the basis of this relationship. To find the optimal values of the parameters of a complex four-terminal network, it is necessary to select some circuit from M≥1 two-terminal network with complex resistance, find its resistance matrix, the elements of which are expressed through the parameters of the complex four-terminal scheme, and substitute them in (6). The equation thus formed must be solved with respect to the resistance of the selected complex bipolar. The values of the parameters of the remaining M-1 complex two-terminal networks can be set arbitrarily or selected from any other physical considerations. In accordance with the described algorithm, the optimal dependence of the resistance of the first complex two-terminal of an overlapped T-shaped connection of four complex two-terminal (Fig. 3) on frequency is obtained according to criterion (5):

$$Z_{\mathrm{one}n}=r_n+j{x}_n=\frac{\left(Z_{3n}+Z_{\mathrm{four}n}\right)\left[Z_{2n}\left(D_{\mathrm{one}}-E_{\mathrm{one}}-z_{0n}\right)-E_{\mathrm{one}}{z}_{0n}\right]-Z_3{Z}_{\mathrm{four}n}\left(Z_{2n}+z_{0n}\right)}{E_{\mathrm{one}}\left(Z_{2n}+Z_{3n}+Z_{\mathrm{four}n}+z_{0n}\right)+\left(Z_{2n}+Z_{3n}\right)\left(z_{0n}-D_{\mathrm{one}}+Z_{\mathrm{four}}{}_n\right)}\left(7\right)$$

; Е₁=z_нn+z_n; n=1, 2 … - номера частот интерполяции. Сопротивления Z_2n, Z_3n, Z_4n могут быть выбраны произвольно или исходя из каких-либо других физических соображений. Индекс n необходимо ввести и в другие обозначения физических величин, явным образом зависящих от частоты.Where $$D_{\mathrm{one}}=\frac{z_{nn}}{m_{21n}\left(\cos {\varphi }_{21n}+j\sin {\varphi }_{21n}\right)}$$

; E ₁ = z _nn + z _n ; n = 1, 2 ... are the numbers of the interpolation frequencies. Resistance Z _2n , Z _3n , Z _4n can be chosen arbitrarily or on the basis of any other physical considerations. The index n must also be introduced in other notation of physical quantities that explicitly depend on the frequency.

Given the frequency response (7) of the first complex two-terminal terminal of a closed T-shaped connection, the given dependences of the module m _{21 of the} phases φ _{21 of} the transmission coefficients on frequency over the entire frequency range would be provided. However, implementation (7) in a continuous, even very narrow frequency band,

impossible.

To implement the optimal approximation (7) on a finite number of frequencies by interpolation, it is necessary to form a two-terminal network with a resistance Z _1n of at least 2N (N is the number of interpolation frequencies) of elements of type R, L, C, find expressions for its resistance, equate their optimal values the two-terminal resistances at the given frequencies determined by formulas (7) and solve the system of 2N equations thus formed with respect to the 2N selected parameters R, L, C. The values of the parameters of the remaining elements can be chosen arbitrarily or on the basis of any other physical considerations, such as the condition of physical realizability. Let the first two-terminal cc with resistance Z _{1n be} formed from a series-connected first resistive two-terminal with resistance R ₁ , a capacitor with capacitance C and a second resistive two-terminal with resistance R ₂ and inductance L connected in parallel to each other (Fig. 4). The complex resistance of the first two-terminal KCH:

$$Z_{\mathrm{one}n}=R_{\mathrm{one}}+\frac{\mathrm{one}}{j{\omega }_{n}C}+\frac{R_{2}j{\omega }_{n}L}{j{\omega }_{n}L+{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000028.png"\; state="\{\{state\}\}">R</figure-callout>}}_2}.\left(8\right)$$

We divide the real and imaginary parts in (8) and for N = 2 we compose a system of four equations:

; r₁, r₂, x₁, x₂ - оптимальные значения действительных и мнимых составляющих сопротивления первого комплексного двухполюсника комплексного четырехполюсника на двух частотах, определенные по формулам (7). $$E=x_2^2\left[{\left(r_{\mathrm{one}}-r_2\right)}^2+x_2^2\right]$$

; r ₁ , r ₂ , x ₁ , x ₂ are the optimal values of the real and imaginary components of the resistance of the first complex two-terminal complex four-terminal at two frequencies, determined by the formulas (7).

The implementation of the optimal approximations of the frequency characteristics of the CN (6) using an overlapped T-shaped connection of four complex two-terminal devices and the frequency characteristics of the first complex two-terminal network (7) of this connection using (8), (10) provides an increase in the frequency band within which with certain deviations the predetermined dependences of the module m ₂₁ and phase φ _{21 of} the transmission coefficient of the high-frequency part of the demodulator on the frequency are provided (5). This allows, with a reasonable choice of the positions of the given frequencies ω ₁ , ω ₂ relative to each other, to expand the frequency band within which a given slope of the frequency response in a given frequency band is provided in the interests of converting the FMS to AFMS or ChMS to AFMS. The frequency characteristics of the resistance of the signal source and the load can be set by any.

The proposed technical solutions are new, since the method and device for the demodulation of phase-modulated and frequency-modulated signals providing the conversion of PMS to AFMS or HMS to AFMS on a given quasilinear slope of the frequency response of the high-frequency part of the demodulator in a given frequency band due to a special choice of the frequency dependence of the element are unknown from publicly available information z _{11 of the} matrix of resistances of the complex four-terminal, implemented by the implementation of this four-terminal in the form of an overlapped T-shaped the unity of the four complex two-terminal networks, the formation of the first complex two-terminal terminal blocking a T-shaped connection from a series of connected first resistive two-terminal network with a resistance R ₁ , a capacitor with a capacitance C and a second resistive two-terminal network with a resistance R ₂ and a coil with inductance L connected in parallel with each other and selecting these parameters according to the corresponding mathematical expressions in the interests of further amplitude demodulation and the allocation of the low-frequency component, am lituda which changes according to the input phase change MBF (PHI input frequency).

The proposed technical solutions have an inventive step, since it does not explicitly follow from the published scientific data and the known technical solutions that the claimed sequence of operations (the execution of a four-terminal complex in the form of the above scheme, the inclusion of a two-pole nonlinear element between the four-terminal and the load in the longitudinal circuit, the implementation of the optimal frequency dependences of the element z _{11 of} the resistance matrix of a complex four-terminal network by performing this four-terminal network in the form open T-shaped connection of four complex two-terminal, the formation of the first complex two-pole closed T-shaped connection from series-connected first resistive two-terminal with resistance R ₁ , capacitor with capacitance C and in parallel connected to each other of the second resistive two-terminal with resistance R ₂ and coil with inductance L and the choice of the indicated parameters according to the corresponding mathematical expressions) provide the given dependences of the module and phase of the transfer function and the high-frequency part of the demodulator from the frequency in a given frequency band, within which a given slope of the frequency response is provided in the interests of converting the FMS to AFMS or HMS to AHMS.

The proposed technical solutions are practically applicable, since semiconductor diodes (parametric diodes, pin diodes, power supply diodes, tunnel diodes, Gunn diodes, etc.), inductances and capacitances formed in the claimed circuit of the demodulation device can be used for their implementation . The frequency characteristics of the RF and the first complex two-pole terminal of an overlapped T-shaped connection, the resistance values of resistive elements, inductances and capacitances can be determined using mathematical expressions given in the claims.

The technical and economic efficiency of the proposed method and device consists in simultaneously providing predetermined dependences of the module and phase of the transfer function of the high-frequency part of the demodulator on frequency, which contributes to the formation of a given quasilinear portion of the slope of the frequency response in the interest of converting the FMS to AFMS or ChMS to AFC in a larger frequency band with increased areas physical feasibility as areas of change in the real and imaginary components of the resistances of the source of the PMS or PMS and the load with the aim of yes neyshey amplitude demodulation and separation of the low frequency component, the amplitude of which varies according to the law of change of phase of the input frequency of the input MBF or PHI, i.e. for demodulation FMS and PHI.

Claims (2)

1. The method of demodulating phase-modulated and frequency-modulated signals, consisting in the fact that the phase-modulated or frequency-modulated signal is fed to a demodulator made of a four-terminal device, a nonlinear element, a low-pass filter, a separation capacitance, and a low-frequency load, the phase-modulated signal is converted to amplitude-phase-modulated a signal, a frequency-modulated signal is converted into an amplitude-frequency-modulated signal, the conversion of a phase-modulated signal into amplitude-phases the modulated signal and the frequency-modulated signal to the amplitude-frequency-modulated signal is carried out by supplying these signals to the left slope of the frequency response of the demodulator, using a nonlinear element, the spectrum of the amplitude-phase-modulated signal and the amplitude-frequency-modulated signal are destroyed on the high-frequency and low-frequency components, the low-frequency component fed to the integrating low-pass filter circuit; using the low-pass filter, an information low-frequency signal is extracted, the amplitude of which changes according to the law of changing the phase of the phase-modulated input signal or according to the law of changing the frequency of the frequency-modulated signal, using a dividing capacitance, the constant component is eliminated, characterized in that the four-terminal device is made of a complex of reactive and resistive elements, the output of the phase-modulated or frequency-modulated signal source is connected to the input of the four-terminal, a nonlinear element is included in the longitudinal circuit between the output of the four-terminal and introduced into the transverse circuit before using a low-frequency filter with a high-frequency load, the given frequency dependences of the transfer function of the high-frequency part of the demodulator for the formation of a given slope of the amplitude-frequency characteristic are provided by selecting the frequency dependence of element z _{11 of} the complex four-terminal resistance matrix using the following mathematical expression:
$$z_{\mathrm{eleven}}=\frac{z_{21}^2+z_{21}{D}_{\mathrm{one}}}{E_{\mathrm{one}}-z_{22}}-z_0,$$

Where $$D_{\mathrm{one}}=\frac{z_n}{m_{21}\left(\cos {\varphi }_{21}+j\sin {\varphi }_{21}\right)}$$

; E ₁ = z _n + z; z ₂₁ , z ₂₂ are the given dependences of the corresponding elements of the resistance matrix of a complex four-terminal network on frequency in a given frequency band; m ₂₁ , φ ₂₁ - the given dependence of the module and phase of the transfer function on frequency in a given frequency band; z is a given dependence of the complex resistance of a bipolar nonlinear element on frequency in a given frequency band; z ₀ , z _n are the given dependences of the complex resistances of the source of the phase-modulated or frequency-modulated signal and the high-frequency load on the frequency in a given frequency band. 2. A device for demodulating phase-modulated and frequency-modulated signals, consisting of a phase-modulated or frequency-modulated signal source, four-terminal, two-electrode nonlinear element, low-pass filter, separation capacitance and low-frequency load, characterized in that the four-terminal is complex in the form of an overlapped T-shaped connection of four complex two-terminal networks, the output of a phase-modulated or frequency-modulated signal source is connected to the four-pole input nick, the nonlinear element is included in a longitudinal path between the output quadripole and introduced into the transverse chain before the lowpass filter high load, the first bipole overlapped T-fitting is formed of serially connected first resistive two-terminal resistance R _1, a capacitor with capacitance C and connected in parallel between a second resistive bipolar with resistance R ₂ and a coil with inductance L, the values of the parameters of the first bipolar blocked T-shaped compounds are defined in accordance with the following mathematical expressions:
$$R_{\mathrm{one}}=\frac{{\omega }_2^3{r}_{\mathrm{one}}{x}_2\left({\omega }_2{x}_2-2{\omega }_{\mathrm{one}}x{}_{\mathrm{one}}\right)-{\omega }_{\mathrm{one}}^3{x}_{\mathrm{one}}{r}_2\left({\omega }_{\mathrm{one}}{x}_{\mathrm{one}}-2{\omega }_{2}x{}_2\right)+{\omega }_{\mathrm{one}}^2{\omega }_2^2{\left(r_{\mathrm{one}}-r_2\right)}^3}{\left({\omega }_2^2-{\omega }_{\mathrm{one}}^2\right){\left({\omega }_2{x}_2-{\omega }_{\mathrm{one}}x{}_{\mathrm{one}}\right)}^2};$$

$$R_2=\frac{{\omega }_{\mathrm{one}}^{\mathrm{four}}A+{\omega }_2{\omega }_{\mathrm{one}}^{3}B+{\omega }_{\mathrm{one}}^2{\omega }_2^2{C}_{\mathrm{one}}+{\omega }_{\mathrm{one}}{\omega }_2^{3}D+{\omega }_2^{\mathrm{four}}E}{(r_2-r_{\mathrm{one}})\left({\omega }_2^2-{\omega }_{\mathrm{one}}^2\right){\left(x_{\mathrm{one}}{\omega }_{\mathrm{one}}-x_2{\omega }_2\right)}^2};$$

$$C=\frac{\left({\omega }_{\mathrm{one}}^2-{\omega }_2^2\right)\left({\omega }_{\mathrm{one}}{x}_{\mathrm{one}}-{\omega }_{2}x{}_2\right)}{{\omega }_{\mathrm{one}}{\omega }_2\left({\omega }_{\mathrm{one}}{\omega }_2\left[{\left(r_{\mathrm{one}}-r_2\right)}^2+x_{\mathrm{one}}^2+x_2^2\right]-x_{\mathrm{one}}{x}_2\left({\omega }_{\mathrm{one}}^2-{\omega }_2^2\right)\right)};$$

$$L=\frac{{\omega }_{\mathrm{one}}^{\mathrm{four}}A+{\omega }_2{\omega }_{\mathrm{one}}^{3}B+{\omega }_{\mathrm{one}}^2{\omega }_2^2{C}_{\mathrm{one}}+{\omega }_{\mathrm{one}}{\omega }_2^{3}D+{\omega }_2^{\mathrm{four}}E}{\left({\omega }_{\mathrm{one}}^2-{\omega }_2^2\right){\left(x_{\mathrm{one}}{\omega }_{\mathrm{one}}-x_2{\omega }_2\right)}^3}$$

where $$A=x_{\mathrm{one}}^2\left[{\left(r_{\mathrm{one}}-r_2\right)}^2+x_{\mathrm{one}}^2\right]$$

;
$$B=-2x_{\mathrm{one}}{x}_2\left[{\left(r_{\mathrm{one}}-r_2\right)}^2+2x_{\mathrm{one}}^2\right]$$

; $$D=-2x_{\mathrm{one}}{x}_2\left[{\left(r_{\mathrm{one}}-r_2\right)}^2+2x_2^2\right]$$

;
$$C_{\mathrm{one}}=\left(x_{\mathrm{one}}^2+x_2^2\right){\left(r_{\mathrm{one}}-r_2\right)}^{{}_2}+6\left(r_{\mathrm{one}}^2{r}_2^2+x_{\mathrm{one}}^2{x}_2^2\right)+r_{\mathrm{one}}^{\mathrm{four}}+r_2^{\mathrm{four}}-\mathrm{four}{r}_{\mathrm{one}}{r}_2\left(r_{\mathrm{one}}^2+r_2^2\right)$$

; $$E=x_2^2\left[{\left(r_{\mathrm{one}}-r_2\right)}^2+x_2^2\right]$$

;
r ₁ , r ₂ , x ₁ , x ₂ are the optimal values of the real and imaginary components of the resistance of the first complex two-terminal terminal of a closed T-shaped connection at two frequencies; $$Z_{\mathrm{one}n}=r_n+j{x}_n=\frac{\left(Z_{3n}+Z_{\mathrm{four}n}\right)\left[Z_{2n}\left(D_{\mathrm{one}}-E_{\mathrm{one}}-z_{0n}\right)-E_{\mathrm{one}}{z}_{0n}\right]-Z_3{Z}_{\mathrm{four}n}\left(Z_{2n}+z_{0n}\right)}{E_{\mathrm{one}}\left(Z_{2n}+Z_{3n}+Z_{\mathrm{four}n}+z_{0n}\right)+\left(Z_{2n}+Z_{3n}\right)\left(z_{0n}-D_{\mathrm{one}}+Z_{\mathrm{four}}{}_n\right)}$$

- the optimal values of the complex resistance of the first complex two-pole terminal of a blocked T-shaped connection at two frequencies; $$D_{\mathrm{one}}=\frac{z_{nn}}{m_{21n}\left(\cos {\varphi }_{21n}+j\sin {\varphi }_{21n}\right)}$$

; E ₁ = z _nn + z _n ; Z _2n , Z _3n , Z _4n - setpoints of the complex resistance of the second, third and fourth complex two-terminal circuits of the closed T-shaped connection at two frequencies; m _21n , φ _21n - given values of the modules and phases of the transfer function of the high-frequency part of the demodulator at two frequencies; z _n - set values of the complex resistance of a bipolar nonlinear element at two frequencies; z _0n , z _nn are the set values of the complex resistances of the source of the phase-modulated or frequency-modulated signal and the high-frequency load at two frequencies; ω _1,2 = 2πf _1,2 ; n = 1, 2 - numbers of the given two frequencies f _1,2 .

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