RU2117960C1 - Method of target tracking by monopulse radar - Google Patents

Abstract

FIELD: radiolocation technique. SUBSTANCE: invention can be used in coherent-pulse radars with monopulse principle of direction finding. Technical problem of invention consists in increase of noise immunity of tracking with regard to active and passive noises with simultaneous increase of tracking accuracy in radars employing complex, for example phase-shift keying of signals and mounted on mobile carriers. EFFECT: increased noise immunity with simultaneous increase of tracking accuracy. 12 dwg

Description

 The invention relates to radar technology, mainly to methods of tracking signals from surface targets in the presence of reflections from local objects-sources of passive interference, and can be used in coherent-pulse radar stations (radar) with a single-pulse direction finding principle, including in radar, using complex, in particular, phase-shifted signals mounted on mobile carriers.

 At present, in monopulse radars, a range and angular coordinate tracking method is used, in which the received high-frequency signals after the sum-difference conversion are fed to the mixers of the total and difference receiving channels, where they are converted into intermediate frequency signals, which are amplified in the intermediate frequency amplifiers (IF) and then fed to the amplitude detector - in the total channel and phase detector, while the intermediate frequency signal of the total channel plays the role of reference fluctuations. As a result of phase detection, an angular mismatch signal is generated, which is used for angular tracking [1, p. 22, fig. 1.9 and 2, p. 20, fig. fifteen].

 The signal of the total channel after amplitude detection is fed to a time discriminator included in the range tracking unit [2, p. 20, Fig. 15], in which a range mismatch (error) signal is generated, which is then used for range tracking. This method [2, p. 20, fig. 15] is closest in its technical essence to the proposed one and is taken as a prototype.

 The disadvantage of the prototype method is its low noise immunity with respect to passive interference - reflections from extended local objects, such as, for example, a coastline when surface ships (NK) are found near the coast, and also a cloud of dipole reflectors (DO), set NK at a relatively low altitude (50 - 100 m) with the task of disrupting escort. In situations where the direction of the coastline (warhead) in the observation area is close to or coincides with the direction of radar radiation propagation, the angle resolution determined by the radiation pattern of the radar antenna may be insufficient for the spatial separation of signals from the NK and from the warhead (or DO) located at the same ranges. Then the application of the prototype method will lead to guidance to the energy center of the NK-BCh or NK-DO system, and in the future, when approaching and with a sufficient intensity of interference, the tracking will be disrupted in range and angular coordinates.

 We illustrate what has been said by the following example.

где
Х - расстояние между НК и береговой чертой в направлении, перпендикулярном направлению распространения;
R - дальность до НК;
φ_0,5 - ширина ДНА по уровню " - 3 дБ".Let the radar installed on the aircraft (LA), monitor the NK located near the coast (see. Fig. 12 below), and the angular resolution due to the real ID is missing, that is

Where
X is the distance between the NK and the coastline in the direction perpendicular to the direction of propagation;
R is the range to the NK;
φ _0,5 - the width of the bottom at the level of "-3 dB".

For example, this takes place at X = 300 m, R = 10 km and φ _0.5 = 0.15 rad.

где
ΔR - разрешение по дальности;
ϑ - угол скольжения;
σ^° - удельная ЭОП берега.For the effective reflecting surface (EOC) of the coastal part falling into the pulsed volume, we obtain

Where
ΔR - range resolution;
ϑ - slip angle;
σ ^° - specific coastal intensifier.

где
Н - высота полета ЛА,
получим при ΔR = 300 м, X = 300 м, R = 10 км, и φ_0,5 = 0,15 рад,
σ^° = 0,03 (-15 дБ)- [3, с. 296, рис. 22],
σ_δ = 1,2•10³•300•0,03 = 1,21•10⁴ м².In particular, at small sliding angles, for example

Where
N - flight altitude,
we obtain at ΔR = 300 m, X = 300 m, R = 10 km, and φ _0.5 = 0.15 rad,
σ ^° = 0.03 (-15 dB) - [3, p. 296, fig. 22]
σ _δ = 1.2 • 10 ³ • 300 • 0.03 = 1.21 • 10 ⁴ m ² .

Thus, even when the provisionally hover radar antenna on NK with GO σ _n ≅ 5 • March ¹⁰ m ² (e.g., information from the navigation system), the energy reflection center is strongly biased towards the coastline as the result is likely to disruption of tracking in the angular coordinate. A similar picture takes place and accompanied by range.

In radars using complex, in particular phase-shift (FM) signals with a large base
N = T _and Δf ≫ 1, (1)
Where
T _AND - signal pulse duration;
f is the signal spectrum width, the prototype method, in addition, has insufficient accuracy in measuring the angular coordinate.

поэтому наиболее рациональным способом оптимальной обработки ФМ сигналов большой длительности является их сжатие по времени на видеочастоте средствами цифровой вычислительной техники.At present, the optimal filtering of complex, in particular, FM signals (time compression) with a long duration (T _AND > 50 μs) at a high or intermediate frequency in monopulse radars is practically unrealizable already because it is not possible to ensure a sufficiently high identity of the compression devices on radio frequencies, so that the stray phase shift between the total and difference channels, in any case, does not exceed

Therefore, the most rational way to optimally process FM signals of long duration is to compress them in time on a video frequency using digital computer technology.

On the other hand, the formation of an angular mismatch signal by using a phase detector connected between the outputs of the IF amplifier of the sum and difference channels and multiplying the FM signals of the sum and difference channels before compression, followed by low-pass filtering, leads to a relatively low accuracy of angle measurement and insufficient noise immunity with respect to noise interference. The fact is that for large signal-to-noise ratios (ρ ≪ 1), the main noise component at the output of the phase detector performing the operation of multiplying the oscillations supplied to its inputs and low-pass filtering the results of multiplication is a component representing the product of the signal voltage in the total channel by noise voltage in the difference channel (near the equal-signal direction). However, in the case of FM signals with a large base, the signal-to-noise ratio to compression ρ _{0 is} usually small, i.e., ρ ₀ ≫ 1, and then the main noise component at the output of the above-mentioned phase detector (PD) is the component representing the product of noise voltage in the total channel noise voltage in the difference channel. It is this circumstance that is the reason for the relatively low accuracy of measurement and noise immunity of the prototype method.

где
σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{ш}}$ - дисперсия шумов в приемных каналах на входах ФД;
U_C - амплитуда сигналов в суммарном приемном канале на входе ФД;
g_Δ - усиление антенны разностного канала, нормированное к усилению антенны суммарного канала (по полю);
k - коэффициент пропорциональности.We illustrate what has been said by the following calculation. Expressions (2) and (3) for the variance of the process σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{out}}$ and the signal voltage S _o at the PD output, taking into account low-pass filtering (with a band of ≈l / T _И ), have the form

Where
σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{w}}$ - dispersion of noise in the receiving channels at the inputs of the PD;
U _C - the amplitude of the signals in the total receiving channel at the input of the PD;
g _Δ is the gain of the antenna of the difference channel, normalized to the gain of the antenna of the total channel (in the field);
k is the coefficient of proportionality.

где

- крутизна пеленгационной характеристики [4, с. 130].The accuracy of a single measurement of angular mismatch is determined by the ratio

Where

- the steepness of the direction-finding characteristic [4, p. 130].

, тогда из (2) - (4) получим

а для ФМ сигналов при N >> 1, наоборот ρ₀ << 1, тогда,

При одинаковой энергии принимаемых сигналов ρ = Nρ₀ так что, как видно из (6), для ФМ сигналов в способе-прототипе дисперсия измерения получается сравнительно большой, а точность, следовательно, сравнительно невысокой.For simple pulse signals, N = 1,

, then from (2) - (4) we obtain

and for FM signals at N >> 1, on the contrary, ρ ₀ << 1, then

For the same energy of the received signals ρ = Nρ ₀ so that, as can be seen from (6), for the FM signals in the prototype method, the dispersion of the measurement is relatively large, and the accuracy, therefore, is relatively low.

не как

- в случае простых сигналов, т.е. помехозащищенность способа-прототипа по отношению к шумовым помехам также является низкой.In addition, when exposed to noise interference, the signal-to-noise ratios ρ and ρ ₀ decrease, while σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{β}}$ increases in the case of FM signals as

not like

- in the case of simple signals, i.e. the noise immunity of the prototype method in relation to noise interference is also low.

 An object of the invention is to increase the noise immunity of tracking in relation to passive and active noise interference while increasing the accuracy of tracking in radars using complex, in particular, FM signals and mounted on mobile carriers.

 To achieve a technical result, it is proposed, after isolating the signal from the true target against the background of interfering reflections from passive interference by narrow-band Doppler filtering, to organize tracking of the target by Doppler frequency, i.e., by the radial velocity of the target relative to the radar, thereby providing adaptive echo filtering in the process of tracking, and the signals filtered from the true target in this way should then be used for tracking by range and angular coordinates, and when tracking along to the angular coordinates to form an angular mismatch signal after coordinated filtering of video pulses and narrow-band Doppler filtering in the total and difference channels by pairwise multiplication of the same quadrature components of the signals with the summation of these products.

, с полосой ΔF и числом каналов

, где F_П - частота повторения зондирующих импульсов, ΔF - ширина спектра межпериодных флюктуаций сигналов от истинных целей, сравнивают мощности спектральных составляющих с пороговым уровнем обнаружения, определяют ширину спектра с наиболее мощной спектральной составляющей по заданному уровню из числа составляющих, превысивших порог обнаружения, сравнивают полученное значение ширины спектра с заданным пороговым значением, при превышении его принимают решение о наличии сигнала от ложной цели и переходят к следующему элементу дальности, а при отсутствии превышения полученным значением ширины спектра заданного порогового значения принимают решение о наличии сигнала от истинной цели в соответствующем элементе дальности, определяют значение доплеровской частоты, соответствующее максимуму спектра, изменяют частоту опорных колебаний на величину измеренного значения доплеровской частоты, определяют сигнал рассогласования по частоте, замыкают контур сопровождения по частоте и подстраивают этим сигналом частоту опорных колебаний, осуществляют узкополосную фильтрацию с полосой ΔF комплексной огибающей импульсных последовательностей сигналов в суммарном и разностном каналах на нулевой доплеровской частоте, выделяют амплитудную огибающую сигналов в суммарном канале, выделяют сигнал ошибки по дальности в суммарном канале, замыкают контур сопровождения по дальности и подстраивают этим сигналом положение строба дальности, выделяют сигнал углового рассогласования путем попарного перемножения результатов фильтрации одноименных квадратурных составляющих сигналов в суммарном и разностном каналах и суммирования этих произведений, замыкают контур сопровождения по углу стробом дальности и подстраивают сигналом углового рассогласования положение антенны.The essence of the invention lies in the fact that in the method of tracking the target of a monopulse radar, which includes emitting pulsed coherent signals in a given direction, receiving high-frequency signals in a given range of ranges, sum-difference conversion of received signals, super-heterodyne conversion of them to an intermediate frequency, amplification of the total and difference signals at an intermediate frequency, after amplification of the sum and difference signals at an intermediate frequency, the spectrum of signals is converted into a region video frequencies by means of phase detection using reference oscillations with the formation of quadrature components of each signal, they carry out a coordinated filtering of video pulses of the quadrature components of signals, for each range element in a given interval, multi-channel Doppler filtering of the complex envelope of the pulse sequence of the total signal in the Doppler frequency range is carried out -

, with ΔF band and number of channels

where F _P is the pulse repetition rate of the probe pulses, ΔF is the spectrum width of the inter-period fluctuations of the signals from the true targets, the powers of the spectral components are compared with the detection threshold level, the spectrum width with the most powerful spectral component is determined from a given level from the number of components that exceeded the detection threshold, compare the obtained value of the width of the spectrum with a given threshold value, when it is exceeded, they decide on the presence of a signal from a false target and proceed to the next element of range and if there is no excess of the specified threshold value by the obtained spectral width, a decision is made about the presence of a signal from the true target in the corresponding range element, the Doppler frequency corresponding to the maximum of the spectrum is determined, the frequency of the reference oscillations is changed by the value of the measured Doppler frequency, the frequency error signal is determined close the frequency tracking loop and adjust the frequency of the reference oscillations with this signal, perform narrow-band filtering with the ΔF band of the complex envelope of the pulse sequences of signals in the total and difference channels at zero Doppler frequency, the amplitude envelope of the signals in the total channel is extracted, the error signal in range in the total channel is isolated, the range tracking loop is closed and the position of the range gate is adjusted with this signal, the signal is isolated angular mismatch by pairwise multiplication of the filtering results of the same quadrature components of the signals in the total and difference channels and mmirovany these works, close the tracking contour along the corner with a range gate and adjust the position of the antenna with the signal of the angular mismatch.

According to the proposed method, the received signals in both the sum and difference receive channels after amplification at an intermediate frequency are converted to a video frequency by phase detection using reference oscillations generated by the transmitter exciter, with the formation of two quadrature components for each signal, then matched filtering (SF ) the quadrature components of the pulsed signals at the video frequency, which in the case of complex, in particular, FM signals, leads to time compression, in As a result, at the outputs of the quadrature channels, the signal-to-noise ratio in power increases N times, and compressed signals are obtained
U _Σ cosφ, U _Σ sinφ, U _Δ cosφ, U _Δ sinφ,
Where
φ is the initial phase of the received signals relative to the reference oscillations,
U _Σ , U _Δ is the amplitude of the signals after the SF in the total and difference receiving channels, respectively, and the value of U _Δ can be either positive or negative (depending on the position of the direction to the target relative to the equal-signal direction).

где
F_П - частота повторения зондирующих импульсов РЛС, - с полосой ΔF, определяемой шириной спектра межпериодных флюктуаций истинной цели, то есть НК и числом частотных каналов

.The next operation is multi-channel Doppler filtering of the envelope of the total signal in the frequency range of the Doppler F _D

Where
F _P - the repetition frequency of the probe pulses of the radar, - with a band ΔF, determined by the width of the spectrum of inter-period fluctuations of the true target, that is, NK and the number of frequency channels

.

соответствующее доплеровское расширение спектра

при скорости ЛА-носителя РЛС - V = 700 м/с и λ = 3,2 см составляет

что существенно меньше ширины спектра межпериодных флюктуаций

. В то же время спектр эхо-сигналов от участка береговой черты, попавшей в импульсный объем, даже при отсутствии собственных флюктуаций составляет (фиг. 12)

Например, при φ_0,5 = 0,15, R = 10 км, X = 300 м, V = 700 м/с, λ = 3,2 см,

•100•0,022 = 480 Гц.It is produced for each of the elements n _i = (i = 1, 2, ... n _R ) of range resolution (in particular, alternately) in the zone of the possible position of the target, determined by the accuracy of target designation - until the target is found. This filtering is possible and effective due to the fact that the spectrum of the complex envelope of the echo signals from the NC is relatively narrow, since the inter-period fluctuations of the signals are slow (the correlation interval is τ ₀ = 0.1 s in the wavelength range λ = 3.2 cm [ 5]), and the Doppler spread of the spectrum when observing the NK with a forward-looking antenna can be neglected, since the angular dimensions of the NK at distances R ≥ 10 km are significantly less than the width of the bottom of the radar. For example, with the transverse size of the LC δX ≤ 30 m and the range R = 10 km, its angular size

corresponding Doppler spreading

when the speed of the LA carrier of the radar - V = 700 m / s and λ = 3.2 cm is

which is significantly less than the width of the spectrum of interperiod fluctuations

. At the same time, the spectrum of echo signals from the coastline plot that fell into the pulse volume, even in the absence of intrinsic fluctuations, is (Fig. 12)

For example, with φ _0.5 = 0.15, R = 10 km, X = 300 m, V = 700 m / s, λ = 3.2 cm,

• 100 • 0.022 = 480 Hz.

Это имеет место в рассматриваемом примере, когда V = 700 м/с, R = 10 км и X = 300 м, и левая часть неравенства составляет 20 Гц.Thus, the spectrum of reflections from the coast is much wider (more than an order of magnitude) of the spectrum of signals from the NK, which determines the possibility of frequency filtering of echo signals from the NK against the background of echo signals from the shore even in the absence of spatial resolution between them. In addition, it should be borne in mind that in many cases - in particular, when the NK is stationary relative to the coast or moves towards the radar - at the location shown in FIG. 12, a complete frequency separation of signals from the NK and from the coast is possible, if the condition is fulfilled (when the NK is stationary)

This takes place in the considered example, when V = 700 m / s, R = 10 km and X = 300 m, and the left side of the inequality is 20 Hz.

After conducting multi-channel Doppler filtering (multi-channel is necessary, since the speed of the ND relative to the radar is not known in advance), it is proposed to determine the spectral width with the most powerful spectral component from a given level from the number of components that exceed the detection threshold determined by the allowable probability of false detection due to noise - and compare it with a pre-selected threshold value, upon exceeding which a decision is made on the presence of a false target and on the transition to the analysis of the next element range and, in the absence of excess, ie, in the presence of a powerful narrowband spectral component -.. the presence of the signal from NC. In this latter case, it is proposed to find the value of the Doppler frequency F _* corresponding to the maximum of the spectrum (that is, the corresponding radial velocity of the NC relative to the radar), and then changing the frequency of the reference oscillations by this value, determine, further, the frequency error signal between the value of this correction and the measured the current value of F _* and, closing the frequency tracking loop, use this signal to fine-tune it. In the process of frequency tracking, the frequency is F _* = 0 (with an accuracy of the order of ΔF), therefore, it is further proposed to filter the complex envelope of the sequences of pulse signals in the total and difference receiving channels at zero Doppler frequency and after this filtering, forming an error signal in the total channel in a known manner , for example, using the method of two half-gates (for example, [2], p. 53, Fig. 41), to close the range tracking loop, adjusting, as usual, the position of the range gate, as usual, with a mismatch signal. Further, it is proposed to generate an angular mismatch signal not by multiplying the signals at an intermediate frequency, as in the prototype, but by pairwise multiplying the signals in the total and difference channels after matched filtering and narrow-band filtering at zero Doppler frequency, followed by summing the multiplication results, and then close the tracking circuit along the angle, adjusting the mismatch signal the position of the antenna, and the contour of the angular tracking is gated as usual, and.

The invention is illustrated by a further description and drawings of the radar that implements this method:
FIG. 1 - structural diagram of the radar;
FIG. 2 is a structural diagram of the pathogen (B) of the transmitter;
FIG. 3 is a block diagram of a frequency bias unit;
FIG. 4 is a block diagram of a code-frequency converter (PCC);
FIG. 5 is a block diagram of a double balanced modulator (DBM);
FIG. 6 is a structural diagram of a block of Doppler filters (BDF);
FIG. 7 is a structural diagram of a frequency discriminator (BH);
FIG. 8 is a structural diagram of a Doppler filter (DF _r ) tuned to frequencies ± rΔF;
FIG. 9 is a structural diagram of a Doppler filter (DF ₀ ) tuned to a zero frequency;
FIG. 10 is a structural diagram of a range finder (D);
FIG. 11 is a program diagram of an analysis and decision unit (BAR);
FIG. 12 is a horizontal arrangement of the radar and observation objects.

In FIG. 1 presents a structural diagram of the radar, where the following notation:
1 - pathogen (B);
2 - power amplifier (PA);
3 - antenna switch (AP);
4 - total difference converter (PSA);
5 - antenna (A);
6 - pulse modulator (IM);
7 - synchronizer (C);
8 - frequency offset unit (BSC);
9 - high frequency amplifier of the total channel (UHF _Σ );
10 - amplifier high frequency difference channel (UHF _Δ );
11 - mixer total channel (cm _Σ );
12 - differential channel mixer (Cm _Δ );
13 - amplifier of the intermediate frequency of the total channel (IFA _Σ );
14 - amplifier intermediate frequency differential channel (IFA _Δ );
15 - 18 - phase detectors (PD);
19 - phase shifter (PV);
20, 21 - matched filters (SF);
22 - the first block of keys (BK ₁ );
23 - block Doppler filters (BDF);
24 - block analysis and decisions (BAR);
25 - Converter "code-time interval"(PCVI);
26 - the first adder ("+1");
27 - the second block of keys (BC ₂ );
28 - frequency discriminator (BH);
29 - Doppler filter of zero frequency of the total channel (DF _oΣ );
30 - Doppler filter of zero frequency differential channel (DF _oΔ );
31, 32 - multipliers;
33 - block combining quadratures (BOK);
34 - second adder ("+2");
35 - range finder (D);
36 - key (C);
37 - integrator (∫);
38 - antenna drive (PrA).

 In the diagram of FIG. 1, a pathogen 1, a power amplifier 2, an antenna switch 3, a sum-difference converter 4, and an antenna 5 are connected in series, a high-frequency amplifier 9 of the total channel, the input of which is connected to the third arm of the antenna switch 3, a mixer 11 of the total channel, and an intermediate amplifier 13 are connected in series frequency of the total channel, a high-frequency amplifier 10 of the differential channel, the input of which is connected to the output of the difference signal of the total-differential converter 4, is connected in series, mixes only 12 differential channels, the heterodyne input of which is combined with the heterodyne input of the mixer 11 of the total channel and connected to the 2nd output of the pathogen 1, and the amplifier 14 of the intermediate frequency of the differential channel are connected in series via two lines to the matched filter 20 of the total channel, the first block of 22 keys and a block of 23 Doppler filters, a matched filter of the difference channel 21 and a Doppler filter 30 of the zero frequency of the difference channel, the outputs of which through the corresponding multiplies Fir-trees 31, 32 are connected to the corresponding inputs of the two-input adder 34, the second two-channel block of keys 27 and the frequency discriminator 28 are connected in series via two lines, the Doppler filter 29 of the zero frequency of the total channel and the block of quadrature combining are connected in series through two lines. The inputs of the matched filters 20 and 21 of the total and difference channels are connected to the outputs of the corresponding amplifiers 13 and 14 of the intermediate frequency through quadrature phase detectors 15 - 18. The inputs of the reference frequency of the phase detectors 15, 17 and 16, 18 of the same quadrature are paired together and connected to the output unit 8 of the frequency offset, the first pair is directly, and the second pair is through the phase shifter 19. The signal inputs of the first and second blocks 22, 27 of the same name and the Doppler filter 29 of the zero frequency of the total channel are combined s with each other, and first and second outputs of the Doppler filter 29 is also connected to second inputs of multipliers 31, 32, respectively.

The multi-channel output of the Doppler filter unit 23 is connected channel-wise to the corresponding multi-channel input of the analysis and decision block 24, the output of the range code (1st output) of which is connected via the code-time interval (PCVI) converter 25 to the control input of the first key block 22, 2 and the output of the Doppler frequency F _* is connected to a first input of a first two-input adder 26, a second input connected to the output of the frequency discriminator 28. The third output (initial target range R ₀ block 24 and the fourth output - to command closure con range tracking enthusiasts - connected to the corresponding second and third inputs of the range finder 35, the output of which is connected to the control inputs of the second block of keys 27 and key 36, the signal input of the latter is connected to the output of the second two-input adder 34, and the output is through the angle mismatch integrator 37 and the drive 38 of the antenna is connected to the control input of the antenna 5.

 The first input of the frequency offset unit 8 is connected to the output of the reference frequency of the exciter 1, the second input of the frequency offset unit 8 is connected to the output of the first two-input adder 26, the third input is with the information output of the radar carrier’s own radial speed meter, and the fourth input is combined with the same input of the range finder 35 and the second input (clock pulses) of the converter 25 "code-time interval", connected to the first output (clock pulses) of the synchronizer 7, the second output of which (clock pulses of the repetition frequency) connected to the input of the pulse modulator 6 and to the combined inputs of the sync pulses of the repetition frequency converter 25 "code-time interval", rangefinder 35, block 23 Doppler filters, frequency discriminator 28, as well as Doppler filters 29 and 30 of zero frequency of the total and difference channels.

In FIG. 2 presents a structural diagram of the pathogen 1, where the following notation:
39 - master oscillator (ZG);
40 - frequency multiplier (UMN);
41 - mixer (SM);
42 - reference frequency generator (GOCH);
43 - 45 - amplifiers (Us).

 In the diagram of FIG. 2, a master oscillator 39, a frequency multiplier 40 and an amplifier 43, the output of which forms the first output (oscillations of the signal frequency) of the exciter 1 are connected in series, the output of the multiplier 40 is also connected through the mixer 41 to the amplifier 44, the output of which is the second output (oscillations of the local oscillator frequency) of the exciter 1, the input of the mixer 41 is connected to the output of the reference frequency generator 42 and the input of the amplifier 45, the output of which is the third output (reference frequency oscillations) of the pathogen 1.

In FIG. 3 is a structural diagram of a frequency offset unit 8, where the following notation is adopted:
46 - the first Converter "code-frequency" (PCC ₁ ),
47 - the second Converter "code-frequency" (PCC ₂ ),
48 - the first double balanced modulator (DBM ₁ ),
49 - the second double balanced modulator (DBM ₂ ).

 In the diagram of FIG. 3, the output of the first code-frequency converter 46 through the first double balanced modulator 48 is connected to the second input of the second double balanced modulator 49, the first input of which is connected to the first input of the frequency offset unit 8 — oscillations of the initial frequency of the reference oscillations — and the output forms the output of the block frequency offset - the corrected frequency of the reference oscillations, the first input of the first code-frequency converter 46 is connected to the second input of the frequency offset unit 8 (target Doppler frequency code), the third input of which (Doppler code radar carrier frequency) is connected to a first input of the second converter 47 "frequency-code", the second inputs of the two converters 46 and 47 "code- frequency" are combined and connected to the fourth input (clock) frequency shift unit 8.

In FIG. 4 is a structural diagram of code-frequency converters 46 (47), where the following notation is used:
50 - decoder (L),
51 - controlled divider (DU),
52 - low-pass filter (low-pass filter).

 In the diagram of FIG. 4, a decoder 50, a controlled divider 51, and a low-pass filter 52 are connected in series, wherein the input of the decoder 50 forms the first code input of the code-frequency converter 46 (47), the second input of the controlled divider 51 forms the second input (clock pulses) of the converter 46 (47 ) "code-frequency", and the output of the low-pass filter 52 is the output of the code-frequency converter 46 (47).

54, 56 - перемножители ("X"),
57 - двухвходовый сумматор ("+").In FIG. 5 is a structural diagram of a dual balanced modulator 48 (49), where the following notation is adopted:
53, 55 - phase shifters on

54, 56 - multipliers ("X"),
57 - two-input adder ("+").

In the diagram of FIG. 5, phase shifters 53 by 90 ^° , multiplier 54 and adder 57 are connected in series, the input of phase shifter 53 by 90 ^° combined with the first input of multiplier 56 forms the first input of a double balanced modulator 48 (49), and its second input is connected to the second input of multiplier 54 and through the phase shifter 55 by 90 ^° with the second input of the multiplier 56, the output of which is connected to the second input of the two-input adder 57, and the output of the latter is the output of the double balanced modulator 48 (49).

In FIG. 6 is a structural diagram of a block of 23 Doppler filters (BDF), where the following notation:
58, 59 - analog-to-digital converters (ADC),
60 ₁ - 60 _{m / 2} - Doppler filters of frequency channels
± ΔF, ... ± rΔF, ... ± m / 2ΔF (DF ₁ , ... DF _r , ... DF _{m / 2} )
In the diagram of FIG. 6, the first and second inputs of the block 23 of Doppler filters are connected to the same inputs of the Doppler filters 60 through analog-to-digital converters 58 and 59, respectively, the third input of the block 23 of the Doppler filters - sync pulses of the repetition frequency - is directly connected to the same inputs of the Doppler filters 60, the outputs of the same Doppler filters ordered in ascending order of frequency channel numbers form an m-channel output of a block of 23 Doppler filters.

In FIG. 7 is a structural diagram of a frequency discriminator 28 (BH), where the following notation is adopted:
61, 62 - analog-to-digital converters (ADC),
63 is a block of Doppler filters of the frequency channels ± ΔF,
64 - subtraction block ("-").

 In the diagram of FIG. 7, the first and second inputs of the Doppler filter unit 63 are connected to the same inputs of the frequency discriminator 28 through analog-to-digital converters 61 and 62, respectively, and the third input is directly, the outputs of the Doppler filter unit 63 are connected to the corresponding inputs of the subtraction unit 64, the output of which forms the frequency output discriminator 28.

In FIG. 8 is a structural diagram of a block 60 _{r of} Doppler filters of frequency channels ± rΔF (r = 1, 2, ..., m / 2), where the following notation is used:
65, 66 - the first and second blocks of n-bit shift registers, respectively, (BSR ₁ , BSR ₂ );
67, 70 - n-input adders with cosine weight function (BC _cos ),
68, 69 - n-input weight combiners with sine weight function (BC _sin ),
71, 74 - two-input adders ("+"),
72, 73 - subtraction blocks ("-"),
75, 76 - blocks combining quadratures (BOK).

In the diagram of FIG. 8, the first and second inputs of the _r -Doppler filter unit 60 are connected to the information inputs of the first and second blocks of the shift registers 65 and 66, respectively, the inputs of the clock pulses of the latter are combined and connected to the third input of the Doppler filter unit 60. The inputs of the same category of n-input weight adders 67 with a cosine weight function and 68 with a sine weight function are pairwise interconnected and connected to the outputs of the corresponding bits of the block 65 of n-bit shift registers, and the inputs of the same category of n-input weight adders 69 with a sine weight function and 70 with the cosine weight function are also pairwise interconnected and connected to the outputs of the corresponding bits of the second block 66 n-bit shift registers, the output of the weight adder 67 is connected to to the first inputs of the two-input adder 71 and the subtracting unit 73, the output of the weight adder 68 is connected to the combined first inputs of the subtracting unit 72 and the two-input adder 74, the output of the weight adder 69 is connected to the second inputs of the two-input adder 71 and the subtracting unit 73, and the output the weight adder 70 is connected to the interconnected second inputs of the subtracting unit 72 and the two-input adder 74. The outputs of the two-input adder 71 and the subtracting unit 72 are connected to the first and second inputs of the combining unit 75 to vadratura respectively, and the outputs of the subtraction unit 73 and the two-input adder 74 are connected to the first and second inputs of the quadrature combining unit 76, respectively, the outputs of the quadrature combining units 75 and 76 are the first and second outputs of the Doppler filter 60 _r, respectively.

In FIG. 9 is a structural diagram of a Doppler filter 29 (30) of zero frequency (DF _{0Σ, Δ} ), where the following notation:
77, 78 - analog-to-digital converters,
79, 80 - the first and second blocks of n-bit shift registers, respectively (BSR ₁ , BSR ₂ );
81, 82 - n-input adders.

 In the diagram of FIG. 9, the first and second inputs of the zero frequency Doppler filter 29 (30) are connected to the information inputs of the first and second blocks of shift registers 79 and 80, respectively, the clock inputs of the BSR 79 and 80 are combined and connected to the third input of the Doppler filter 29 (30). The outputs of all n bits of the first and second blocks of shift registers 79 and 80 are connected bitwise to the corresponding inputs of the n-input adders 81, 82, respectively, the outputs of the n-input adders 81, 82 are connected respectively to the outputs 1 and 2 of the Doppler filter 29 (30).

In FIG. 10 presents a functional diagram of the range finder (D) 35, where the following notation is accepted:
83 - temporary discriminator (VD),
84 - key (C),
85 - reverse counter (PC),
86 - Converter "code-time interval" (PCVI).

 In the diagram of FIG. 10, a time discriminator 83, a key 84, a reverse counter 85, and a code-time interval converter 86 are connected in series, the output of which is the output of the range finder 35 and the second input (strobe pulse) of the temporary discriminator 83, the input of the latter being the first (signal) input of the range finder 35, the second input (initial range code) of the range finder 35 is connected to the second input (initial setting) of the reverse counter 85. The third input (command) of the range finder is connected to the control input of the key 84, and the fourth and fifth inputs (clock pulse s and clock) -, respectively, the second and third inputs of the converter 86 "code-time slot".

The analysis and decision unit 24 (BAR) can be made in the form of a programmable microprocessor, the program circuit of which is presented in FIG. 11. It consists of the following blocks:
87 - unit assigning the initial values of variables, namely: range counter i takes the values of the number of the initial range i ₀ , the code of the selected range R is the minimum value of the beginning of the zone R ₀ stored when exiting the program in the absence of real targets,
88 is a block for checking the achievement of the number of the final range i _N (end of the zone). If the condition i ≤ i _{N is} not met, the program exits,
89 is a block comparing signals proportional to the powers of the spectral components S _ri in the frequency range from - F _{P / 2} to F _{P / 2} , with a detection threshold S ₀ for the i-th range element, that is, checking the condition
S _ri <S ₀
for
r = - m / 2, ..., m / 2 (m is even).

затем переход к блоку 91,
91 - блок обнуления параметров циклов κ₁,κ₂,, затем переход к блоку 92,
92 - проверка условия выхода из цикла по κ₁ при достижении максимального значения κ. При выполнении условия κ₁≥κ производится переход к следующему элементу дальности (блок 93), при невыполнении - к блоку 94. Этой проверкой исключается возможность зацикливания программы при неограниченном увеличении параметра цикла κ₁,
94 - блок нахождения нижней граничной частоты спектра с максимальной спектральной составляющей путем определения минимального целого числа κ₁, не удовлетворяющему условию
S(F_*-κ₁-1)≥aS(F_*),
где
а < 1 - выбранный заранее относительный уровень отсчета ширины спектра (например, а = 0,5). При выполнении этого условия производится переход к блоку 95, при невыполнении - к блоку 96,
95 - блок увеличения значения параметра цикла κ₁ на 1 с последующим переходом к проверке его значения в блоке 92,
96 - блок проверки условия выхода из цикла по κ₂ при достижении им максимального значения κ. При выполнении условия κ₂≥κ производится переход к следующему элементу дальности (блок 93), при невыполнении - к блоку 97,
97 - блок нахождения верхней граничной частоты спектра с максимальной спектральной составляющей путем определения минимального целого числа κ₂ не удовлетворяющего условию
S(F_*+κ₂+1)≥aS(F_*),
При выполнении этого условия производится переход к блоку 98, при невыполнении - к блоку 99.When this condition is met, the transition to the next element of the range (block 93), if not fulfilled - to block 90,
90 - block finding the maximum value of the spectral component

then go to block 91,
91 - block zeroing parameters of the cycles κ ₁ , κ ₂ ,, then go to block 92,
92 - checking the conditions for exiting the cycle by κ ₁ upon reaching the maximum value of κ. When the condition κ ₁ ≥κ is fulfilled, the transition to the next range element is performed (block 93), if not fulfilled, to block 94. This check eliminates the possibility of the program looping with an unlimited increase in the cycle parameter κ ₁ ,
94 - block finding the lower cutoff frequency of the spectrum with the maximum spectral component by determining the minimum integer κ ₁ that does not satisfy the condition
S (F _* -κ ₁ -1) ≥aS (F _* ),
Where
and <1 is the pre-selected relative level of the spectrum width reference (for example, a = 0.5). When this condition is met, a transition is made to block 95, if not fulfilled, to block 96,
95 - block increasing the value of the parameter of the cycle κ ₁ by 1, followed by the transition to checking its value in block 92,
96 - block checking the conditions for exiting the cycle by κ ₂ when it reaches the maximum value of κ. When the condition κ ₂ ≥κ is fulfilled, the transition to the next element of the range is performed (block 93), if not fulfilled - to block 97,
97 is a block for finding the upper cutoff frequency of the spectrum with the maximum spectral component by determining the minimum integer κ ₂ not satisfying the condition
S (F _* + κ ₂ +1) ≥ aS (F _* ),
When this condition is met, a transition is made to block 98, if not fulfilled, to block 99.

98 - block increase the value of the parameter of the cycle by 1 with the subsequent transition to checking its value in block 96,
99 is a block comparing the width of the spectrum with the maximum spectral component with a threshold value of "κ" by checking the conditions
κ ₁ + κ ₂ + 1≥κ
If this condition is met, the transition to the next element of the range (block 93), otherwise, the transition to block 100,
100 is a unit for assigning a value R = R _{* to a} code selected to accompany a range element i _* , outputting an R _* value to a rangefinder 35, and outputting a frequency value F _* through a two-input adder to a frequency offset unit 8. The initial value of R ₀ , remaining after assignment in block 87 in the absence of a real target, means the absence of the target for tracking either due to its non-detection, or due to the classification of all targets in a given range of ranges as interference.

 In accordance with the diagrams of FIG. 1 - 11 radar that implements the proposed method, works as follows.

 Before starting the radar in tracking mode, the antenna 5 is installed in the direction of the target and the size and position of the viewing area in range are determined. This is done according to target designation data obtained, for example, from a navigation system or as a result of the operation of the same radar in viewing mode.

Microwave oscillations generated by the pathogen 1 at a frequency f _c pass into a power amplifier 2, in which they are amplified, and probing pulses are formed under the influence of a pulse modulator 6 controlled by clock pulses from a synchronizer 7, following with a repetition frequency F _P. They pass the antenna switch 3, the total-differential converter 4 and are emitted by the antenna 5 into space.

The reflected signals from the antenna 5 pass through the total channel through the sum-difference converter 4 and the antenna switch 3 to the high-frequency amplifier 9 of the total channel, where the amplification occurs at the frequency of the received signals, and enter the mixer 11 of the total channel. At the same time, the signals from the antenna 5 pass through the difference channel (in the radar circuit in Fig. 1, one differential receive channel is shown, if necessary, the second differential receive channel is implemented in the same way) through the total-differential converter 4 to the high-frequency amplifier 10 of the differential channel, where amplification occurs according to the frequency of the received signals, and fall into the mixer 12 of the differential channel. In mixers 11 and 12, the superheterodyne conversion of the received signals to an intermediate frequency occurs, and the oscillations of the microwave frequency f _r generated in the exciter 1 and coming from the output 2 of the exciter 1 to the heterodyne inputs of the mixers 11 and 12 are used as heterodyne oscillations.

 Pathogen 1 (Fig. 2) works as follows.

The master oscillator 39 generates continuous oscillations of a stable frequency, from which, by multiplying in the multiplier 40 and amplification in the amplifier 43, oscillations are generated with the carrier frequency of the probing signals f _c , which are supplied to the first output of the exciter. The reference frequency generator 42 generates stable oscillations of the intermediate frequency f of the _inverter , which are fed to the input of the mixer 41, to the other input of which the oscillations of the signal frequency come. After mixing at the mixer output, oscillations of the local oscillation frequency f _{r are formed} (for example, f _r = f _c + f _IF ), which, after amplification in the amplifier 44, go to the second output of the pathogen 1. Finally, the oscillations of the intermediate frequency after amplification in the amplifier 45 go to the third pathogen output 1.

 After the superheterodyne conversion of the signals in the mixers 11 and 12, the signals in the total and difference channels are amplified by the intermediate frequency in the amplifiers 13 and 14 of the intermediate frequency, respectively, and then they are fed to the first (signal) inputs of the corresponding phase detectors 15 and 16 in the total channel, 17 and 18 - in the difference.

The second inputs - the inputs of the reference frequency of the phase detectors - receive reference oscillations from the output of the frequency offset unit 8, and directly to the first phase detectors 15 and 17 in the total and difference channels, and to the second phase detectors through a phase shifter 19 by 90 ^° , that both quadrature and difference channels form two quadratures, which, as you know, eliminates the influence of the unknown initial phase of the received signals.

 Block 8 frequency offset operates as follows (Fig. 3).

The first input of the frequency offset unit 8 receives intermediate frequency oscillations from the third output of the pathogen 1, the second and third inputs of unit 8 receive the Doppler frequency codes F of the target _DC and Doppler frequency F _DO corresponding to the component of the radar carrier speed in the direction to the target (for example, radar carrier navigation system). These codes are supplied to the first inputs of the code-frequency converters 46 and 47, in which they are converted into oscillations of the corresponding frequencies.

 Converters 46 and 47 "code-frequency" work as follows (Fig. 4).

 The second inputs of the code-frequency converters 46 (47) receive stable frequency clock pulses from the first output of the synchronizer 7 through the fourth input of the frequency offset unit 8.

 These pulses arrive at the information input of a controlled frequency divider 51, made on the basis of a counter, to the control input of which control signals determining the frequency division coefficient are received through the first input of the converter 46 (47) and the decoder 50. Square-wave pulses in the form of a meander with the required frequency are supplied from the output of the frequency amplifier 51 to the low-pass filter 52, which isolates the first harmonic, sinusoidal oscillations of the required frequency are supplied to the output of the code-frequency converter 46 (47), and the minimum frequency value in absolute value corresponds to the unit of the least significant bit of the code.

The oscillations of the frequencies F _DC and F _DO in block 8 are fed to the inputs of the dual balanced modulator 48, the output of which is formed by the oscillations of the total frequency F _DC + F _DO received at the second input of the dual balanced modulator 49, the first input of which through input 1 of block 8 oscillations of the intermediate frequency f _IF , at the output of the double balanced modulator 49, oscillations of the total frequency f _IF + F _DC + F _{DO are formed} , which pass to the output of the frequency offset unit 8 as oscillations of the reference frequency.

 Dual balanced modulator 48 (49) works as follows (Fig. 5).

Let the oscillations a ₁ cos (2πf ₁ t-φ ₁ ) and the oscillations a ₂ cos (2πf ₂ t-φ ₂ ) come to the first input of the double balanced modulator, then the first input of the multiplier 54 after a delay of 90 ^o in phase in the phase shifter 53, vibrations a ₁ sin (2πf ₁ t-φ ₁ ) come, and the second input of the oscillations a ₂ cos (2πf ₂ t-φ ₂ ), after multiplication, the product comes to the first input of adder 57
κa ₁ a ₂ sin (2πf ₁ t-φ ₁ ) cos (2πf ₂ t-φ ₂ )
κ is the coefficient of proportionality.

The first input of the multiplier 56 receives oscillations a ₁ cos (2πf ₁ t-φ ₁ ), and the second input of the multiplier 56, after a delay of 90 ^o in phase in the phase shifter 55, receives oscillations a ₂ sin (2πf ₂ t-φ ₂ ).
After multiplying the second input of the adder 57 comes the product
κa ₁ a ₂ cos (2πf ₁ t-φ ₁ ) sin (2πf ₂ t-φ ₂ ).
As a result of the summation at the output of the adder 57, oscillations are formed
κa ₁ a ₂ sin [2π [f ₁ + f ₂ ) t-φ ₁ -φ ₂ ],
those. total frequency f ₁ + f ₂ .

Видеоимпульсы с квадратур согласованного фильтра 20 в суммарном канале поступают на блок 22 ключей, представляющий собой пару одинаковых ключевых каскадов, которые открываются лишь на время действия на их управляющие входы прямоугольных видеоимпульсов - стробов с длительностью

(где c - скорость света), соответствующей разрешению по дальности ΔR. Эти импульсы приходят с выхода преобразователя 25 "код-временной интервал", на вход которого поступает код дальности с блока 24 анализа и решений, определяющий их задержку относительно синхроимпульсов, поступающих со второго выхода синхронизатора 7 на 3-ий вход преобразователя 25 "код-временной интервал". Пройдя блок 22 ключей, видеоимпульсы сигналов от целей поступают на блок 23 доплеровских фильтров.After phase detection, the video signals of the quadrature of the sum and difference channels are fed to the inputs of the matched filters 20 and 21, respectively, which in the case of simple pulse sounding signals - rectangular radio pulses with a duration of τ _AND - are pairs of identical video amplifiers with a cutoff frequency equal to -

Video pulses from the quadrature of the matched filter 20 in the total channel are sent to the key block 22, which is a pair of identical key stages that open only for the duration of the action on their control inputs of rectangular video pulses - strobes with a duration of

(where c is the speed of light) corresponding to the range resolution ΔR. These pulses come from the output of the transducer 25 "code-time interval", the input of which receives a range code from block 24 analysis and solutions, determining their delay relative to the clock pulses from the second output of the synchronizer 7 to the 3rd input of the transducer 25 "code-time interval". Having passed block 22 keys, the video pulses of the signals from the targets are sent to block 23 Doppler filters.

 Block 23 Doppler filters, which can be made in the form of a combination of m / 2 two-channel digital filters (Fig. 6), operates as follows.

The bipolar video pulses of the signals of the quadrature channels pass through the first and second inputs to analog-to-digital converters 58 and 59, where they are converted into multi-bit numbers, which are quantized signals that are sent to m / 2 of parallel-connected 60 _r filters. The third input of block 23 Doppler filters receives clock pulses with a repetition frequency F _P from the synchronizer 7. Two-channel Doppler filter 60 _r operates as follows (Fig. 8).

где

После объединения квадратурных составляющих на выходах блоков 75, 76 образуются соответственно сигналы

которые проходят на выходы +rΔF,-rΔF блока 60_r соответственно.Quantized Signals in Quadrature Channels X $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{l}}$ , X $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{l}}$ get respectively through the first and second inputs 1 and 2 of the two-channel Doppler filter 60 _r to the blocks of the shift registers 65, 66, respectively, filled by the action of advance pulses with a repetition frequency F _P coming from the third input of the block 23 of the Doppler filters to the third input of the two-channel Doppler filter 60 _r . In blocks 65, 66 of the shift n-bit registers, the signal samples are stored for n repetition periods. The stored signals from all n digits go through the taps to the inputs of the n-input weight adders 67 and 68 for the first quadrature - and 69 and 70 - for the second quadrature, after multiplying by the weighting factors and summing the signals are fed to the inputs of the addition-subtraction blocks 71 - 74, at the outputs of which the following quadrature components are formed:

Where

After combining the quadrature components at the outputs of blocks 75, 76, signals are formed respectively

which go to the outputs + rΔF, -rΔF of the block 60 _r, respectively.

The above transformations are, in essence, algorithms for multichannel optimal filtering of the process X _l = X $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{l}}$ + iX $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{l}}$ with discrete time.

The signals from the outputs of the two-channel Doppler filters 60 _r go to the output of the block 23 of Doppler filters, where they are ordered in order of increasing frequency tuning, and then to the input of block 24 analysis and solutions. The operation of the analysis and decision block 24 consists in executing the program that has been described above, and the circuit of which is shown in FIG. eleven.

The analysis and decision block 24 analyzes the signals from the range elements, starting from the range R _{0 of the} beginning of the zone to which the number i ₀ corresponds. If at this range in none of the m frequency channels there are signals exceeding the detection threshold, or if they are, but their spectral width exceeds a predetermined value, a decision is issued from the first output of the analysis unit 24 and decisions about the transition to the analysis of the next range element - with the number i ₀ + 1, the corresponding code is fed to the converter 25 "code-time interval", the inputs of which receive clock pulses with a repetition frequency F _P - from output 2 of the synchronizer 7 - and pulses with a clock frequency f _T = 1 / τ _And , carrying out with a range of the time clock. As a result, at the output of the code-time interval converter 25, a strobe pulse is delayed relative to the previous position by a value of τ _And corresponding to the resolution element, this strobe pulse is incident on the control input of the key block 22 and analysis of the next range element begins, and so until a signal with a spectral width smaller than the threshold value κΔF is detected in some range element with number i _* . In this case, the analysis and decision block 24 measures the frequency F _* corresponding to the maximum of the Doppler spectrum of the signal from the target, and transfers the code of this frequency through the second output and the first two-input adder 26 to the second input of the frequency offset unit 8, due to which the frequency changes by F _* reference oscillations coming from the output of the frequency offset unit 8 to the phase detectors 15-18, so that the resulting Doppler frequency of the signals from the target becomes close to zero. At the same time, according to the signals following from the analysis and decision block 24, the range finder 35 is opened (at input 3) and its initial value 2 receives the value of the target's initial range R _* .

 After that, through the Doppler filters 29 and 30 of zero frequency in the total and difference channels, respectively, signals from the target begin to pass.

 The Doppler filter 29 (30) of zero frequency operates as follows (Fig. 9).

которые соответственно, поступают на выходы 1 и 2 доплеровского фильтра 29 (30) нулевой частоты.The video pulses of the signals of the quadrature channels from the outputs of the matched filters 20 and 21 - in the total and difference receiving channels, respectively - get through inputs 1 and 2 to the analog-to-digital converters 77 and 78, respectively, which quantize the signals at several levels. The quantized signals from the outputs of the analog-to-digital converters 77 and 78 X $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{l}}$ , X $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{l}}$ fall respectively on the blocks of n-bit shift registers 79 and 80, filled under the action of advancing clock pulses with a repetition frequency F _P , fed to the third input of the Doppler filter 29 (30) of zero frequency from the second output of the synchronizer 7. In blocks of shift registers 79 and 80, which in the simplest case of quantization of signals in analog-to-digital converters to 2 levels (0,1) can be made in the form of single shift registers, the signal samples are stored for n repetition periods. The stored signals from all n bits go through taps to the inputs of the n-input adders 81 and 82 for the first and second quadrature, respectively, the following signals are formed at their outputs, respectively

which, respectively, are supplied to the outputs 1 and 2 of the Doppler filter 29 (30) of zero frequency.

не зависящий от неизвестной начальной фазы принимаемых сигналов. Блок 33 объединения квадратурных каналов, как и аналогичные блоки 75 и 76, может быть выполнен в виде двухадресного постоянного запоминающего устройства, содержащего значения функции

, от двух аргументов.Signals Y $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{0}}$ , Y $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{0}}$ from the outputs 1 and 2 of the Doppler filter 29 of zero frequency in the total channel are fed to block 33 combining quadrature, which forms a signal

independent of the unknown initial phase of the received signals. Block 33 combining quadrature channels, like similar blocks 75 and 76, can be made in the form of a dual-address read-only memory containing function values

, from two arguments.

After the signal from the target Y ₀ to the input 1 of the range finder 35, automatic tracking of the target in range begins.

 The range finder 35 operates as follows (Fig. 10).

 The signal from the target through the first input of the range finder 35 goes to the first input of the time discriminator 83, a pulse from the code-time interval converter 86 is supplied to its second input (the operation of the converter 86 occurs as well as the code-time interval converter 25 described above ), the input of which comes with a code of the initial range to the target - through the second input of the range finder 35. At the output of the temporary discriminator, a bipolar code signal of time inconsistency is generated, passing through the key 84, opened by the command that came from the tre rd input rangefinder 35, this signal goes to the count input of down counter 85, where the sum (with sign) from the initial code range, the new value of the code range is supplied to the first input transducer 86 "code-time interval", etc.

 The strobe pulse, the position of which relative to the clock corresponds to the distance to the target, enters through the output of the rangefinder 35 to the control inputs of the second block 27 of the key and key 36. The block 27 of the keys opens while the pulses of the signal from the target pass, passing them to the frequency discriminator 28, forming the signal Doppler frequency mismatch.

 Frequency discriminator 28 works as follows (Fig. 7).

The video pulses of the quadrature components of the signal from the tracked target through the key block 27 pass from the matched filter 20 of the total channel to the inputs 1 and 2 of the frequency discriminator 28, then they are quantized into several levels in the analog-to-digital converters 61 and 62, respectively, the quantized signals are fed to a two-channel Doppler filter 63, the channels of which are tuned to the frequencies "ΔF" and "-ΔF" located symmetrically with respect to the zero frequency. The device and operation of the two-channel Doppler filter 63 completely coincides with the device and operation of the two-channel Doppler filter 60 _r (Fig. 8) at, r = 1. The signals from the outputs ΔF and -ΔF of the filter 63 go to the inputs of the subtraction block 64, where a bipolar encoded mismatch signal of the Doppler frequency of the received signal with respect to the zero frequency is generated, this mismatch signal from the output of the frequency discriminator 28 goes to the second input of the first two-input adder 26, where it is added to the frequency code F _* he previously from the analysis and decision block 24, the resulting Doppler frequency code falls on the second input of the frequency offset unit 8 and changes the frequency of the reference oscillations, due to which the signals from the target are tracked by the Doppler frequency, the value of which is kept near zero after phase detection.

где
c - коэффициент пропорциональности.Simultaneously with the key block 27, the strobe from the rangefinder 35 also opens the key 36, to the output of which the angle mismatch signal is generated after multiplying the quadrature components of the same name from the target, which passed through the total and difference receiving channels through the corresponding zero frequency Doppler filters 29 and 30, and summing these products in a two-input adder 34 according to the scalar product rule, namely

Where
c is the coefficient of proportionality.

 As is known, this rule, up to a normalizing factor, coincides with the optimal rule for estimating angular mismatch during monopulse direction finding (for example, [2], p. 20, (1)).

 The angular error signals obtained in this way, passing through the key 36, are accumulated in the integrator 37, the resulting signal from the output of the integrator 37 controls the antenna 1 using the antenna drive 38.

, поступающие с первого выхода синхронизатора 7 на четвертый вход блока 8 смещения частоты и на счетные входы преобразователей 25 и 86 "код-временной интервал" (последний в составе дальномера 35), а также синхроимпульсы с частотой повторения F_П со второго выхода - для запуска импульсного модулятора 6 и для управления работой преобразователей 25 и 86 "код-временной интервал". Синхронизатор 7 может быть построен на основе задающего генератора импульсов с частотой f_Т и счетчика-делителя с формирователем - для формирования синхроимпульсов с частотой F_П<f_Т.The synchronizer 7, providing overall control of the radar, generates clock pulses with a frequency

coming from the first output of the synchronizer 7 to the fourth input of the frequency offset unit 8 and to the counting inputs of the code-time interval converters 25 and 86 (the last in the range finder 35), as well as clock pulses with a repetition frequency F _P from the second output - to start pulse modulator 6 and to control the operation of the converters 25 and 86 "code-time interval". The synchronizer 7 can be built on the basis of a master pulse generator with a frequency f _T and a counter divider with a shaper - for the formation of clock pulses with a frequency F _P <f _T.

 Thus, a radar station that implements the claimed method performs target tracking in angular coordinate, in range and in Doppler frequency at a higher noise immunity compared to the prototype in relation to passive interference, both organized - like dipole clouds, and natural - reflections from coastline, and for radars using FM signals, there is also an increase in tracking accuracy and noise immunity with respect to active noise interference.

 The effect of increasing noise immunity with respect to passive interference can be estimated as follows.

что при σ_ц= 5•10³м²,ϑ≤0,1,σ⁰= 0,03,ΔR = 300 м, R = 10 км, φ_0,5 = 0,15, Х = 300 м составляет ρ₁ 0,4 (-4 дБ).In the conditions of the above example (Fig. 12) we have for the signal-to-noise ratio in the prototype, as follows from the above, even in the absence of frequency resolution

that when σ _C = 5 • 10 ³ m ² , ϑ≤0.1, σ ⁰ = 0.03, ΔR = 300 m, R = 10 km, φ _0.5 = 0.15, X = 300 m is ρ ₁ 0.4 (-4 dB).

 This value of the signal / noise ratio is not enough for reliable detection of the signal from the target (NK) against the background of interference, especially for capture and tracking.

и при ширине спектра отражений от участка берега внутри импульсного объема, определяемого реальной ДНА и разрешением по дальности R (см. выше)

получим, что отношение сигнал/помеха возрастает в

раз, так как сигналы от НК целиком попадают в полосу ΔF = 10 Гц (при интервале корреляции флюктуаций сигналов от НК, не превосходящем 0,1 с в диапазоне длин волн λ≅ 3,0 см), а отражения от берега отфильтровываются, как показано выше.In the proposed method in the same conditions at a time of coherent processing

and with the width of the spectrum of reflections from the coastal section inside the pulse volume, determined by the real BOTTOM and range resolution R (see above)

we get that the signal / noise ratio increases in

times, since the signals from the NC completely fall into the band ΔF = 10 Hz (with a correlation interval of fluctuations of signals from the NC not exceeding 0.1 s in the wavelength range λ≅ 3.0 cm), and reflections from the shore are filtered out, as shown above.

т. е. 20 раз (13 дБ), что достаточно для уверенного захвата и сопровождения по дальности и угловым координатам (при моноимпульсной пеленгации).Thus, the signal-to-noise ratio in the proposed method increases, for example, by 17 dB compared with the prototype and is

i.e., 20 times (13 dB), which is sufficient for confident capture and tracking in range and angular coordinates (with monopulse direction finding).

По мере сближения носителя РЛС с целью вероятности захвата и сопровождения возрастают как ввиду повышения отношений сигнал/шум и сигнал/помеха (см. (8), (9)), так и благодаря улучшению условия разрешения (величина R в (10) уменьшается). Наконец, в ситуациях, когда направление БЧ перпендикулярно направлению РЛС - НК, при условии нахождения НК от БЧ на расстояниях, превосходящих ΔR - разрешение по дальности, имеет место разделение сигналов от НК и от БЧ по разным каналам дальности (благодаря разрешению по дальности) и предлагаемый способ работает в облегченных условиях, а когда расстояние НК от БЧ меньше ΔR, осуществляется частотная селекция, как описано выше, с достаточно высокой эффективностью.Moreover, this takes place in the presence of frequency resolution, that is, when the signals from the NK and from the warhead are separated by frequency channels - when the condition

As the carrier approaches the radar with the aim of the probability of capture and tracking, they increase both due to an increase in the signal-to-noise and signal-to-noise ratios (see (8), (9)) and due to an improvement in the resolution condition (the value of R in (10) decreases) . Finally, in situations where the direction of the warhead is perpendicular to the direction of the radar - NK, provided that the NK from the warhead at distances exceeding ΔR - range resolution, there is a separation of signals from the NK and from the warhead at different distance channels (due to range resolution) and the proposed method works in light conditions, and when the distance of the NC from the warhead is less than ΔR, frequency selection is carried out, as described above, with a sufficiently high efficiency.

 Thus, the proposed method is effective for any relative locations of the radar-NK-warhead.

и выражения для дисперсии для заявляемого способа, которое имеет вид

Сравнение формул (6) и (11) показывает, что в заявляемом способе дисперсия в 1/ρ₀ меньше (обычно ρ₀≪ 1 - отношение сигнал/шум до сжатия, так как N > 1), а точность соответственно выше, чем в способе-прототипе.The effect of increasing noise immunity with respect to active noise interference and the accuracy of angular tracking in radars using complex, in particular phase-shifted signals, is determined by comparing the expression for the dispersion of the angle measurement with monopulse direction finding, which for the prototype method has the form (at ρ ₀ <1) ( see above (6))

and expressions for dispersion for the proposed method, which has the form

A comparison of formulas (6) and (11) shows that in the claimed method the dispersion in 1 / ρ _{0 is} less (usually ρ ₀ ≪ 1 is the signal-to-noise ratio before compression, since N> 1), and the accuracy is correspondingly higher than in prototype method.

Under the influence of noise interference, the signal-to-noise ratio decreases, while σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{β}}$ increases in the claimed method as 1 / ρ ₀ , and in the prototype as 1 / ρ $\genfrac{}{}{0ex}{}{\text{2}}{\text{0}}$ , i.e., to a greater extent, so that the claimed method has greater noise immunity compared to the prototype. In particular, when ρ ₀ = 0.1, the dispersion in the claimed method is 10 times less than in the prototype, respectively, the accuracy is 3.15 times higher.

 Thus, the technical effect in the industrial use of the proposed method is to achieve higher noise immunity in relation to passive interference, and in radars using complex signals, also in relation to active interference while achieving higher accuracy compared to the prototype.

 Using the information presented in the application materials, a device that implements the proposed method can be manufactured and used in radars with a single-pulse direction finding principle, including those using complex, in particular phase-shifted signals, which proves the industrial applicability of the subject invention.

 In accordance with the application materials, a prototype of the device was manufactured, the tests of which confirmed the achievement of the technical result indicated in the application materials.

Sources of information
1. Leonov A.I., Fomichev K.I. Monopulse radar. - M.: Soviet Radio, 1970.

 2. Guide to radar / Ed. M. Skolnik, T. 4.- M .: Soviet Radio, 1978, p. 20, fig. 15, prototype.

 3. Handbook of radar / Ed. M. Skolnik, vol. 1.-M .: Soviet Radio, 1976.

 4. Falkovich S.E. Reception of radar signals against fluctuation interference. -M .: Soviet Radio, 1961.

 5. Krasyuk N.P., Rosenberg V.I. Shipborne radar and meteorology. - L .: Shipbuilding, 1970.

Claims (1)

A method of tracking a target with a monopulse radar station, including emitting pulsed coherent signals in a given direction, receiving high-frequency signals in a given range of distances, sum-difference converting the received signals, superheterodyne converting them to an intermediate frequency, amplifying the sum and difference signals at an intermediate frequency, characterized in that after amplification of the sum and difference signals at an intermediate frequency, the spectrum of signals is converted into a region in ideofrequencies, through phase detection using reference oscillations with the formation of quadrature components of each signal, carry out a coordinated filtering of video pulses of the quadrature components of signals, for each range element in a given interval, carry out multi-channel Doppler filtering of the complex envelope of the pulse sequence of the total signal in the Doppler frequency range

with ΔF band and number of channels

where F _p is the pulse repetition rate of the probe pulses, ΔF is the spectrum width of inter-period fluctuations of the signals from the true targets, the powers of the spectral components are compared with the detection threshold level, the spectrum width with the most powerful spectral component is determined from the set level from the number of components that exceeded the detection threshold, and the obtained the value of the width of the spectrum with a given threshold value, when it is exceeded, they decide on the presence of a signal from a false target and go to the next element of range, and if there is no excess of the specified threshold value by the obtained spectral width, a decision is made on the presence of a signal from the true target in the corresponding range element, the Doppler frequency corresponding to the maximum of the spectrum is determined, the frequency of the reference oscillations is changed by the value of the measured Doppler frequency, the frequency error signal is determined, close the frequency tracking loop and adjust with this signal the frequency of the reference oscillations, carry out narrow-band filtering with band ΔF of the complex envelope of the pulse sequences of signals in the total and difference channels at zero Doppler frequency, the amplitude envelope of the signals in the total channel is extracted, the error signal in range in the total channel is isolated, the range tracking loop is closed and the position of the range gate is adjusted with this signal, the angular signal is isolated discrepancies by pairwise multiplication of the filtering results of the same quadrature signal components in the total and difference channels and sum Migrations of these works, close the tracking loop along the angle with a range gate and adjust the signal of the angular mismatch of the antenna position.

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