RU2099739C1 - Radar - Google Patents

Abstract

FIELD: radiolocation, radars of autonomous and command and control systems. SUBSTANCE: given radar is related to radars meant for detection of signals from targets, for measurement of their coordinates and automatic tracking by range and angular coordinates in presence of natural and organized interference. EFFECT: increased noise immunity of radar both under modes of detection and tracking with simultaneous increase of tracking accuracy. 1 cl, 15 dwg

Description

 The invention relates to the field of radar technology and can be used in radar stations (autonomous radars) of autonomous and command control systems designed to detect signals from targets, measure their coordinates and automatically track in range and angular coordinates in the presence of natural and organized radio interference.

 At present, the use of complex signals, that is, signals with intrapulse modulation, is an important factor determining the increase in the basic quality indicators of the radar, especially the detection range, resolution and noise immunity with respect to natural and organized radio interference. The most promising, especially for radars of autonomous and telecontrolled control systems, are signals with intrapulse phase shift keying (FM) binary multi-bit code that allow the organic use of digital methods for generating and processing signals.

 Known radar [1] which contains a transceiver capable of emitting and receiving signals with intrapulse phase shift keying or frequency modulation, a signal limiting block, a pulse compression device and a threshold block. In the indicated radar, the signals reflected from the target after reception are limited in amplitude, are compressed in time in a matched filter constructed on the basis of a delay line with taps, and then are detected in the threshold block. The disadvantage of this radar is the operation at one frequency and, as a result, insufficient noise immunity with respect to impact noise interference.

 Known radar [2] which is the closest in technical essence to the proposed device and adopted as a prototype. The prototype device allows the tuning of the carrier frequency from pulse to pulse according to an arbitrary law and uses signals with intrapulse phase shift keying binary multi-bit code. The radar is built on a coherent principle and contains a serially connected synchronizer, transmitter, antenna switch and antenna, a receiver and an information processing device connected to the third arm of the antenna switch, the transmitter being made on the basis of series-connected exciter, phase manipulator and power amplifier controlled by a frequency tuning unit, a code generator and a pulse modulator, respectively, and the receiver contains a high-frequency amplifier, a decoding device, a second mixer and a second mixer (phase detector), the frequency tuning unit being connected to the control input of the exciter, the code generator connected to the control inputs of the phase manipulator and the decoding device, the output of the local oscillator frequency of the exciter connected to the heterodyne input of the mixer, and the reference frequency output of the exciter with the reference input frequency detector phase.

 Due to the use of complex FM signals with the tuning of the carrier frequency from pulse to pulse, this radar has a higher noise immunity with respect to response and aiming frequency interference than a radar that uses simple pulse signals with the same energy, without tuning the carrier frequency from pulse to pulse .

 The disadvantage of the radar prototype is the low accuracy of coordinate measurement with not enough stealth and noise immunity, which is a consequence of the inclusion of the FM signal compression device (the "decoding device", as it is called in the description) directly behind the receiver’s high-frequency amplifier, which means that the compression The FM signals in the prototype are performed at the frequency of the received signals, which limits the possibility of its implementation by the relatively short durations of complex signals (up to 10-20 μs). This, in turn, leads to low accuracy of coordinate measurement and insufficiently high stealth and noise immunity of the radar in relation to response interference. In particular, this disadvantage is manifested in radars designed not only to detect signals from targets and measure their coordinates in the viewing mode, but also to automatically track targets in range and in angular coordinates based on monopulse direction finding. The low stability of the parameters of complex signal compression devices (see, for example, [10, p. 175]) operating at ultrahigh carrier frequencies of the reflected signals limits the possibility of their implementation in monopulse receivers, which require high identical phase-frequency characteristics of the sum and difference receiving channels, Comparatively short durations of complex signals (up to 10 μs).

 In addition, the noise immunity and secrecy of the radar prototype in tracking mode also become insufficient for another reason due to an increase in the power of the probing radiation at the input of reconnaissance receivers included in electronic countermeasures (RE) (for example, type AN SLQ-32, see [11] ), as the radar approaches the target, and the increase in power increases inversely with the square of the distance between the radar and the target covered by the REP (for example, a surface ship). The loss of secrecy leads to an increase in the possibility of creating interference in the tracking mode.

, соответствующей длительности T_и зондирующих сигналов (C скорость света), и длительность принимаемых сигналов

становится соответственно меньше длительности зондирующих (из-за бланкирования приемника), устройство сжатия оказывается несогласованным с принимаемыми сигналами, что также ведет к потерям в точности измерения координат и помехозащищенности.Further, these features of the construction of the radar prototype do not allow you to adjust the duration and width of the spectrum of the probing signals as the distance to the target decreases, and in conditions where the distance to the circuit R is less than

corresponding to the duration of T _and sounding signals (C is the speed of light), and the duration of the received signals

becomes accordingly shorter than the duration of the probing (due to receiver blanking), the compression device is inconsistent with the received signals, which also leads to losses in the accuracy of coordinate measurements and noise immunity.

 An object of the invention is to increase the noise immunity of the radar both in the detection mode and, in particular, in the tracking mode while improving the accuracy of tracking.

 To achieve the claimed technical result in the proposed radar, the compression of complex FM signals is carried out not on the carrier frequency of the received signals, but on the video frequency in digital matched filters (CSF) after heterodyne conversion, intermediate frequency amplification, phase detection and signal amplification on the video frequency in two quadrature channels both in the total and in the difference receiving channel with a single-pulse direction finding method in tracking mode. In this case, the video signals at the input of the CSF are subjected to amplitude binary quantization with a zero threshold level and time quantization, the angle mismatch signals are generated by the scalar product of ternary quantized compressed signals from the outputs of the CSF of the total and difference channels, to control the steepness of the direction-finding characteristic, the power of the probe pulses is controlled depending on the measured signal-to-noise ratio by signals from the target being tracked, to eliminate the measurement error ti tracking target due to quantization of time input signals changing moments CSF is administered in time with changes in the quantization target range. In addition, as the duration of the received signals decreases, the range to the tracking target decreases to increase the accuracy of coordinate measurement and noise immunity, the CSF bandwidth is changed while the primary detection threshold is adjusted at the CSF output, as well as an abrupt decrease in the duration of the probing signals by switching to signals with more high resolution in range while maintaining the FM code with the simultaneous adjustment of the CSF in the total and difference receiving channels and bandwidth expansion of both receiving channels.

 The essence of the invention lies in the fact that in a radar station containing a serially connected synchronizer, a frequency tuner, an exciter, a phase manipulator and a power amplifier, a serially connected antenna switch and an antenna, a serially connected receiver and a first block of quadrature phase detectors, as well as a code generator, a pulse modulator and an information processing device, the first input of the receiver being connected to the third arm of the antenna switch, the generator outputs are diodes and a pulse modulator are connected to the control inputs of the phase manipulator and the power amplifier, respectively, the output of the local oscillator frequency of the pathogen is connected to the local oscillator input of the exciter, the output of the reference frequency of the exciter with the input of the reference frequency of the block of quadrature phase detectors, a threshold generation unit is introduced, connected in series through two input-outputs the quadrature components of the total signal is the first video amplifier, the signal inputs of which are connected to the outputs of the first block of quadrature phases output detectors, a first amplitude quantizer, a first digital matched filter and a quadrature combining unit, the first to third outputs of which are connected to the first, second and third inputs of the information processing device, respectively, connected in series through the two inputs and outputs of the quadrature components of the difference signal of the second quadrature phase block detectors, the signal input of which is connected to the second output of the receiver, which consists of two identical channels of total and difference, and the input of the reference part the frequencies to the corresponding output of the exciter, the second video amplifier, the second amplitude quantizer, the second digital matched filter and the angular discriminator, the third and fourth inputs of which are connected respectively to the first and second outputs of the first matched filter, connected in series to the first switch, the information input of which is connected to the output of the angular discriminator , and the integrator is connected in series to the second switch, the information input of which is connected to the third output of the unit a quadrature unit, an intensity measuring unit, a power gradation decoder, a code-voltage converter and a power regulator, the second input of which is connected to the output of the power amplifier, and the output to the second arm of the antenna switch, the antenna contains an antenna unit connected to a sum-difference converter, and connected with it, the kinematic drive of the antenna, the first input of which is connected to the output of the integrator, and the second input to the output of the azimuth signal of the target of the information processing device, the output of the difference signal of the antenna under it is connected to the second input of the receiver, the control inputs of the switches are combined and connected to the output of the range gate of the information processing device, the third output of which is connected to the control inputs of the video amplifiers and the input of the synchronizer, and the fourth output with the control input of the frequency tuning unit, the second output of the synchronizer is connected to the input of the pulse modulator and to the first input of the code generator, the second clock input of which is connected to the third output of the synchronizer, and the output is connected to the control inputs of digital matched filters, the inputs of the zeroing signal of which, the fourth input of the information processing device, as well as the inputs of the zeroing signals of the intensity measurement and threshold generation units are connected to the fourth output of the synchronizer, the third output of which is also connected to the fifth input of the information processing device, the sixth input of which is connected to the information output antenna drive, and the seventh input is connected to the output of the threshold generation unit, the clock input of which is combined with the clock inputs of the digital matched ltrov and connected to the fifth output of information processing apparatus.

 The information processing device (UOI) contains a series-connected comparison unit with a threshold, a primary processing unit and a secondary processing unit, a key and a buffer storage device, as well as a range finder connected in series at two input-outputs, the first input of the range finder being connected to the output of the comparison unit with a threshold , and its first output, the third and fourth inputs are connected respectively to the second input, the first and second outputs of the secondary processing unit, the third input of which is connected to the output of the buffer a device, the control input of which is connected to the control input of the key, the output of the range strobe of the range finder and is the output of the range strobe of the information processing device, the outputs of the secondary processing unit from third to fifth are the second, third and fourth outputs of the information processing device, respectively, and the third output of the range finder, combined with the third input of the primary processing unit, is the fifth clock output of the information processing device, the first and second inputs of which are connected to the information by the key inputs, the third input is connected to the first input of the comparison unit with a threshold, the fourth input is to the inputs of the range start signal of the primary processing unit and the range finder, the second clock input of which is connected to the fifth input of the information processing device, the sixth and seventh inputs of which are formed respectively by the second inputs of the block primary processing and comparison unit with a threshold.

 The digital matched filter (CSF) consists of two identical filters, each of which contains a shift register, random access memory (RAM), a multi-input adder and a two-input adder, the signal inputs of the shift registers connected respectively to the first and second signal inputs of the CSF, the outputs of the shift registers bitwise connected via random access memory codes with the inputs of the corresponding multi-input adders, the outputs of which are connected to the first inputs of the two-input adders c, the outputs of which are the outputs of the CSF, and the second inputs are connected together and connected to the numerical bus, to which the number N / 2 is supplied, where N is the bit depth of the phase-shift code, the control inputs of the RAM codes are connected together and connected to the third input of the CSF, the zeroing signal inputs and clock pulses of the shift registers are also pairwise interconnected and connected to the corresponding 4th and 5th inputs of the CSF.

The invention is illustrated by a further description and drawings, which show:
FIG. 1 structural diagram of the radar;
FIG. 2 block diagram of the CSF;
FIG. 3 block diagram of the OOI;
FIG. 4 block diagram of a block combining quadratures;
FIG. 5 is a structural diagram of a threshold generation unit;
FIG. 6 is a block diagram of an intensity measuring unit;
FIG. 7 structural diagram of the range finder;
FIG. 8 is a block diagram of a frequency tuning unit;
FIG. 9 block diagram of the synchronizer;
FIG. 10 timing diagrams of voltages at the outputs of the synchronizer;
FIG. 11 is a flow chart of a secondary processing unit;
FIG. 12 the dependence of the power of the probe pulses of the radar at the input of the reconnaissance receiver on the distance between them;
FIG. 13 dependence of the steepness of the direction-finding characteristic of the angular discriminator on the signal-to-noise ratio;
FIG. 14 the dependence of the slope of the characteristics of the range discriminator on the signal-to-noise ratio;
FIG. 15 equivalent escort circuit.

In FIG. 1 the following notation is accepted:
1 antenna (A);
2 antenna switch (AP);
3 power regulator (RM);
4 power amplifier (PA);
5 phase manipulator (FM);
6 pathogen (B);
7 pulse modulator (IM);
8 code generator (GK);
9 frequency tuning block (BCH);
10 synchronizer (C);
11 two-channel total-difference receiver (Pr);
12, 13, the first and second blocks of quadrature phase detectors (PD ₁ ), (PD ₂ ), respectively;
14, 15, the first and second video amplifiers (WU ₁ ), (WU ₂ ), respectively;
16, 17 first and second amplitude quantizers (AK ₁ ), (AK ₂ ), respectively;
18, 19, the first and second digital matched filters (DSP ₁ ), (DSP ₂ ), respectively;
20 block combining quadratures (BOC);
21 angle discriminator (DU);
22 information processing device (UOI);
23 block threshold formation (BFP);
24, 25, the first and second switches (K ₁ ), (K ₂ ), respectively;
26 integrator;
27 intensity measurement unit (BII);
28 power gradation decoder (DGM);
29 code-voltage converter (PCN);
30 antenna unit (AB);
31 total difference converter (PSA);
32 antenna drive (PA);
33 high-frequency amplifier (UHF);
34 mixer (CM);
35 intermediate frequency amplifier (IFA).

 In the diagram of FIG. 1, a synchronizer 10, a frequency adjustment unit 9, an exciter 6, a phase manipulator 5 and a power amplifier 4, an antenna 1 and an antenna switch 2 are connected in series, the second output of the antenna 1 being connected to the input of the difference signal of the receiver 11, the input of the total signal of which is connected to the third arm antenna switch 2, and the heterodyne input to the heterodyne output of the pathogen 6, the first two-channel block 12 of the quadrature phase detectors, a video amplifier are connected in series through two inputs-outputs of the quadrature channels 14, an amplitude quantizer 16, a digital matched filter 18, and also a quadrature combining unit 20, the second two-channel block of quadrature phase detectors 13, a video amplifier 15, an amplitude quantizer 17, a digital matched filter 19, and also an angular, are connected in series via two inputs-outputs of the quadrature channels discriminator 21, the third and fourth inputs of the angular discriminator 21 are connected respectively to the first and second outputs of the first digital matched filter 18, the first inputs of the first and second blocks 1 2, 13 quadrature phase detectors are connected respectively to the outputs of the total and difference channels of the receiver 11, the inputs of the reference frequency of blocks 12, 13 of the quadrature phase detectors are combined and connected to the output of the reference frequency of the exciter 6, and the outputs of the unit 20 combining quadratures from the first to the third are connected to the corresponding the inputs of the information processing device 22.

 The output of the angular discriminator 21 through the series-connected first switch 24 and the integrator 26 is connected to the first input of the drive 32 of the antenna 1, and the third output of the block 20 combining quadratures through the series-connected second switch 25, the unit 27 of the intensity measurement, the decoder 28 gradations of power, the Converter 29 code the voltage and power controller 3 are connected to the second arm of the antenna switch 2, and the second input of the power controller 3 is connected to the output of the power amplifier 4, the control input of which is via a pulse The modulator 7 is connected to the second output of the synchronizer 10, which is also connected to the first input of the code generator 8, the output of which is connected to the control input of the phase manipulator 5, and to the third (control) inputs of the first and second digital matched filters 18, 19.

 The fourth inputs (zeroing) of the digital matched filters 18, 19 are interconnected, connected to the second input of the intensity measuring unit 27, the second input of the threshold forming unit 23, the fourth input of the information processing device 22 and connected to the fourth output of the synchronizer 10. The first output (range gate ) information processing device 22 is connected to interconnected control inputs of the first and second switches 24, 25, the second output (target azimuth signal) of information processing device 22 is connected to the second the input of the drive 32 of the antenna 1, the third output of the information processing device 22 is connected to the input of the synchronizer 1 and with the combined control inputs of the first and second video amplifiers 14, 15, the fourth output of the information processing device 22 is connected to the second (control) input of the frequency tuning unit 9 and the fifth output (clock pulses) of the information processing device 22 is connected to the first input of the threshold generation unit 23 and to the fifth matched digital clock inputs (clock pulses) interconnected This filter 18, 19.

 The fifth input of the information processing device 22 is connected to the third output (clock pulses) of the synchronizer 10, to which the second (clock) input of the code generator 8 is connected, the sixth input of the information processing device 22 is connected to the information output of the antenna 32 drive 32, and its seventh input is connected with the output of the block 23 of the formation of thresholds.

 Antenna 1 contains an antenna unit 30, consisting of a mirror with a 4-horn feed, connected by waveguides to the sum-difference converter 31, the first and second outputs of the antenna 1 are formed by the outputs of the sum-difference converter 31 of the same name. The kinematic output of the drive is connected to the control input of the antenna block 30 32 antenna, the information output of which is the third output of antenna 1, and the first and second inputs of the drive 32 of antenna 1 are respectively the same inputs of antenna 1.

The receiver 11 contains connected in series at two inputs / outputs
the sum and difference channels a two-channel high-frequency amplifier 33, a balanced mixer 34, the heterodyne input of which is the third heterodyne input of the receiver, the first and second inputs of the high-frequency amplifier 33 are inputs, and the first and second outputs of the intermediate-frequency amplifier 35 are the outputs of the total and difference channels of the receiver 11 respectively.

In the diagram of FIG. 2 digital matched filter 18 (19) the following notation:
36 shift register of cosine quadrature (CP _c );
37 shift register of sinus quadrature (CP _s );
38 random access memory cosine quadrature codes (RAM _c );
39 random access memory sine quadrature codes (RAM _s );
40 multi-input adder of cosine quadrature (CM _c );
41 multi-input sine-wave adder (CM _s );
42 two-input cosine quadrature adder (CM _c );
43 two-input sine-wave adder (CM _s ).

In the digital matched filter 18 (19) (Fig. 2), the signal inputs of the shift registers 36 and 37 of the quadrature channels are connected respectively to the first and second signal inputs of the CSF, the outputs of CP _c 36, CP _s 37 are bitwise connected via random access memory 38, 39 codes with inputs of the corresponding multi-input adders 40, 41, the outputs of which are connected to the first inputs of the corresponding two-input adders 42, 43, the outputs of which are the outputs of the DSP 18 (19), and the second inputs are interconnected and connected to the N / 2 digital bus, control The RAM inputs _c 38, RAM _s 39 are combined and connected to the third input of the CSF 18 (19), the inputs of the zeroing signal and the clock pulses CP _c 36 and CP _s 37 are also paired together and connected respectively to the 4th and 5th the inputs of the CSF 18 (19).

On the diagram of the information processing device 22 (Fig. 3), the following notation:
44 two-channel key scheme (C);
45 buffer storage device (BZU);
46 block comparison with a threshold (BSP);
47 primary processing unit (BPO);
48 rangefinder (D);
49 secondary processing unit (BWO).

 In the diagram of the information processing device 22 of FIG. 3, the threshold comparison unit 46, the primary processing unit 47 and the secondary processing unit 49 are connected in series, the output 46 being connected via a range finder 48 also to the second input of the BWO 49, the first and second outputs of which are connected respectively to the third and fourth inputs D 48, the second the output of which is connected to the first output of the UOI 22 and to the third control inputs connected in series via the two CL 44 and BZU buses 45, the output of which is connected to the third input of the BWO 49, and the first and second inputs of the Cl 44 are the inputs of the UOI 22 of the same name, the third input which is connected to the signal input of the BSP 46, the fourth input of the UOI 22 is connected to the same input BPO 47 and the fifth input D 48, the fifth, sixth and seventh inputs of the UOI 22 are connected to the second inputs respectively D 48, BPO 47 and BSP 46. The third, fourth and the fifth outputs of the BWO 49 are connected respectively to the second, third and fourth outputs of the UOI 22, and the third output D 48 is connected to the same input of the BPO 47 and the fifth output of the UOI 22.

In the diagram of FIG. 4 blocks 20 combining quadratures adopted the following notation:
50 squarer cosine quadrature (q _c);
51 quadrature sine quadrature (kV _s );
52 quadrature adder ((Σ)).

In FIG. 4 BOK 20, the first and second inputs of the BOK are connected respectively with the outputs of the BOK 20 and the inputs Kv _c 50 and Kv _s 51, the outputs of which are connected to the inputs of the two-input adder ((Σ)) 52, the output of the latter is connected to the third output of the BOK 20.

In the diagram of FIG. 5 block 23 of the formation of thresholds (BFP) adopted the following notation:
53 counter (Mid);
54 element of read-only memory (EPP);
55 decoder (L);
56 trigger (Tg).

 In the circuit of the block 23 of the formation of thresholds BFP in FIG. 5 counter counter input (SCH) 53 is connected to the first input of the block, zero input SCh 53 to the second input of the block and the first input of the trigger (Tg) 56. The outputs of the counter 53 are connected bitwise to the inputs of the decoder (L) 55 and the inputs of the least significant bits of the address element 54 of permanent memory (EPP), the output Дш 55 is connected to the second input Tg 56, the output of the latter is connected to the input of the senior bit EPP 54, the output of which is the output of the BFP 20.

In the diagram of FIG. 6 block 27 intensity measurement (BII) adopted the following notation:
57 two-input adder (DS);
58 first register (Rg ₁ );
59 second register (Rg ₂ );
60 frequency divider (DF);
61 first delay element (EZ ₁ );
62 second delay element (EZ ₂ );
63 multi-input element "OR" (OR).

In a block 27 for measuring the intensity of the BII in FIG. 6, the input of the frequency divider 60 (DC) is connected to the second input of the unit, and the output with the input of the read signal directly and the input of the setup to zero register (Рг ₁ ) 58 through the delay element 61 (ЭЗ ₁ ), the first input of the two-input adder (ДС) 57 is connected with the first input of the BII 27, and bitwise with the inputs of the OR 63 element, the output of which is connected to the input of the recording Pr ₁ 58 through the delay element 62 (EZ ₂ ). The input Rg ₁ 58 is connected to the output of the DS 57, and the output is connected to the second input of the DS 57 and the input of the register (WG ₂ ) 59, the output of which is connected to the output of the BII 27.

In the diagram of the range finder 48 (D) of FIG. 7 the following notation is accepted:
64 range discriminator (DD);
65 switch (Km);
66 reverse counter (PC);
67 code-time interval converter (PCVI).

 In FIG. 7, the first input of the range finder (D) 48 is the information input of the range discriminator 64 (DD), the output of which is connected to the first input of the switch (Km) 65, the second input of the switch 65 is connected to the third input of the range finder 48, and the output with the input of the reversible counter (PC) 66. The second input of the reverse counter 66 is connected to the fourth input of the range finder 48, the output of the reverse counter 66 is connected to the input of the converter 67 code-time interval (PCVI) and connected to the output 1 of the range finder 48, the second input of the converter 67 is the second input of the range finder 48, and the third input of the converter is the fifth input of the range finder. The first output of the converter 67 is connected to the output 2 of the range finder 48 and to the second input of the range discriminator 64, the second output of the converter 67 with the output 3 of the range finder 48.

In the circuitry of the frequency tuning unit 9 (BFC) of FIG. 8 the following notation is accepted:
68 noise generator (GS);
69 limiting amplifier (UO);
70 switch (Km);
71 counter (Mid);
72 element "And"(And);
73 register (P);
74 decoder (L).

 In FIG. 8, the control input of the And element 72 is the first input of the frequency tuning unit (BFC) 9, the signal inputs of the And element are connected to the corresponding outputs of the counter (MF) 71, to the input of which a noise generator 68 is connected in series through the switch (Km) 70 (GS) and amplifier-limiter (UO) 69, the control input of the switch 70 is connected to the second input of the frequency tuning unit 9, and the outputs of the element "I" 72 are bitwise connected to the register (P) 73, the outputs of which are connected to the decoder (L) 74 , the outputs of the latter form the output of block 9 p frequency tuning.

In the synchronizer circuit 10 of FIG. 9 the following notation is accepted:
75 master oscillator (ZG);
76 frequency divider (DF);
77 counter (Mid);
78 decoder (Dsh);
79 block of triggers (BTR).

 In FIG. 9, the input of the synchronizer (C) 10 is the control input of the frequency divider 76 (DF), to the signal input of which the output of the master oscillator (ZG) 75 is connected, and the output of the frequency divider 76 is connected to the counter input (MF) 77 and to the output 3 of the synchronizer 10. The outputs of the counter 77 through the decoder (L) 78 are connected to the inputs of the block 79 RS-flip-flops (BTR), the outputs of which are connected to the outputs 1, 2, 4 of the synchronizer 10.

с длительностью τ₁ ≪ τ_и на выходе 1 синхронизатора, с упреждением на время перестройки τ_f относительно начала периода, предназначенные для управления блоком 9 перестройки частоты;
81 импульсы с частотой

с длительностью τ₂ = T_и на выходе 2 синхронизатора, предназначенные для управления генератором 8 кода и импульсным модулятором 7;
82 импульсы с частотой

и длительностью τ₃ ≪ τ_и на выходе 3 синхронизатора, предназначенные для управления генератором 8 кода и устройством 22 обработки информации (дальномером 48);
83 импульсы с частотой F_п с длительностью τ₄ ≪ τ_и на выходе 4 синхронизатора с задержкой на T_и относительно начала периода повторения, предназначенные для управления цифровыми согласованными фильтрами 18, 19, блоком 23 формирования порогов и устройством 22 обработки информации (блоком 47 первичной обработки и дальномером 48).In FIG. 10 shows the timing diagram of the voltages at the outputs of the synchronizer 10, namely:
80 pulses, with frequency

with a duration of τ ₁ ≪ τ _and at the output of 1 synchronizer, anticipated by the tuning time τ _f relative to the beginning of the period, intended for controlling the frequency tuning block 9;
81 pulses with a frequency

with a duration of τ ₂ = T _and at the output 2 of the synchronizer, designed to control the code generator 8 and the pulse modulator 7;
82 pulses with a frequency

and duration τ ₃ ≪ τ _and at the output 3 of the synchronizer, designed to control the code generator 8 and the information processing device 22 (range finder 48);
83 pulses with a frequency of F _p with a duration of τ ₄ ≪ τ _and at the output of 4 synchronizers with a delay of T _and relative to the beginning of the repetition period, designed to control digital matched filters 18, 19, threshold generation unit 23 and information processing device 22 (primary 47 processing and range finder 48).

 Antenna switch 2 can be made, for example, in the form of a three-arm ferrite Y-circulator.

при требуемой высокой кратковременной стабильности частот f_г и f_с.Power controller 3 can be implemented, for example, according to the scheme given in [3]
Power amplifier (UM) 4 microwave amplifier with pulse modulation, implemented, depending on the required power and the band of amplified frequencies, based on an electrovacuum device (amplitron, traveling wave lamp, multipath klystron, etc.) or a semiconductor device [5]
The phase manipulator (FM) 5 can be performed, for example, according to the scheme [2], and a segment of a strip waveguide switched by microwave diodes that are controlled by pulses from a code generator 8 can be used as microwave delay lines. The causative agent (B) 6 can be performed, for example, according to the scheme shown in [2] as a part of the carrier frequency generator f _c , the frequency of which can be step-wise tuned from period to period under the action of a control signal from the frequency tuning unit 9, the reference frequency generator, equal to the intermediate frequency of the receiver f _op f _pch , a local oscillator generating oscillations with a frequency f _g , tunable synchronously with a frequency f _s , and a phase locked loop, ensuring equality

with the required high short-term stability of the frequencies f _g and f _s .

 The pulse modulator (IM) 7, depending on the parameters of the AM, is implemented according to well-known schemes (see [5, p. 103-107, Fig. 43-45]).

 The code generator (GK) 8 can be implemented according to a scheme containing a serially-activated clock valve and a shift register with feedback through modulo 2 adders to form a pseudo-random M-sequence (see, for example, [5, p. 421, Fig. 17] or [6, pp. 147-153]).

где

комплексные сигналы на выходе суммарного и разностного каналов соответственно (после ЦСФ), а символ * означает "комплексно-сопряженный".The angle discriminator 21, in meaning, performs the operation of generating the angular mismatch signal by forming the scalar product of the total and difference signals after they are compressed in the corresponding CSF ₁ , CSF ₂ . To this end, it contains two multipliers and an adder block combining quadratures in accordance with the rule

Where

complex signals at the output of the sum and difference channels, respectively (after the CSF), and the symbol * means "complex conjugate."

This rule, as is known, differs from the optimum only in the absence of normalization [7]
Sum-difference Converter (SRP) 31 in the antenna (A) 1 can be performed, for example, based on waveguide T-bridges.

 Amplifier 33 high frequency (UHF) in the receiver (Pr) 11 can be implemented in the form of a transistor two-channel microwave amplifier.

 The mixer (Cm) 34 in the receiver 11 is in the form of a two-channel balanced mixer (see, for example, [5, p. 144]).

An intermediate frequency amplifier (IFA) 35 in the receiver 11, for the purpose of identifying the amplitude-phase characteristics of the sum and difference receiving channels, is performed by a combined two-channel, for example, based on the frequency separation of channels according to the scheme given in [8]
Block 47 primary testing (BPO) in the device 22 information processing (UOI) can be performed according to the scheme shown in the monograph [9, p. 255, fig. 3.17] Block 49 of the secondary information processing (BWO), included in UOI 22, can be implemented on the basis of a digital computer (digital computer), the work of which consists in sequentially solving the problems presented in the block diagram (Fig. 11):
1. After switching on the review mode, an inter-review information processing is performed, consisting of the accumulation of discrete signals corresponding to the decisions made according to the results of each of the reviews in the BPO after their identification by coordinate signs taking into account the movement of the radar carrier, and the final decision on the presence or absence of signals from goals, for example, by the criterion "K from M" (at least K decisions about the presence of a signal from the target in a series of M reviews).

Next, the number of detected targets I is calculated and, if it is not less than a predetermined threshold number I ^* depending on the type of detected target (in particular, for a single target I ^* 1), the BVI proceeds to the next program item, otherwise, t. e. when I <I ^{*, the} review is turned off for a given time, then it is turned back on and item 1 is executed.

2. When the condition I> I ^{* is} fulfilled, tracking along the range and angular coordinates is carried out alternately for each of the detected targets i 1, 2 I, for which a command is issued from output 1 of BVO 49 to input 3 D 48 to close the rangefinder circuit and the values of R _i with output 2 of BVI 49 to input 4 D 48 (initial tracking range), as well as values of j _i (initial angular coordinate) from output 3 of BVI 49 through output 2 of UOI 22 to input 2 A1, and from output 2 D 48 of output 1 of UOI 22 gating signals are supplied to the control inputs K ₁ 24 and K ₂ 25, closing the tracking contour along the angular th coordinate and intensity.

 3. At the input 2 of the BPC 9 from the output 5 of the BWO 49 through the output 4 of the UOI 22, a command is received to stop the random tuning of the transmitter carrier frequency.

 4. The accompanied i-th target is classified according to the spectral-correlation characteristics of inter-period fluctuations of the received signals, for which a spectral analysis of the complex envelope of the signal packet is performed, for example, by the fast Fourier transform (FFT) method, measurement of the width of the obtained spectrum, for example, by 3 dB ", comparing it with a threshold value and deciding on the target class (true or false) depending on the results of this comparison (in particular, if the spectrum width is less than the threshold value true goal, otherwise false). After the classification of the i-th goal 2-4 are repeated for i i + 1 and so on to i I, while false targets are excluded from further consideration.

где L ≅ I число истинных целей.5. The choice of target for tracking, for example, by determining the coordinates of the target R _C , ψ _C , which is closest to the point with the coordinates

where L ≅ I is the number of true goals.

6. The output of the coordinate R _{c of the} selected target from the output 2 of the BVI 49 to the input 4 D 48 and the coordinates ψ _{c of} the output 3 of the BVI 49 through the output 2 of the UOI 22 to the input 2 A1 simultaneously with the command to close the tracking loop in range from the output of 1 BVO 49 at the entrance 3 D 48.

где T_и1- длительность сигнала вида 1) с выдачей команды с выхода 4 БВО 49 через выход 3 УОИ 22 на вход синхронизатора 10, по которой происходит переключение С 10 на режим с повышенной тактовой частотой

и на управляющие входы видеоусилителей (ВУ) 14, 15 для изменения полосы видеоусилителей с целью согласования ее с дискретом τ_и фазовой манипуляции.7. Comparison of the current range R _c (t) coming from the output 1 D 48 to the input 2 of the BVO 49 and switching to signal type 2 when the condition R _c (t) ≅ 0.5R _о , where R _о is the predetermined range ( eg,

where T _{and 1} is the duration of a signal of the form 1) with the issuance of a command from the output 4 of the BWO 49 through the output 3 of the UOI 22 to the input of the synchronizer 10, by which the C 10 switches to the mode with increased clock

and to the control inputs of video amplifiers (VU) 14, 15 to change the band of video amplifiers in order to match it with the discrete τ _and phase manipulation.

 In accordance with the considered structural diagram, the proposed radar operates as follows.

Перестройка частот f_сi, и f_гi, производится с помощью блока 9 перестройки частоты (БПЧ), работающего следующим образом (фиг. 8).The exciter 6 of the transmitter generates continuous oscillations of the signal frequency f _si , local oscillator f _gi , and reference oscillations of the intermediate frequency f _pch , while the frequencies f _ci , f _{gi are} highly stable during one repetition period T _{p of the} probe pulses, but can change stepwise from period to the period according to a random law under the action of signals from the frequency adjustment unit 9, taking one of n _f values (i 1, 2, 3, n _f ), and so that the relation always holds

Restructuring frequency f _{c i} and f _{plaster Gi,} is performed using frequency adjustment unit 9 (BORO) operating as follows (FIG. 8).

The noise voltage generator (GS) 68, built, for example, on the basis of a noise diode, generates noise fluctuations with a spectral width significantly exceeding the repetition frequency (Δf _w ≫ F _p ), then these oscillations are amplified and limited in the limiting amplifier (UO) 69 and they arrive through the switch (Km) 70 to the counter (MF) 71, which counts, for example, positive edges modulo n _f and thus has n _f equiprobable states. At the moment determined by the clock pulses arriving through the repetition period T _p from the output of 1 synchronizer 10 with a lead time of τ _f relative to the beginning of the next period, the readings of the counter 71 through the element "And" 72 are recorded in the register (P) 73 and converted into a decoder (Дш ) 74 into a parallel code with the number of bits n _f , which is stored during one repetition period and determines the values of the frequencies f _сi , and f _Гi in the next repetition period.

Fluctuations in the frequency of the signal f _сi enter the phase manipulator (FM) 5, where they are phase-shifted to levels 0, π, by a binary multi-bit code (number of bits N) generated in the code generator 8 (GC), amplified by power in the power amplifier 4 ( UM), which generates, under the action of a pulse modulator (IM) 7 probing pulses with a duration T _and with intrapulse phase shift keying (FM) an N-bit binary code. The probe pulses generated in the transmitter, passing the power regulator 3 (PM), which has a minimum attenuation in the viewing mode (switch 25 is open), and the antenna switch (AP) 2, enter the antenna 1 (via the sum-difference converter (SRP) 31 into the antenna unit (AB) 30) and are emitted into space. Moreover, in the detection mode, the antenna drive (PA) 32 operates autonomously, performing a survey of the space (switch 24 is open).

The received signals go to the inputs 1, 2 of the receiver (Pr) 11, where, passing through a two-channel amplifier 33 high-frequency (UHF), they are fed to a two-channel mixer (SM) 34, which carries out oscillations of the local oscillation frequency f _gi in each receiving channel from the transmitter exciter 8, conversion to an intermediate frequency is amplified in a two-channel intermediate frequency amplifier (IFA) 35 and fed to the inputs of the blocks of quadrature phase detectors (PD _1.2 ) 12, 13 of the total and difference channels, respectively. The second inputs of the quadrature phase detectors 12, 13 are supplied as reference oscillations of the frequency f _pc , which are generated in the exciter 8 of the transmitter, and they go directly to the inputs of the reference oscillations of the first channels of the two-channel phase detectors (for example, cosines of the sum and difference channels, respectively) , and at the inputs of the reference oscillations of the second channels of the quadrature two-channel phase detectors, they come through phase shifters that rotate the phase of the reference oscillations by 90 ^o (sine quadratures of the total and difference channels, respectively).

At the outputs of blocks 12 and 13 of quadrature phase detectors, the quadrature components of bipolar video signals are formed, the polarity of which changes in accordance with the FM code of the received signals, these signals are transmitted through amplitude quantizers (AK ₁ , AK ₂ ) 16, 17 to digital matched filters (CSF ₁ , CSF ₂ ) 18, 19, in which a coordinated filtering (compression) of signals is carried out.

 Digital matched filters (see Fig. 2) work as follows.

, поступающими через входы 5 ЦСФ на соответствующие входы сдвиговых регистров 36 (37). Сигналы со всех N разрядов сдвиговых регистров попадают в многовходовые сумматоры СМ_c 40 (СМ_s 41) через оперативные запоминающие устройства кодов (ОЗУК_с, ОЗУК_s) 38 (39), в которые перед началом приема в данном периоде повторения записываются кодовые символы V_j, (0 или 1), поступающие из генератора 8 кодов через вход 3 ЦСФ 18 (19).The inputs 1, 2 of the identical quadrature channels of the CSF 18 (19) receive from the outputs of the amplitude quantizers 16 (17) binary quantized signals x $\genfrac{}{}{0ex}{}{\text{(c, s)}}{\text{i}}$ which can take values 1 or 0. They pass to the signal inputs of the shift registers CP _c 36 (CP _s 37), filling them at a pace determined by clock pulses with a frequency

coming through the inputs 5 CSF to the corresponding inputs of the shift registers 36 (37). Signals from all N bits of the shift registers fall into the multi-input adders СМ _c 40 (СМ _s 41) through random access memory codes (RAM, _s , RAM _s ) 38 (39), into which code symbols V _{j are} written before reception in this repetition period , (0 or 1), 8 codes coming from the generator through input 3 of CSF 18 (19).

то есть осуществляется согласованная фильтрация двоично-квантованных ФМ видеосигналов. Сжатые сигналы с выходов цифрового согласованного фильтра суммарного канала (ЦСФ₁) 18, пропорциональные (при малых отношениях сигнал/шум до сжатия a) величинам N•a•g_ΣcosΦ, N•a•g_ΣsinΦ (Φ - случайная начальная фаза, g_Σ усиление антенны суммарного канала), поступают на блок 20 объединения квадратур, в котором эти сигналы объединяются по правилу "сумма квадратов", то есть

в результате сигнал на выходе 3 блока 20 объединения квадратур (см. фиг. 4) не зависит от начальной фазы Φ. Этот сигнал Y_i, а также сигналы y $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{i}}$ , y $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{i}}$ поступают в устройство 22 обработки информации (УОИ), которое работает следующим образом.In blocks 38 (39), bitwise summation of signals with code symbols modulo 2 occurs, and in blocks 40 (41), all N bits are added together. These total signals are then fed to the first inputs of the two-input adders SD _c , SD _s 42 (43), the combined second inputs of which receive the number N / 2. As a result, at the outputs 1, 2 of the CSF 18 (19), the quadrature components of the compressed signals of the total (difference) channels appear in accordance with the expression

that is, matched filtering of binary-quantized FM video signals is performed. The compressed signals from the outputs of the digital matched filter of the total channel (CSF ₁ ) 18 are proportional (for small signal-to-noise ratios to compression a) to the values of N • a • g _Σ cosΦ, N • a • g _Σ sinΦ (Φ is the random initial phase, g _Σ antenna gain of the total channel), are fed to the quadrature combining unit 20, in which these signals are combined according to the “sum of squares” rule, that is

as a result, the signal at the output 3 of the quadrature combining unit 20 (see Fig. 4) is independent of the initial phase Φ. This signal Y _i , as well as signals y $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{i}}$ , y $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{i}}$ enter the device 22 information processing (UOI), which operates as follows.

The signal Y _i from the output 3 of the quadrature combining unit 20 is fed through the input 3 (Σkv) of the UOI to the input 1 of the comparison unit 46 with a threshold (BSP), and if there is an excess of the threshold level input to the input 2 of the comparison unit 46 with a threshold through the input 7 of the UOI 22 from the output of the threshold generation unit 23, a normalized signal is generated.

 The normalized signals are fed to the inputs 1 of the primary processing unit 47 (BPO) and the range finder (D) 48. In the primary processing unit 47, the signals are periodically accumulated, packets are detected by the criterion "k of n" and the coordinates of the detected targets (range and azimuth) are measured in review process. The coordinates of the detected targets for each review cycle are fed to input 1 of the secondary processing unit 49, where inter-review processing is performed, as described above.

 In the selection mode, the radar performs tracking in range and in angular coordinates alternately for each of the detected targets, as described above, while the tuning of the carrier frequency of the probing signals stops and the work is done at a fixed frequency according to the command received from output 5 of BVO 49 through output 4 of UOI 22 to input 2 of the BFCH 9. At the same time, the gate pulse from output 2 of D 48 is fed to a two-channel key circuit (C) 44, to the signal inputs of which are the quadrature components of the compressed signals from outputs 1, 2 of BOK 20, By strobing compressed signals from a selectable target. These signals are recorded in a buffer storage device (BZU) 45, from where they arrive in the BWO 49 at a convenient pace for it, in the BWO 49 a spectral analysis of the envelopes of inter-period fluctuations of the signals and selection of false targets along the width of the spectrum, as well as target selection, as described above, are performed and the transition to tracking the chosen goal.

 Let us consider in more detail the operation of the radar in tracking mode.

In the tracking mode, according to the signal from the third output of the BWO 49 (2nd output of the UOI 22) to the second input of the drive 32 of the antenna 1, the latter is transferred from the offline mode to the tracking mode and the azimuth value of the selected target is fed to it, simultaneously by the command received from the first output of the BWO 49 to the third input of the range finder 48, the range tracking circuit is closed via the switch (Km) 65 (Fig. 7), at the same time, the range code of the selected target R _c comes from the second output of the BWO 49 through the fourth input of the range finder 48 to the reverse counter (RS) 66 and from it to the converter 67 the code is the time interval that forms the range gate. This gate through the second output of the range finder and the first output of the UOI passes to the control inputs of the switches 24 and 25, closing the loop of the angular tracking and adjusting the power of the probing signals, and to the second input of the range discriminator (DD) 64, at the output of which a mismatch signal is received through the switch 65 to the reverse counter 66, the output of the latter generates a code of the current range, it, in turn, passes the code-time interval to the first input of the converter 67, shifting the position of the strobe at the second output of the range omer 48 and the second input of the range discriminator, etc.

-интервал квантования по времени, равный длительности одного разряда кода ФМ, то есть

поступает с третьего выхода синхронизатора 10 через вход 5 УОИ 22 и вход 2 дальномера 48 на младшие разряды преобразователя 67 код временной интервал (вход 2 ПКВИ 67) и, сдвигаясь в соответствии с поправкой дальности, образующейся в дискриминаторе 64 дальности и реверсивном счетчике 66, на величины, измеряемые долями τ_и, поступает затем через выход 3 дальномера 48 на вход 3 БПО 47, а также через выход 5 УОИ 22 на входы 5 цифровых согласованных фильтров 18, 19, это обеспечивает перемещение сжатых сигналов по оси времени с малой дискретностью

(например, m 4-16), то есть высокую точность сопровождения цели по дальности.The sequence of clock pulses with a frequency

-time quantization interval equal to the duration of one bit of the FM code, i.e.

comes from the third output of the synchronizer 10 through the input 5 of the UOI 22 and the input 2 of the range finder 48 to the lower bits of the converter 67 the time interval code (input 2 PKVI 67) and, shifting in accordance with the range correction generated in the range discriminator 64 and the counter 66, to the quantities measured by the fractions τ _and then passes through the output 3 of the range finder 48 to the input 3 of the BPO 47, and also through the output 5 of the UOI 22 to the inputs 5 of the digital matched filters 18, 19, this ensures the movement of the compressed signals along the time axis with low resolution

(for example, m 4-16), that is, high accuracy tracking the target in range.

 In the discriminator 21 of the angle by digitally multiplying the compressed signals from the outputs of the same quadrature CSF 18, 19 of the total and difference channels and summing, a bipolar signal of angular mismatch is formed, which, in the simplest case, can take one of three values: +1, 0, -1 in depending on the position of the direction to the target relative to the equal-signal direction. The output signal of the angle discriminator 21, which is, for example, a two-bit code, is fed through the switch 24 to the integrator 26 of the angular error signals (DUT). After accumulation in the integrator 26, a control signal is generated, which is fed through the input 1 of the antenna to the input 1 of the drive 32 of the antenna, which controls the position of the movable mirror of the antenna unit 30 and provides tracking along the angular coordinate.

 The compressed signal from the tracking target from the output 3 of the quadrature combining unit 20 passes through the switch 25 in the tracking mode to the information input of the intensity measurement unit 27 (BII). The intensity measurement in block 27 (see Fig. 6) is as follows.

 The accumulation interval is determined by dividing by the divider 60 the repetition rate of the pulse signal supplied to the input 2 of the block 27 from the output 4 of the synchronizer 10. The signal from the output of the frequency divider 60, which is a binary counter, overwrites the contents of register 58 to the output register 59, and then sets it to zero register 58 through delay element 61. The accumulated signal from input 1 of block 27 is fed to the first input of the accumulating adder, formed by a two-input adder (DS) 57 and register 58. The write signal to register 58 is formed by combining by OR and delaying all bits of the accumulated signal so that if there is a unit at least in one bit of the input signal causes recording through the OR element 63 and the delay element 62. In this case, a result equal to the sum of the previously accumulated value and the next value of the input signal is formed in the accumulating adder. Once during the accumulation interval, the accumulated result is transferred to the register 59 and is output to the intensity measuring unit 27, and then the register 58 is reset.

 The code of the intensity of the compressed signals of the target being sent to the input of the decoder 28 gradations of power, which is a parallel connection of the decoders (the number of bits of the code intensity). At the output of the decoder 28 power gradations, a parallel code of the power of the probing signals corresponding to the value of the intensity of the received signals is generated. This code is supplied to the code-voltage converter (PCN) 29, which generates a corresponding current value that controls the attenuation of the power controller 3 (PM). The power controller 3, changing the microwave attenuation, sets the necessary power of the emitted signals, corresponding to the required power of the echo signals. This, in turn, leads to regulation of the power of the echo signals from the target being followed, stabilizing it, and, therefore, the quality factor of tracking contours in range and in angular coordinates at a given level.

 The stabilization of the quality factor of the circuits leads to an increase in the accuracy of tracking. In addition, adjusting the radiation power at a given level leads to an increase in stealth, and therefore to an increase in the radar noise immunity with respect to the means of response interference.

(T_и длительность зондирующих импульсов, c скорость света), часть отраженного сигнала, приходящая на вход приемного устройства с задержкой в интервале от 0 до T_и ("мертвая зона") относительно начала периода повторения, не будет приниматься вследствие бланкирования импульсом передатчика, длительность же принимаемой части сигнала при этом составляет величину

то есть уменьшается пропорционально уменьшению дальности до сопровождаемой цели. Для того чтобы в этих условиях осуществить согласованную фильтрацию, необходимо в соответствии с уменьшением длительности принимаемых сигналов изменять и число разрядов регистров 36, 37 сдвига (фиг. 2), участвующих в образовании сжатого сигнала. С этой целью в начале очередного периода повторения на входы 4 цифровых согласованных фильтров 18, 19 поступает сигнал обнуления от синхронизатора 10, который проходит на установочные шины сдвигающих регистров 36, 37 и устанавливает их в нулевое положение. Сигнал от сопровождаемой цели на дальности R <R₀, поступающий на входы 1, 2 ЦСФ 18 (19) сразу после окончания зондирующего импульса передатчика, занимает

разрядов регистров 36, 37, остальные же разряды не участвуют в образовании выходного сигнала. В соответствии с изложенным дисперсия шумов на выходе ЦСФ в этом режиме при R <R₀ также уменьшается пропорционально уменьшению дальности до цели, поэтому для повышения вероятности правильного обнаружения при фиксированной вероятности ложного обнаружения порог первичного обнаружения на выходе 3 блока 20 объединения квадратур в блоке 46 сравнения с порогом, входящем в устройство 22 обработки информации снижается с уменьшением дальности до цепи. Это достигается с помощью блока 23 формирования порогов. На первый вход этого блока поступает последовательность тактирующих импульсов с частотой f_T от дальномера 48 (с выхода 3 Д 48 через выход 5 УОИ 22), сдвигающаяся по оси времени в соответствии с изменением дальности до цели ("подвижная сетка"), необходимая для формирования порога, меняющегося в функции от текущей дальности до сопровождаемой цели. Подвижная сетка, как уже упоминалось выше, поступает одновременно также на входы 5 цифровых согласованных фильтров 18, 19 (на сдвигающие шины регистров 36, 37), обеспечивая перемещение сжатого сигнала по оси времени с малой дискретностью.With a decrease in the range to the followed target R to values smaller than

(T _{and the} duration of the probe pulses, c is the speed of light), the part of the reflected signal arriving at the input of the receiving device with a delay in the range from 0 to T _and (“dead zone”) relative to the beginning of the repetition period will not be accepted due to blanking by the transmitter pulse, the duration the received part of the signal in this case is

that is, decreases in proportion to the decrease in range to the target being followed. In order to carry out consistent filtering under these conditions, it is necessary, in accordance with the reduction in the duration of the received signals, to change the number of bits of the shift registers 36, 37 (Fig. 2) involved in the formation of the compressed signal. To this end, at the beginning of the next repetition period, a zeroing signal from the synchronizer 10 is received at the inputs 4 of the digital matched filters 18, 19, which passes to the installation buses of the shift registers 36, 37 and sets them to the zero position. The signal from the tracking target at a range of R <R ₀ , arriving at the inputs 1, 2 of the CSF 18 (19) immediately after the end of the probe pulse of the transmitter, takes

bits of the registers 36, 37, the remaining bits do not participate in the formation of the output signal. In accordance with the above, the noise variance at the output of the CSF in this mode at R <R ₀ also decreases in proportion to the decrease in the range to the target, therefore, to increase the probability of correct detection with a fixed probability of false detection, the threshold of primary detection at the output 3 of block 20 combining quadrature in block 46 comparison with a threshold included in the information processing device 22 decreases with decreasing range to the circuit. This is achieved using the block 23 of the formation of thresholds. The first input of this unit receives a sequence of clock pulses with a frequency f _T from the range finder 48 (from the output 3 D 48 through the output 5 of the UOI 22), shifted along the time axis in accordance with the change in the distance to the target ("moving grid"), necessary for the formation threshold changing in function from the current range to the target being followed. The moving grid, as already mentioned above, simultaneously arrives at the inputs of 5 digital matched filters 18, 19 (on the shift buses of the registers 36, 37), ensuring the movement of the compressed signal along the time axis with low resolution.

(информация о дальности в режиме сопровождения передается с первого выхода дальномера 48 (фиг. 3) на второй вход блока 49 вторичной обработки), длительность излучаемых импульсов снижается, например, в 2 раза путем уменьшения соответственно в 2 раза длительности дискретов и при сохранении разрядности N кодов ФМ.At the moment when the range to the target is reduced to a range

(information about the range in tracking mode is transmitted from the first output of the range finder 48 (Fig. 3) to the second input of the secondary processing unit 49), the duration of the emitted pulses is reduced, for example, by 2 times by decreasing, respectively, 2 times the duration of the samples and while maintaining the bit depth N FM codes.

например, в 2 раза с одновременным уменьшением длительности зондирующих импульсов T_и = Nτ_и и расширением (в рассматриваемом примере в 2 раза) полосы пропускания видеоусилителей 14, 15, определяющей общую полосу пропускания приемных каналов (суммарного и разностного), выбираемую из условия

Таким образом, достигается при сохранении энергии принимаемых сигналов (с учетом бланкирования), во-первых, снижение уровня боковых лепестков, который при дальнейшем снижении разрядности принимаемой части эхо-сигналов оказался бы чрезмерно большим, во-вторых, повышение разрешающей способности по дальности, что позволяет повысить точность сопровождения по дальности и помехозащищенность по отношению к пассивным помехам.This is achieved by the command of the BWO 49 coming from the output 4 of the BWO through the output 3 of the UOI 22 to the input of the synchronizer 10 and to the control inputs of the video amplifiers 14, 15 by increasing the frequency of the clock pulses

for example, by 2 times with a simultaneous decrease in the duration of the probe pulses T _and = Nτ _and and by expanding (in the considered example by 2 times) the passband of video amplifiers 14, 15, which determines the total passband of the receiving channels (total and differential), chosen from the condition

Thus, while maintaining the energy of the received signals (taking into account blanking), firstly, a decrease in the level of the side lobes, which with a further decrease in the bit depth of the received part of the echo signals, would be excessively large, and secondly, an increase in the range resolution, which allows you to increase the accuracy of tracking in range and noise immunity in relation to passive interference.

 The technical advantage of the proposed radar over the prototype device is that it allows you to increase stealth and noise immunity while improving the tracking accuracy in range and angular coordinates.

 The increase in stealth and noise immunity with respect to response interference in the detection and tracking modes is determined by the possibility of a significant increase in the duration of complex FM signals compared to the prototype while maintaining their energy required to achieve a given target detection range due to the use of CSF at the video frequency.

The duration of the probe pulses T _and in the proposed radar can be increased by at least an order of magnitude compared with the prototype, that is, from 10-20 μs (in the prototype) to 200-500 μs. While maintaining the pulse energy equal to
E _and = T _and P _and
and determined by a given detection range, this means the possibility of a corresponding decrease (10-50 times) in the pulse power P _and, consequently, stealth and noise immunity with respect to response interference induced by executive intelligence (IR) receivers.

 The increase of the noise immunity of the claimed radar in comparison with the prototype at ranges shorter than the limit, including when accompanied, is determined by the following.

где R_max дальность начала наблюдения, в частности, сопровождения цели.The power of the radar probe pulses at the input of the IR receiver of the jammer at a distance R from the radar in the absence of transmitter power control (in the prototype) is determined from the ratio

where R _{max the} range of the beginning of observation, in particular, tracking the target.

(2)
Благодаря применению регулирования имеем выигрыш в скрытности в заявляемой РЛС по сравнению с прототипом в

раз (при R > R₀).In the presence of regulation (in the claimed radar) we get

(2)
Thanks to the application of regulation, we have a gain in stealth in the claimed radar compared to the prototype in

times (for R> R ₀ ).

In FIG. 14, dependences (1) and (2) are represented by curves 1 and 2, respectively. As you can see, with RR ₀ 0.25R _max gain in stealth is 24 dB. Let us show by example that an increase in stealth also leads to an increase in noise immunity with respect to response interference.

где P_пер мощность передатчика РЛС,
G, G_n усиление антенны РЛС и станции помех соответственно,
λ длина волны.Conditions response creating interference is exceeded, the power _{P, etc.} on its IR receiver input sensitivity threshold P _pr.min. . We have the expression

where P _lane radar transmitter power,
G, G _{n the} gain of the radar antenna and the jamming station, respectively,
λ wavelength.

3•10^-8 Вт, получим из (3), что при отсутствии регулирования ответные помехи будут действовать на дальностях R < R^* (при этом кривая 1 выше прямой 3 P_пр P_пр.min фиг. 11), а при наличии регулирования этого нигде не произойдет (кривая 2 всегда ниже прямой 3 фиг. 12).Assuming the values G = 500, G _p = 10, P _per = 1 kW, R _max = 40 km, λ = 3 • 10 ^-2 m, P _{per min} = 10 ^-7.5 W

3 • 10 ^-8 W, from (3) that in the absence of reciprocal interference regulation will operate at ranges R <R ^* (wherein curve 1 above _straight line P 3 P _pr.min FIG. 11), and in the presence of regulatory this will not happen anywhere (curve 2 is always below line 3 of Fig. 12).

приводит к эквивалентному повышению отношения сигнал/шум в

раз по сравнению с прототипом (благодаря уменьшению дисперсии шумов, действующих при сжатии), в частности, при

это повышение составляет в 2 раза (по мощности), то есть 3 дБ. Этот выигрыш означает соответствующий дополнительный выигрыш в скрытности и помехозащищенности по отношению к ответным помехам.Zeroing the shift registers of the CSF at the beginning of the repetition period simultaneously with the regulation of the primary detection threshold at ranges

results in an equivalent increase in signal to noise ratio in

times compared with the prototype (due to a decrease in the dispersion of noise acting during compression), in particular, when

this increase is 2 times (power), that is, 3 dB. This gain means a corresponding additional gain in stealth and noise immunity with respect to response interference.

вероятность селекции эхо-сигналов от помехи и от истинной цели при разрешающей способности по дальности ΔR ограничена величиной

.The increase in range resolution at ranges R <0.5R ₀ , in particular, by 2 times, contributes to increased noise immunity with respect to passive means of creating interference (dipole clouds, corner reflectors). So, if the interference is placed at a distance l from the protected target at an angle to the direction of propagation

the probability of selecting echoes from interference and from the true target with a range resolution ΔR is limited to

.

 In particular, at l = 200 m, ΔR = 150 we have p = 0.46, and at ΔR = 75 m p = 0.75, thus, in the considered example, the probability of selection increases by 1.6 times.

 The stabilization of the quality factor of automatic tracking loops in angular coordinates and in range due to maintaining the power of the received signals at a constant level by adjusting the power of the probe pulses leads to an increase in tracking accuracy.

где ρ_э эквивалентное отношение сигнал/шум после ЦСФ,
ψ_0,5 ширина диаграммы направленности антенны по уровню половинной мощности.The steepness of the direction-finding characteristic of the angular discriminator is approximated by the dependence

where ρ _{e the} equivalent signal-to-noise ratio after CSF,
ψ _{0.5 the} width of the antenna pattern at half power.

As the analysis of this dependence, constructed in FIG. 13 by the method of mathematical modeling (in units of 1 / ψ _0.5 ), a change in the signal-to-noise ratio ρ _e by 6 dB (from 3 dB to 9 dB) leads to a change in the slope k _ψ (ρ _e ) by 2 times. An analysis of the dependence of the slope of the characteristic of the range discriminator k _R (ρ _e ) obtained by mathematical modeling (Fig. 14) leads to a similar result, in addition, for large signal-to-noise ratios (ρ _e > 10 dB), the range discriminator becomes substantially nonlinear and the steepness k _R (ρ _e ) decreases with increasing ρ _e .

где DU_ψ дисперсия сигнала рассогласования на выходе дискриминатора,
a коэффициент передачи интегратора,
w угловая скорость перемещения линии визирования на сопровождаемую цель.Consideration of the equivalent circuit of the tracking contour in angular coordinates and in range in the linear approximation (Fig. 15) allows us to obtain expressions for tracking errors. In particular, the variance of the fluctuation error and the value of the dynamic error of the angular accompaniment are expressed as

where DU _{ψ is the} variance of the error signal at the discriminator output,
a integrator transmission coefficient,
w the angular velocity of the line of sight on the tracked target.

 The expressions for range tracking errors are similar to (4), (5), and are obtained from them by substituting j _ → R, ω _ → v and v is the component of the target’s speed relative to the radar along the propagation direction.

As can be seen from expressions (4), (5), the steepness k _ψ (ρ _e ) leads to an increase in angular tracking errors, while an increase in the steepness with increasing signal-to-noise ratio leads to a loss of circuit stability.

 A similar situation exists for tracking along range.

 In the inventive radar by controlling the power of the emitted pulses, stabilization of the signal-to-noise ratio of the received signals is achieved, and therefore, the steepness of the characteristics of the discriminators and the quality factor of the tracking contours in angular coordinates and in range. As follows from the foregoing, this leads to an increase in tracking accuracy by 1.5-2 times and ensures the stability of tracking contours.

с изменением дальности до сопровождаемой цели ведет к снижению ошибок сопровождения по дальности вследствие дискретности вплоть до их практического устранения. Ошибка из-за дискретности при этом уменьшается в m раз и становится меньше ошибки за счет шумов. Так, например, при τ_и = 1 мкс и σ_R = 4,5 м и неподвижной сетке получим для стандартного отклонения ошибки из-за дискретности величину

применение же "подвижной сетки" способно уменьшить эту ошибку в m=16 раз, доведя ее до 2,5 м, так что результирующая ошибка составляет около 5 м, то есть влияние дискретности практически устраняется.The use of a "moving grid" of the movement of a sequence of clock pulses along the time axis with low resolution

with a change in the range to the followed target, it leads to a decrease in range tracking errors due to discreteness up to their practical elimination. The error due to the discreteness decreases m times and becomes smaller due to noise. So, for example, for τ _and = 1 μs and σ _R = 4.5 m and a fixed grid, we obtain for the standard deviation of the error due to the discreteness

the use of a "moving grid" is able to reduce this error by m = 16 times, bringing it to 2.5 m, so that the resulting error is about 5 m, that is, the effect of discreteness is practically eliminated.

 Thus, the implementation of the proposed measures will lead to a significant increase in stealth, noise immunity and accuracy of tracking radar autonomous and command control systems.

 The effectiveness of the proposed radar is highest in terms of organized radio countermeasures.

Sources of information
1. Pulse radar. UK application N 1514158, CL G 01 S 9/233 H 4 D, publ. 06/14/1978.

 2. Pulse coherent Doppler radar with frequency tuning. U.S. Patent No. 4,338,604, cl. G 01 S 13/24, publ. 07/06/1982 (prototype).

 3. Guided microwave attenuator. Auth. St. USSR N 456330, class H 01 P 1/32, H 01 P 1/22, 1975.

 4. Handbook of Radar, ed. M. Skolnik, vol. 1. M. Sov. Radio, 1976, p. 421.

 5. Handbook of Radar, ed. M. Skolnik, T. 3. M. Sov. Radio, 1979, p. 19-52, 103-107, 144.

 6. Yakovlev V. V. Fedorov R. F. Stochastic computers. L. Mechanical Engineering, 1974.

 7. Rhodes, D. R. Introduction to monopulse radar, trans. from English M. Sov. Radio, 1960.

 8. US patent, NKI 343-16, N 3162851, publ. 12/22/1964.

 9. Likharev V. A. Digital methods and devices in radar. M. Sov. Radio, 1983.

 10. Shirman Y.D. Manzhos V.N. The theory and technique of processing radar information against the background of interference. M. Radio and Communications, 1981.

 11. The SLQ-32 Story. Electronic Warfare Defense Electronics, May 1978.

Claims (2)

 1. A radar station containing a serially connected synchronizer, a frequency tuner, a driver, a phase shifter and a power amplifier, a serially connected antenna switch and an antenna, a serially connected receiver and a first block of quadrature phase detectors, as well as a code generator, a pulse modulator and an information processing device moreover, the first input of the receiver is connected to the third arm of the antenna switch, the outputs of the code generator and pulse modulator are connected to with the input inputs of the phase manipulator and the power amplifier, respectively, the output of the local oscillator frequency of the pathogen is connected to the heterodyne input of the receiver, the output of the reference frequency of the exciter with the input of the reference frequency of the block of quadrature phase detectors, characterized in that a threshold forming unit is introduced into it, connected in series through two input-outputs the quadrature components of the total signal, the first video amplifier, the signal inputs of which are connected to the outputs of the first block of quadrature phase detectors, are not a first amplitude quantizer, a first digital matched filter and a quadrature combining unit, the first to third outputs of which are connected respectively to the first third inputs of the information processing device, connected in series through two inputs and outputs of the quadrature components of the difference signal of the second block of quadrature phase detectors, the signal input of which is connected to the second output of the receiver, consisting of two identical channels, total and difference, and the input of the reference frequency to the corresponding output the pathogen has a second video amplifier, a second amplitude quantizer, a second digital matched filter and an angular discriminator, the third and fourth inputs of which are connected respectively to the first and second outputs of the first matched filter, the first commutator connected in series, the information input of which is connected to the output of the angular discriminator, and an integrator connected in series to a second switch, the information input of which is connected to the third output of the quadrature combining unit, a measurement unit intensities, a power gradation decoder, a voltage code converter and a power regulator, the second input of which is connected to the output of the power amplifier, and the output to the second arm of the antenna switch, the antenna contains an antenna unit connected to a sum-difference converter, and an antenna drive kinematically connected to it, the first input of which is connected to the output of the integrator, and the second input to the output of the azimuth signal of the target of the information processing device, the output of the difference signal of the antenna is connected to the second input of the receiver a, the control inputs of the switches are combined and connected to the output of the range gate of the information processing device, the third output of which is connected to the control inputs of the video amplifiers and the input of the synchronizer, and the fourth output with the control input of the frequency tuning unit, the second output of the synchronizer is connected to the input of the pulse modulator and to the first input code generator, the second, clock, whose input is connected to the third output of the synchronizer, and the output is connected to the control inputs of digital matched filters, inputs the zeroing signal of which, the fourth input of the information processing device, as well as the inputs of the zeroing signals of the intensity measurement and threshold generation units are connected to the fourth output of the synchronizer, the third output of which is also connected to the fifth input of the information processing device, the sixth input of which is connected to the information output of the antenna drive, and the seventh input is connected to the output of the threshold forming unit, the clock input of which is combined with the clock inputs of digital matched filters and connected to the fifth output do information processing devices.  2. The station according to claim 1, characterized in that the information processing device comprises a series-connected comparison unit with a threshold, a primary processing unit and a secondary processing unit, a key and a buffer storage device, as well as a range finder, and the first range-measuring device, connected in series via two inputs and outputs the input of the range finder is connected to the output of the comparison unit with a threshold, and its first output, third and fourth inputs are connected respectively to the second input, the first and second outputs of the secondary processing unit, the third input of which is connected the output of the buffer storage device, the control input of which is connected to the control input of the key, the output of the range strobe of the range finder and is the output of the range strobe of the information processing device, the outputs of the secondary processing unit from third to fifth are the second fourth outputs of the information processing device, respectively, and the third output of the range finder combined with the third input of the primary processing unit is the fifth clock output of the information processing device, the first and second inputs of which are are dined with the information inputs of the key, the third input is connected to the first input of the comparison unit with a threshold, the fourth input is to the inputs of the start signal of the range of the primary processing unit and the range finder, the second, clock, whose input is connected to the fifth input of the information processing device, the sixth and seventh inputs of which are formed respectively, the second inputs of the primary processing unit and the comparison unit with a threshold.

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