RU2114444C1 - Target tracking monopulse radar - Google Patents

Abstract

FIELD: radars, applicable in coherent-pulsed target tracking radars. SUBSTANCE: radar uses series-connected synchronizer, transmitter, receive-transmit selector switch, summary- difference converter and antenna, mixer, intermediate- frequency amplifier, phase detectors with video filters in summary channel, mixer, intermediate- frequency amplifier, phase detectors with video filters in difference channel, phase shifter, unit, analysis and solution unit, two adders, two multipliers, frequency detector, analysis and solution unit, quadrature unification unit, range finder, low-pass Doppler filters of summary and difference channels, code-to-time interval converter and integrator. EFFECT: enhanced tracking noise immunity with respect to passive and active noise, enhanced accuracy of tracking. 12 dwg

Description

 The invention relates to radar technology and can be used in coherent-pulse radar stations (radar) with a single-pulse direction finding principle, in particular in radars using complex, in particular, phase-shifted signals, and mounted on mobile carriers.

 Currently, in mono-pulse radars, a tracking method in range and angular coordinates is used, in which the received high-frequency signals after the sum-difference conversion are fed to the mixers of the total and difference receiving channels, where they are converted into intermediate-frequency signals, which are amplified in the intermediate-frequency amplifiers (IFA) and then fed to the amplitude detector - in the total channel and to the phase detector, while the intermediate frequency signal of the total channel plays the role of reference to hesitation. As a result of phase detection, an angular mismatch signal is generated, which is used for angular tracking [1, p.22, Fig. 1.9].

 The closest in technical essence to the proposed one is a monopulse radar target tracking [2, p. 20, fig. fifteen]. This radar contains a serially connected synchronizer, transmitter, receive-transmit switch, a sum-difference converter and an antenna, a series-connected mixer and an intermediate frequency amplifier in the total channel, a series-connected mixer, an intermediate frequency amplifier and a phase detector with a video filter in the difference channel, and video filter in the total channel, range finder and series-connected integrator of angular error errors and antenna drive, kinematically coupled an antenna, wherein the signal input of the mixer of the total channel is connected to the third arm of the receive-transmit switch, the signal input of the mixer of the difference channel is connected to the second output of the sum-difference converter, the heterodyne inputs of the mixers and difference channels are combined and connected to the output of the heterodyne frequency generator, and the synchronizer output of the synchronizer also connected to the corresponding input of the range finder.

 In the prototype device, the signal of the total channel after amplitude detection is fed to a temporary discriminator included in the range tracking unit [2, p.20, Fig. 15], in which a range mismatch (error) signal is generated, which is then used for range tracking.

 The disadvantage of the prototype device is its low noise immunity with respect to passive interference - reflections from extended local objects, such as, for example, a coastline when detecting surface ships (NK) located near the shore, as well as a cloud of dipole reflectors (DO) set by the NK at a relatively low altitude of 50 - 100 m with the task of disrupting escort. In situations where the direction of the coastline (warhead) in the observation area is close to or coincides with the direction of radar radiation propagation, the angle resolution determined by the radiation pattern of the radar antenna may be insufficient for the spatial separation of signals from the NK and from the warhead (or DO) located at the same ranges. Then the use of the prototype device will lead to guidance to the energy center of the NK-BCH or NK-DO system, and later, when approaching and with sufficient interference intensity, the tracking will be disrupted in range and angular coordinates.

 We illustrate what has been said by the following example.

где
X - расстояние между НК и береговой чертой в направлении, перпендикулярном направлению распространения;
R - дальность до НК;
φ_0,5 - ширина ДНА по уровню "-3 дБ".Let the radar installed on the aircraft (LA), monitor the NK located near the coast (Fig. 12), and the angular resolution due to the real DND is absent, that is

Where
X is the distance between the NK and the coastline in the direction perpendicular to the direction of propagation;
R is the range to the NK;
φ _0,5 - the width of the bottom at the level of "-3 dB".

For example, this occurs at X = 300 m, R = 10 km, and φ _0.5 = 0.15 rad.

где
ΔR - разрешение по дальности;
ϑ - угол скольжения;
σ^° - удельная ЭОП берега.For the effective reflecting surface (EOC) of the coastal part falling into the pulsed volume, we obtain

Where
ΔR - range resolution;
ϑ - slip angle;
σ ^° - specific coastal intensifier.

где
H - высота полета ЛА,
получим при ΔR = 300 м, X = 300 м, R = 10 км и φ_0,5= 0,15 рад, σ^° = 0,03 (-15дБ- [3, с. 296, рис. 22)], σ_δ = 1,2•10³•300•0,03=1,2•10⁴ м²
Таким образом, даже при предварительном наведении антенны РЛС на НК с ЭОП σ_ц ≅ 5•10³м² (например, по информации от навигационной системы), энергетический центр отражений будет сильно смещен в сторону береговой черты, что и приведет с большой вероятностью к срыву сопровождения по угловой координате. Аналогичная картина имеет место и с сопровождением по дальности.In particular, at small sliding angles, for example,

Where
H is the flight altitude of the aircraft
we obtain at ΔR = 300 m, X = 300 m, R = 10 km and φ _0.5 = 0.15 rad, σ ^° = 0.03 (-15dB- [3, p. 296, Fig. 22)], σ _δ = 1.2 • 10 ³ • 300 • 0.03 = 1.2 • 10 ⁴ m ²
Thus, even when the provisionally hover radar antenna on NK with GO σ _n ≅ 5 • March ¹⁰ m ² (e.g., information from the navigation system), the energy reflection center is strongly biased towards the coastline as the result is likely to disruption of tracking in the angular coordinate. A similar picture takes place and accompanied by range.

In radars using, in particular, phase-shifted (FM) signals with a large base
N = T _and Δf ≫ 1, (1)
Where
T _AND - signal pulse duration;
Δf is the signal spectrum width,
in addition, the prototype device has insufficient accuracy in measuring the angular coordinate.

. Поэтому наиболее рациональным способом оптимальной обработки ФМ сигналов большой длительности является их сжатие по времени на видеочастоте средствами цифровой вычислительной техники.At present, the optimal filtering of complex, in particular, FM signals (time compression) with a long duration (T _AND > 50 μs) at a high or intermediate frequency in monopulse radars is practically unrealizable already because it is not possible to ensure a sufficiently high identity of the compression devices on radio frequencies so that the stray phase shift between the total and difference channels, in any case, does not exceed

. Therefore, the most rational way to optimally process FM signals of long duration is to compress them in time on a video frequency using digital computer technology.

On the other hand, the formation of an angular mismatch signal by using a phase detector connected between the outputs of the IF amplifier of the sum and difference channels and multiplying the FM signals of the sum and difference channels before compression, followed by low-pass filtering, leads to a relatively low accuracy of angle measurement and insufficient noise immunity with respect to noise interference. The fact is that for large signal-to-noise ratios (ρ ≫ 1), the main noise component at the output of the phase detector performing the operation of multiplying the oscillations supplied to its inputs and low-pass filtering of the multiplication results is a component representing the product of the signal voltage in the total channel by noise voltage in the difference channel (near the equal-signal direction). However, in the case of FM signals with a large base, the signal-to-noise ratio to compression ρ _{0 is} usually small, i.e., ρ ₀ ≪ 1, and then the main noise component at the output of the above-mentioned phase detector (PD) is a component representing the product of the noise voltage in the total channel noise voltage in the difference channel. It is this circumstance that is the reason for the relatively low measurement accuracy and noise immunity of the prototype device.

имеют вид

где
σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{ш}}$ - дисперсия шумов в приемных каналах на входах ФД;
U_С -амплитуда сигналов в суммарном приемном канале на входе ФД;
g_Δ - усиление антенны разностного канала, нормированное к усилению антенны суммарного канала (по полю);
K - коэффициент пропорциональности.We illustrate what has been said by the following calculation. Expressions (2) and (3) for the variance of the process σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{out}}$ and signal voltage S _o at the PD output taking into account low-pass filtering (with a band ≈

have the form

Where
σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{w}}$ - dispersion of noise in the receiving channels at the inputs of the PD;
U _{With the} amplitude of the signals in the total receiving channel at the input of the PD;
g _Δ is the gain of the antenna of the difference channel, normalized to the gain of the antenna of the total channel (in the field);
K is the coefficient of proportionality.

где

- крутизна пеленгационной характеристики [4, с. 130].The accuracy of a single measurement of angular mismatch is determined by the ratio

Where

- the steepness of the direction-finding characteristic [4, p. 130].

тогда из (2) - (4) получим

а для ФМ сигналов при N>>1, наоборот ρ₀≪ 1,, тогда

При одинаковой энергии принимаемых сигналов ρ = Nρ₀, как видно из (6), для ФМ сигналов в устройстве-прототипе дисперсия измерения получается сравнительно большой, а точность, следовательно, сравнительно невысокой.For simple pulse signals, N = 1,

then from (2) - (4) we get

and for FM signals at N >> 1, on the contrary, ρ ₀ ≪ 1, then

For the same energy of the received signals ρ = Nρ ₀ , as can be seen from (6), for the FM signals in the prototype device, the dispersion of the measurement is relatively large, and the accuracy, therefore, is relatively low.

, не как

- в случае простых сигналов, то есть помехозащищенность устройства-прототипа по отношению к шумовым помехам также является низкой.In addition, when exposed to noise interference, the signal-to-noise ratios ρ and ρ ₀ decrease, while σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{β}}$ increases in the case of FM signals as

, not like

- in the case of simple signals, that is, the noise immunity of the prototype device with respect to noise interference is also low.

 An object of the invention is to increase the noise immunity of tracking in relation to passive and active noise interference while increasing the accuracy of tracking in radars using complex, in particular, FM signals and mounted on mobile carriers.

 To achieve the proposed technical result, it is proposed, after isolating the signal from the true target against the background of interfering reflections from passive interference, by narrow-band Doppler filtering, then to organize tracking of the target by Doppler frequency, that is, by the radial velocity of the target relative to the radar, thereby ensuring adaptive filtering of echo signals in the process tracking, and the signals from the true target filtered in this way are then used for tracking in range and in angular coordinates, and at accompanied by angular coordinates to form an angular mismatch signal after matched filtering of video pulses and narrow-band Doppler filtering in the total and difference channels by pairwise multiplication of the same quadrature signal components with the summation of these products.

The essence of the invention lies in the fact that in a monopulse target tracking radar, which contains a serially connected synchronizer, transmitter, a receive-transmit switch, a sum-difference converter and an antenna, a series-connected mixer of an intermediate frequency in the total channel, a series-connected mixer, an intermediate frequency amplifier, and a phase a detector with a video filter in the difference channel, as well as a video filter in the total channel, a range finder and series-connected error integrator the main mismatch and the antenna drive kinematically connected to the antenna, the signal input of the channel mixer connected to the third arm of the receive-transmit switch, the signal input of the difference channel mixer connected to the second output of the sum-difference converter, the heterodyne inputs of the mixers of the sum and difference channels are combined and connected to the output of the heterodyne frequency generator, and the output of the clock pulses of the clock signal is also connected to the corresponding input of the range finder, a frequency offset unit, a phase detector and a video filter in the total channel additional phase detector with a video filter in the difference channel, the phase shifter 90 ^o, consecutively connected along two lines first dual key block and a block of Doppler filters serially connected analysis unit and solutions and a first adder connected in series by two lines, a second two-channel block of keys and a frequency discriminator, connected in series along two lines with a Doppler low-pass filter of the total channel and the block is combined a quadrature connected in series along two lines through the respective multipliers, a Doppler low-pass filter of the difference channel and a second adder, as well as a code-time interval converter and a key, the transmitter comprising a series-connected exciter, including signal, heterodyne, and reference frequency generators, and a power amplifier whose output is the output of the transmitter, as well as a pulse modulator, the input of which is the input of the transmitter, and the output is connected to the control input of the force ator power output reference frequency of agent through the frequency shift unit is connected to the combined input reference frequency of the first phase detector in the sum and difference channels directly, with the combined input reference frequency the second phase detectors in the sum and difference channels - through the phase shifter 90 ^o, the combined inputs the first and second phase detectors of the total channel are connected to the output of the intermediate frequency amplifier of the total channel, and their outputs are connected through the corresponding video filters with connected by the same inputs of the first and second block of keys and the Doppler low-pass filter of the total channel, the combined inputs of the first and second phase detectors of the differential channel are connected to the output of the intermediate-frequency amplifier of the difference channel, and their outputs are connected through the corresponding video filters to the corresponding inputs of the Doppler low-pass filter a difference channel, a multi-channel output of a block of Doppler filters is connected channel-by-channel with a multi-channel input of a block of analysis and The second input of the first adder is connected to the output of the frequency discriminator, and the output of the first adder is connected to the second information input of the frequency offset unit, the output of the quadrature combining unit is connected to the signal input of the range finder, the input of the initial range and command input of which are connected respectively to the outputs of the analysis unit and solutions, the output of the range code of which is connected through the code-time interval converter with the control input of the first block of keys, the first and second inputs of the block combining quadra the tour is also connected to the second inputs of the first and second multipliers, respectively, the output of the range finder is connected to the control inputs of the second block of keys and the key, the signal input of which is connected to the output of the second adder, and the output is connected to the input of the integrator, the output of the clock pulses of the clock repetition frequency is also connected with the combined corresponding code-time interval converter inputs, rangefinder, Doppler filter unit, frequency discriminator, as well as total and value channels, and the synchronizer clock output is connected to the combined clock inputs of the frequency offset unit, code-time interval converter, and range finder.

According to the invention, the received signals in both the sum and difference receive channels after amplification at an intermediate frequency are converted to a video frequency by phase detection using the reference oscillations generated by the transmitter exciter, with the formation of two quadrature components for each signal, then matched filtering (SF) the quadrature components of the pulsed signals at the video frequency, which leads to time compression, and as a result, at the outputs of the quadrature channels in the signal-to-noise ratio, the power increases N times, and compressed signals are obtained
U _Σ cosφ, U _Σ sinφ, U _Δ cosφ, U _Δ sinφ,
Where
φ is the initial phase of the received signals relative to the reference oscillations;
U _Σ , U _Δ is the amplitude of the signals after the SF in the total and difference receiving channels, respectively, and the value of U _Δ can be either positive or negative (depending on the position of the direction to the target relative to the equal-signal direction).

где
F_П - частота повторения зондирующих импульсов РЛС, - с полосой ΔF , определяемой шириной спектра межпериодных флюктуаций истинной цели, то есть НК, и числом частотных каналов

, которая производится для каждого из элементов n_i (i = 1, 2, ... n_R) разрешения по дальности (в частности, поочередно) в зоне возможного положения цели, определяемой точностью целеуказания до нахождения цели.Next, the envelope of the total signal is subjected to multichannel Doppler filtering in the frequency range of the Doppler F _D

Where
F _P - the repetition frequency of the probe pulses of the radar, - with a band ΔF, determined by the width of the spectrum of inter-period fluctuations of the true target, that is, NK, and the number of frequency channels

, which is performed for each of the elements n _i (i = 1, 2, ... n _R ) of range resolution (in particular, alternately) in the zone of the possible position of the target, determined by the accuracy of target designation until the target is found.

After conducting multi-channel Doppler filtering (multi-channel is necessary, since the speed of the ND relative to the radar is not known in advance), it is proposed to determine the spectral width with the most powerful spectral component from the number of components that exceed the detection threshold, determined by the allowable probability of false detection due to noise, and compare it with a pre-selected threshold value, upon exceeding which a decision is made on the presence of a false target and on the transition to the analysis of the next element range, and in the absence of excess, i.e. in the presence of a powerful narrowband spectral component, - the presence of the signal from NC. In this latter case, it is proposed to find the value of the Doppler frequency F _* corresponding to the maximum of the spectrum (that is, corresponding to the radial velocity of the NC relative to the radar), and then, changing the frequency of the reference oscillations by this value, then determine the frequency inconsistency signal between the value of this correction and the measured the current value of F _* and, closing the frequency tracking loop, use this signal to fine-tune it. In the process of frequency tracking, the frequency is F _* = 0 (with an accuracy of the order of ΔF), therefore, it is further proposed to filter the complex envelope of the sequence of pulse signals in the total and difference receiving channels at zero Doppler frequency and after this filtering, forming a mismatch signal in the total channel in a known manner , for example, using the method of two half-gates (for example, [2, p. 53, Fig. 41]), close the range tracking loop, adjusting, as usual, the position of the range gate, as usual, with a mismatch signal. Further, it is proposed to generate an angular mismatch signal not by multiplying the signals at an intermediate frequency, as in the prototype, but by pairwise multiplying the signals of the same quadrature in the total and difference channels after matched filtering and narrow-band filtering at zero Doppler frequency, followed by summing the multiplication results, and then close the angle tracking contour, adjusting the mismatch signal to position the antenna, and the angular tracking contour is gated as usually a range gate.

In FIG. 1 shows a structural diagram of a radar; in Fig.2 - structural diagram of the pathogen (B) of the transmitter; in FIG. 3 is a block diagram of a frequency bias unit; in FIG. 4 is a block diagram of a code-frequency converter (PCC); in FIG. 5 is a block diagram of a double balanced modulator (DBM); in FIG. 6 is a structural diagram of a block of Doppler filters (BDF); in FIG. 7 is a structural diagram of a frequency discriminator (BH); in FIG. 8 is a structural diagram of a Doppler filter (DF _r ) tuned to frequencies ± rΔF; in FIG. 9 is a structural diagram of a Doppler filter (DF ₀ ) tuned to a zero frequency; in FIG. 10 is a structural diagram of a range finder (D); in FIG. 11 is a program diagram of an analysis and decision unit (BAR); in FIG. 12 is a horizontal arrangement of the radar and observation objects.

In FIG. 1 presents a structural diagram of the radar, on which the notation:
1 - pathogen (B); 2 - power amplifier (PA); 3 - receive-transmit switch (RFP); 4 - total difference converter (PSA); 5 - antenna (A); 6 - pulse modulator (IM); 7 - synchronizer (C); 8 - frequency offset unit (BSC); 9 and 10 - video filters of the total channel; 11 - mixer total channel (cm _Σ ); 12 - differential channel mixer (Cm _Δ ); 13 - amplifier of the intermediate frequency of the total channel (IFA _Σ ); 14 - amplifier intermediate frequency differential channel (IFA _Δ ); 15 - 18 - phase detectors (PD); 19 - phase shifter (PV); 20 and 21 - difference channel video filters; 22 - the first block of keys (BC _I ); 23 - block Doppler filters (BDF); 24 - block analysis and decisions (BAR); 25 - code-time interval converter (PCVI); 26 - the first adder (+ ₁ ); 27 - the second block of keys (BC ₂ ); 28 - frequency discriminator (BH); 29 - Doppler low-pass filter of the total channel (DF _0Σ ); 30 - Doppler filter of the low frequencies of the differential channel (DF _0Δ ); 31 and 32 are multipliers; 33 - block combining quadratures (BOK); 34 - second adder (+ ₂ ); 35 - range finder (D); 36 - key (K _l ); 37 - integrator of errors of angular mismatch (∫); 38 - antenna drive (PrA);
In the diagram of FIG. 1, a pathogen 1, a power amplifier 2, a receive-transmit switch 3, a sum-difference converter 4 and an antenna 5 are connected in series, a total channel mixer 11 is connected in series, the input of which is connected to the third arm of the switch 3, an intermediate frequency amplifier 13, a phase detector 15 and video filter 9, a phase detector 16, the input of which is connected to the output of the amplifier 13, is connected in series, and video filter 10, a mixer 12, the input of which is connected to the output of the difference signal a differential converter 4, and the heterodyne input is combined into the heterodyne input of the mixer 11 and connected to the output of the local oscillator frequency of the pathogen 1, an amplifier 14 of the intermediate frequency of the difference channel, a phase detector 17 and a video filter 20, a phase detector 18 is connected in series, the input of which is connected to the output of the amplifier 14, and a video filter 21. The outputs of the first and second video filters 9 and 10 of the total channel are connected to the corresponding inputs of the first block 22 keys, the outputs of which are connected to the corresponding inputs of the block 23 Doppler filters, the second block of keys 27, the outputs of which are connected to the corresponding inputs of the frequency discriminator 28, the Doppler filter 29, the outputs of which are connected to the corresponding inputs of the block 33 combining quadratures. The outputs of the first and second difference channel filters 20 and 21 are connected to the corresponding inputs of the Doppler filter 30, the outputs of which are connected through the respective multipliers 31 and 32 to the corresponding inputs of the two-input adder 34. The reference frequency inputs of the phase detectors 15, 17 and 16, 18 of the same quadrature are paired interconnected and connected to the output of the frequency offset unit 8, the first pair is directly, and the second pair is through the phase shifter 19. The first and second outputs of the Doppler filter 29 are also connected to the second inputs of multipliers 31 and 32 respectively.

The multi-channel output of block 23 of the Doppler filters is connected channel-by-channel with the corresponding multi-channel input of block 24 of analysis and decisions, the output of 1 range code (1st output) of which is connected via the converter 25 code-time interval to the control input of the first block of 22 keys, and the output is 2 Doppler frequencies F _{* is} connected to the first input of the first two-input adder 26, the second input of which is connected to the output of the frequency discriminator 28. The third output (initial distance to the circuit R ₀ ) of block 24 and the fourth output are commands to close the circuit with range checking - connected respectively to the second and third input of the range finder 35, the output of which is connected to the control inputs of the second block of keys 27 and key 36, the signal input of which is connected to the output of the second two-input adder 34, and the output through the integrator 37 of the angular mismatch is connected to the drive 38 kinematically associated with the antenna 5.

 The first input of the frequency offset unit 8 is connected to the output of the reference frequency of the exciter 1, the second input of the frequency offset unit 8 is connected to the output of the first two-input adder 26, the third input is the input of the radial carrier’s own radial speed, and the fourth input is combined with the same input of the range finder 35 and the second the input (clock pulses) of the converter 25 is a code-time interval, connected to the first output (clock pulses) of the synchronizer 7, the second output of which (clock pulses of the repetition frequency) is connected to the input of the imp snogo modulator 6, and inputs to the combined clock repetition frequency converter 25, a code-time interval, the range finder 35, the Doppler filter unit 23, a frequency discriminator 28, as well as Doppler filters 29 and 30, lowpass sum and difference channels. Blocks 1, 2, and 6 form a transmitter.

In FIG. 2 presents a structural diagram of the pathogen 1, where the following notation:
39 - master oscillator (ZG); 40 - frequency multiplier (UMN); 41 - mixer (SM); 42 - reference frequency generator (GOCH); 43 - 45 - amplifiers (Us).

 In the diagram of FIG. 2, a master oscillator 39, a frequency multiplier 40 and an amplifier 43, the output of which forms the first output (oscillations of the signal frequency) of the exciter 1 are connected in series, the output of the multiplier 40 is also connected through the mixer 41 to the amplifier 44, the output of which is the second output (oscillations of the local oscillator frequency) of the exciter 1, the input of the mixer 41 is connected to the output of the reference frequency generator 42 and the input of the amplifier 45, the output of which is the third output (reference frequency oscillations) of the pathogen 1.

In FIG. 3 is a structural diagram of a frequency offset unit 8, where the following notation is adopted:
46 - the first code-frequency converter (PCC ₁ ); 47 - the second code-frequency converter (PCC ₂ ); 48 - the first double balanced modulator (DBM ₁ ); 49 - the second double balanced modulator (DBM ₂ ).

 In the diagram of FIG. 3, the output of the first code-frequency converter 46 through the first double balanced modulator 48 is connected to the second input of the second double balanced modulator 49, the first input of which is connected to the first input of the frequency offset unit 8 — oscillations of the initial frequency of the reference oscillations, and the output forms the output of the frequency offset unit, corrected reference oscillation frequency, the first input of the first code-frequency converter 46 is connected to the second input of the frequency offset unit 8 (target Doppler frequency code), the third input of which (Doppler often radar carrier) is connected to the first input of the second code-frequency converter 47, the second inputs of both code-frequency converters 46 and 47 are combined and connected to the fourth input (clock pulses) of the frequency offset unit 8.

In FIG. 4 is a block diagram of code-frequency converters 46 (47), where the following notation is used:
50 - decoder (Dsh); 51 - controlled divider (DU); 52 - low-pass filter (low-pass filter).

 In the diagram of FIG. 4, a decoder 50, a controlled divider 51 and a low-pass filter 52 are connected in series, and the input of the decoder 50 forms the first code input of the code-frequency converter 46 (47), the second input of the controlled divider 51 forms the second input (clock pulses) of the code converter 46 (47) -frequency, and the output of the low-pass filter 52 is the output of the code-frequency converter 46 (47).

; 54 и 56 - перемножители (X); 57 - двухвходовый сумматор (+).In FIG. 5 is a structural diagram of a dual balanced modulator 48 (49), where the following notation is adopted:
53 and 55 - phase shifters on

; 54 and 56 - multipliers (X); 57 - two-input adder (+).

In the diagram of FIG. 5, phase shifters 53 by 90 ^° , multiplier 54 and adder 57 are connected in series, the input of phase shifter 53 by 90 ^° , combined with the first input of multiplier 56, forms the first input of a double balanced modulator 48 (49), and its second input is connected to the second input of multiplier 54 and through the phase shifter 55 by 90 ^° with the second input of the multiplier 56, the output of which is connected to the second input of the two-input adder 57, and the output of the latter is the output of the double balanced modulator 48 (49).

In FIG. 6 is a structural diagram of a block of 23 Doppler filters (BDF), where the following notation:
58 and 59 - analog-to-digital converters (ADC); 60 ₁ - 60 _{m / 2} - Doppler filters of frequency channels.

± ΔF, ... ± rΔF, ... ± m / 2 ΔF (DF ₁ , ... DF _r , ... DF _{m / 2} )
In the diagram of FIG. 6, the first and second inputs of the block 23 of Doppler filters are connected to the same inputs of the Doppler filters 60 through analog-to-digital converters 58 and 59, respectively, the third input of the block 23 of the Doppler filters - sync pulses of the repetition frequency - is directly connected to the same inputs of the Doppler filters 60, the outputs of the same Doppler filters ordered in ascending order of frequency channel numbers form an m-channel output of a block of 23 Doppler filters.

In FIG. 7 is a structural diagram of a frequency discriminator 28 (BH), where the following notation is adopted:
61 and 62 - analog-to-digital converters (ADC); 63 - block Doppler filters of frequency channels ± ΔF; 64 - subtraction block (-).

 In the diagram of FIG. 7, the first and second inputs of the Doppler filter unit 63 are connected to the same inputs of the frequency discriminator 28 through analog-to-digital converters 61 and 62, respectively, and the third input is directly, the outputs of the Doppler filter unit 63 are connected to the corresponding inputs of the subtraction unit 64, the output of which forms the frequency output discriminator 28.

In FIG. 8 is a structural diagram of a block 60 _{r of} Doppler filters of frequency channels ± rΔF (r = 1, 2, ..., m / 2), where the following notation is used:
65 and 66 - the first and second blocks of n-bit shift registers, respectively (BSR ₁ , BSR ₂ ); 67 and 70 - n-input adders with cosine weight function (BC _cos ); 68 and 69 - n-input weight combiners with sine weight function (BC _sin ); 71 and 74 - two-input adders (+); 72 and 73 - subtraction blocks (-); 75 and 76 are blocks combining quadratures (BOK).

In the diagram of FIG. 8, the first and second inputs of the block 60 _{r of} Doppler filters are connected to the information inputs of the first and second blocks of the shift registers 65 and 66, respectively, the inputs of the clock pulses of the latter are combined and connected to the third input of the block 60 of the Doppler filters. The inputs of the same name n-input weight adders 67 with a cosine weight function and 68 with a sine weight function are paired together and connected to the outputs of the corresponding bits of the block 65 of n-bit shift registers, and the inputs of the same name n-input weight adders 69 with a sine weight function and 70 with a cosine weight function are also pairwise interconnected and connected to the outputs of the corresponding bits of the second block 66 of n-bit shift registers, the output of the weight adder 67 is connected to interconnected the first inputs of the two-input adder 71 and the subtracting unit 73, the output of the weight adder 68 is connected to the combined first inputs of the subtracting unit 72 and the two-input adder 74, the output of the weight adder 69 is connected to the second inputs of the two-input adder 71 and the subtracting unit 73, and the output of the weight adder 70 connected to the interconnected second inputs of the subtracting unit 72 and the two-input adder 74. The output of the two-input adder 71 and the subtracting unit 72 are connected to the first and second inputs of the quadrature combining unit 75 continuously, and outputs the subtraction unit 73 and a two-input adder 74 connected to first and second inputs of the quadrature combining unit 76 respectively, outputs of blocks 75 and 76 are the quadrature combining the first and second outputs of the Doppler filter 60 _r respectively.

In FIG. 9 shows the structure of the Doppler filter 29 (30) of the low frequencies (DF _{0Σ, Δ} ), where the following notation is accepted:
77 and 78 - analog-to-digital converters; 79 and 80 - the first and second blocks of n-bit shift registers, respectively (BSR ₁ , BSR ₂ ); 81 and 82 are n-input adders.

 In the diagram of FIG. 9, the first and second inputs of the Doppler filter 29 (30) are connected to the information inputs of the first and second blocks 79 and 80 of the shift registers, respectively, the inputs of the clock pulses of the BSR 79 and 80 are combined and connected to the third input of the Doppler filter 29 (30). The outputs of all n bits of the first and second block 79 and 80 of the shift registers are connected bitwise to the corresponding inputs of the n-input adders 81, 82, respectively, the outputs of the n-input adders 81 and 82 are connected respectively to the outputs 1 and 2 of the Doppler filter 29 (30).

In FIG. 10 presents a functional diagram of the range finder (D) 35, where the following notation is accepted:
83 - temporary discriminator (VD); 84 - key (C); 85 - reverse counter (PC); 86 - code-time interval converter (PCVI).

 In the diagram of FIG. 10, a time discriminator 83, a key 84, a reverse counter 85, and a converter 86 are connected in series to a time-code interval whose output is the output of the range finder 35 and the second input (strobe pulse) of the temporary discriminator 83, the input of the latter being the first signal) input of the range finder 35, the second the input (initial range code) of the range finder 35 is connected to the second input (initial setting) of the reverse counter 85. The third input (command) of the range finder is connected to the control input of the key 84, and the fourth and fifth inputs (clock pulses) and sync pulses), respectively, with the second and third inputs of the converter 86 code-time interval.

Analysis and decision unit 24 (BAR) can be made in the form of a programmable microprocessor, the program circuit of which is presented in FIG. 11. It consists of the following blocks:
87 - block assigning the initial values of the variables, the range counter i takes the values of the number of the initial range i ₀ , the code of the selected range R is the minimum value of the beginning of the zone R ₀ , stored when exiting the program in the absence of real targets;
88 is a block for checking the achievement of the number of the final range i _N (end of the zone). If the condition i≤i _{N is} not fulfilled, the program exits;
89 is a block comparing signals proportional to the powers of the spectral components S _ri in the frequency range from - F _P / 2 to F _{P / 2} , with a detection threshold S ₀ for the i-th range element, that is, checking the condition
S _ri <S ₀
for
r = -m / 2, ..., m / 2 (m is the quotient).

затем переход к блоку 91;
91 - блок обнуления параметров циклов к₁, к₂, затем переход к блоку 92;
92 - проверка условия выхода из цикла по к₁ при достижении максимального значения к. При выполнении условия к₁ ≥ к производится переход к следующему элементу дальности (блок 93), при невыполнении к блоку 94. Этой проверкой исключается возможность зацикливания программы при неограниченном увеличении параметра цикла к₁;
94 - блок нахождения нижней граничной частоты спектра с максимальной спектральной составляющей путем определения минимального целого числа к₁, не удовлетворяющему условию
S(F_* - к₁ - 1)≥aS(F_*),
где
a < 1 - выбранный заранее относительный уровень отсчета ширины спектра (например, a = 0,5). При выполнении этого условия производится переход к блоку 95, при невыполнении - к блоку 96;
95 - блок увеличения значения параметра цикла к₁ на 1 с последующим переходом к проверке его значения в блоке 92;
96 - блок проверки условия выхода из цикла по к₂ при достижении им максимального значения к. При выполнении условия к₂ ≥ к производится переход к следующему элементу дальности (блок 93), при невыполнении - к блоку 97;
97 - блок нахождения верхней граничной частоты спектра с максимальной спектральной составляющей путем определения минимального целевого числа к₂, не удовлетворяющего условию
S(F_* + к₂ + 1 )≥aS(F)
При выполнении этого условия производится переход к блоку 98, при невыполнении - к блоку 99;
98 - блок увеличения значения параметра цикла на 1 с последующим переходом к проверке его значения в блоке 96;
99 - блок сравнения спектра с максимальной спектральной составляющей с пороговым значением "к" путем проверки условия.When this condition is met, the transition to the next element of the range (block 93), if not fulfilled - to block 90;
90 - block finding the maximum value of the spectral component

then go to block 91;
91 - block zeroing parameters of the cycles to ₁ , to ₂ , then go to block 92;
92 -. Verification output conditions from cycle to ₁ when the maximum value of k When the condition for ≥ ₁ to proceed to the next item in the range (block 93), the case of violation to the block 94. This test eliminated the possibility program loop with an indefinite increase parameter cycle to ₁ ;
94 - block finding the lower cutoff frequency of the spectrum with the maximum spectral component by determining the minimum integer to ₁ that does not satisfy the condition
S (F _* - to ₁ - 1) ≥aS (F _* ),
Where
a <1 - pre-selected relative level of the spectrum width reference (for example, a = 0.5). When this condition is met, a transition is made to block 95, if not fulfilled, to block 96;
95 - block increase the value of the parameter of the cycle to ₁ by 1 with the subsequent transition to checking its value in block 92;
96 is a block for checking the conditions for exiting the cycle with respect to k ₂ when it reaches the maximum value of k. When the condition k ₂ ≥ k is fulfilled, the system proceeds to the next range element (block 93), and if it is not fulfilled, go to block 97;
97 - block finding the upper cutoff frequency of the spectrum with the maximum spectral component by determining the minimum target number k ₂ that does not satisfy the condition
S (F _* + k ₂ + 1) ≥ aS (F)
When this condition is met, a transition is made to block 98, if not fulfilled, to block 99;
98 - block increase the value of the parameter of the cycle by 1 with the subsequent transition to checking its value in block 96;
99 is a block comparing the spectrum with the maximum spectral component with a threshold value of "k" by checking the conditions.

k ₁ + k ₂ +1 ≥ k.

If this condition is met, a transition to the next range elements is performed (block 93); otherwise, a transition to block 100;
100 is a unit for assigning a value R = R _{* to a} code selected to accompany a range element i _* , outputting an R _* value to a rangefinder 35, and outputting a frequency value F _* through a two-input adder to a frequency offset unit 8. The initial value of R ₀ , remaining after assignment in block 87 in the absence of a real target, means the absence of the target for tracking either due to its non-detection, or due to the classification of all targets in a given range of ranges as interference.

 The proposed radar operates as follows.

 Before starting the radar in tracking mode, the antenna 5 is installed in the direction of the target and the size and position of the viewing area in range are determined. This is done according to target designation data obtained, for example, from a navigation system or as a result of the operation of the same radar in viewing mode.

Microwave oscillations generated by the pathogen 1 at a frequency f _c pass into a power amplifier 2, in which they are amplified, and probing pulses are formed under the influence of a pulse modulator 6 controlled by clock pulses from the synchronizer 7, following with a repetition frequency F _p . They pass the switch 3, the total-differential converter 4 and are emitted by the antenna 5 into space.

The reflected signals from the antenna 5 pass through the total channel through the sum-difference converter 4 and switch 3 to the mixer 11 of the total channel. At the same time, the signals from the antenna 5 pass through the difference channel (in the radar diagram in Fig. 1, one differential receive channel is shown, if necessary, the second differential receive channel is implemented in the same way) through the total-difference converter 4 to the mixer 12 of the differential channel. In mixers 11 and 12, the superheterodyne conversion of the received signals to an intermediate frequency occurs, and the oscillations of the microwave frequency f _G generated in the exciter 1 and coming from the output 2 of the exciter 1 to the heterodyne inputs of the mixers 11 and 12 are used as heterodyne oscillations.

 Pathogen 1 (figure 2) works as follows.

The master oscillator 39 generates continuous oscillations of a stable frequency, from which, by multiplying in the multiplier 40 and amplification in the amplifier 43, oscillations are generated with the carrier frequency of the probing signals f _c , which are supplied to the first output of the exciter. The reference frequency generator 42 generates stable oscillations of the intermediate frequency f of the _inverter , which are fed to the input of the mixer 41, to the other input of which the oscillations of the signal frequency come. After the bias at the mixer output, oscillations of the local oscillation frequency f _{G are formed} (for example, f _G = F _c + F _IF ), which, after amplification in the amplifier 44, go to the second output of the pathogen 1. Finally, the oscillations of the intermediate frequency after amplification in the amplifier 45 go to the third pathogen output 1.

 After the superheterodyne conversion of the signals in the mixers 11 and 12, the signals in the total and difference channels are amplified by the intermediate frequency in the amplifiers 13 and 14 of the intermediate frequency, respectively, and then they are fed to the first (signal) inputs of the corresponding phase detectors 15 and 16 in the total channel, 17 and 18 - in the difference.

The second inputs - the inputs of the reference frequency of the phase detectors - receive reference oscillations from the output of the frequency offset unit 8, and directly to the first phase detectors 15 and 17 in the total and difference channels, and to the second phase detectors through a phase shifter 19 by 90 ^° , that both quadrature and difference channels form two quadratures, which, as you know, allows you to eliminate the influence of the unknown initial phase of the received signals.

 Block 8 frequency offset operates as follows (Fig. 3).

The first input of frequency offset unit 8 receives intermediate frequency oscillations from the third output of the pathogen 1, the second and third inputs of unit 8 receive the Doppler frequency codes F of the target _DC and Doppler frequency F _DO corresponding to the component of the radar carrier speed towards the target (for example, radar carrier navigation system). These codes are supplied to the first inputs of the code-frequency converters 46 and 47, in which they are converted into oscillations of the corresponding frequencies.

 Converters 46 and 47 code-frequency operate as follows (Fig. 4).

 The second inputs of the converters 46 (47) code-frequency receive clock pulses of a stable frequency from the first output of the synchronizer 7 through the fourth input of the frequency offset unit 8.

 These pulses arrive at the information input of the control frequency divider 51, made on the basis of a counter, to the control input of which control signals determining the frequency division coefficient come through the first input of the converter 46 (47) and the decoder 50. Square-wave pulses in the form of a meander with the required frequency are supplied from the output of the frequency amplifier 51 to the low-pass filter 52, which isolates the first harmonic, sinusoidal oscillations of the required frequency are sent to the output of the code-frequency converter 46 (47), and the minimum frequency value in terms of absolute value corresponds to the unit of the lowest discharge code.

The oscillations of the frequencies F _DC and F _DO in block 8 are fed to the inputs of the dual balanced modulator 48, the output of which is formed by the oscillations of the total frequency F _DC + F _DO received at the second input of the dual balanced modulator 49, the first input of which through input 1 of block 8 oscillations of the intermediate frequency f _IF at the output of the double balanced modulator 49, oscillations of the total frequency f _IF + F _DC + F _{DO are formed} , which pass to the output of the frequency offset unit 8 as oscillations of the reference frequency.

 Dual balanced modulator 48 (49) works as follows (Fig. 5).

Let the oscillations a ₁ cos (2πf ₁ t-φ ₁ ) and the oscillations a ₂ cos (2πf ₂ t-φ ₂ ) come to the first input of the double balanced modulator, then the first input of the multiplier 54 after a delay of 90 ^o oscillations a ₁ sin (2πf ₁ t-φ ₁ ) come in phase in phase shifter 53, and vibrations a ₂ cos (2πf ₂ t-φ ₂ ) come to the second input, after multiplication the product comes to the first input of adder 57
ka ₁ a ₂ sin (2πf ₁ t-φ ₁ ) cos (2πf ₂ t-φ ₂ ),
Where
k is the coefficient of proportionality.

The first input of the multiplier 56 receives oscillations a ₁ cos (2πf ₁ t-φ ₁ ), and the second input of the multiplier 56, after a delay of 90 ^o in phase in the phase shifter 55, receives oscillations a ₂ sin (2πf ₂ t-φ ₂ ).

After multiplying the second input of the adder 57 comes the product
ka ₁ a ₂ cos (2πf ₁ t-φ ₁ ) sin (2πf ₂ t-φ ₂ )
As a result of the summation at the output of the adder 57, oscillations are formed
ka ₁ a ₂ sin [2π (f ₁ + f ₂ ) t-φ ₁ -φ ₂ )],
that is, the total frequency f ₁ + f ₂ .

. Видеоимпульсы с видеофильтров 9 и 10 квадратур в суммарном канале поступают на блок 22 ключей, представляющий собой пару одинаковых каскадов, которые открываются лишь на время действия на их управляющие входы прямоугольных видеоимпульсов - стробов с длительностью

(где C - скорость света), соответствующей разрешению по дальности ΔR . Эти импульсы приходят с выхода преобразователя 25 код-временной интервал, на вход которого поступает код дальности с блока 24 анализа и решений, определяющий их задержку относительно синхроимпульсов, поступающих с второго выхода синхронизатора 7 на 3-й вход преобразователя 25 код-временной интервал.After phase detection, the video signals of the quadrature of the sum and difference channels are fed to the inputs of the video filters 9, 10 and 20, 21, respectively, which in the case of simple pulsed sounding signals - rectangular radio pulses with a duration of τ _and - are pairs of identical video amplifiers with a cutoff frequency equal to

. Video pulses from video filters of 9 and 10 quadrature in the total channel are sent to the key block 22, which is a pair of identical cascades that open only for the duration of the action on their control inputs of rectangular video pulses - strobes with a duration of

(where C is the speed of light) corresponding to the range resolution ΔR. These pulses come from the output of the converter 25 code-time interval, the input of which receives the range code from the block 24 analysis and decisions, determining their delay relative to the clock pulses from the second output of the synchronizer 7 to the 3rd input of the converter 25 code-time interval.

 After passing block 22 keys, the video pulses of the signals from the targets arrive at block 23 Doppler filters, which can be made in the form of a combination of m / 2 two-channel digital filters (Fig. 6), works as follows.

The bipolar video pulses of the signals of the quadrature channels pass through the first and second inputs to analog-to-digital converters 58 and 59, where they are converted into multi-bit numbers, which are quantized signals that are sent to m / 2 of parallel-connected 60 _r filters. The third input of block 23 Doppler filters receives clock pulses with a repetition frequency F _P from the synchronizer 7. Two-channel Doppler filter 60 _r operates as follows (Fig. 8).

где

После объединения квадратурных составляющих на выходах блоков 75 и 76 образуются соответственно сигналы

которые проходят на выходы +rΔF,-rΔF блока 60_r соответственно.Quantized Signals in Quadrature Channels X $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{l}}$ , X $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{l}}$ get respectively through the first and second inputs 1 and 2 of the two-channel Doppler filter 60 _r to the blocks of the shift registers 65 and 66, respectively, filled under the action of advancing pulses with a repetition frequency F _P coming from the third input of the block 23 of the Doppler filters to the third input of the two-channel Doppler filter 60 _r . In blocks 65 and 66 of the shift n-bit registers, the signal samples are stored for n repetition periods. The stored signals from all n digits go through the taps to the inputs of the n-input weight adders 67 and 68 for the first quadrature - and 69 and 70 - for the second quadrature, after multiplying by the weight coefficients and, summing the signals go to the inputs of the addition-subtraction blocks 71 - 74, at the outputs of which the following quadrature components are formed:

Where

After combining the quadrature components at the outputs of blocks 75 and 76, signals are formed, respectively

which go to the outputs + rΔF, -rΔF of the block 60 _r, respectively.

с дискретным временем.The above transformations are, in essence, algorithms for multichannel optimal filtering of a process

with discrete time.

The signals from the outputs of the two-channel Doppler filters 60 _r go to the output of the block 23 of Doppler filters, where they are ordered in order of increasing frequency tuning, and then to the input of block 24 analysis and solutions. The operation of the analysis and decision block 24 consists in executing the program that has been described above, and the circuit of which is shown in FIG. eleven.

The analysis and decision block 24 analyzes the signals from the range elements, starting from the range R _{0 of the} beginning of the zone to which the number i ₀ corresponds. If at this range in none of the m frequency channels there are signals exceeding the detection threshold, or if they are, but their spectral width exceeds a predetermined value, a decision is issued from the first output of the analysis unit 24 and decisions about the transition to the analysis of the next range element - with number i ₀ + 1, the corresponding code is fed to the converter 25 code-time interval, the inputs of which receive clock pulses with a repetition frequency F _P - from output 2 of the synchronizer 7 - and pulses with a clock frequency f _t = 1 / τ _and distance from the time of the sync pulse. As a result, a gate pulse appears at the output of the converter 25 code-time interval, which is delayed relative to the previous position by a value of τ _and , corresponding to the resolution element, this gate pulse arrives at the control input of the key block 22 and the analysis of the next range element begins, and so on until a signal with a spectrum width smaller than the threshold value kΔF is detected in some range element with number i _* . In this case, the analysis and decision block 24 measures the frequency F _* corresponding to the maximum of the Doppler spectrum of the signal from the target, and transfers the code of this frequency through the second output and the first two-input adder 26 to the second input of the frequency offset unit 8, due to which the frequency changes by F _* reference oscillations coming from the output of the frequency offset unit 8 to the phase detectors 15-18, so that the resulting Doppler frequency of the signals from the target becomes close to zero. Simultaneously, according to the signals following from the analysis and decision block 24, the range finder 35 is opened (by input 3), and the value of the target’s initial range R _{* is} received at its input 2.

 After that, through the Doppler filters 29 and 30 low frequencies in the total and difference channels, respectively, signals from the target begin to pass.

 The Doppler filter 29 (30) operates as follows (Fig. 9).

которые соответственно поступают на выходы 1 и 2 доплеровского фильтра 29 (30).The video pulses of the signals of the quadrature channels from the outputs of the matched filters 20 and 21 - in the total and difference receiving channels, respectively - get through inputs 1 and 2 to the analog-to-digital converters 77 and 78, respectively, which quantize the signals at several levels. The quantized signals from the outputs of the analog-to-digital converters 77 and 78 X $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{l}}$ , X $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{l}}$ fall respectively on the blocks of n-bit shift registers 79 and 80, filled under the action of advancing clock pulses with a repetition frequency F _P , coming to the third input of the Doppler filter 29 (30) from the second output of the synchronizer 7. In the block of shift registers 79 and 80, which in the simplest case of quantization of signals in analog-to-digital converters at 2 levels (0,1) can be performed in the form of single shift registers, the signal samples are stored for n repetition periods. The stored signals from all n bits go through taps to the inputs of the n-input adders 81 and 82 for the first and second quadrature, respectively, the following signals are formed at their outputs, respectively

which respectively arrive at the outputs 1 and 2 of the Doppler filter 29 (30).

,
не зависящий от неизвестной начальной фазы принимаемых сигналов. Блок 33 объединения квадратурных каналов, как и аналогичные блоки 75 и 76, может быть выполнен в виде двухадресного постоянного запоминающего устройства, содержащего значения функции

от двух аргументов.Signals Y $\genfrac{}{}{0ex}{}{\text{(c)}}{\text{0}}$ , Y $\genfrac{}{}{0ex}{}{\text{(s)}}{\text{0}}$ from the outputs 1 and 2 of the Doppler filter 29 in the total channel arrive at block 33 combining quadratures, which forms a signal

,
independent of the unknown initial phase of the received signals. Block 33 combining quadrature channels, like similar blocks 75 and 76, can be made in the form of a dual-address read-only memory containing function values

from two arguments.

After the signal from the target Y ₀ to the input 1 of the range finder 35, automatic tracking of the target in range begins.

 The range finder 35 operates as follows (Fig. 10).

 The signal from the target through the first input of the range finder 35 goes to the first input of the time discriminator 83, the second signal receives a pulse from the converter 86 code-time interval (operation of the converter 86 occurs as well as the above-described converter 25 code-time interval), to the input which receives the code of the initial range to the target through the second input of the range finder 35. At the input of the temporary discriminator, a bipolar code signal of a time mismatch is generated, passing through the key 84, opened by the command that came from the third Log rangefinder 35, this signal goes to the count input of down counter 85, where the sum (with sign) from the initial code range, the new value range is supplied to a first input 86 code-interval time code converter and the like

 The strobe pulse, the position of which relative to the clock corresponds to the distance to the target, enters through the output of the rangefinder 35 to the control inputs of the second block 27 of the key and key 36. The block 27 of the keys opens while the pulses of the signal from the target pass, passing them to the frequency discriminator 28, forming the signal Doppler frequency mismatch.

 Frequency discriminator 28 works as follows (Fig. 7).

The video pulses of the quadrature components of the signal from the tracked target through the key block 27 go from the video filters 9 and 10 of the total channel to the inputs 1 and 2 of the frequency discriminator 28, then they are quantized into several levels in the analog-to-digital converters 61 and 62, respectively, the quantized signals are fed to the two-channel Doppler filter 63, the channels of which are tuned to the frequencies ΔF and -ΔF located symmetrically with respect to the zero frequency. The device and operation of the two-channel Doppler filter 63 completely coincides with the device and operation of the two-channel Doppler filter 60 _r (Fig. 8) at r = 1. The signals from the outputs ΔF and -ΔF of the filter 63 go to the inputs of the subtraction unit 64, where a bipolar encoded error signal is generated the Doppler frequency of the received signal relative to the zero frequency, this error signal from the output of the frequency discriminator 28 goes to the second input of the first two-input adder 26, where it is added to the frequency code F _* previously, from the analysis and decision block 24, the resulting Doppler frequency code falls on the second input of the frequency offset unit 8 and changes the frequency of the reference oscillations, due to which the signals from the target are tracked by the Doppler frequency, the value of which after phase detection is kept near zero.

где
C - коэффициент пропорциональности.At the same time as the key block 27, the strobe from the range finder 35 also opens the key 36, the output of which passes the angle mismatch signal generated after multiplying the quadrature components of the same name from the target transmitted through the total and difference receiving channels through the corresponding Doppler filters 29 and 30, and summing of these products in a two-input adder 34 according to the scalar product rule

Where
C is the coefficient of proportionality.

 As is known, this rule, up to a normalizing factor, coincides with the optimal rule for estimating angular mismatch during monopulse direction finding (for example, [2, p. 20 (1)]).

 The angular error signals obtained in this way, passing through the key 36, are accumulated in the integrator 37, the resulting signal from the output of the integrator 37 controls the antenna 1 using the antenna drive 38.

, поступающие с первого выхода синхронизатора 7 на четвертый вход блока 8 смещения частоты и на счетные входы преобразователей 25 и 86 код-временной интервал (последний в составе дальномера 35), а также синхроимпульсы с частотой повторения F_Пс второго выхода - для запуска импульсного модулятора 6 и для управления работой преобразователей 25 и 86 код-временной интервал. Синхронизатор 7 может быть построен на основе задающего генератора импульсов с частотой f_Т и счетчика-делителя с формирователем - для формирования синхроимпульсов с частотой F_П << f_Т.The synchronizer 7, performing general control of the RSL, generates clock pulses with a frequency

coming from the first output of the synchronizer 7 to the fourth input of the frequency offset unit 8 and to the counting inputs of the converters 25 and 86 code-time interval (the last in the range finder 35), as well as clock pulses with a repetition frequency F _P from the second output - to start the pulse modulator 6 and to control the operation of the transducers 25 and 86 code-time interval. The synchronizer 7 can be built on the basis of a master pulse generator with a frequency f _T and a counter divider with a shaper - for the formation of clock pulses with a frequency F _P << f _T.

 Thus, the proposed radar provides target tracking in angular coordinate, in range and in Doppler frequency at a higher noise immunity compared to the prototype in relation to passive interference, both organized - like dipole clouds, and natural - reflections from the coastline, and for Radars using FM signals, there is also an increase in tracking accuracy and noise immunity with respect to active noise interference.

 The effect of increasing noise immunity with respect to passive interference can be estimated as follows.

что при
σ_ц= 5•10³м², ϑ ≤ 0,1, σ^°= 0,03, ΔR = 300 м, R = 10 км, φ_0,5= 0,15, X = 300 м составляет ρ₁ ≅ 0,4(-4 дБ).
Это значение отношения сигнал/помеха недостаточно для уверенного обнаружения сигнала от цели (НК) на фоне помехи, тем более для захвата и сопровождения.In the conditions of the above example (Fig. 12) we have for the signal-to-noise ratio in the prototype, as follows from the above, even in the absence of frequency resolution

what with
σ _c = 5 • 10 ³ m ² , ϑ ≤ 0.1, σ ^° = 0.03, ΔR = 300 m, R = 10 km, φ _0.5 = 0.15, X = 300 m is ρ ₁ ≅ 0.4 (-4 dB).
This value of the signal / noise ratio is not enough for reliable detection of the signal from the target (NK) against the background of interference, especially for capture and tracking.

и при ширине спектра отражений от участка берега внутри импульсного объема, определяемого реальной ДНА и разрешением по дальности R (выше)

получим, что отношение сигнал/помеха возрастает в

, так как сигналы от НК целиком попадают в полосу ΔF = 10 Гц (при интервале корреляции флюктуаций сигналов от НК, не превосходящем 0,1 с в диапазоне длин волн λ ≅ 3,0 см , а отражения от берега отфильтровываются, как показано выше.In the proposed device in the same conditions at a time of coherent processing

and with the width of the spectrum of reflections from the coastal section inside the pulse volume, determined by the real BOTTOM and range resolution R (above)

we get that the signal / noise ratio increases in

since the signals from the NC completely fall into the band ΔF = 10 Hz (with a correlation interval of fluctuations of signals from the NC not exceeding 0.1 s in the wavelength range λ ≅ 3.0 cm, and reflections from the shore are filtered out, as shown above.

то есть 20 раз (13 дБ), что достаточно для уверенного захвата и сопровождения по дальности и угловым координатам (при моноимпульсной пеленгации).Thus, the signal-to-noise ratio in the proposed device increases, for example, by 17 dB compared with the prototype and is

that is, 20 times (13 dB), which is sufficient for confident capture and tracking in range and angular coordinates (with monopulse direction finding).

По мере сближения носителя РЛС с целью вероятности захвата и сопровождения возрастают как ввиду повышения отношений сигнал/шум и сигнал/помеха ((8), (9)), так и благодаря улучшению условия разрешения (величина R в (10) уменьшается). Наконец, в ситуациях, когда направление БЧ перпендикулярно направлению РЛС-НК, при условии нахождения НК от БЧ на расстояниях, превосходящих ΔR - разрешение по дальности, имеет место разделение сигналов от НК и от БЧ по разным каналам дальности (благодаря разрешению по дальности), и предлагаемая РЛС работает в облегченных условиях, а когда расстояние НК от БЧ меньше ΔR , осуществляется частотная селекция, как описано выше, с достаточно высокой эффективностью.Moreover, this takes place in the presence of frequency resolution, that is, when the signals from the NK and from the warhead are separated by frequency channels, when the condition

As the carrier approaches the radar with the aim of the probability of capture and tracking, they increase both due to an increase in the signal-to-noise and signal-to-noise ratios ((8), (9)) and due to an improvement in the resolution condition (the value of R in (10) decreases). Finally, in situations where the direction of the warhead is perpendicular to the direction of the radar-NK, provided that the NK from the warhead at distances exceeding ΔR - range resolution, there is a separation of signals from the NK and from the warhead at different distance channels (due to range resolution), and the proposed radar operates in light conditions, and when the distance of the NK from the warhead is less than ΔR, frequency selection is performed, as described above, with a sufficiently high efficiency.

 Thus, the proposed radar is more effective for any mutual location of the radar-NK-warhead.

и выражения для дисперсии для предлагаемого устройства, которое имеет вид

Сравнение формул (6) и (11) показывает, что в предлагаемой РЛС дисперсия в 1/ρ₀ меньше (обычно ρ₀ ≪ 1 - отношение сигнал/шум до сжатия, так как N > > 1), а точность соответственно выше, чем в устройстве-прототипе.The effect of increasing noise immunity with respect to active noise interference and the accuracy of angular tracking in radars using complex, in particular phase-shifted signals, is determined by comparing the expression for the dispersion of the angle measurement with monopulse direction finding, which for the prototype device has the form (at ρ ₀ ≪ 1) (above (6))

and expressions for dispersion for the proposed device, which has the form

A comparison of formulas (6) and (11) shows that the dispersion in the proposed radar is 1 / ρ ₀ less (usually ρ ₀ ≪ 1 is the signal-to-noise ratio before compression, since N>> 1), and the accuracy is correspondingly higher than in the prototype device.

Under the influence of noise interference, the signal-to-noise ratio decreases, while σ $\genfrac{}{}{0ex}{}{\text{2}}{\text{β}}$ increases in the proposed radar as 1 / ρ ₀ , and in the prototype as 1 / ρ $\genfrac{}{}{0ex}{}{\text{2}}{\text{0}}$ , that is, to a greater extent, so that the proposed radar has greater noise immunity compared to the prototype. In particular, when ρ ₀ = 0.1, the dispersion in the proposed radar is 10 times less than in the prototype, respectively, the accuracy is 3.15 times higher.

 Thus, the technical effect in the proposed radar is to achieve higher noise immunity with respect to active interference while achieving higher accuracy compared to the prototype.

 Using the information presented in the application materials, it is possible to manufacture and use the proposed radar to track targets in range and angular coordinates, which proves the industrial applicability of the object of the invention.

 In accordance with the application materials, a prototype of the device was manufactured, the tests of which confirmed the achievement of the technical result indicated in the application materials.

Sources of information:
1. Leonov A.I., Fomichev K.I. Monopulse radar. - M .: Owls. Radio, 1970.

 2. Reference radar. / Ed. M. Skolnik. - M .: Owls. Radio, 1978, v. 4, p. 20, fig. 15 (prototype).

 3. Reference radar. / Ed. M. Skolnik. - M .: Owls. Radio, 1976, v. 1.

 4. Falkovich S.E. Reception of radar signals against fluctuation interference. - M .: Owls. Radio, 1961.

 5. Krasyuk NP, Rosenberg V.I. Shipborne radar and meteorology. - L .: Shipbuilding, 1970.

Claims (1)

A monopulse target tracking radar, which contains a serially connected synchronizer, transmitter, a receive-transfer switch, a sum-difference converter and an antenna, a series-connected mixer and an intermediate frequency amplifier in the total channel, a series-connected mixer, an intermediate frequency amplifier, and a phase detector with video filters in the difference channel, as well as a video filter in the total channel, a range finder and series-connected error integrator angular ra mismatch and drive of the antenna kinematically connected to the antenna, the signal input of the mixer of the total channel connected to the third shoulder of the receive-transmit switch, the signal input of the mixer of the differential channel connected to the second output of the sum-difference converter, the heterodyne inputs of the mixers of the total and difference channels are combined and connected to the output of the transmitter heterodyne frequency generator, and the synchronizer clock output is also connected to the corresponding input of the range finder, which differs in m, that a frequency offset block, a series-connected decision analysis block and a first adder, two phase detectors and a video filter in the total channel, an additional phase detector with a video filter in the difference channel, a 90 ^o phase shifter, the first and second two-channel key blocks, outputs are introduced into it which are connected to the corresponding input of the block of Doppler filters and the frequency discriminator, respectively, Doppler low-pass filters of the sum and difference channels, the outputs of the first of which are connected to the corresponding the corresponding inputs of the quadrature combining unit, and the outputs of the second are connected to the inputs of the second adder through the corresponding multipliers, the second inputs of which are connected respectively to the first and second outputs of the Doppler low-pass filter of the total channel, as well as a code converter - a time interval and a key, and the transmitter contains series-connected an exciter, including signal, local oscillator and reference frequency generators, and a power amplifier, the output of which is the output of the transmitter, and a pulse modulator, the input of which is the input of the transmitter, and the output is connected to the control input of the power amplifier, the output of the reference frequency of the pathogen through the frequency offset unit is connected directly to the combined inputs of the reference frequency of the first phase detectors in the total and difference channels, and to the combined inputs of the second frequency reference detectors in the total and differential channels through a 90 ^o phase shifter, the combined inputs of the first and second phase detectors of the total channel are connected to the output the intermediate channel frequency of the total channel, and their outputs are connected through the corresponding video filters with the same inputs of the first and second two-channel key blocks and the Doppler low-pass filter of the total channel, the combined input of the first and second phase detectors of the differential channel are connected to the output of the intermediate frequency amplifier of the difference channel, and their the outputs are connected through the corresponding video filters to the corresponding inputs of the Doppler low-pass filter of the difference channel, multi-channel the output of the Doppler filter unit is connected channel by channel with the multi-channel input of the analysis and decision unit, the second input of the first adder is connected to the output of the frequency discriminator, and the output of the first adder is connected to the second information input of the frequency offset unit, the output of the quadrature combining unit is connected to the signal input of the range finder, the input of the initial range and the command input of which is connected to the outputs of the analysis and decision block of the same name, the output of the range code of which is connected through the code-time interval converter with with the input input of the first two-channel block of keys, the output of the range finder is connected to the control inputs of the second two-channel block of keys and the key, the signal input of which is connected to the output of the second adder, and the output is connected to the input of the integrator of the error of angular mismatch, the output of the synchronizer pulses is connected to the corresponding inputs of the code-time converter interval, rangefinder, block of Doppler filters, frequency discriminator, as well as Doppler filters of low frequencies of the total and difference channel And an output clock synchronizer is connected to the clock inputs of the frequency shift block code converter - time slot and rangefinder.

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