RU2278397C2 - Method and device for selection of signals from above-water target in mono-impulse radiolocation station - Google Patents

Abstract

FIELD: engineering of mono-impulse coherent radiolocation systems, operating on moveable carriers, meant for detecting signals from above-water targets and dispensing coordinates thereof to controlling system, under conditions of natural, organized active and passive interference.

SUBSTANCE: at first stage method includes determining width of spectrum with most powerful spectral component on basis of given level from number of components, exceeding first detection threshold(first criterion), received value of spectrum width is compared to second threshold value, in case of fulfillment of condition for exceeding of spectrum width of second threshold value(second false target criterion) - signal is removed from further analysis as false. At second stage for resolved signals, not dropped on basis of first or second criterion and positioned at same distance, dispersion of difference is calculated between estimates of angular position of frequency-resolved elements, produced by two methods: in accordance to Doppler frequency shift and mono-impulse method. If found dispersion exceeds third threshold(third criterion), signal, received at analyzed distance, is classified as interference signal and is also removed from analysis.

EFFECT: increased trustworthiness of classification between target and passive interference (excluding interference of corner reflector type) with view-finding angles and distances, when spectrum width if target signal is comparable to and even greater than width of dipole reflector cloud spectrum.

2 cl, 17 dwg

Description

The present invention relates to monopulse coherent radar systems (radar) operating on mobile carriers designed to detect signals from surface targets and output their coordinates to the control system in conditions of natural, organized active and passive interference.

When targets are selected against the background of passive interference, differences in the Doppler frequencies of signals from targets and interference sources are used, which are a consequence of differences in their radial velocities [1, p. 147]. The information sign used to select the signal from the target is the Doppler frequency f _{D of the} received signals - the decision on the presence of the target signal is made if | f _D | ≫0 | and goes beyond the band of the Doppler interference spectrum.

The disadvantage of this method is erroneous selection in those cases when the object being selected (true target) is stationary or moving in a direction perpendicular to the direction of radiation propagation (f _D ≈ 0), and also when the interference source has a Doppler signal spectrum (center frequency and Doppler spectrum width ) close to the Doppler spectrum of the target.

There is another method for selecting a signal from interference caused by reflections from the ground, considered in the description of the patent [2], which is based on Doppler filtering of signals received by the airborne radar into two antennas spaced apart along the fuselage. The outputs of the Doppler filters of signals received at spaced antennas are fed to a filter where the power of each frequency component P _i , the phase difference of the same frequency components of the spectra of signals received by spaced antennas Δφ _i , and the deviations of the measured phase differences from the calculated values for reflections from the Earth are determined

Where:

Δ _i is the deviation of the measured phase difference Δφ _i from the calculated for f _Di

d is the base of the antenna system,

f _Di - Doppler shift of the i-th frequency component of the signal,

V is the speed of the aircraft (LA) relative to the Earth when measured,

Δφ _i is the phase difference of the i-th frequency components of the spectrum received via the antenna spaced at a distance d.

The output of the Doppler filter is fed to a threshold device in which the signal of a moving target is selected from the signal of reflections from the Earth (coast, sea) according to two criteria. The first is the excess of signal power P _{i of a} given threshold signal-to-noise ratio. The second is the excess of the deviation Δ _{i of the} threshold nσ _φ (σ _φ is the calculated standard deviation Δ _i for reflections from the coast (sea), n is the weight coefficient of the threshold). For reflections from the Earth, Δ _i does not exceed the threshold.

The disadvantage of this method is that it distinguishes only moving and stationary targets, respectively, does not distinguish between interference in the form of a cloud of dipole reflectors and the target. In addition, the method is applicable when the target is within ± 20 ° relative to the normal to the base of the antenna system, while the base of the antenna system should be located along the fuselage.

The method and device according to the description of the patent [3] allows the pulse-Doppler radar to select the target against the background of passive interference using two criteria. According to the first criterion, the signal in the time-frequency region should exceed the first high threshold of the signal-to-noise ratio. According to the second criterion, the time-frequency target size, determined by the lower detection threshold, should be within 1≤K _f ≤K _fpor (K _f is the width of the Doppler spectrum of the signal, referred to the radar resolution by Doppler frequency) and 1≤K _r ≤ K _rpoor (K _r is the time size of the reflected signal, referred to the radar resolution in time).

At the same time, in a narrow frequency-time sliding window of M _f × M _r resolvable in frequency and range elements should be no more than one target.

With the simultaneous fulfillment of both criteria for the analyzed points of a time-frequency sliding window of size M _f × M _r , a conclusion is made about the presence of a target signal. All signals that do not meet this criterion are rejected.

This method reliably separates true targets from false passive reflections from the Earth and reflections from clouds of dipole reflectors (ODO), if the time-frequency size of the passive interference with an acceptable probability exceeds the size of the useful signal. In some cases, this is not performed. For example, if the angle of observation of a surface ship (NK) and its removal from the radar is such that the width of its Doppler spectrum is comparable to the width of the spectrum of a cloud of dipole reflectors (ODO) and even more. We illustrate this by calculating the spectrum width of the resolved NK element located at a distance of R = 6 km, observed at an angle β = 45 ° to the aircraft velocity vector. Let the linear azimuthal size of the range-resolving element of the spacecraft be equal to ΔL = 70 m, the radar wavelength λ = 3 cm and the speed of the aircraft V = 300 m / s.

The width of the spectrum of the signal reflected from the resolved NK element for these conditions is equal to:

which is wider than the expected spectral width of the ODO of 52 Hz [6, p. 272] taking into account the symmetry of the Gaussian envelope of the spectrum with respect to the central frequency [7, p. 85-88].

There is a method of tracking the target of a monopulse radar in conditions of passive and active interference [4], the closest in technical essence to the proposed one and adopted as a prototype. According to this method, in a monopulse radar, a coherent pulse signal is emitted in a given direction. The received reflected signals after the sum-difference conversion are transferred to the intermediate frequency and amplified. Amplified sum and difference signals using quadrature-phase detection are transferred to the video frequency. Further, a consistent and multi-channel Doppler filtering of signals in a given range of ranges is sequentially performed at the video frequency. The results of filtering in the total channel are used for threshold detection of signals and the subsequent determination of the width of the spectrum of the signal at the analyzed range. Then the found spectrum width is compared with a given threshold. If the spectral width of a given threshold is exceeded, a decision is made that the signal belongs to a false target, and if there is no excess, it belongs to a true target. In addition, due to narrow-band Doppler filtering of the total and difference signal, the radar resolution is increased in angle, respectively, the signal / background ratio, and the target is distinguished in the Doppler fluctuation band of reflections from the coast (sea) when the signal exceeds the threshold signal / background value. The detected target is taken for auto tracking by Doppler frequency, range and angle. In this case, to determine the error signal in range, the difference in the power of the signals at the outputs of the similar Doppler filters of the total channel tuned to the monitored Doppler frequency of the target at spaced distances is used. The angle error signal for a target tracked by Doppler frequency and range is determined by pairwise multiplication of the results of narrow-band Doppler filtering of the same quadrature signal components in the total and difference channels and summing these products.

The disadvantage of the method [4] is that it works reliably only if the observed frequency size of the spectrum of passive interference (reflections from the sea, coast, ODO) exceeds the maximum expected frequency size of the NK, which, as shown above, does not hold for all conditions of sight (observed angular dimensions of the target).

A device that implements this method is a coherent monopulse radar, described in [5]. For radiation, a pulsed (simple or complex) signal is used. After the combined filtering of the total and difference signals at the video frequency, threshold detection of the total signal is performed, the radar is switched to auto tracking mode in range and angle. The angle error signal at the followed range is defined as the scalar product of the quadrature components of the signals at the outputs of the digital matched filters of the sum and difference channels. In the tracking mode of the estimated target signal by range, Doppler spectral analysis and determination of the signal spectrum width are performed. If the width of the spectrum of the signal at the followed range does not exceed the calculated boundary, then a decision is made that the analyzed signal belongs to the target, otherwise it is an interference and is excluded from further consideration.

The disadvantage of the method implemented in a coherent monopulse radar [5] is that it reliably classifies the target / passive interference only provided that the observed width of the spectrum of passive interference (reflections from the sea, coast, ODO) exceeds the maximum estimated frequency size of the NK, which , as shown above, does not occur under all conditions of sight.

The technical task of the invention is to increase the reliability of the target / passive interference classification (with the exception of angular reflector type interference) at viewing angles and ranges when the calculated spectral width of the target signal is comparable to and even greater than the expected ODO spectrum width.

до плюс

фильтрами с полосой δf, число фильтров

где f_п - частота повторения зондирующих импульсов, ширина полосы пропускания доплеровских фильтров δf выбирается как половина ширины спектра межпериодных флюктуации сигналов от ОДО (Δf_ОДО), сравнивают мощности спектральных составляющих с первым пороговым уровнем обнаружения, определяют ширину спектра с наиболее мощной спектральной составляющей по заданному уровню из числа составляющих, превысивших порог обнаружения, сравнивают полученное значение ширины спектра со вторым заданным пороговым значением, при превышении ширины спектра второго порогового значения (второй критерий ложной цели) принимают решение о наличии сигнала от ложной цели, который как ложный отбрасывается, и переходят к следующему элементу дальности, а при отсутствии превышения вычисляют скалярное произведение квадратурных сигналов, полученных в результате фильтрации в одноименных доплеровских фильтрах суммарного и разностного каналов путем попарного перемножения одноименных квадратурных составляющих сигналов с суммированием этих произведений введено последовательное вычисление "пеленгов" (γ_i), "доплеровских углов" (β_i) и дисперсии (D_ψ) угловой разности между ними (ψ_i=γ_i-β_i) для всех разрешаемых частотно-временных элементов сцены, не забракованных по первому и второму критерию и находящихся на одинаковых дальностях, "пеленг" (γ_i) разрешаемого элемента сцены определяется делением (нормировкой) вычисленного для него скалярного произведения на мощность суммарного сигнала для того же разрешаемого элемента сцены и умножением полученного результата на масштабирующий коэффициент, "доплеровский угол" разрешаемого элемента сцены (β_i) определяется как произведение углового разноса между соседними доплеровскими направлениями (δβ_v) на номер доплеровского фильтра (i), в котором на анализируемой дальности обнаружен сигнал, найденная на каждой разрешаемой дальности дисперсия разности углов сравнивается с третьим порогом, при превышении дисперсией расчетного третьего порогового значения (третий критерий ложной цели) - сигнал сцены на этой дальности также отбрасывается, угловой разнос между соседними направлениями (δβ_v), разрешаемыми по частоте и находящимися в пределах узкой суммарной ДНА, определяется по известной связи доплеровского сдвига частоты отраженного сигнала с параметрами излучения, движения ЛА и условиями визирования по формуле:This goal is achieved by the fact that in the method described in [4], which includes emitting coherent radio pulses with a constant carrier frequency in a given direction, receiving reflected signals in a given range of ranges, sum-difference converting the received signals, converting them to an intermediate frequency, amplification the sum and difference signals at an intermediate frequency, the conversion of the spectrum of the sum and difference signals into the region of video frequencies by means of a quadrature phase detector Using reference oscillations with the formation of the quadrature components of each signal, the matched filtering of the video pulses of the quadrature components of the signals in the total and difference channels for each range element in a given time interval performs multichannel Doppler filtering of the complex envelope of the pulse sequence in the Doppler frequency range from minus

up plus

filters with band δf, number of filters

where f _p is the probe pulse repetition rate, the bandwidth of the Doppler filters δf is selected as half the width of the spectrum of inter-period fluctuations of the signals from the ODO (Δf _ODO ), the power of the spectral components is compared with the first detection threshold level, the spectrum width with the most powerful spectral component is determined from the level of the number of components that exceeded the detection threshold, compare the obtained value of the width of the spectrum with a second predetermined threshold value, when exceeding the width of the spectrum of the threshold value (the second criterion of a false target), they decide on the presence of a signal from a false target, which is rejected as a false one, and proceed to the next range element, and if there is no excess, the scalar product of quadrature signals obtained by filtering in the Doppler filters of the same total and differential channels by pairwise multiplication of the same quadrature components of the signals with the summation of these products introduced sequential calculation of "bearings" ( _i), "Doppler angle" (β _i) and the dispersion (D _ψ) angular difference therebetween _{(ψ i = γ i -β i} ) for all resolvable time-frequency elements of the scene is not rejected by the first and the second criterion and being on of the same ranges, the "bearing" (γ _i ) of the resolved scene element is determined by dividing (normalizing) the scalar product calculated for it by the power of the total signal for the same resolved scene element and multiplying the result by a scaling factor, the "Doppler angle" of the resolved scene element ( β _i ) is defined as the product of the angular separation between adjacent Doppler directions (δβ _v ) and the Doppler filter number (i), in which a signal is detected at the analyzed range, the dispersion of the angle difference found at each resolved range is compared with the third threshold, when the dispersion exceeds the calculated third the threshold value (the third criterion decoy) - signal at this stage is also discarded range, the angular spacing between adjacent directions (δβ _v), resolvable frequency and located in the pre ah overall narrow beam is determined by the known association of the Doppler shift of the reflected signal with the frequency of the radiation parameters, and conditions for the aircraft motion of sight by the formula:

Where:

λ is the wavelength of the emitted signal,

F _P - repetition frequency of the probe pulses,

β is the angle between the longitudinal axis of the aircraft and the equal signal direction (RSN) of the antenna system,

α is the drift angle of the aircraft.

As a prototype of a device that implements the proposed method, taken coherent monopulse radar [5].

The goal is also achieved by the fact that in a monopulse radar device containing a series-connected antenna switch, a sum-difference converter and an antenna, a power amplifier, an antenna drive, a two-channel receiver, the first output of which is connected to the input of the video amplifier of the total channel through a quadrature-phase detector of the total channel , the second output of the two-channel receiver through the quadrature-phase detector of the differential channel is connected to the input of the video amplifier of the differential channel, digital matched fi liters of total and difference channels, quadrature combining unit, angle discriminator, primary processing unit and secondary processing unit, while the antenna drive is kinematically connected to the third input of the antenna, the second output of the sum-difference converter is connected to the second input of the two-channel receiver, the third output of the antenna switch is connected with the first input of the two-channel receiver, the synchronizer is connected via a pulse modulator to the second input of the power amplifier, the second output of the exciter is connected to the third input a two-channel receiver house, the third output of the pathogen is connected to the second inputs of the quadrature phase detectors of the total and difference channels, the second output of the secondary processing unit is connected to the input of the antenna drive, the second output of which is connected to the seventh input of the primary processing unit, the input of the quadrature combining unit is connected to the first input of the discriminator angle, the output of the block combining quadrature is connected to the second input of the primary processing unit, the fourth output of the pathogen is connected to the input of the synchronizer, the first output d of the secondary processing unit is connected to the fourth input of the primary processing unit, the first output of which is connected to the first input of the secondary processing unit, the two-channel receiver contains a two-channel high-frequency amplifier, a two-channel balanced mixer, and a third (heterodyne) amplifier connected in series at the two inputs-outputs of the total and difference channels the input of which is the third heterodyne input of the two-channel receiver, the first and second outputs of the high-frequency amplifier are inputs of a balanced mixer and the first and second outputs of a two-channel intermediate frequency amplifier — by the outputs of the total and difference channels of the two-channel receiver, respectively, multi-channel Doppler filters of the total and difference channels, a phase shifter, a divider and a velocity and drift meter are introduced, while the third exciter output is connected to the third quadrature inputs through a phase shifter phase detectors of the total and difference channels, the output of the video amplifier of the total channel through a digital matched filter of the total channel It is connected to the first input of the multi-channel Doppler filter of the total channel, the output of the video amplifier of the difference channel through a digitally matched filter of the differential channel is connected to the first input of the multi-channel Doppler filter of the differential channel, the output of the quadrature combining unit is connected to the first input of the divider, the output of which is connected to the fifth input of the primary processing unit, the first pathogen output through the power amplifier is connected to the first input of the antenna switch, the fourth pathogen output is connected about the second inputs of digital matched filters of the total and difference channels, the second inputs of the multi-channel Doppler filters of the total and difference channels and the first input of the primary processing unit, the first synchronizer output is additionally connected to the pathogen input and the fifth inputs of the multi-channel Doppler filters of the total and difference channels, the output of the multi-channel Doppler filter the total channel is connected to the input of the quadrature combining unit, the output of the multi-channel Doppler filter is different the channel is connected to the second input of the angle discriminator, the output of which is connected to the second input of the divider, the second output of the synchronizer is connected to the third input of the primary processing unit; the fourth synchronizer output is connected to the third inputs of the multi-channel Doppler filters of the total and differential channels and the sixth input of the secondary processing unit, the third synchronizer output is connected to the fourth inputs of the multi-channel Doppler filters of the total and differential channels, the seventh input of the secondary processing unit and the sixth input of the primary processing unit, the third output the secondary processing unit is connected to the second input of the pulse modulator, the second output of the primary processing unit is connected to the fifth input ohm of the secondary processing unit, the first and second outputs of the speed and drift meter are connected to the third and fourth inputs of the secondary processing unit, respectively, the second output of the antenna drive is additionally connected to the second input of the secondary processing unit, the fourth output of the secondary processing unit is connected to the eighth input of the primary processing unit, the primary processing unit contains a series-connected threshold detector, a first circuit AND, a threshold target size selector, a series-connected integrator, a circuit in difference numbers, a dispersion calculator, a threshold device and an information output device, a target Doppler frequency meter connected in series, a second AND circuit, and an information output device, the first and second outputs of which are the second and first outputs of the primary processing unit, an adder connected to the second input of the device information, and the second input of the primary processing unit is connected to the second inputs of the threshold detector and meter Doppler frequency of the target, the fourth input of the calculator of mathematical expectation and the sixth input of the dispersion calculator, the output of the threshold detector is connected to the first inputs of the dispersion calculator and the second circuit AND, the third input of the primary processing unit is connected to the third inputs of the threshold detector, integrator, Doppler frequency meter, calculator, the sixth input of the output device information, the second input of the threshold target size selector and the first input of the variance calculator, the first input of the primary processing unit is connected to the fourth input the house of the dispersion calculator and the first inputs of the threshold detector, integrator and target Doppler frequency meter and the second input of the first circuit And, the fourth input of the primary processing unit is connected to the second input of the integrator, the second output of the threshold target size selector is connected to the fifth input of the information output device, the output of the calculation circuit the difference is connected to the second input of the calculator, the output of the first circuit And is additionally connected to the first input of the calculator, the fifth input primary processing unit connected to the first input of the adder and a second input of the difference calculation circuit, the seventh input of the primary processing unit connected to the second input of the adder, a sixth input of the primary processing unit coupled to a fourth input of the device issuing information.

According to the proposed method, the received signals in both the sum and difference channels after amplification at an intermediate frequency are converted to a video frequency by means of quadrature-phase detection using the reference oscillations generated by the transmitter exciter with the formation of two quadrature components for each signal, then the filtering of the quadrature components of the pulse signals on the video frequency, which in the case of complex, in particular chirp signals, leads to compression of the signal in time audio, whereby at the outputs of the quadrature channels of the signal / noise power increases in the _{compression channel} N times (N _SJ - compression ratio), the obtained despread signals:

U _Σ cosφ, U _Σ sinφ and U _Δ cosφ, U _Δ sinφ,

Where:

φ is the initial phase of the received oscillations relative to the reference oscillations,

U _Σ and U _Δ are the amplitude of the signals after the matched filters in the total and difference channels, respectively, and the value of U _Δ can be either positive or negative (depending on the position of the direction to the target relative to the equal-signal direction).

The next operation is multichannel Doppler filtering of the envelope of the total signal in the frequency range of the Doppler f _D :

where F _P - the repetition frequency of the probe pulses of the radar.

The probe pulse repetition frequency F _P is selected for reasons of measuring both the maximum delay of the reflected target signal and the unambiguous measurement of the maximum width of the Doppler spectrum of the passive interference signal Δf _Дmax :

которое обеспечивает, при априорно неопределенной радиальной скорости цели, возможность определить ширину доплеровского спектра сигнала.The bandwidth of the Doppler filters δf _D is chosen to be twice as low as the minimum band Δf of the _{ODO of the} spectrum of inter-period fluctuations of the ODO to enable the correct measurement of the width of the Doppler spectrum of passive interference from discrete samples and to determine the center of gravity of the spectrum to set the initial conditions for the subsequent auto-tracking of the target signal in range. The total number of Doppler filters is equal to

which provides, with a priori uncertain radial velocity of the target, the ability to determine the width of the Doppler spectrum of the signal.

Multichannel Doppler filtering (coherent signal accumulation from fixed directions) in the entire range of working signal delays allows increasing the radar resolution in the azimuthal angle, which makes it possible to separate the target signal from interference, if their spectra do not overlap, and increase the signal / noise ratio from the coast (sea). If the spectra of the target signal and interference overlap, then an increase in the radar resolution in the angle in some cases makes it possible to detect the target signal if the effective scattering area (ESR) of the target is greater than the ESR of the resolved surface element (interference).

Show it. Let the linear azimuthal size of the resolved surface element at the analyzed range be equal to δL _az . When the angle between the velocity vector of the aircraft and the RSN axis β _{VO is} ensured by the choice of the bandwidth of the Doppler filters δf _D equal to

where: β is the angle between the axis of the aircraft and the RSN of the antenna system,

α is the drift angle of the aircraft.

With a linear azimuthal size of the PC ΔL <δL _az the signal-to-noise ratio is

Where:

σ _C - effective target area,

σ _O is the specific effective area of reflection from the Earth’s surface,

- угол между нормалью к разрешаемому элементу поверхности Земли и направлением разрешаемый элемент-РЛС,

- the angle between the normal to the resolved element of the Earth’s surface and the direction of the resolved radar element,

δR - radar resolution in range.

значение С/П>20 дБ. Этого вполне достаточно для надежной селекции полезного сигнала ПК на фоне отражений от моря (берега). При ухудшении разрешения по углу (доплеровской частоте) в 10 раз это соотношение ухудшается на 10 дБ, и вероятность правильной селекции сигнала НК на фоне отражений от моря становится недопустимо малой.It is seen that the C / P ratio increases with increasing radar resolution on the Earth's surface (decreasing δL _az and δR). When σ _C = 200 m ² , σ _O = 0.03, δR = 20 m, ΔL == 20 m and

S / N value> 20 dB. This is quite enough for reliable selection of a useful PC signal against reflections from the sea (coast). If the resolution of the angle (Doppler frequency) deteriorates by 10 times, this ratio deteriorates by 10 dB, and the probability of correct selection of the NK signal against reflections from the sea becomes unacceptably small.

After conducting multi-channel Doppler filtering, a determination is made by a given level of the spectrum width Δf _D with the most powerful spectral component from among the components that exceed the first detection threshold (P _por1 ), which is determined by the allowable probability of false detection due to noise.

Where:

Δj _D is the number of Doppler cells occupied by the target signal at the analyzed range,

j _max is the number of the Doppler filter corresponding to the maximum frequency of the signal spectrum at the analyzed range,

j _min is the number of the Doppler filter corresponding to the minimum frequency of the signal spectrum at the analyzed range,

F _P - repetition frequency of the probe pulses,

N is the number of coherently accumulated signal samples at the analyzed range.

The found spectrum width Δj _{D is} compared with a preselected second threshold value (K _por2 ) corresponding to the maximum spectral width of the signal reflected from the target for the same viewing conditions.

Where

δβ _v - azimuthal angular resolution on the Earth’s surface, corresponding to the realized frequency resolution Δf _D ,

Δf _ODO - the estimated width of the Doppler spectrum of the signal reflected from the ODO (depends only on λ),

β is the angle between the axis of the aircraft and the RSN of the antenna system,

α is the drift angle of the aircraft.

If Δj _D > K _pore2 is _exceeded , a decision is made on the presence of a false target and on the transition to the analysis of the next range element. In the presence of a powerful narrow-band spectral component, the correspondence of the received signal to the third criterion of the false target is determined: the variance of the difference between the bearing and the Doppler angle (ψ) at the analyzed distance is greater than the third threshold (D _ψ > D _pore3 ). To clarify it, we consider the differences in the signals reflected by the NK and the ODO, which are revealed during multichannel Doppler filtering and subsequent monopulse estimation of the bearing of the resolved scene element.

The hull of the ship (NK) is a rigid structure, therefore, dipoles reflecting the probing signal located within the element resolved by range and Doppler frequency have a close radial velocity relative to the aircraft, respectively, the angular azimuthal separation between the resolved elements (Δβ _VT ), determined by the Doppler the shift is equal to the angular separation (Δγ _C ), determined by monopulse bearings of the same resolved elements.

Where:

Δf _DC - Doppler frequency separation of the reflected signals for the resolved elements of the SC.

The width of the NK spectrum at the analyzed range does not depend on the NK speed, it depends only on the angular azimuthal size of the NK element permitted by the range:

Where:

Δβ _NK - the angular azimuthal size of the NK at the resolved range R,

ΔL _NK - the linear azimuthal size of the NK at the resolved range R.

The radial velocity of the NK leads to a shift in the Doppler shift of the reflected signal relative to the Doppler frequency of the signal from the same resolvable element with a fixed NK by

where: V _RNA - the projection of the speed of the aircraft on the direction of LA-NC.

From the model of reflection of the signal from the NK, it follows that for the difference between the "bearing" and the "Doppler" angles, the resolved elements of the NK at the analyzed range and determined by the expression:

ψ _i = γ _i -β _i ,

D _ψ = M {(ψ _i -M _ψ ) ² } <D _por3 ,

Where:

(A _Σ (i), A _Δ (i)) is the scalar product of the total A _Σ (i) and difference A _Δ (i) signals from the ith frequency-resolved NK element,

k1 is the scaling factor,

M _ψ is the mathematical expectation of the difference of angles,

D _ψ is the variance of the difference of angles,

γ _i - bearing of the i-th frequency-resolved NK element from RSN,

β _i - "Doppler angle" of the i-th frequency-resolved NC element,

δf _D is the bandwidth of Doppler filters (radar resolution in frequency).

In contrast to the ND, the ODL consists of many dipoles randomly located in the cloud, each of which moves in a turbulent medium with its speed and direction, respectively, the radar speed of a radar - a single dipole has a large spread relative to the average value. Accordingly, the spectrum of reflections from the ODO relative to the central frequency of the Doppler spectrum has the form [6, p.272], where only half is shown due to symmetry.

In the range-resolved volume element of the ODL, dipoles with the same radial velocities reflect a probing signal with a random amplitude and phase. The total signal from these dipoles corresponds to the reflection of the signal from the energy center, randomly located within the range of the allowed ODO range. Accordingly, its bearing is randomly located within the range of the allowed ODO volume. The angular positions of the energy centers of reflection from dipoles with other radial velocities at the current time are independent within the same volume-resolved range. From the model of signal reflection from the ODL, it follows that the geometric angular size of the ODL does not coincide with both the angular size of the ODL calculated by the width of the spectrum, and the bearings resolved in range and frequency of the ODL elements. Wherein:

ψ _i = γ _i -β _i ≠ const,

Δγ _i ≠ Δβ _i ,

Δi _D = i _max -i _min ,

Δγ _i = γ _imax -γ _imin,

D _ψ > D _por3

Where:

Δi _D is the difference in the numbers of Doppler filters corresponding to the boundaries of the Doppler spectrum of the ODO,

Δγ _i is the angular size of the ODO calculated from bearings of the resolved elements at the boundaries of the Doppler spectrum,

Δβ _i is the angular size of the ODO calculated from the difference of the Doppler frequencies at the boundaries of the Doppler spectrum,

D _ψ is the variance of the estimate of the difference in the angles ψ _i at the analyzed range.

For all goals remaining after selection according to three criteria, their range (R), Doppler signal shift (f _D ), azimuthal angle, measured from the velocity vector (β _V ) as the sum of the current RSN angle (β), counted from the longitudinal axis, are recorded LA, drift angle (α) and bearing (γ):

β _V = β + γ-α.

The invention is illustrated by a further description and drawings of the radar that implements this method:

Figure 1 - structural diagram of the device of the proposed monopulse radar,

Figure 2 - structural diagram of the pathogen (B),

Figure 3 is a structural diagram of a programmable signal generator (GPS),

Figure 4 is a structural diagram of a synchronizer (C),

5 is a structural diagram of a digital matched filter (CSF),

6 is a structural diagram of a multi-channel Doppler filter (MDF),

7 is a structural diagram of a package of Doppler filters (PDF),

Fig. 8 is a block diagram of a quadrature combining unit (BOC),

Fig.9 is a structural diagram of an angle discriminator (DU),

Figure 10 - algorithm of the secondary processing unit (BWO),

11 is a structural diagram of a threshold detector (ON),

12 is a structural diagram of a threshold target size selector (PSRC),

Figure 13 is a structural diagram of a dispersion calculator (VDO),

Fig - structural diagram of an integrator (INT),

Fig is a structural diagram of a calculator mathematical expectation (WMO),

Fig is a structural diagram of a meter of the Doppler frequency of the target (IDC),

Fig is a structural diagram of a device for issuing information (UVI).

Figure 1 presents the structural diagram of the device of the proposed monopulse radar, where the following notation:

1 - antenna drive (PA),

2 - antenna (A),

3 - total differential Converter (PSA),

4 - antenna switch (AP),

5 - power amplifier (PA),

6 - pathogen (B),

7 - synchronizer (C),

8 - pulse modulator (IM),

9 - two-channel receiver (PR),

10 - quadrature phase detector of the total channel (KFD-S),

11 - quadrature phase difference channel detector (KFD-R),

12 - phase shifter (PV),

13 - video amplifier total channel (VU-S),

14 - video amplifier differential channel (VU-R),

15 - digital matched filter of the total channel (SF-S),

16 - digital matched filter differential channel (SF-R),

17 - multi-channel Doppler filter of the total channel (MDV-S),

18 - multi-channel Doppler filter of the difference channel (MDV-R),

19 - block combining quadratures (BOK),

20 - angle discriminator (DU),

21 - divider (Affairs),

22 - primary processing unit (BPO),

23 - block secondary processing (BVI),

24 - speed and drift meter (ASC),

25 - two-channel high frequency amplifier (UHF),

26 - two-channel balanced mixer (BSM),

27 - two-channel intermediate frequency amplifier (UPCH),

28 - threshold detector (ON),

29 - the first circuit And (I1),

30 - threshold target size selector (PSRTS),

31 - variance calculator (VDO),

32 - integrator (INT),

33 is a difference calculation circuit (CBP),

34 - calculator mathematical expectation (WMO),

35 - threshold device (PU),

36 - meter Doppler frequency of the target (IDC),

37 - the second scheme And (And 2),

38- adder (Sum),

39 - information output device (UVI).

The proposed monopulse radar device of figure 1 contains a series-connected antenna switch 4, a sum-difference converter 3 and an antenna 2, a power amplifier 5, an antenna drive 1, a two-channel receiver 9, the first output of which is connected to the input through a quadrature-phase detector 10 of the total channel the video amplifier 13 of the total channel, the second output of the two-channel receiver 9 through the output of the quadrature-phase detector 11 of the differential channel is connected to the input of the video amplifier 14 of the differential channel, digital matched e filters 15 of the total and 16 difference channels, the quadrature combining unit 19, the angle discriminator 20, the primary processing unit 22 and the secondary processing unit 23, while the drive of the antenna 1 is kinematically connected to the third input of the antenna 2, the second output of the sum-difference converter 3 is connected to the second input of the two-channel receiver 9, the third output of the antenna switch 4 is connected to the first input of the two-channel receiver 9, the synchronizer 7 through the pulse modulator 8 is connected to the second input of the power amplifier 5, the second output of the exciter 6 is connected to the third input of the two-channel receiver 9, the third output of the pathogen 6 is connected to the second inputs of the quadrature phase detectors 10 of the total and 11 differential channels, the first output of the two-channel receiver 9 is connected to the first input of the quadrature phase detector 10 of the total channel, the second output of the two-channel receiver 9 is connected to the first input of the quadrature phase detector 11 of the difference channel, the second output of the secondary processing unit 23 is connected to the input of the drive of the antenna 1, the second output of which is connected to the seventh input the primary processing unit 22, the input of the quadrature combining unit 19 is connected to the first input of the angle discriminator 20, the output of the quadrature combining unit 19 is connected to the second input of the primary processing unit 22, the fourth output of the pathogen 6 is connected to the input of the synchronizer 7, the first output of the secondary processing unit 23 is connected with the fourth input of the primary processing unit 22, the first output of which is connected to the first input of the secondary processing unit 23, the two-channel receiver 9 contains in total connected in series at the two inputs-outputs o and differential channels, a two-channel high-frequency amplifier 25, a balanced mixer 26, the third (heterodyne) input of which is the third heterodyne input of the receiver 9, the first and second outputs of the high-frequency amplifier 25 are inputs, and the first and second outputs of the intermediate-frequency amplifier 27 are outputs of the total and differential channels of the receiver 9, respectively, introduced multi-channel Doppler filters total 17 and differential 18 channels, a phase shifter 12, a divider 21 and a speed and drift meter 24, while the third output the wake-up device 6 through a phase shifter 12 is connected to the third inputs of the quadrature phase detectors of the total 10 and difference 11 channels, the output of the video amplifier of the total channel 13 through a digitally matched filter of the total channel 15 is connected to the first input of a multi-channel Doppler filter of the total channel 17, the output of the video amplifier of the differential channel 14 through digitally matched the differential channel filter 16 is connected to the first input of the multi-channel Doppler filter of the differential channel 18, the output of the block combining quadrature 19 s is single with the first input of the divider 21, the output of which is connected to the fifth input of the primary processing unit 22, the first output of the pathogen 6 through the power amplifier 5 is connected to the first input of the antenna switch 4, the fourth output of the pathogen 6 is connected to the second inputs of the digital matched filters of total 15 and differential 16 channels, the second inputs of multi-channel Doppler filters total 17 and differential 18 channels and the first input of the primary processing unit 22, the first output of the synchronizer 7 is additionally connected to the excitation input Pouring 6 and the fifth inputs of the multi-channel Doppler filters of the total 17 and the differential 18 channels, the output of the multi-channel Doppler filter of the total channel 15 is connected to the input of the quadrature combining unit 19, the output of the multi-channel Doppler filter of the differential channel 18 is connected to the second input of the angle discriminator 20, the output of which is connected to the second the input of the divider 21, the second output of the synchronizer 7 is connected to the third input of the primary processing unit 22, the fourth output of the synchronizer 7 is connected to the third inputs of the multi-channel Doppler filters of total 17 and differential 18 channels and the sixth input of the secondary processing unit 23, the third output of the synchronizer 7 is connected to the fourth inputs of multi-channel Doppler filters of total 17 and differential 18 channels, the seventh input of the secondary processing unit 23 and the sixth input of the primary processing unit 22, third output the secondary processing unit 23 is connected to the second input of the pulse modulator 8, the second output of the primary processing unit 22 is connected to the fifth input of the secondary processing 23, the first and second outputs are speed and drift booster 24 are connected to the third and fourth inputs of the secondary processing unit 23, respectively, the second output of the antenna drive 1 is additionally connected to the second input of the secondary processing unit 23, the fourth output of the secondary processing unit 23 is connected to the eighth input of the primary processing unit 22, the primary processing unit 22 comprises a threshold detector 28 connected in series, a first AND circuit 29, a target size threshold selector 30, an integrator 32 connected in series, a difference calculation circuit 33, a disperser calculator these 31, the threshold device 35 and the information output device 39, the doppler frequency meter of the target 36 connected in series, the second circuit AND 37, and the information output device 39, the first and second outputs of which are the second and first outputs of the primary processing unit 22, the adder 38 connected with the second input of the information output device 39, and the second input of the primary processing unit 22 is connected to the second inputs of the threshold detector 28 and the Doppler frequency meter of the target 36, the fourth input of the mathematical calculator liquid 34 and the sixth input of the dispersion calculator 31, the output of the threshold detector 28 is connected to the first inputs of the dispersion calculator 31 and the second circuit And 37, the third input of the primary processing unit 22 is connected to the third inputs of the threshold detector 28, the integrator 32, the Doppler frequency meter of the target 36, the calculator mathematical expectation 34, the sixth input of the information output device 39, the second input of the threshold selector of the size of the target 30 and the first input of the variance calculator 31, the first input of the primary processing unit 22 is connected to the fourth the input of the dispersion calculator 31 and the first inputs of the threshold detector 28, the integrator 32 and the Doppler frequency meter of the target 36 and the second input of the first circuit And 29, the fourth input of the primary processing unit 22 is connected to the second input of the integrator 32, the output of the threshold selector of the size of the target 30 is connected to the fifth input information output devices 39, the output of the difference calculation circuit 33 is connected to the second input of the mathematical calculator 34, the output of the first circuit And 29 is additionally connected to the first input of the mathematical calculator 34, the fifth input of the primary processing unit 22 is connected to the first input of the adder 38 and the second input of the difference calculation circuit 33, the seventh input of the primary processing unit 22 is connected to the second input of the adder 38, the sixth input of the primary processing unit 22 is connected to the fourth input of the information output device 39 .

Figure 2 presents the structural diagram of the pathogen (B) 6, where the following notation:

40 - the first frequency multiplier (UCH 1),

41 - the first programmable signal generator (GPS 1),

42 - the first generator of the shifted signal (GSS 1),

43 - the second frequency multiplier (UCH 2),

44 - master oscillator (ZG),

45 - the third frequency multiplier (UCH 3),

46 - the fourth frequency multiplier (UCH 4),

47 - the second programmable signal generator (GPS 2),

48 - the second generator of the shifted signal (GSS 2),

49 - the fifth frequency multiplier (UCH 5).

In the driver circuit 6 (Fig. 2), the master oscillator 44, the first frequency multiplier 40, the first shifted signal generator 42 and the second frequency multiplier 43, the output of which is the second output of the driver 6, are connected in series, the output of the first programmable signal generator 41 is connected to the second input of the generator shifted signal 42, the input of the exciter 6 is connected to the first inputs of the first 41 and second 47 generators of the programmable signal, the output of the master oscillator 44 is additionally connected to the inputs of the fourth frequency multiplier 46, the third frequency multiplier 45, the second inputs of the first 41 and second 47 programmable signal generators and is the fourth output of the exciter 6, the output of the second programmable signal generator 47 through the second shifted signal generator 48 connected in series and the fifth frequency multiplier 49 is connected to the first output of the exciter 6, the fourth the frequency multiplier 46 is connected to the first input of the second generator of the shifted signal 48, the output of the third frequency multiplier 45 is the third output of the pathogen 6.

Figure 3 presents the structural diagrams of the programmable signal generators (GPS) 41 and 47, which are similar. In figure 3, the following notation:

50 - the first counter (SI 1),

51 - the first read-only memory (ROM 1),

52 - digital-to-analog converter (DAC),

53 - the first low-pass filter (low-pass filter).

In the diagram of the programmable signal generator 41 (Fig. 3), the first and second inputs of the first counter 50 are the first and second input of the programmable signal generator 41, respectively, the output of the first counter 50 through the first ROM 51 connected in series, the digital-to-analog converter 52, and the low-pass filter 53 are connected to the output of the programmable signal generator 41.

Figure 4 presents the structural diagram of the synchronizer (C) 7, where the following notation:

54 - the first code bus (KSh 1),

55 - the second counter (SI 2),

56 - the first trigger (TG 1),

57 - the second code bus (KSh 2),

58 - the third counter (SI 3),

59 - the fourth counter (SI 4),

60 - the second trigger (TG 2),

61 is the fifth counter (SI 5),

62 - the third code bus (KSh 3),

63 - the fourth code bus (KSh 4),

64 - decoder (DS).

In the synchronizer circuit 7 (Fig. 4), the input of the synchronizer is connected to the first inputs of the second counter 55, the third counter 58 and the fifth counter 61, the output of the second counter 55 through the first trigger 56 is connected to the second input of the third counter 58, the output of the third counter 58 is connected to the second the input of the first trigger 56, the second output of the first trigger 56 is connected to the second and first inputs of the second 55 and fourth 59 counters, respectively, and is the first output of the synchronizer 7, the outputs of the first code bus 54, second code bus 57, third code bus 62 and the fourth code bus 63 is connected to the third inputs of the second 55, third 58, fourth 59 and fifth 61 counters, respectively, the output of the fourth counter 59 is connected to the first input of the second trigger 60, the first output of which is connected to the second input of the fifth counter 61, the second output of the fifth counter 61 connected to the input of the decoder 64 and is the third output of the synchronizer 7, the first output of the fifth counter 61 is connected to the second input of the second trigger 60, the second output of which is connected to the second input of the fourth counter 59, the output of the decoder 64 is is the second output of the synchronizer 7.

Figure 5 presents the structural diagram of a digital matched filter (CSF) 15 (16), where the following notation:

65 - the first analog-to-digital Converter (ADC 1),

66 - the first shift register (SDR 1),

67 - the first weight multiplier (VUM 1),

68 - the second adder (Sum 2),

69 - the third adder (Sum 3),

70 - the second weight multiplier (Vum 2),

71 - the second analog-to-digital multiplier (ADC 2),

72 - the second shift register (SDR 2).

In the diagram (Fig. 5), the quadrature cosine component of the signal supplied to the first input of the digital matched filter 15 is connected to the first input of the first ADC 65, and the sine to the first input of the second ADC 71, the second input of the digital matched filter 15 is connected to the second inputs of the first 65 and second 71 ADCs, the first 66 and second 72 shift registers, the output of the first ADC 65 is connected to the first (information) input of the first shift register 66, the outputs of which are from the first to the Qth through the first weight amplifiers from the first to the Qth (pos. . 67 ₁ ... 67 _Q) under lyucheny eponymous to inputs of the second adder 68, the output of the second ADC 71 is connected to first (information) input of the second shift register 72, which outputs the first through Q-th through the second weighting amplifier of the first through Q-th (pos. ₁ ... 70 70 _Q ) are connected to the inputs of the same name of the third adder 69, the outputs of the second 68 and the third adder 69 are quadrature components of the output of the digital matched filter 15.

Figure 6 presents the structural diagram of a multi-channel Doppler filter (MDF) 17 (18), where the following notation:

73 - the third shift register (SDR 3),

74 _k - k-th block of Doppler filters (BDF k),

75 _k - the first register (WP 1k),

76 _k - the first packet of the k-th Doppler filter (PDF 1k),

77 - multiplexer.

According to the scheme of Fig.6, the second input of the third shift register 73 is the second input of the multi-channel Doppler filter 17, the first input of which is connected to the second inputs of the blocks of Doppler filters from the first (item 74 ₁ ) to the m-th (item 74 _m ), the third input a multi-channel Doppler filter 17 is connected to the third inputs of the blocks of Doppler filters from the first (item 74 ₁ ) to the m-th (item 74 _m ), the fifth input of the multi-channel Doppler filter 17 is connected to the first input of the third shift register 73 and the fourth inputs of the blocks of Doppler filters from the first (pos. 74 ₁ ) to the mth (pos. 74 _m ), the outputs of the third shift register 73 from the first to the mth are connected to the first inputs of the Doppler filter blocks from the first to the mth (pos. 74 ₁ to 74 _m ), respectively, N outputs of each of the blocks of Doppler filters from the first to the mth (pos. 74 ₁ to 74 _m ) are connected to the inputs from the second to (m · N + 1) th multiplexer 77, and the input number of the multiplexer 77 for the i-th output of the k-th block of Doppler filters (where i = 1 ... N, k = 1 ... m) is (k-1) · N + 1 + i, the first input of multiplexer 77 is the fourth input multi-channel Doppler filter 17, output d multiplexer 77 is the output of the multi-channel Doppler filter 17, a first input of each of m Doppler filter units (74 ₁ ... 74 _m) is coupled to a first input of the first register 75 _k, the output of which is connected to the input of the Doppler filter package 76 _k, a second input block Doppler filters 74p are connected to the second input of the first register 75 _k , the third input of the block of Doppler filters 74 _{k is} connected to the second input of the packet of Doppler filters 76 _k , the fourth input of the block of Doppler filters 74k is connected to the third input of the packet of Doppler filters liters 76 _k , the outputs of the packet of Doppler filters 76 _k from the first to the Nth are the outputs of the block of Doppler filters 74 _k from the first to the Nth, respectively.

Figure 7 presents the structural diagram of a package of Doppler filters (PDF) 76, where the following notation:

78 - the third trigger (TG 3),

79 - the third scheme And (And 3),

80 - the fourth scheme And (And 4),

81 - sixth counter (SI 6),

82 _i - i-th Doppler filter (DF _i ),

83 _i - second read-only memory (ROM 2 _i ),

84 _i - the first multiplier (Smart 1 _i ),

85 _i - the second multiplier (Smart 2 _i ),

86 _i - the third multiplier (Smart 3 _i ),

87 _i - the fourth multiplier (Smart 4 _i ),

88 _i - the second scheme for calculating the difference (SVR 2 _i ),

89 _i - the fourth adder (Sum 4 _i ),

90 _i - second integrator (Int 2 _i ),

91 _i is the third integrator (Int 3 _i ).

According to the diagram of Fig. 7, the second input of the packet of Doppler filters 76 is connected to the first inputs of the third trigger 78, the third 79 and the fourth 80 of circuit I, the third input of the packet of Doppler filters 76 is connected to the second inputs of the third trigger 78 and the fourth circuit And 80, the output of the third trigger 78 connected to the second input of the third And 79 circuit, the output of which is connected to the second input of the sixth counter 81 and the second inputs of the Doppler filters from the first to the Nth (pos. 82 ₁ ... 82 _N ), the output of the fourth And 80 circuit is connected to the first input sixth counter 81 and the first inputs d first-to-Nth Opler filters (pos. 82 ₁ ... 82 _N ), the output of the sixth counter 81 is connected to the third inputs to the first to N-Doppler filters (pos. 82 ₁ ... 82 _N ), the first input the packet of Doppler filters 76 is connected to the fourth inputs of the Doppler filters from the first to the Nth (pos. 82 ₁ ... 82 _N ), each Doppler filter 82; contains the second ROM 83 _i , multipliers from the first 84 _i to the fourth 87 _i , the second difference calculation circuit 88 _i , the fourth adder 89 _i , the second 90 _i and the third integrator 91 _i , while the cosine components of the filtered signal from the fourth input of the Doppler filter 82 _i connected to the first inputs of the first 84 _i and third 86 _i multipliers, the sine components of the filtered signal from the fourth input of the Doppler filter 82 _{i are} connected to the first inputs of the second 85 _i and fourth 87 _i multipliers, the second inputs of the first 84 _i and fourth 87 _i multipliers the sine output of the second ROM 83 _i , the cosine output of which is connected to the second inputs of the second 85 _i and the third 86 _i multipliers, the outputs of the first 84 _i and the second 85 _i multipliers are connected to the first and second input of the second difference calculation circuit 88 _i , the output of which is connected to the second the input of the second integrator 90 _i , the outputs of the third 86 _i and the fourth 87 _i multipliers are connected to the first and second input of the fourth adder 89 _i , the output of which is connected to the second input of the third integrator 91 _i , the first input of the Doppler filter 82 _{i is} connected to the first inputs and the second 90 _i and the third integrators 91 _i , the second input of the Doppler filter 82 _{i is} connected to the third inputs of the second 90 _i and the third integrators 91 _i , the outputs of the second 90 _i and the third 91 _i integrators are quadrature outputs of the Doppler filter 82 _i , the outputs of N Doppler filters (pos. 82 ₁ ... 82 _N ) are the outputs of the packet of Doppler filters 76 from the first to the Nth.

On Fig presents a structural diagram of a block combining quadratures (19), where the following notation:

92 - the fifth multiplier (Smart 5),

93 - the sixth multiplier (Smart 6),

94 - the fifth adder (Sum 5).

In Fig. 8, the cosine quadrature component of the input signal of the quadrature combining unit 19 is connected to the first and second inputs of the sixth multiplier 93, the output of which is connected to the first input of the fifth adder 94, the sine component of the quadrature input signal of the quad combining unit 19 is connected to the first and second inputs of the fifth multiplier 92, the output of which is connected to the second input of the fifth adder 94, the output of the fifth adder 94 is the output of the quadrature combining unit 19.

Figure 9 presents the structural diagram of the angle discriminator (DU) 20, where the following notation:

95 - the seventh multiplier (Smart 7),

96 - the eighth multiplier (Smart 8),

97 - the sixth adder (Sum 6).

In the diagram of Fig. 9, the sine quadrature component from the first input of the angle discriminator 20 is connected to the first input of the seventh multiplier 95, the output of which is connected to the second input of the sixth adder 97, the cosine quadrature component from the first input of the angle discriminator 20 is connected to the first input of the eighth multiplier 96, the output which is connected to the first input of the sixth adder 97, the sine quadrature component from the second input of the angle discriminator 20 is connected to the second input of the seventh multiplier 95, the cosine quadrature component Ascending from the second input of the angle discriminator 20 is connected to the second input of the eighth multiplier 96, the output of the sixth adder 97 is the output of the angle discriminator 20.

Figure 10 shows the algorithm of the secondary processing unit (BWO) 23, where the following notation:

PC1 - the first permission to retrieve information,

PC2 - the second permission to retrieve information.

Figure 11 presents the structural diagram of the threshold detector (28), where the following notation:

98 - the third shift register (SDR 3),

99 _k - the third weight multiplier (VUM 3 _k ),

100 - fifth code bus (KSh 5),

101 - seventh adder (Sum 7),

102 - the first comparator (Comp 1).

11, the first, second and third inputs of the threshold detector (28) are connected to the second, first and third inputs of the third shift register 98, respectively, the outputs of the shift register 98 from the first to the qth through third weight multipliers (pos. 99 ₁ .. .99 _q ) are connected to the inputs from the first to the qth seventh adder 101, respectively, the output of the seventh adder 101 is connected to the first input of the first comparator 102, the output of the fifth code bus 100 is connected to the second input of the first comparator 102, the output of which is the output of the threshold detector 28 .

On Fig presents a structural diagram of a threshold selector size target (PSRTS) 30, where the following notation:

103 - seventh counter (SI 7),

104 - second comparator (Comp 2),

105 - the fourth trigger (TG 4).

In the diagram of Fig. 12, the first input of the threshold target size selector 30 is connected to the first (counting) input of the seventh counter 103, the output of which is connected to the first input of the second comparator 104, the third input of the threshold size selector of the target 30 is connected to the second input of the second comparator 104, the output of which connected to the second input of the fourth trigger 105, the second input of the threshold selector size target 30 is connected to the second input of the seventh counter 103 and the first input of the fourth trigger 105, the output of which is the output of the threshold selector size and 30.

On Fig presents a structural diagram of a calculator of dispersion (VDO) 31, where the following notation:

106 - the fourth shift register (SDR 4),

107 is a third difference calculation scheme (CBP 3),

108 is a diagram of the calculation of the square (KB),

109 - fifth shift register (SDR 5),

110 - the fifth scheme And (And 5),

111 - the ninth multiplier (Smart 9),

112 - fourth integrator (Int 4),

113 - the second divider (Div. 2).

13, the fourth input of the dispersion calculator 31 is connected to the second inputs of the fourth 106 and fifth 109 shift registers and the fifth AND circuit 110, the first input of the dispersion calculator 31 is connected to the third inputs of the fourth 106 and fifth 109 shift registers and the fourth integrator 112, the output of which is connected with the first input of the second divider 113, the second input of the dispersion calculator 31 is connected to the first input of the fourth shift register 106, the output of which is connected to the first input of the third difference calculation circuit 107, the fifth input of the dispersion calculator and 31 is connected to the second input of the third difference calculation circuit 107, the output of which is connected to the input of the square calculation circuit 108, the output of which is connected to the second input of the ninth multiplier 111, the sixth input of the dispersion calculator 31 is connected to the first input of the ninth multiplier 111, the output of which is connected to the second the input of the fourth integrator 112, the third input of the dispersion calculator 31 is connected to the first input of the fifth shift register 109, the output of which is connected to the first input of the fifth circuit And 110, the output of the fifth circuit And 110 is connected to the first input of even ertogo integrator 112, the seventh input variance calculator 31 is connected to a second input of the second divider 113, whose output is the output of the calculator dispersion.

On Fig presents a structural diagram of the integrator (Int) 32, (90, 91) where the following notation:

114 - the eighth adder (Sum 8),

115 - the second register (WP 2).

In the diagram of Fig. 14, the second input of the integrator 32 is connected to the first input of the eighth adder 114, the output of which is connected to the first input of the second register 115, the output of the second register 115 is connected to the second input of the eighth adder 114 and is the output of the integrator 32, the second and third inputs of the second register 115 are the third and first input of integrator 32, respectively.

On Fig presents a structural diagram of a calculator mathematical expectation (WMO) 34, where the following notation:

116 - fifth integrator (Int 5),

117 - the third register (WP 3),

118 - the third divider (Del 3),

119 - the tenth multiplier (Smart 10),

120 - sixth integrator (Int 6),

121 - the fourth register (WP 4).

On Fig the third input of the math calculator 34 is connected to the third inputs of the fifth 116 and the sixth 120 integrators, the second inputs of the third 117 and the fourth 121 registers, the first input of the math calculator 34 is connected to the first inputs of the fifth 116 and the sixth 120 integrators, the second input of the calculator mathematical expectation 34 through series-connected tenth multiplier 119, fifth integrator 116, third register 117 and third divider 118 connected to the first output of the calculator 34, the output of the sixth in egratora 120 through the fourth register 121 is coupled to a second input of the third divider 118 and a second output expectation calculator 34, the second inputs of the tenth multiplier 119 and sixth integrator 120 are coupled to a fourth input of the calculator 34, expectation.

On Fig presents a structural diagram of a meter of the Doppler frequency of the target (IDC) 36, where the following notation:

122 - sixth shift register (SDR 6),

123 - the fourth weight multiplier (VUM 4),

124 - ninth adder (Sum 9),

125 - zero detector (DO).

In Fig.16, the first, second and third inputs of the sixth shift register 122 are the second, first, and third input of the Doppler frequency meter of the target 36, respectively, the outputs of the sixth shift register 122 from the first to the qth through q of the fourth weight multipliers 123 ₁ ... l23 _{q is} connected to the inputs of the ninth adder 124 with the same name, the output of which through the zero detector 125 is connected to the output of the target Doppler frequency meter 36.

On Fig presents a structural diagram of a device for issuing information (UVI) 39, where the following notation:

126 - fifth register (WP 5),

127 - sixth register (WP 6),

128 - seventh register (WP 7),

129 - eighth register (WP 8),

130 - ninth register (WP 9),

131 - tenth register (WP 10),

132 - OR circuit,

133 - eleventh register (WP 11),

134 - the sixth scheme And (And 6),

135 - Shaper permission information retrieval (Fed).

In the diagram of FIG. 17, the first input of the information output device 39 is connected to the second inputs of the fifth 126, sixth 127 and seventh 128 registers, the second input of the information output device 38 is connected to the first input of the fifth register 126, the output of which is connected to the first input of the eighth register 129, code the address of the interrogated cell (i, j) is input into the information output device 39 through the fourth input, and the line code i from the fourth input of the information output device 39 is connected to the first input of the sixth register 127, the output of which is connected to the first input of the ninth register 130, the column code j from the fourth input of the information output device 39 is connected to the first input of the seventh register 128, the output of which is connected to the first input of the tenth register 131, the third and fifth inputs of the information output device 39 are connected to the first and second inputs of the OR circuit 132, output which through serially connected the eleventh register 133 and the sixth circuit And 134 is connected to the input of the imaging permissions former information 135, the second inputs of the eleventh register 133 and the sixth circuit And 134 are connected to the sixth input of the issuing device Info 39, output driver authorization information reading 135 is coupled to second inputs of the eighth 129, ninth 130 and tenth 131 registers and a second output device issuing information 39, the outputs of the eighth 129, ninth 130 and tenth 131 registers jointly represent the first output of the device issuing information 39.

Antenna 2 can be constructed in the form of a mirror with a 2-horn feed connected by waveguides to a sum-difference converter 3.

Sum-difference Converter 3 can be built on the basis of waveguide T-bridges.

Antenna switch 4 can be made in the form of a three-arm ferrite Y-circulator.

The power amplifier 5 can be implemented depending on the required power and the band of amplified frequencies based on the amplitron, klystron, traveling wave lamp or semiconductor device [11, p.19, 20, 25, 42 ... 47].

The pulse modulator 8, depending on the parameters of the power amplifier 5, is implemented according to well-known schemes [11, p. 103 ... 107, Fig. 43 ... 45].

The secondary processing unit 23 can be built on the basis of a digital processor, the operation of which consists in sequentially solving the tasks presented on the block diagram of the algorithm (Fig. 10).

The Doppler speed and drift angle meter 24 can be a speed and drift angle meter [10, p. 368 ... 370, Fig. 11.16].

A two-channel high-frequency amplifier 25 in a two-channel receiver 9 can be implemented as a transistor two-channel microwave amplifier

A two-channel balanced mixer 26 in a two-channel receiver 9 can be performed according to the scheme [11, p. 144].

The threshold detector 28 can be performed according to a scheme similar to [9, p.181 Fig 2.36].

The Doppler frequency meter of target 36 can be performed according to a scheme similar to [9, p. 255 Fig. 3.17].

In accordance with the diagrams of FIGS. 1-17, a radar that implements the proposed method works as follows.

Before starting work in the secondary processing unit 23 of the radar, the operation algorithm of which is shown in Fig. 10, data from the control system β _n and the end β _to the scanning sector are input from the control system, then information about the initial position of the scanning sector β _n from the secondary information unit 23 is entered to the antenna drive 1. The antenna drive 1 from this moment rotates the antenna 2 to a predetermined position and gives to the secondary processing unit 23 data on the current angle between the rotatable axis of the PCN antenna 2 and the axis of the aircraft. The secondary processing unit 23 compares this information with a predetermined initial position. When the current angle β coincides with the given β _n, the secondary processing unit 23 turns on the radiation by applying the corresponding command to the second input of the pulse modulator 8. The microwave signals generated by the exciter 6 at a frequency F _c pass into the power amplifier 5, in which they are amplified, and under the action pulse modulator 8, controlled by pulses from the synchronizer 7, following with a repetition frequency F _P , probing pulses are formed. They pass the antenna switch 4, the sum-difference converter 3 and are emitted by the antenna 2 into a given space. The reflected signals are received by antenna 2, pass to the sum-difference converter 3, where the sum and the difference signals are formed at the first output. The total signal from the first output of the sum-difference converter 3 through the antenna switch 4 is fed to the first (total) input of the two-channel high-frequency amplifier 25. At the same time, the difference signals from the second output of the sum-difference converter 3 are fed to the second (difference) input of the high-frequency amplifier 25. The sum and difference signals are amplified in a two-channel high-frequency amplifier 25 and get from a two-channel balanced mixer 26. In the mixer 26, superheterodyne conversion occurs signals to the intermediate frequency, and the oscillations of the microwave frequency F _r generated in the exciter 6 and coming from the second output of the exciter 6 to the third (heterodyne) input of the two-channel balanced mixer 26 are used as heterodyne oscillations.

Pathogen 6 (figure 2) works as follows. The master oscillator 44 generates continuous oscillations of a stable frequency F _O , from which, by multiplying in three frequency multipliers 40, 45 and 46, frequencies F1, F2 and F3 equal to

F1 = n1 · F _O - frequency at the output of the first multiplier 40,

F2 = n2 · F _O is the frequency at the output of the fourth multiplier 46,

F3 = (n1-n2) · n3 · F _O = F _ol - the frequency at the output of the third multiplier 45.

Oscillations with a frequency of F1 are shifted in frequency by the amount of frequency generated by the first programmable generator (GPS1) 41 using the first shifted signal generator (GCC1) 42. Next, the frequency from the output of the shifted signal generator 42 is multiplied n3 times in the second frequency multiplier 43 and becomes equal to heterodyne F _g :

F _g = n3 · F _GCCl = n3 · (F1 + F _GPS1 ) = n3 · (n1 · F _O + f _cm ),

Where

f _cm - the offset frequency is selected as possible, based on the capabilities of the programmable signal generator 41 (47) to generate a signal in a given frequency band,

F _GCCl - the frequency at the output of the first shear generator 42,

F _GPS1 - the frequency at the output of the first programmable generator 41,

n3 is the frequency multiplier in the second multiplier 43.

Similarly, the carrier frequency F _{c is formed} by shifting the frequency F2 by the frequency generated by the second programmable generator (GPS2) 47 and then multiplying it in the multiplier 49 by a factor n3 equal to the multiplication factor in the second multiplier 43. Unlike GPS1 41, the GPS2 generator 47 generates a chirp signal with a frequency F _GPS2 = f _cm + k _O · t, therefore, the carrier frequency is:

F _c = n · F _GCC2 = n3 · (F2 + F _GPS2 ) = n3 · (F2 + f _cm + k _O · t) = n3 · (n2 · F _O + f _cm + k _O · t).

The intermediate frequency when mixing the reflected signal with the local oscillator will be equal to

F _CR = F _c -F _G = (n2-n1) · n3 · F _O

and corresponds to the frequency at the output of the third frequency multiplier.

The frequency of the master oscillator 44 F _O without multiplication enters the fourth output of the pathogen 6 and is used to chronize all signal processing processes.

The programmable signal generator 41 (47) can be made according to the scheme of FIG. 3. A timing frequency F _O arrives at its second input, a modulating pulse from the first output of the synchronizer 7 comes to the first input. The period and duration of the modulating pulse correspond to the probing one. When the modulating pulse arrives at the first input of the first counter 50, the ban on counting the timing pulses arriving at its second input is lifted. From this moment on, the code of the counter 50 changes linearly and goes to the address input of the first read-only memory (ROM) 51, in the memory of which, with a step of a period of timing pulses, digitized samples of the signal with a frequency of either F _GPS1 or frequency F _{GPS2 are recorded} depending on the version. The digitized samples of the output signal at the output of the first ROM 51 are converted into analog form using digital-to-analog conversion 52, and after filtering the high-frequency components, the analog signal at the output of the low-pass filter 53 is the output of the programmable frequency generator 41 (47). The timing frequency is selected on the basis of providing 4 · F _GPS <F _O (conditions for the correct restoration of the signal from its discrete samples and the possibility of suppressing undesirable harmonics of the frequency F _O in the analog signal F _GPS using filtering.

Signal shift generators 42 and 48 can be implemented using a phase-locked loop [7, p.190 fig. 4.10, 4.11]. A similar variant of obtaining a signal shifted in frequency provides an output signal with the best spectral characteristics in terms of amplitude and phase noise.

After the superheterodyne signal conversion in the two-channel balanced mixer 26, the signals in the total and difference channels are amplified by the intermediate frequency in the two-channel amplifier of the intermediate frequency 27, then they are fed to the first (signal) inputs of the corresponding quadrature phase detectors 10 in the total channel and 11 in the difference channel.

The second inputs (reference frequency inputs) of the quadrature phase detectors 10 and 11 receive reference oscillations from the third output of the exciter 6. The third inputs of the quadrature phase detectors 10 and 11 receive the reference frequency from the third output of the exciter 6 through a 90-degree phase shifter 12. After the quadrature -phase detection, video pulses are formed, which, through the corresponding video amplifiers of the total 13 and difference channels 14, are fed to matched digital filters 15 and 16. The bandwidth of the video amplifiers 13 and 14 is equal to the width probing signal spectrum. Digital matched filters 15 and 16 are similar and can be performed according to the scheme of figure 5. The cosine component of the input quadrature signal of the digital matched filter 15 (16) arrives at the first input of the first analog-to-digital converter (ADC 1) 65 and is digitized with a period of timing pulses To arriving at its second input. Similarly, the sine component of the input signal passes to the first input of the second ADC 71 and is digitized with the same period. The output signals of the first 65 and second 71 ADCs are fed to two quadrature compression channels constructed in a similar manner. The compression channel of the cosine component consists of the first shift register 66, the first weight multipliers 67 ₁ ... 67 _Q and the second adder 68. The compression channel of the sine component is constructed similarly with the difference in the values of the multiplication coefficients of the weight multipliers 70 ₁ ... 70 _Q ) from the weight multiplier coefficients 67 ₁ ... 67 _Q. In the cosine quadrature channel, the reference function of the chirp signal is written in the form of coefficients

Where:

- номер выборки сигнала, взятого с периодом хронирующих импульсов T_O=1/F_O,

- the sample number of the signal taken with the period of the timing pulses T _O = 1 / F _O ,

Q - the length of the reference signal (corresponds to the length of the probing), expressed in the number of samples with a period T _O ,

k _O and n3 are the coefficients used in the formation of the carrier frequency F _C ,

p is a previously known constant.

In the sinus quadrature channel, the support function in the form of weighting coefficients of weighting factors 70 ₁ ... 70 _Q is - sin (2π · k _O · n3 · q ² · T ² _O ) = cos (p · q) ² . The quadrature components of the compressed signal are removed from the outputs of the second 68 and third 69 adders and represent the output signal of the digital matched filter 15 (16). The sweeps of the time-compressed signals of the total and difference channels from the matched digital filters 15 and 16 enter the multichannel Doppler filters of the total 17 and difference 18 channels, respectively.

Multichannel Doppler filters 17 and 18 are similar and can be performed according to the scheme of Fig.6, the work of each of them occurs in the following sequence.

Значения найденных спектральных квадратур последовательно в соответствии с кодом опрашиваемой разрешаемой ячейки частота-дальность, приходящем с третьего выхода синхронизатора 7 на первый вход мультиплексора 77, подаются на выход многоканального доплеровского фильтра 17 (18).The signal modulating the period and duration of the probe signal T _P from the first output of the synchronizer 7 is fed to the first (information) input of the third shift register 73, the shift pulses with the sampling frequency of the signal F _O are fed to the second input of the third shift register 73. The information signal passing through the shift register 73, delayed and removed from m outputs. The value of the signal delay relative to the input at the kth output is k · T _O. Thus, at the m outputs of the third shift register 73, we have pulses used to synchronize samples from the scan of the input compressed signal at m ranges. From the first output of the third shift register 73, the signal is supplied to the first (control) input of the register 1 _l (pos. 75 _l ) of the first block of Doppler filters 74]. Using the signal received at the control input of the register 1 _l (pos. 75 _l ), a quadrature signal is sampled from the first range from the time-sweep signal received at the first input of the multi-channel Doppler filter 17 (18). The contents of the register 1 _l (pos. 75 _l ) is updated in each period of the probe pulses and fed to the package of Doppler filters 76 _l , in which multi-frequency Doppler filtering of the signal arriving from the first analyzed range is performed. Similarly, multi-frequency Doppler filtering is performed at the remaining (m-1) ranges in Doppler filtering units 74 ₂ ... 74 _m . All packages of Doppler filters 76 _{k are} similar and provide quadrature components of the spectrum at N frequencies with a uniform step equal to F _P / N. The bandwidth of each packet filter is

The values of the found spectral quadrature sequentially in accordance with the code of the interrogated resolved frequency-range cell coming from the third output of the synchronizer 7 to the first input of the multiplexer 77 are fed to the output of the multi-channel Doppler filter 17 (18).

The structural diagram of a variant of constructing a package of Doppler filters 76 is shown in Fig.7. The operation of the package of Doppler filters 76 is as follows. A sample of the digitized compressed signal from the analyzed range comes from the first input of the packet of Doppler filters 76 to N Doppler filters 82 ₁ ... 82 _N (due to the identity of the Doppler filter circuits in Fig. 7, the 1st and the i-th ones are shown, where i = 1. ..N), the information inputs of which are the first inputs of the first 84 ₁ ... 84 _N , the second 85 ₁ ... 85 _N , the third 86 ₁ ... 86 _N , and the fourth multipliers 87 ₁ ... 87 _N. The repetition period of the probe pulses T _P = 1 / F _P (clock cycle of the signal at the analyzed range) is selected from the condition of providing an unambiguous determination of both the maximum delay of the reflected signal of the target R _max and the maximum width of the Doppler spectrum of the signal Δf _Дmax :

Each of the filters 82 _i consists of a reference frequency quadrature generator executed on a second read-only memory (ROM 2 _i ) pos. 83 _i , reference quadrature multipliers with squared digitized compressed signal at the analyzed range Smart 1 _i , Smart 2 _i , Smart 3 _i and Smart 4 _i (pos. 84 _i , 85 _i , 86 _i and 87 _i ), the second scheme for calculating the difference 88 _i , the fourth adder 89 _i , the integrators Int 2 _i (pos. 90 _i ) and Int 3 _i (pos. 91 _i ). The outputs of the integrators Int 2 _i (pos. 90 _i ) and Int 3 _i (pos. 91 _i ) are the quadrature outputs of the Doppler filter 82 _i , issued at the i-th output of the packet of Doppler filters 76. The signal accumulates during T _kg = N · T _P , where N is the number of accumulated samples of the signal, taken into account by flashing the second ROM 83 _i . In this case, the values of quadratures (cosine C (i) and sine S (i) components at a frequency f _i ) of the signal filtered in the 1 / T _kg band are equal to:

Where:

Ac (q) and As (q) are the values of the cosine and sine quadratures of the input signal in the qth sample received at the first inputs of the multipliers 84 _i , 85 _i , 86 _i and 87 _i ,

- номер выборки сигнала.

- signal sample number.

The time base of the quadrature reference signal with a frequency f _i = i / T _kg ; recorded in the ROM 83 _i , is provided by a linearly increasing code of the sixth counter 81, received at the input of the ROM 83 _i . The sixth counter 81, with a zero value set by a pulse arriving at its second input, counts the number of emitted pulses with a period T _P arriving at its first input from the fourth AND 80 circuit during the signal accumulation time (a single value of the PC1 signal at the second input of the Doppler packet filters 76 coming from the fourth output of the synchronizer 7). The output pulses of the fourth circuit And 80, in addition, are fed to the first inputs of the second 90 _i and third 91 _i integrators, synchronizing the reception of input information in them. The input pulses of the third circuit And 79 are the positive pulse PC1 and it is also delayed by the period T _P from the output of the third trigger 78. The third circuit And 79 forms the pulse of the installation of the sixth counter 81 and integrators Int 2 _i (pos. 90 _i ) and Int 3 _i (pos. 91 _i ) to the starting position. The pulse duration from the output of the third circuit And 79 is equal to T _P and precedes the beginning of the next accumulation interval (PC1 = 1), due to this, the filtering result during the polling interval, when PC1 = 0, remains until its end.

Integrator circuits 32, 90 and 91 are similar and are shown in Fig. 14. An integrable input signal arrives at the first input of the eighth adder 114. At its second input, an integrator output 32 is received from the output of the second register 115. The summation result from the output of the eighth adder 114 is recorded in the second register 115 with the input signal clock (pulses arriving at third input of the second register 115). The integrator is reset to zero before the start of its operation by a pulse arriving at the second input of the third register 115.

For the case of the integrator working in the Doppler filter, the information acquisition cycle is equal to the probe pulse period T _P , and its zeroing is performed by the positive pulse front PC1 = 0 (corresponds to the end of the survey of allowed cells in the scene and the beginning of coherent accumulation) generated at the second output of the synchronizer 7.

The outputs of multichannel Doppler filters total 17 and differential 18 channels are connected to the angle discriminator 20 (Fig.9), which calculates the scalar product of the input signals:

W (i, j) = C _Σ (i, j) -C _Δ (i, j) + S _Σ (i, j) -S _Δ (i, j),

Where:

i and j are the numbers of resolved elements in frequency and range, respectively.

S _Δ (i, j) is the sine quadrature component at the output of the multi-channel Doppler filter of the difference channel for (i, j) the resolved scene element,

S _Σ (i, j) is the sine quadrature component at the output of the multi-channel Doppler filter of the total channel for (i, j) the resolved scene element,

C _Δ (i, j) is the cosine quadrature component at the output of the multi-channel Doppler filter of the difference channel for (i, j) the resolved element of the scene,

C _Σ (i, j) is the cosine quadrature component at the output of the multi-channel Doppler filter of the total channel for (i, j) the resolved scene element.

The output of the quadratures of the resolved scene elements of the multi-channel Doppler filter of the total channel is also supplied to the quadrature combining unit 19 (Fig. 8), which calculates the power of the total signal for each scene element according to the formula:

P (i, j) = | A _Σ (i, j) | ² = (C _Σ (i, j)) ² + (C _Δ (i, j)) ²

The result of the calculation of P (i, j) is further used to detect the signal and obtain the bearing of the resolved scene element γ (i, j) - the deviation of the angular position of the resolved (i, j) scene element from the RSN. The bearing is calculated in divider 21 and is equal to

where k1 is the scaling factor.

The outputs of the divider 21 and the block combining quadrature 19 are received in the primary processing unit 22, the operation of which occurs in the following sequence.

When the accumulation of the signal ends, the PC1 signal at the output of the synchronizer 7 goes to a logical zero and goes to the secondary processing unit 23, preparing it to receive information about the measured coordinates of the targets from the primary processing unit 22, as well as to the third inputs of the multi-channel Doppler filters of channels of the sum 17 and difference 18, stopping the coherent accumulation of signals. During the time when PC1 = 0, the values of the measured total and difference signals in the resolved frequency-time cells of the multi-channel filters of the sum and difference 17 and 18 are interrogated. The code of the interrogated cells is generated by the synchronizer 7 on the third output during the time when PC1 = 0. At the end of the survey, the trailing edge of the signal PC1 = 0 resets the information measured during the accumulation time on the resolved time-frequency scene elements in the multi-channel Doppler filters 17 and 18. The code of the time-frequency cells being polled changes from zero up line by line, starting from the cell polling in frequency with a minimum number at the minimum operating range. The transition to a neighboring range is made after polling the frequency cell with the maximum number. At the end of the survey of each frequency line, the synchronizer 7 generates an impulse T _p supplied to the primary processing unit 22 to zero the results of the analysis of the signals on the line.

The first stage of selection of the signals analyzed in the primary processing unit 22 — the detection of signals when the first threshold is exceeded, determined by the calculated signal-to-noise ratio, is performed by the threshold detector 28, the second input of which from the quadrature combining unit 19 receives a frequency line scan of the signal in the total channel on the current channel range. The rate of information reception is synchronized by the synchronization pulses with a frequency F _O received at the first input of the BPO 22 from the fourth output of the pathogen 6. A variant of the threshold detector 28 is shown in Fig. 11, the operation of which is described in [9, p.181, Fig 2.36] and occurs in the following sequence.

The frequency line scan of the signal from the output of the BOC 19 comes to the first (information) input of the third shift register 98 and is recorded with a clock pulse T _O arriving at its second (shift) input in a sliding window with a length of q elements. The outputs of the shift register 98 through the weight amplifiers 99 ₁ ... 99 _{q are} connected to the seventh adder 101, the output of which is supplied to the first comparator 102, the second input of which from the fifth code bus 100 is supplied with the detection threshold value. When the detection threshold is exceeded, a logical unit appears at the output of the first comparator 102. The number of bits q connected to the seventh adder 101 is determined by the width of the spectrum of the true target, the weighting coefficients of the weight multipliers 99 ₁ ... 99 _q describe the expected spectrum of the useful signal. At the end of the line before receiving the next, the contents of the third register 98 are reset to zero by the pulse T _p coming to its third input.

In parallel with the detection of the signal in the frequency line in the Doppler frequency meter of target 36 (Fig. 16), the position of the energy center of the Doppler spectrum is determined (the corresponding Doppler filter number). The meter 36 circuit is similar to the meter described in [9, p. 255, Fig. 3.17]. The same signals arrive at its input as at the threshold detector 28. The structural diagram of the Doppler frequency meter of target 36 is close to that of the detector 28. Weighing signals from the outputs of the sixth shift register 122 arrive at the ninth adder 124. The weighting coefficients are selected so that the output of the ninth adder 124 corresponded to the output of the differential correlator, which determines the deviation of the center of the window from the energy center of the spectrum in a sliding window of q elements. When the mismatch signal of the energy center of the spectrum of the signal passes through zero at the output of adder 124, the sign that will be detected by the zero detector 125 changes. At the same time, a pulse of duration T0 appears in the form of a logical unit, which will allow the target Doppler frequency code to be calculated in the information output device 39 (position code of the resolved cell in frequency). The output of the zero detector 125 is the output of the Doppler frequency meter target 36.

The threshold target size selector 30 (Fig. 12) determines the width of the Doppler spectrum of the intended target (target frequency size) based on the number of pulses That passed through the first circuit AND 29 and coincided with the detection signal from the output of the threshold detector 28. This number is determined on each line being polled signal frequency limited by the period of the signal T _p . The threshold target size selector 30 operates as follows.

The signal T _page at the end of each line (before the start of line scanning) resets the seventh counter 103. When pulses arrive from the first circuit AND 29 at the counting input of the seventh counter 103, they are counted, increasing the output code of the counter. The trailing edge of the pulse T _page at the end of the line code of the seventh counter 103 is reset. In the process of counting (measuring the frequency size of the target), the code of the seventh counter 103 is compared with the second threshold (K _por2 ) coming to the second input of the second comparator 104 from the fourth output of the secondary processing unit 23. When the counter 103 exceeds the second threshold, the second comparator 104 is activated, the output the pulse of which sets the fourth trigger 105 to one. This state of the trigger 105 is held until the trailing edge of the pulse T _p at the end of the line. The output of the fourth trigger 105 is the second output of the threshold target size selector 30, a single state of which means that the information output device 39 does not provide the coordinates of this target to the control system, since its signal across the width of the spectrum corresponds to a passive interference signal.

Prior to the start of the polling of signals in multichannel Doppler filters 17 and 18, when at the sixth input of the secondary processing unit (BWO) 23, the signal PC1 = 1, according to the algorithm of Fig. 10, the BWO 23 receives information from the speed meter and drift angle 24 about the current drift angle α and the speed of the aircraft V and the angle of sight of the RSN relative to the axis of the aircraft β from the antenna drive 1. Based on this, the BWO 23 determines the current angular separation of directions to neighboring frequency-resolved scene elements δβ _V and provides it to the fourth input of the primary processing unit 22. In the process of changing addresses about the millet of the resolved cells of the scene in range (arrives at its seventh input, with PC1 = PC2 = 0 at the sixth and first input, respectively), the secondary processing unit 22 calculates the second threshold K _por2 and issues it from the fourth output to the eighth input of the primary processing unit 22 (third input threshold selector size target).

R _j = j · ΔR = j · c · To / 2

Where:

] [ - the integer part of number,

ΔL is the maximum expected tangential linear size of the NK at the analyzed range in terms of sight,

ΔR is the price of the least significant bit of the code of the interrogated range (the distance between adjacent interrogated cells of the scene in range),

j is the address of the interrogated cell in range (time). Code j arrives at the seventh input of the secondary processing unit 23 from the third output of the synchronizer 7.

The value δβ _V c of the fourth input of the primary processing unit 22 is supplied to the integrator 32, the output code of which with the clock cycle of the cells in frequency To increases by the value of the input signal and corresponds to the "Doppler angle" β _i = i · δβ _V ,

the current ith frequency-polled permission element. At the end of the line scan before starting to receive new information, the integrator 32 is reset to zero by the signal T _p coming to its third input. Information about the target / ODO is contained in the variance of the difference ψ _{i of the} "Doppler angle" β _i and the "bearing" γ _i of the frequency-resolved elements at the same range. For its calculation, the difference between the “Doppler angle” at the output of the integrator 32 and the “bearing” from the output of the divider 21 is first determined in the calculation circuit of the difference 33.

Next, for the difference in the angles ψ _i calculated in the current row, the mathematical expectation M _ψ is determined in the mathematical expectation computer 34. The circuit of the computer 34 is shown in Fig. 15. His work is as follows. Information about the difference in angles ψ _i from the circuit of the calculator of the difference 33 is fed to the first input of the tenth multiplier 119, the second input of which receives a signal from the block combining quadrature 19 as a weight. The result of the multiplication is fed to the second (information) input of the fifth integrator 116. The values of the weights (signals from the quadrature combining unit) are sent to the second (information) input of the sixth integrator 120, which works similarly to the fifth integrator 116 with the same receive and reset clocks. The results of integration with the outputs of the fifth 116 and sixth 120 integrators are read into the third 117 and fourth register 121, respectively, at the end of the line with a pulse T _p arriving at their second inputs. The outputs of the third 117 and the fourth registers 121 go to the third divider 118, as a result of which we have the calculated expectation equal to

where P _i is the output of the block combining quadratures at the i-th frequency. The result of the accumulation of weights from the fourth register 121 is supplied to the second output of the mathematical expectation calculator 34 and is used in the dispersion calculator 31 (Fig.13), which operates as follows.

The angle difference ψ _i , current in the line, comes to the first (informational) input of the fourth shift register 106 from the difference calculation circuit 33, and the mathematical expectation M _ψ to the second input of the third difference calculation circuit 107 from the mathematical expectation calculator 34. The fourth shift register 106 has N bits equal to the number of filters polled by the frequency line, therefore, provides a delay of the input signal ψ _i by the length of the line. This delay corresponds to the delay in the formation of M _ψ in the calculator 34. The synchronization of the reception of input information in the fourth shift register 106 is performed by pulses with a frequency Fo coming from the synchronizer 7 to its second input. The output of the third difference calculation circuit 107 is (ψ _i -M _ψ ) and, through the square calculation circuit 108, is supplied to the ninth multiplier 111. At the first input of the ninth multiplier 111, the signal P _i comes from the output of the quadrature combining unit 19 as a weight. The result of the multiplication is integrated by the fourth integrator 112 and fed to the second divider 113. The sum of the weights ΣP _i found earlier in the calculator 34 comes to the second input of the second divider 113. As a result of the division, we obtain the desired signal variance:

Information is synchronized from the ninth multiplier 111 to the fourth integrator 112 by delaying the output signal of the threshold detector 28 by the length of the line and by passing pulses of the polling cycle of the signals along the line (pulses with frequency Fo) through the fifth AND 110 circuit with a single value of the delayed detection signal at the fifth output Resetting shift register 109. The shift registers 106, 109 and the integrator 112 is made at the end of line pulse T _p, coming to the first input of the calculator 31 with dispersion synchronizer 7. O d of the second divider 113 is the output of the variance calculator 31.

The found dispersion value D _{ψ is} supplied to a threshold device 35, which, when the calculated third threshold is exceeded, generates a second inhibit signal output to the third input of the information output device 39 (Fig. 17), which operates as follows.

The coordinates (codes) of the current interrogated time-frequency cells (i, j) come from the synchronizer 7 to the fourth input of the UVI 39. The corresponding azimuthal angle β _Tsi = β + γ _{i of the} interrogated resolved scene element is determined by the adder 38 and comes to the second input of the output device information 39. (β is the angle between the RSN and the axis of the aircraft, comes from the antenna drive 1, γ _i is the bearing of the resolved cell, comes from the divider 21).

The listed input information (i, j, β _Цi ) goes to the fifth 126th, sixth 127th and seventh 128 registers and is written to them by the signal from the output of the second And 37 circuit (a signal indicating the presence in the current interrogated cell of the energy center of the signal spectrum). The information recorded in registers 126, 127, and 128 is transferred to the corresponding registers 129, 130, and 131 and is transmitted from them to the output at the end of the line along the leading edge of the PC2 pulse generated by the pulse shaper 135 in the absence of prohibition signals on the interrogated line at the inputs of the OR circuit 132. pulses prohibition reset at the end of each line, but are stored for a period T _p in the eleventh register 133 on the rising edge of pulse T _p input at its second input, so the presence of a logic one at the output of the eleventh register 133 n sixth output of AND gate 134 passes pulse T _p, respectively, does not update information about the found target but not introduced false. The trailing edge of the pulse at the output of the sixth circuit AND 134 starts the information retrieval enabler 135, which generates a duration-normalized RS2 information retrieval pulse, the duration of which is selected sufficient for reliable detection by the secondary processing unit 22, but not more than half the pulse period T _page

The secondary processing unit 23 represents a processor operating according to the algorithm shown in FIG. 10, designed to receive source data from a control system for viewing a scene in a given angular sector from β _n to β _to output initial data about the position of the beginning of the scanning sector β _n to the antenna drive 1, turning the radiation on and off, calculating the angular separation between adjacent directions to the frequency-resolved scene elements δβ _V , threshold K _por2 , and issuing them to the primary processing unit 22, receiving the result from the primary processing unit 22 Tata measuring the coordinates of targets and issuing them to the control system.

Individual fragments of the operation algorithm of the secondary processing unit 23 in interaction with other elements of the circuit have been described above. Below is described its operation algorithm (figure 10) in more detail. The beginning of the work, as noted above, is the arrival in block 23 of the source data management system about the boundaries of the scene viewing sector β _n and β _k . The secondary processing unit 23, based on the source data, provides the required initial position of the viewing sector β _n to the antenna 1 drive. The antenna drive 1 processes the received information and moves the PCH axis in the desired direction. The secondary processing unit 23 proceeds to the analysis of current data on the angular position of the RSN β coming from the second output of the antenna 1 drive to its second input. Upon reaching β the required β _n from the secondary processing unit 23, a command is issued to the second input of the pulse modulator 8, allowing the radiation of probe pulses. Next, the PC1 signal generated by the synchronizer 7 is analyzed. With PC1 = 1 (during coherent signal accumulation), the secondary processing unit 23 receives the current value of the aircraft speed V V and drift angle α from the drift meter and drift angle 24 and the current value of the azimuthal orientation of the RSN relative to the longitudinal axis of the aircraft (3 from the antenna drive 1. Based on these data, δβ _V is calculated and output to the fourth input of the primary processing unit 22. This cycle, when PC1 = 1, can be repeated until the value of PC1 changes to PC1 = 0 (then if the accumulation is completed, the interrogation of the time-frequency cells of multichannel Doppler filters has begun.) Next, the secondary processing unit 23 analyzes the logical level of the PC2 signal arriving at its first input from the primary processing unit 22. When PC2 = 0, the secondary processing unit 23 reads the address code of the polled cells issued by the synchronizer 7 and arriving at the seventh input of the secondary processing unit 23, selects a range address (j) from it and calculates K _por2 . The calculation result K _por2 is issued to the eighth input of the primary processing unit 22. After this, the secondary processing unit 22 expects a change in the logic level of signal PC2 to unity. When PC2 = 1 (permission to retrieve information from the primary processing unit 22), the secondary processing unit 23 receives the coordinates of the detected target (i, j, β _Tsi ) from the first output of the information output device 39 and determines the azimuthal coordinate of the target relative to the velocity vector, taking into account the drift angle

β _Vi = β _{i i} -α.

The next reception of the coordinates of the detected targets is performed at PC1 = 0 (the interrogation of the frequency-time cells of the scene continues) and the appearance of a new signal PC2 = 1 (target detection signal on a new poll line). When PC1 = 1 (the interrogation of the frequency-time cells of the scene is completed, coherent signal accumulation has begun), the current azimuthal angle of orientation of the antenna β is compared with the specified upper boundary of the viewing sector β _k . If the boundary is not reached (β ≠ β _k ), the secondary processing unit 23 starts repeating the cycle associated with the calculation of δβ _V , thresholds K _por2 , signal accumulation, reading of the measured coordinates of the targets in the new azimuthal direction of the antenna β, discussed above and starting with checking the status PC1 signal. The end of the cycle is associated with checking the achievement of the angle between the RSN antenna and the longitudinal axis of the aircraft β border β _to . Upon reaching the angle β of the boundary β _to the secondary processing unit 23 turns off the radiation (removes the resolution from the pulse modulator 8) and provides the coordinates of the targets (i, j, β _Vi ) in the control system.

Synchronizer 7 (Fig. 4) is intended for generating pulses of modulation of probe pulses by the repetition period and duration (pulse T _P ), signal accumulation intervals and polling of the results of signal accumulation from resolved elements of the time-frequency scene (PC1 signal), address code of the scene elements being polled (i, j) and the polling interval of the line of the time-frequency array (pulse T _p ). The operation of the synchronizer 7 is as follows. Chroning pulses with a period To arrive at the counting (first) inputs of the second 55 and third 58 counters. In counting mode, they work in turn and are controlled by the first trigger 56. When the logical unit arriving at the second (installation) input of the third counter 58 is not counted at the first output of the first trigger 56, it sets the code in accordance with the code coming from the second code bus 57. At the same time, a logical zero at the second output of the first trigger 56, entering the second installation input of the second counter 55, allows it to count pulses arriving at its first input. The transfer pulse at the output of the second counter 55 occurs when the counter code passes through the maximum value. This pulse, arriving at the first input of the first trigger 56, changes its state to the opposite, in which the third counter 58 is allowed to count, the second counter 55 is prohibited, and the initial code of the counter 55 is set in accordance with the code coming from the first code bus 54 to third entrance. By setting the corresponding codes on buses 54 and 57 at the second output of the first trigger 56, a calculated pulse of modulation of the period and duration of the probe pulses T _{P is formed} . The PC1 signal and the address code (i, j) of the interrogated time-frequency cells of the scene are generated in the same way as the considered pulse formation T _P using the fourth counter 59, the second trigger 60, the fifth counter 61 and the code buses 62 and 63. The difference is that the fourth counter 59 instead of pulses That counts the pulses T _P. The volume of the fifth binary counter 61 (V _cf = 2 ^S ) is selected as a multiple of twice the maximum address code of the scene cell being polled K _max = j _max · i _max = 2 ^S-1 , where s is the number of bits of the counter 61, i _max is the maximum value of the Doppler number filter (in our case, i _max = 2 ^s-1 = N), j _max is the maximum value of the code of the scene-resolved range cell. In this case, the initial value of the least significant (s-1) bits of the counter code 61 (the minimum value of the address code of the interrogated cells) is zero. When you arrive at the fifth counter 61 in the counting mode K _max pulses (generating the maximum code of the scene cell being polled at the second output), a transfer pulse will appear at the first output, which prohibits this count through the second trigger 60 and sets the code of the counter 61 to its initial state in accordance with the fourth code code bus 63. The output of the second flip-flop 60 (the signal PC1) is a synchroniser fourth output 7. a single value PC1 corresponding coherent accumulation interval equal to NT _n where N - number of Doppler filters in packets d plerovskih filter 76 and zero - polling interval resolved scene cells. The frequency line polling interval T _p is formed by a decoder 64, to the input of which the current address code of the polled cells is supplied from the second output of the fifth counter 61. The pulse period T _p is NTo.

The technical advantage of the proposed method and device over prototypes [4] and [5] is to increase the reliability of the target / passive interference classification (with the exception of angular reflector type interference) by expanding the ability to select ODO signals in conditions when the width of its spectrum is close and even less than expected the width of the spectrum of the target signal.

We will carry out a calculation confirming the effectiveness of the proposed solution. Let the target be observed in the R range from 6 to 18 km in the radiation range λ = 3 cm, at an angle β = 45 ° relative to the velocity vector. We assume that the linear azimuthal dimensions of the aircraft depending on the conditions of sighting from the aircraft and the orientation of the aircraft are in the range of 20 ... 70 m, the speed of the aircraft is V = 300 m / s. We find the width of the spectrum of the target signal by the formula

Calculation result R, km ΔL, m Δfds, Hz 6 twenty 53 6 70 160 eighteen twenty fifteen eighteen 70 46

From the analysis of the table, the width of the ODO spectrum for λ = 3 cm, equal to 53 Hz [6, p.272], and the task of screening out passive interference signals (ODO, shore), it is advisable to choose a frequency resolution of δf _ds = 25 Hz. This allows us to have the number of spectral samples of the ODO 2 ... 3, find the variance D _ψ , compare with the threshold and make the correct classification in all the conditions R and DE mentioned above. In the prototype, the correct classification at R = 18 km and ΔL = 70 m at the selected frequency resolution is impossible, since it turned out that the width of the NK spectrum is equal to the width of the ODO spectrum and the probability of a correct solution and false alarm is 50%. At R = 6 km and ΔL = 20 m, the calculated NK spectrum width is equal to the ODO spectrum width, respectively, the probability of the correct solution and false alarm is also 50%. Continuing the analysis, we find that at a distance of R = 6 km at Δ L = 70 m in the prototype the NK will be erroneously classified as a shore, since the width of its spectrum is higher than the spectral width of the ODL, while the signal of the ODL as narrowband will be classified as a target. Modeling shows that this does not happen in the proposed method and device, which confirms the achievement of the technical result indicated in the application materials.

Thus, the technical effect in using the proposed method and device consists in increasing the reliability of the target / passive interference classification (with the exception of angular reflector type interference) by expanding the ability to select ODO signals in conditions when the width of its spectrum is close to or even less than the expected width of the signal spectrum goals. At the same time, the unconditionally useful capabilities of the prototypes were preserved: 1) selection of signals from the coastline and ODO when the spectrum width of the calculated spectrum of the NK signal was exceeded, 2) increased noise immunity to active impact-like noise interference due to the use of a complex signal and narrow-band Doppler filtering.

Based on the above description and drawings, the proposed device can be manufactured using known components, known in the electronic industry of technological equipment and used on mobile carriers as a radar for target selection against passive reflections from the Earth and ODO, determining their initial coordinates for output to control system and subsequent auto tracking.

LITERATURE

1. P.A. Bakulev. Radar of moving targets. - M.: Soviet Radio, 1964 (p. 147).

2. US patent 6137439 from 10.24.2000, CL G 01 S 13/52. Continuous wave Doppler system with suppression of ground clutter.

3. US patent 4119966 from 10.10.1978, cl. G 01 S 13/52. Clutter discriminating apparatures for use with pulsed Doppler radar systems and like.

4. Patent of Russia 2117960 from 08.20.1998, cl. G 01 S 13/44. A method of tracking a monopulse radar station (prototype method).

5. Patent of Russia 2099739 from 12.20.1997, cl. G 01 S 13/42. Radar station, (prototype device)

6. S.A. Vakin, L.N. Shustov. Fundamentals of radio countermeasures and electronic intelligence. - M.: Soviet Radio, 1968 (p. 272).

7. Protection against radio interference, ed. Maximova M.V. - M: Soviet Radio, 1976 (p. 85-88).

8. V. Manasevich. Frequency synthesizers, theory and design. - M.: Communication, 1979 (p. 190 Fig. 4.10, 4.11).

9 V.A. Likharev. Digital methods and devices in radar. - M.: Soviet Radio, 1973. (p. 181, fig. 2.36; p. 255 fig. 3.17).

10. E.Kolchinsky, I.A. Mandurovsky, M.I. Konstantinovsky. Doppler devices and navigation systems. - M .: Soviet Radio, 1975. (p. 368-370 fig. 11.16).

11 Radar Handbook, Ed. M. Skolnik, v. 3. - M .: Soviet Radio, 1979 (from 19 ... 47, 103 ... 107, 144).

Claims (2)

1. A method for selecting a surface target with a monopulse radar station, which includes emitting coherent radio pulses with a constant carrier frequency in a given direction, receiving reflected signals in a given range of ranges, sum-difference converting the received signals, superheterodyne converting them to an intermediate frequency, amplifying the total and difference signals by intermediate frequency, converting the spectrum of the sum and difference signals into the region of video frequencies by means of a quadrature call detection using reference oscillations with the formation of the quadrature components of each signal, the coordinated filtering of the video pulses of the quadrature components of the signals, in the total and difference channels for each range element in a given time interval, carries out multi-channel Doppler filtering of the complex envelope of the pulse sequence in the Doppler frequency range from minus F _P / 2 to plus F _P / 2 filters with a band of δf, the number of filters N = F _P / δf, where F _P is the pulse repetition rate, the bandwidth of the Doppler filters δf is selected as half the spectrum width of the inter-period fluctuations of the signals from the ODO Δf _D , the powers of the spectral components are compared with the first detection threshold level, the spectrum width with the most powerful spectral component is determined from a given level from the number of components that exceed the detection threshold, and the obtained the value of the spectrum width with a second predetermined threshold value, when the spectrum width of the second threshold value is exceeded (the second criterion is false spruce) make a decision on the presence of a signal from a false target, which is rejected as a false one, and proceed to the next element of range, and if there is no excess, the scalar product of quadrature signals obtained by filtering the same and total channels in the Doppler filters by multiplying the same name is calculated quadrature components of the signals with the summation of these products, characterized in that the sequential calculation of "bearings" (γ _i ), "Doppler angles" (β _i ) and variance (D _ψ ) of the angular difference between them (ψ _i = γ _i -β _i ) for all resolvable time-frequency elements of the scene that are not rejected according to the first and second criteria and are at the same ranges, the bearing (γ _i ) of the resolvable element scene is determined by dividing (normalizing) the calculated scalar product on the total signal power to it for the same scene resolvable element and multiplying the result by a scaling factor, "Doppler angle" scene resolvable element (β _i) is defined as the produ ix angular separation between adjacent Doppler lines (δβ _V) to the number of Doppler filter (i), wherein the analyzed range detected signal found at each resolvable range dispersion angle difference (D _ψ) is compared with a third threshold, if exceeded variance calculated third threshold values (the third criterion of a false target) at any of the analyzed ranges - the scene signal at this range is also discarded, the angular separation between adjacent directions (δβ _V ), resolved in frequency and found target within a narrow total DND, is determined by the known relationship of the Doppler frequency shift of the reflected signal with the parameters of radiation, the movement of the aircraft and the conditions of sight according to the formula:

Where λ is the wavelength of the emitted signal; F _P - the repetition frequency of the probe pulses; V is the speed of the aircraft relative to the Earth; β is the angle between the longitudinal axis of the aircraft and the RSN of the antenna system; α is the drift angle of the aircraft. 2. A single-pulse radar that implements the method according to claim 1, comprising a series-connected antenna switch, a sum-difference converter and an antenna, a power amplifier, an antenna drive, a two-channel receiver, the first output of which is connected to the input of a video amplifier through a quadrature-phase detector of the total channel the total channel, the second output of the two-channel receiver through the quadrature-phase detector of the differential channel is connected to the input of the video amplifier of the differential channel, digital matched filters of the total and difference channel, quadrature combining unit, angle discriminator, primary processing unit and secondary processing unit, while the antenna drive is kinematically connected to the third input of the antenna, the second output of the sum-difference converter is connected to the second input of the two-channel receiver, the third output of the antenna switch is connected to the first input a two-channel receiver, the synchronizer is connected via a pulse modulator to the second input of the power amplifier, the second output of the exciter is connected to the third input of the two-channel about the receiver, the third output of the pathogen is connected to the second inputs of the quadrature phase detectors of the total and difference channels, the second output of the secondary processing unit is connected to the input of the antenna drive, the second output of which is connected to the seventh input of the primary processing unit, the input of the quadrature combining unit is connected to the first input of the angle discriminator , the output of the quadrature combining unit is connected to the second input of the primary processing unit, the fourth output of the pathogen is connected to the input of the synchronizer, the first output of the secondary unit the processing unit is connected to the fourth input of the primary processing unit, the first output of which is connected to the first input of the secondary processing unit, the two-channel receiver contains a two-channel high-frequency amplifier, a two-channel balanced mixer, and the third (heterodyne) input of which is a series input at the two inputs-outputs of the total and difference channels the third heterodyne input of the two-channel receiver, the first and second inputs of the high-frequency amplifier are the inputs, and the first and second outputs of the two-channel an intermediate frequency amplifier — by the outputs of the total and difference channels of a two-channel receiver, respectively, characterized in that multichannel Doppler filters of the total and difference channels, a phase shifter, a divider and a speed and drift meter are introduced into it, while the third output of the driver through a phase shifter is connected to the third inputs of the quadrature phase detectors of the total and difference channels, the output of the video amplifier of the total channel through a digital matched filter of the total channel is connected to to the input of the multi-channel Doppler filter of the total channel, the output of the video amplifier of the differential channel through a digitally matched filter of the differential channel is connected to the first input of the multi-channel Doppler filter of the differential channel, the output of the quadrature combining unit is connected to the first input of the divider, the output of which is connected to the fifth input of the primary processing unit, the first output the pathogen through the power amplifier is connected to the first input of the antenna switch, the fourth output of the pathogen is connected to the second inputs by the digital matched filters of the total and difference channels, the second inputs of the multi-channel Doppler filters of the total and difference channels and the first input of the primary processing unit, the first synchronizer output is additionally connected to the pathogen input and the fifth inputs of the multi-channel Doppler filters of the total and difference channels, the output of the multi-channel Doppler filter of the total channel connected to the input of the block combining quadratures, the output of the multi-channel Doppler filter differential channel connected to the second input of the angle discriminator, the output of which is connected to the second input of the divider, the second synchronizer output is connected to the third input of the primary processing unit, the fourth synchronizer output is connected to the third inputs of the multichannel Doppler filters of the total and difference channels and the sixth input of the secondary processing unit, the third synchronizer output connected to the fourth inputs of the multichannel Doppler filters of the total and difference channels, the seventh input of the secondary processing unit and the sixth input ohm of the primary processing unit, the third output of the secondary processing unit is connected to the second input of the pulse modulator, the second output of the primary processing unit is connected to the fifth input of the secondary processing unit, the first and second outputs of the speed and drift meter are connected to the third and fourth inputs of the secondary processing unit, respectively, the second the output of the antenna drive is additionally connected to the second input of the secondary processing unit, the fourth output of the secondary processing unit is connected to the eighth input of the primary processing unit, bl to the primary processing comprises a threshold detector, a first AND circuit, a target size threshold selector, an integrator, a difference calculation circuit, a dispersion calculator, a threshold device and an information output device, a series connected Doppler frequency meter of the target, a second AND circuit, and an output device information, the first and second outputs of which are the second and first outputs of the primary processing unit, an adder connected to the second input of the output device and information, moreover, the second input of the primary processing unit is connected to the second inputs of the threshold detector and the Doppler frequency meter of the target, the fourth input of the calculator and the sixth input of the dispersion calculator, the output of the threshold detector is additionally connected to the third input of the dispersion calculator and the first input of the second circuit the input of the primary processing unit is connected to the third inputs of the threshold detector, integrator, meter Doppler frequency of the target, the calculator mathematically waiting state, the sixth input of the information output device, the second input of the threshold selector of the target size and the first input of the dispersion calculator, the first input of the primary processing unit is connected to the fourth input of the dispersion calculator and the first inputs of the threshold detector, integrator and Doppler frequency meter of the target and the second input of the first AND circuit , the fourth input of the primary processing unit is connected to the second input of the integrator, the second output of the threshold selector of the target size is connected to the fifth input of the information output device ii, the output of the difference calculation circuit is connected to the second input of the mathematical calculator, the output of the first circuit And is additionally connected to the first input of the mathematical calculator, the fifth input of the primary processing unit is connected to the first input of the adder and the second input of the difference calculation circuit, the seventh input of the primary processing unit is connected with the second input of the adder, the sixth input of the primary processing unit is connected to the fourth input of the information output device.

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