JP5396741B2 - Control device for permanent magnet type synchronous motor - Google Patents

Description

  The present invention relates to a control device for a permanent magnet type synchronous motor that does not have a magnetic pole position detector. More specifically, the control constant is optimally set even when an electrical constant such as an inductance value of the permanent magnet type synchronous motor is unknown. Thus, the present invention relates to a control device for a permanent magnet synchronous motor that can accurately calculate the magnetic pole position of the motor.

In order to reduce the cost of a control device for a permanent magnet type synchronous motor (hereinafter also referred to as PMSM), a so-called sensorless control technique that operates without using a magnetic pole position detector has been put into practical use.
Many of the sensorless control techniques calculate the magnetic pole position using an induced voltage induced between the motor terminals by the permanent magnet of the rotor, and are applied to the operation in the medium to high speed region. However, since this magnetic pole position calculation method cannot be applied to a low speed region including when the motor is stopped, it is necessary to calculate the magnetic pole position by another method.

  By the way, PMSM is roughly classified into a surface magnet structure permanent magnet type synchronous motor (hereinafter also referred to as SPMSM) and an embedded magnet structure permanent magnet type synchronous motor (hereinafter also referred to as IPMSM) according to the structure of the rotor. Among these, in IPMSM, a technique for calculating the magnetic pole position using the saliency of the rotor has been put into practical use.

On the other hand, as a technique for calculating the magnetic pole position of the IPMSM having a small saliency of the SPMSM or the rotor, a method using the magnetic saturation characteristic of the electric motor core is known.
For example, Patent Document 1 gives a minute change to the voltage command values of the dc axis and qc axis, which are the estimated magnetic flux axes of PMSM, and the current change rate of the dc axis and qc axis at this time, or the inductance that is the reciprocal thereof. A method for calculating the magnetic pole position from the change is described.
According to this method, the magnetic pole position can be calculated in a short time by a relatively simple process. Furthermore, it is also possible to calculate the frequency command value so that the position calculation error obtained from the current change rates of the dc axis and the qc axis becomes zero, and to calculate the magnetic pole position by integrating the frequency command value.

JP 2002-78392 A (paragraphs [0079] to [0082], [0100] to [0112], [0120] to [0121], FIGS. 16 to 19, FIGS. 23 to 25, FIG. 28, etc.)

In order to realize the magnetic pole position calculation described in Patent Document 1, it is necessary to determine the magnitude of the high-frequency voltage so that a sufficiently high-frequency current flows due to the limitation of the detection accuracy of the current detector. In order to perform the magnetic pole position calculation stably and with high response, it is necessary to optimally design the control constant.
In order to realize these, information on electrical characteristics such as inductance values and magnetic saturation characteristics of PMSM is indispensable. In particular, in order to perform magnetic pole position calculation for PMSM whose electrical characteristics are unknown, these electrical characteristics are required. A means for measuring the characteristics and adjusting the control constant based on the measured characteristics is required.
That is, conventionally, when the electrical constant of PMSM is unknown, in order to reliably calculate the magnetic pole position, much effort and cost are required for the design and adjustment of the control constant.

  Therefore, the problem to be solved by the present invention is that, even if the electrical constant of the motor is unknown, the magnetic pole position calculation using the magnetic saturation characteristics of the motor is reliably realized by automatically adjusting the amplitude of the high-frequency voltage applied to the motor. It is an object of the present invention to provide a permanent magnet type synchronous motor control device that is made possible.

In order to solve the above-mentioned problem, a control device according to claim 1 is a control device for a permanent magnet synchronous motor without a magnetic pole position detector,

  Control equipped with magnetic pole position calculation means for calculating the magnetic pole position of the motor from the high-frequency current that flows when a high-frequency alternating voltage is applied to the motor, using the motor's terminal voltage and current as vectors and using the magnetic saturation characteristics of the motor In the device

  Amplitude command value automatic adjustment means for automatically adjusting the amplitude command value of the high-frequency alternating voltage,

  The amplitude command value automatic adjustment means is

  Means for applying a high-frequency alternating voltage of a rectangular wave composed of two pulse voltages having the same amplitude and pulse width and different polarities in a plurality of vector directions;

  Means for detecting the amplitude of a high-frequency current having the same frequency as the high-frequency alternating voltage;

  Means for detecting a d-axis high-frequency current amplitude, which is a high-frequency current amplitude when the high-frequency alternating voltage is applied in a direction parallel to the magnetic pole position of the motor, from the high-frequency current amplitude detection value detected by the means;

  high-frequency current adjusting means for calculating an amplitude command value of the high-frequency alternating voltage so that the d-axis high-frequency current amplitude detection value matches the d-axis high-frequency current amplitude command value;

  Have
The high-frequency current adjusting means includes

  High-frequency current amplitude estimating means for calculating an estimated value of the d-axis high-frequency current amplitude from the amplitude command value and gain estimated value of the high-frequency alternating voltage;

  Gain estimation means for amplifying a deviation between the estimated value of the d-axis high-frequency current amplitude and the detected value of the d-axis high-frequency current amplitude to calculate the gain estimated value;

  Means for calculating an amplitude command value of the high-frequency alternating voltage from the d-axis high-frequency current amplitude command value and the gain estimation value;

  It consists of

The control device according to claim 2 is the control device according to claim 1,

  Means for extracting from the high-frequency current amplitude detection value a Fourier series first-order component that is a component that changes in a cycle of an electrical angle of 360 degrees depending on the vector direction of the high-frequency alternating voltage;

  Means for determining whether or not to calculate the magnetic pole position from the magnitude of the Fourier series primary component;

  It is equipped with.

  According to the present invention, even when the electrical constant such as the inductance value of the motor is unknown, the optimum value of the amplitude of the high-frequency alternating voltage applied to the motor can be automatically adjusted, and the magnetic pole utilizing the magnetic saturation characteristics of the motor Position calculation can be realized reliably.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. First, the principle of calculating the magnetic pole position using the magnetic saturation characteristic will be described.
The PMSM can achieve highly accurate torque control by performing current control according to the d-axis of the rotor (the magnetic pole direction of the rotor) and the q-axis advanced 90 degrees from the d-axis. However, since the d and q axes cannot be directly detected without the magnetic pole position detector, γ and δ of the orthogonal rotation coordinate system that rotates at the angular velocity ω ₁ (= speed calculation value) corresponding to the d and q axes. The control calculation is performed by estimating the axis to the control device side.

Here, FIG. 4 is a diagram showing the definition of the coordinate axes used for the magnetic pole position calculation.
In FIG. 4, d and q axes are rotational coordinates that rotate in synchronization with the rotor, and the magnetic pole direction of the rotor is defined as the d axis, and the advance direction of 90 [deg] from the d axis is defined as the q axis. The γ and δ axes are the estimated axes of the d and q axes, and the current control and voltage control of the PMSM are performed by the control device using various quantities on the γ and δ axes. The rotational speeds of the d and q axes and the rotational speeds of the γ and δ axes are ω _r and ω ₁ , respectively.
Further, the angle difference θ _err between the γ-axis (γ, δ-axis) and the d-axis (d, q-axis), that is, the position calculation error is defined by Equation 1.

On the other hand, for the x and y axes in FIG. 4, the vector direction of the high-frequency alternating voltage applied to the stator winding of the permanent magnet synchronous motor to measure the inductance when calculating the magnetic pole position is 90 [ deg] The advance direction is defined as the y-axis. If the angle difference between the x-axis relative to the γ-axis (also referred to as angle) is defined as [delta] _x, the angle error theta _xerr the x-axis (x, y-axis) and the d-axis (d, q-axis) , Expressed by Equation 2.

Next, FIG. 5 shows the relationship between the d-axis current and the flux linkage, and the relationship between the d-axis current and the d-axis inductance.
The linkage flux when the d-axis current I _d is zero is equal to the permanent magnet flux Ψ _m . When the d-axis current I _d is controlled to the plus side, the permanent magnet magnetic flux and the magnetic flux generated by the d-axis current I _d are combined to increase the magnetic flux, and the iron core of the motor is magnetically saturated to reduce the d-axis inductance. .
On the other hand, if the d-axis current I _d is controlled to the minus side, the permanent magnet magnetic flux and the magnetic flux generated by the d-axis current I _d are canceled out, so that the magnetic saturation of the iron core of the motor is reduced and the d-axis inductance increases. To do. That is, the inductance is changed by the value of d-axis current I _d.
From the above, the magnetic saturation characteristics change depending on the angle error between the d-axis and the current vector, and the inductance changes accordingly.

FIG. 6 shows the relationship between the angle error θ _xerr between the x-axis and the d-axis and the x-axis inductance when a constant current is passed in the x-axis plus direction. FIG. 6 shows the case of SPMSM in which the rotor does not have saliency (d-axis inductance and q-axis inductance are equal) for easy understanding of the principle.
As shown in FIG. 6, the x-axis inductance changes in one electrical angle cycle with respect to the angle error θ _xerr . By utilizing this fact, the angle δ _x that minimizes the x-axis inductance with the γ-axis angle θ ₁ being constant is obtained from the relational expressions of Formula 1 and Formula 2 to obtain the d-axis angle (magnetic pole position). Can be calculated.

Here, the x-axis inductance can be measured from an x-axis high-frequency alternating current flowing at this time by applying a high-frequency alternating voltage in the x-axis direction.
FIG. 7 shows waveforms of the x-axis high-frequency alternating voltage v _xh , the x-axis current i _x, and the x-axis high-frequency current amplitude I _xh . In order to understand the basic principle, attention is paid to the high frequency components of voltage and current in FIG. 7, and the fundamental wave component is zero.

The high-frequency alternating voltage v _xh applied to the x-axis is a rectangular wave obtained by combining two pulse voltages having the same amplitude and pulse width but different polarities. Note that one cycle of the high frequency alternating voltage _{v xh} and _{T h.}
When measuring the inductance at the time was passed a constant current to the x-axis positive direction, as shown, in the period T _h, control positively in the first half cycle amplitude V _h of x-axis high frequency voltage v _xh In the next half cycle, it is controlled to be negative (hereinafter, the application of the high-frequency voltage is defined as “application of a high-frequency voltage in the x-axis positive direction”). In this case, the x-axis current i _x, first as shown changes in the x-axis positive direction, followed by returning to the value of the applied immediately before the start of the high-frequency alternating voltage v _xh. As is clear from FIG. 7, a high-frequency current i _xh having an amplitude I _xh having a DC bias component in the positive direction of the x-axis flows through the x-axis.

Similarly, the inductance when a current through the constant current to the x-axis negative direction, in the period T _h, the amplitude V _h of x-axis high frequency voltage v _xh controlled to negative in the first half cycle, the next half The period is controlled to be positive (hereinafter, application of this high-frequency voltage is defined as “application of a high-frequency voltage in the negative direction of the x axis”), and measurement can be made from the high-frequency current i _xh flowing at this time.
From FIG. 7, the high-frequency current amplitude I _xh can be calculated from the current deviation at the sample point where the polarity of the x-axis high-frequency voltage v _xh changes. Here, in order not to be influenced by the fundamental wave voltage or the fundamental wave current, the high-frequency current amplitude I _xh is calculated by Equation 3.

Next, the relationship between the high frequency current and the magnetic pole position is derived.
The amplitude vector I _xyh of the high-frequency current that flows when the rectangular-wave high-frequency alternating voltage v _xh as shown in FIG.

FIG. 8 shows the relationship between the angle error θ _xerr and the x-axis high-frequency current amplitude I _xh shown in Equation 4, where the d-axis inductance and the q-axis inductance are equal to make the principle easy to understand. The case of SPMSM is illustrated. FIG. 9 is for IPMSM.
The x-axis high-frequency current amplitude I _xh pulsates at an electrical angle of 360 degrees and becomes maximum when the angle error θ _xerr is zero. Thus, the magnetic pole position can be calculated from the x and y axis angles at which the x-axis high-frequency current amplitude I _xh is maximized.

Next, the principle of magnetic pole position calculation will be described.
A component that changes at an electrical angle of 360 degrees with respect to the angle δ _x of the x-axis high-frequency current can be calculated by Equation 5 as a Fourier series.

Here, in each angle [delta] _x, when applying a high frequency voltage v _xh in the x-axis positive direction and the x-axis negative direction as shown in Figure 7, instead of Equation 5, the angle [delta] _x from zero 180 [deg ] From the Fourier series up to [6].

The position calculation error θ _err can be calculated by Equation 7 from the Fourier series obtained by Equation 6.

Next, a specific magnetic pole position calculation method will be described.
From Equation 5, the Fourier series I _xhb1 of the x-axis high-frequency current amplitude is a sine wave function of the position calculation error θ _err . Therefore, when the position calculation error θ _err is close to zero, the Fourier series I _xhb1 and the position calculation error θ _err can be approximated to be in a proportional relationship, and by configuring a PLL circuit that _receives the Fourier series I _xhb1 as an input, The magnetic pole position and angular velocity can be calculated.
In order to obtain the Fourier series I _xhb1 from Equation 6, it is only necessary to have information on the x-axis high-frequency current amplitude I _xh when the angle δ _x is 90 [deg]. Since the x axis coincides with the δ axis when the angle δ _x is 90 [deg], the Fourier series I _xhb1 can be calculated from the δ axis high frequency current amplitude I _δh when a high frequency voltage is applied to the δ axis. The voltage waveform and current waveform at this time are shown in FIG.
The x-axis high-frequency current amplitude I _xh is calculated by Equation 8.

The Fourier series I _xhb1 is calculated by Equation 9.

The angle error calculation value θ _errest is calculated by Equation 10 _{assuming that} it is proportional to the Fourier series I _xhb1 .

The speed calculation value ω ₁ is _obtained by proportional-integral calculation of the angle error calculation value θ _errest as shown in Equation 11.

The magnetic pole position calculation value θ ₁ is calculated by integrating the speed calculation value ω ₁ described above.
By these operations, a PLL circuit that converges the Fourier series I _xhb1 to zero is configured, and the magnetic pole position θ ₁ can be calculated.

FIG. 11 is a control block diagram for performing magnetic pole position calculation using the magnetic saturation characteristics of the electric motor in this embodiment.
First, the main circuit for driving the permanent magnet type synchronous motor 80 having no magnetic pole position detector will be described. 50 is a three-phase AC power source, and the rectifier circuit 60 rectifies the three-phase AC voltage of the power source 50 to generate a DC voltage. Convert to This DC voltage is supplied to a power converter 70 composed of a PWM inverter, and is converted into a predetermined three-phase AC voltage for driving the electric motor 80.

Next, the configuration and operation of the control device are as follows.
The current coordinate converter 14 detects the phase current detection values i _u and i _w detected by the u-phase current detector 11u and the w-phase current detector 11w, respectively, based on the magnetic pole position calculation value θ ₁ and detects γ and δ-axis currents. Coordinates are converted to values i _δ and i _γ .
The notch filter 23 removes the high-frequency current that flows due to the high-frequency alternating voltage superimposed for the magnetic pole position calculation from the γ and δ-axis current detection values i _γ and i _δ , and the γ and δ-axis fundamental wave currents i _γf and i _δf. Is detected.

The deviation between the γ-axis current command value i _γ ^* and the γ-axis fundamental wave current i _γf is calculated by the subtractor 19a, and this deviation is amplified by the γ-axis current regulator 20a to be amplified by the γ-axis fundamental wave voltage command value v _γf ^*. Is calculated. On the other hand, a deviation between the δ-axis current command value i _δ ^* and the δ-axis fundamental wave current i _δf is calculated by the subtractor 19b, and this deviation is amplified by the δ-axis current regulator 20b to be amplified by the δ-axis fundamental wave voltage command value v. _δf ^* is calculated. The γ and δ axis current command values i _γ ^* and i _δ ^* are both zero.

Frequency voltage calculator 21, the amplitude is equal to the high-frequency voltage amplitude command value V _h ^*, cycle calculates the T _h at a rectangular wave of δ-axis high frequency alternating voltage command value v _.delta.h ^*. The waveform of the δ-axis high-frequency voltage command value v _δh ^* is as shown in FIG.
The γ-axis fundamental wave voltage command value v _γf ^* is directly input to the voltage coordinate converter 15 as the γ-axis voltage command value v _γ ^* . On the other hand, the adder 22 superimposes the δ-axis high-frequency alternating voltage command value v _δh ^{* on the} δ-axis fundamental wave voltage command value v _δf ^* , and the result is converted to the voltage coordinate converter 15 as the δ-axis voltage command value v _δ ^*. Is input.
The γ and δ-axis voltage command values v _γ ^* and v _δ ^* are converted into phase voltage command values v _u ^* , v _v ^* , and v _w ^* based on the magnetic pole position calculation value θ ₁ by the voltage coordinate converter 15 and PWM. It is given to the circuit 13.

The PWM circuit 13 generates a gate signal from the phase voltage command values v _u ^* , v _v ^* , v _w ^* and the DC input voltage E _dc of the power converter 70 detected by the voltage detector 12. The power converter 70 controls the internal semiconductor switching element based on the gate signal, thereby controlling the terminal voltage of the permanent magnet type synchronous motor 80 to the phase voltage command values v _u ^* , v _v ^* , v _w ^* . To do.

On the other hand, the bandpass filter 24 calculates the x-axis high-frequency current amplitude I _xh from the detected δ-axis current value i _{δ according} to Equation 8. The Fourier series calculator 27 calculates a sin component I _xhb1 of the x-axis high-frequency current amplitude I _xh by Equation 9. The speed calculator 25 calculates the speed calculation value ω ₁ using various gains K _{θerr and the} like according to Expressions 10 and 11. The electrical angle calculator 26 integrates the speed calculation value ω ₁ to obtain the magnetic pole position calculation value θ ₁ .

Next, examples of the present invention will be described.
FIG. 1 is a control block diagram showing a first embodiment of the present invention, which corresponds to the invention described in claims 1 and 2. In the first embodiment, the relationship between the angle error θ _xerr and the x-axis high-frequency current amplitude I _xh shown in FIGS. 8 and 9 and the high-frequency voltage for controlling the x-axis high-frequency current amplitude I _xh to a command value are shown. The amplitude V _h ^** is measured, and an optimal control constant for calculating the magnetic pole position is calculated from these pieces of information. The control constants to be obtained are the high frequency voltage amplitude command value V _h ^* input to the high frequency voltage calculator 21 shown in FIG. 11 and the angle error calculation gain K _θerr input to the speed calculator 25.
FIG. 1 and FIG. 11 are substantially the same _{except for the} way of giving the δ-axis high-frequency voltage command value v _δh ^* and the angular velocity ω ₁ of the γ and δ axes. The description of the control block diagram of FIG. 1 is performed mainly on the different points from FIG. 11, and the same portions are omitted.

That is, in FIG. 1, gamma, and input to the electrical angle calculator 26 controls the angular velocity omega ₁ of the δ-axis at a constant value _ω Lθ, γ, the angle theta ₁ of the δ-axis rotating at a constant speed _ω Lθ. The bandpass filter 24 calculates the x-axis high-frequency current amplitude I _xh from the δ-axis current detection value i _{δ according} to Equation 8.

The Fourier series calculator 31 receives the x-axis high-frequency current amplitude I _xh and the angle θ _1, and the Fourier series calculator 31 has a DC component I _xha0 that does not depend on the angle θ ₁ of the x-axis high-frequency current amplitude I _xh. , The component I _xhc1 pulsating at a period of 360 degrees electrical angle and the component I _xhc2 pulsating at a period of 180 degrees electrical angle are _expressed by Equations 12 to 11, respectively, by Fourier series from zero to 180 [deg] with respect to the angle θ _1. 14 according to the calculation.
In these formulas 12 to 14, the subscripts (θ ₁ ) and (θ ₁ + π) mean the x-axis plus direction and the x-axis minus direction, respectively.

Using the calculation results of Expressions 12 to 14, a d-axis high-frequency current amplitude detection value I _dhP that is a high-frequency current amplitude when a high-frequency voltage is applied in the d-axis positive direction is obtained by Expression 15.

The high-frequency current regulator 32 adjusts the first high-frequency current regulator 32 so that the d-axis high-frequency current amplitude detection value I _dhP obtained by the Fourier series calculator 31 matches a d-axis high-frequency current amplitude command value I _dh ^{* (} not shown). The voltage amplitude command value V _h ^** is calculated.
Frequency voltage calculator 21, the amplitude is equal to the first high frequency voltage amplitude command value V _h ^**, cycle calculates the T _h at a rectangular wave of δ-axis high frequency alternating voltage command value v _.delta.h ^*. The waveform of the δ-axis high frequency alternating voltage command value v _δh ^* is as shown in FIG.

Further, the control constant calculator 33 obtains a second high-frequency voltage amplitude command from the first high-frequency voltage amplitude command value V _h ^** and the component I _xhc1 pulsating at an electrical angle of 360 degrees of the x-axis high-frequency current amplitude. A value V _h ^* and an angle error calculation gain K _θerr are obtained. Here, ^* the second high frequency voltage amplitude command value V _h, which corresponds to the high frequency voltage amplitude command value V _h ^* being input to the high frequency voltage calculator 21 in FIG. 11.

When the detection errors of the current detectors 11u and 11w are taken into consideration, the accuracy of the magnetic pole position calculation increases as the component I _xhc1 pulsating with an electrical angle of 360 degrees of the x-axis high-frequency current amplitude increases.
For this reason, the control constant calculator 33 determines that the accuracy of the magnetic pole position calculation is sufficient when the component I _xhc1 is larger than a predetermined value, and sets the second high-frequency voltage amplitude command value V _h ^* as the second high-frequency voltage amplitude command value V _h ^* . The high frequency alternating voltage command value v _δh ^* is calculated by the high frequency voltage calculator 21 using the high frequency voltage amplitude command value V _h ^** of 1 as it is.

Of course, the second constant high-frequency voltage amplitude command value V _h ^* is set to be greater than the first high-frequency voltage amplitude command value V _h ^** by the control constant calculator 33 so that the size of I _xhc1 is minimized. The noise due to the high-frequency current may be reduced by calculating a high-frequency alternating voltage command value v _δh ^* with a small setting.
On the other hand, when I _xhc1 is smaller than a predetermined value, the control constant calculator 33 causes the second high-frequency voltage amplitude command to flow so that sufficient I _xhc1 flows to obtain the required magnetic pole position calculation accuracy. it is desirable to calculate the high-frequency alternating voltage command value v _.delta.h ^* values V _h ^* first high-frequency voltage is greater than the amplitude command value V _h ^**.

Further, the control constant calculator 33 _{sets the} angle error calculation gain K _θerr as shown in Expression 16 by linear approximation in the vicinity of zero of θ _err in Expression 5.

In FIG. 1, the bandpass filter 24, the Fourier series calculator 31, the high frequency current regulator 32, the high frequency voltage calculator 21 and the like constitute the amplitude command value automatic adjusting means in claim 1, and the Fourier series calculator 31, The high-frequency current regulator 32, the control constant calculator 33, and the like constitute the control constant automatic adjusting means in claim 2.
As described above, in the first embodiment, the optimum high-frequency voltage amplitude command value V _h ^* and angle for the magnetic pole position calculation according to the component I _xhc1 that pulsates with a period of 360 degrees of the electrical angle of the x-axis high-frequency current amplitude. The error calculation gain K _θerr is calculated, and using these control constants, the magnetic pole position θ ₁ of the electric motor 80 can be accurately calculated according to the control block diagram shown in FIG.

Next, a second embodiment of the present invention will be described.
In the second embodiment, the high-frequency current regulator 32 in the first embodiment is configured by an integral regulator like the high-frequency current regulator 32A shown in FIG. Equivalent to.
That is, in FIG. 2, the deviation between the d-axis high-frequency current amplitude command value I _dh ^* and the d-axis high-frequency current amplitude detection value I _dhP is calculated by the subtractor 101, and this deviation is amplified by the integration controller 102 to be the first The high-frequency voltage amplitude command value V _h ^** is calculated.
As a result, the d-axis high-frequency current amplitude detection value I _dhP can be controlled to the command value I _dh ^* with a simple configuration.

  Furthermore, in the third embodiment of the present invention, the high-frequency current regulator 32 in the first embodiment is configured as a high-frequency current regulator 32B shown in FIG. Equivalent to. According to the third embodiment, responsiveness and stability can be ensured even when the characteristics of the controlled object are unknown, and the response of the high-frequency current can be accelerated even when the inductance of the motor 80 is unknown. it can.

In FIG. 3, the d-axis high-frequency current amplitude command value I _dh ^* is input to the divider 201 and divided by the gain estimated value Θ _est1 to calculate the first high-frequency voltage amplitude command value V _h ^** . Further, by multiplying the high frequency voltage amplitude command value _V ^{h **} and gain estimate theta _est1 by the multiplier 202 calculates the d-axis high frequency current amplitude estimate _{I dhPest.}
Further, a subtractor 203 calculates a deviation ε between the d-axis high-frequency current amplitude estimated value I _dhPest and the d-axis high-frequency current amplitude detected value I _dhP, and the gain estimator 204 amplifies the deviation ε to obtain a gain estimated value Θ. _est1 is calculated.
The specific calculation content is as the following Expression 17.

As a result of the above calculation processing, the gain estimation value Θ _est1 converges to a true value in order to make the deviation ε zero, and the high frequency current regulator 32B converts the d-axis high frequency current amplitude detection value I _dhP to the command value I _dh ^* . The high-frequency voltage amplitude command value V _h ^** for control can be output.

Finally, in the fourth embodiment of the present invention, it is determined in advance whether or not the magnetic pole position calculation using the magnetic saturation characteristics of FIG. This corresponds to the invention according to claim 5.
In actual devices, there are constraints on the maximum output voltage and maximum output current of the power converter. For this reason, due to these restrictions, the component I _xhc1 pulsating at a period of 360 ° electrical angle of the high-frequency current amplitude may not be controlled to a value sufficient to realize the magnetic pole position calculation.

Therefore, in the fourth embodiment, when I _xhc1 cannot be controlled to be larger than a predetermined value, it is determined that the magnetic pole position calculation is impossible.
When it is determined that the magnetic pole position calculation is impossible in this way, other methods such as a magnetic pole position calculation by another method or a method of starting the rotor after drawing the rotor into the current vector by controlling the current vector to be constant You can drive using the starting method.

It is a control block diagram which shows 1st Example of this invention. It is a control block diagram of the high frequency current regulator in 2nd Example of this invention. It is a control block diagram of the high frequency current regulator in 3rd Example of this invention. It is a figure which shows the definition of the coordinate axis used for a magnetic pole position calculation. It is a figure which shows the relationship between d-axis current and a linkage flux, and the relationship between d-axis current and d-axis inductance. It is a figure which shows the relationship between the angle error and x-axis inductance in SPMSM. It is a wave form diagram of x axis high frequency alternating voltage, x axis current, and x axis high frequency current amplitude in an embodiment of the present invention. It is a figure which shows the relationship between the angle error and the x-axis high frequency current amplitude in SPMSM. It is a figure which shows the relationship between the angle error in IPMSM, and x-axis high frequency current amplitude. FIG. 4 is a waveform diagram of a δ-axis high-frequency alternating voltage, a δ-axis current, an x-axis high-frequency current amplitude, and the like in the embodiment of the present invention. It is a control block diagram for performing magnetic pole position calculation using the magnetic saturation characteristic in the embodiment of the present invention.

Explanation of symbols

50 Three-phase AC power supply 60 Rectifier circuit 70 Power converter 80 Permanent magnet synchronous motor (PMSM)
11u u-phase current detector 11w w-phase current detector 12 voltage detector 13 PWM circuit 14 current coordinate converter 15 voltage coordinate converters 19a and 19b subtractor 20a γ-axis current regulator 20b δ-axis current regulator 21 high-frequency voltage calculation Unit 22 Adder 23 Notch filter 24 Band pass filter 25 Speed calculator 26 Electrical angle calculator 31 Fourier series calculator 32, 32A, 32B High frequency current regulator 33 Control constant calculator 101 Subtractor 102 Integrator regulator 201 Divider 202 Multiplier 203 Subtractor 204 Gain estimator

Claims (2)

A control device for a permanent magnet type synchronous motor having no magnetic pole position detector,

  Control equipped with magnetic pole position calculation means for calculating the magnetic pole position of the motor from the high-frequency current that flows when a high-frequency alternating voltage is applied to the motor, using the motor's terminal voltage and current as vectors and using the magnetic saturation characteristics of the motor In the device

  Amplitude command value automatic adjustment means for automatically adjusting the amplitude command value of the high-frequency alternating voltage,

  The amplitude command value automatic adjustment means is

  Means for applying a high-frequency alternating voltage of a rectangular wave composed of two pulse voltages having the same amplitude and pulse width and different polarities in a plurality of vector directions;

  Means for detecting the amplitude of a high-frequency current having the same frequency as the high-frequency alternating voltage;

  Means for detecting a d-axis high-frequency current amplitude, which is a high-frequency current amplitude when the high-frequency alternating voltage is applied in a direction parallel to the magnetic pole position of the motor, from the high-frequency current amplitude detection value detected by the means;

  high-frequency current adjusting means for calculating an amplitude command value of the high-frequency alternating voltage so that the d-axis high-frequency current amplitude detection value matches the d-axis high-frequency current amplitude command value;

  Have
The high-frequency current adjusting means includes

  High-frequency current amplitude estimating means for calculating an estimated value of the d-axis high-frequency current amplitude from the amplitude command value and gain estimated value of the high-frequency alternating voltage;

  Gain estimation means for amplifying a deviation between the estimated value of the d-axis high-frequency current amplitude and the detected value of the d-axis high-frequency current amplitude to calculate the gain estimated value;

  Means for calculating an amplitude command value of the high-frequency alternating voltage from the d-axis high-frequency current amplitude command value and the gain estimation value;

  A control device for a permanent magnet type synchronous motor, comprising: The control device according to claim 1,

  Means for extracting from the high-frequency current amplitude detection value a Fourier series first-order component that is a component that changes in a cycle of an electrical angle of 360 degrees depending on the vector direction of the high-frequency alternating voltage;

  Means for determining whether or not to calculate the magnetic pole position from the magnitude of the Fourier series primary component;

  A control device for a permanent magnet type synchronous motor, comprising:

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