JP7196469B2 - Controller for synchronous reluctance motor - Google Patents

Description

The present invention provides a synchronous reluctance motor using an alternating current supplied to the synchronous reluctance motor without a position sensor for detecting the magnetic pole position of the rotor of the synchronous reluctance motor and a speed sensor for detecting the rotational speed of the rotor. The present invention relates to a control device for a synchronous reluctance motor that controls rotation.

In Patent Document 1, the salient pole ratio of an Interior Permanent Magnet Synchronous Motor (IPMSM) is used to estimate the magnetic pole position of the rotor by applying a rectangular high-frequency alternating voltage to the IPMSM. A method is disclosed. The salient pole ratio is the ratio of the q-axis inductance component to the d-axis inductance component. The d-axis is parallel to the direction of the magnetic flux produced by the magnetic poles of the rotor, and the q-axis is parallel to the direction perpendicular to the d-axis.

JP-A-2009-273254

In the method for estimating the magnetic pole position disclosed in Patent Document 1, the magnetic pole position can be estimated when the rotor having a salient pole ratio is rotating at a low speed or when the rotor is stopped. However, in the IPMSM, when the rotor is rotating, an induced voltage is generated in the windings of the stator due to the magnetic flux generated by the permanent magnets provided in the rotor. An alternating voltage cannot be applied to the IPMSM if it is rotating at . Therefore, the method for estimating the magnetic pole position disclosed in Patent Document 1 cannot be applied to the IPMSM in which the rotor is rotating at high speed, and it is difficult to estimate the magnetic pole position. For this reason, in the magnetic pole position estimation method of Patent Document 1, in order to start a motor having a rotor when the rotor is rotating at high speed, the rotation speed of the rotor is caused by air resistance and the like. By natural deceleration, it is necessary to wait until the rotational speed is reduced or until the rotor stops rotating. Therefore, in the magnetic pole position estimation method of Patent Document 1, there is a problem that the starting of the electric motor is delayed.

SUMMARY OF THE INVENTION The present invention has been made in view of the above, and provides a synchronization method that can estimate the rotation speed of a rotor during high-speed rotation without waiting until the rotation speed of the rotor becomes low or until the rotation of the rotor stops. The object is to obtain a control device for a reluctance motor.

In order to solve the above-described problems and achieve the object, a control device for a synchronous reluctance motor according to the present invention includes a position sensor for detecting the magnetic pole position of the rotor of the synchronous reluctance motor and a speed sensor for detecting the rotational speed of the rotor. and a control unit for controlling the rotation of the synchronous reluctance motor using an alternating current supplied to the synchronous reluctance motor. When a high-frequency voltage is applied to the synchronous reluctance motor, the control unit of the synchronous reluctance motor controller controls a dq-axis orthogonal rotating coordinate system consisting of the d-axis in the magnetic flux axis direction of the rotor and the q-axis orthogonal to the d-axis. γ-axis high-frequency current amplitude indicating the amplitude of the high-frequency current on the γ-axis in the γδ orthogonal rotating coordinate system on the control different from the δ-axis high-frequency current amplitude indicating the amplitude of the high-frequency current on the δ-axis orthogonal to the γ-axis , an estimating unit for estimating a second rotational speed of the rotor in the dq-axis orthogonal rotating coordinate system based on the first rotational speeds of the γ-axis and the δ-axis in the γδ orthogonal rotating coordinate system.

According to the present invention, it is possible to estimate the rotation speed of the rotor during high-speed rotation without waiting until the rotation speed of the rotor becomes low or until the rotation of the rotor stops.

1 is a configuration diagram of a control device 200 for a synchronous reluctance motor according to an embodiment of the present invention; Diagram for explaining dq axis and γδ axis 7 is a diagram showing the γ-axis high-frequency alternating voltage v _γh , the γ-δ-axis current detection values i _γ and i _δ , and the γ-δ-axis high-frequency current amplitudes I _γh and I _δh with respect to the relative phase θ _err ; FIG. 1 showing enlarged high-frequency alternating voltage _vγh and _δ -axis current detection value iδ shown in FIG. FIG. 2 is an enlarged view of the high-frequency alternating voltage _vγh and the _δ -axis current detection value iδ shown in FIG. Time chart for explaining the operation of the relative velocity/phase calculator 15 A diagram showing the state of the δ-axis high-frequency current amplitude _{I δh} when the γδ-axis rotation speed ω1 is changed while the rotor is stopped. FIG. 4 is a diagram showing a configuration example of the relative velocity/phase calculator 15 when controlling the γδ _- axis rotation speed ω1 by feedback control; A diagram showing waveforms of γδ-axis high-frequency current information Iγ _{(HPF) and Iδ(HPF) obtained} as a result of passing the γδ-axis high-frequency current amplitudes _Iγh and _Iδh shown in FIGS. 1 and 3 through a high-pass filter. A diagram showing a first configuration example of the relative velocity/phase calculator 15 that calculates the relative velocity ω _err and the relative phase θ _err using the γδ-axis high-frequency current information I _{γ (HPF)} and I _{δ (HPF)} shown in FIG. A diagram showing a second configuration example of the relative velocity/phase calculator 15 that calculates the relative velocity ω _err and the relative phase θ _err using the γδ-axis high-frequency current information I _{γ (HPF)} and I _{δ (HPF)} shown in FIG. FIG. 2 is a diagram showing a hardware configuration example of the control unit 100 shown in FIG. 1;

A control device for a synchronous reluctance motor according to an embodiment of the present invention will be described below in detail with reference to the drawings. In addition, this invention is not limited by this embodiment.

Embodiment 1.
FIG. 1 is a configuration diagram of a control device 200 for a synchronous reluctance motor according to an embodiment of the present invention. The control device 200 for the synchronous reluctance motor shown in FIG. The control device includes a control unit 100 for controlling the rotation of the synchronous reluctance motor 1 using an alternating current supplied to the . Hereinafter, the control device 200 of the synchronous reluctance motor may be simply referred to as the control device 200 in some cases. Also, the synchronous reluctance motor 1 may be simply referred to as the motor 1 in some cases.

The control device 200 rectifies the AC power supplied from the three-phase AC power supply 4 and outputs the rectified power as DC power, and the DC power supplied from the rectification circuit 3 drives the motor 1. and a power converter 2 that converts the AC power into AC power for output. Control device 200 also includes a control unit 100 that controls on/off operations of a plurality of semiconductor switching elements provided in power converter 2, a U-phase current detector 6u, and a W-phase current detector 6w.

One end of the three-phase AC wiring 50 is connected to the power converter 2 , and the other end of the three-phase AC wiring 50 is connected to the motor 1 . Of the three-phase AC wiring 50, the U-phase wiring is provided with a U-phase current detector 6u for detecting the current flowing through the U-phase wiring, and the W-phase wiring is provided with a W-phase current detector 6u for detecting the current flowing through the W-phase wiring. A current detector 6w is provided. The U-phase current detector 6u and the W-phase current detector 6w are, for example, CTs (Current Transformers). Each of the U-phase current detector 6u and the W-phase current detector 6w converts the detected current into a signal indicating the value of the current and outputs the signal. iu represents a signal output from the _U -phase current detector 6u. In the following, it will be referred to as _U -phase current iu. iw represents a signal output from the _W -phase current detector 6w. In the following, it will be referred to as _W -phase current iw.

The reason why the two U-phase current detectors 6u and W-phase current detectors 6w are used is that the relationship of U-phase current+V-phase current+W-phase current=0 is established, and the remaining This is because the current of one phase can also be obtained. Therefore, the control device 200 may be configured to detect currents flowing through at least two of the U-phase wiring, the V-phase wiring, and the W-phase wiring. Note that the control device 200 detects the current flowing through the positive or negative bus (not shown) inside the power converter 2 instead of the U-phase current detector 6u and the W-phase current detector 6w, or detects the current flowing through the power converter. A current detector that detects a current flowing through a switching element (not shown) that constitutes 2 may be used. Since the respective configurations of the rectifier circuit 3 and the power converter 2 are known, the description thereof is omitted.

The configuration of the control unit 100 will be described in detail below. First, the rotating coordinate system will be explained. FIG. 2 is a diagram for explaining the dq axis and the γδ axis. The dq-axes are orthogonal axes that define a dq-axis orthogonal rotating coordinate system that rotates in synchronization with the rotor of the motor 1 . The d-axis is parallel to the direction of the magnetic flux produced by the magnetic poles of the rotor. The q-axis is an axis that is 90 degrees ahead of the d-axis. Further, in sensorless control in which the magnetic pole position of the rotor and the rotation speed of the rotor are not directly detected, the γδ axes are defined separately from the dq axes. The γδ axes are orthogonal axes that define a control γδ orthogonal rotating coordinate system different from the dq-axis orthogonal rotating coordinate system. The δ-axis is an axis advanced by 90 degrees from the γ-axis.

ω1 is the γδ axis rotation speed indicating the _first rotation speed of the γδ axis of the γδ orthogonal rotating coordinate system. _ωr is the dq-axis rotation speed indicating the rotation speed of the dq-axes. θ1 is the γδ-axis phase indicating the phase of the γδ-axis with respect to the position of the U-phase winding provided in the motor ₁ . _θr is the dq-axis phase indicating the phase of the dq-axis with respect to the position of the U-phase winding provided in the motor 1 . ω _err is the relative velocity between the γ-axis and the d-axis. The relative speed ω _err is equal to the speed difference between the dq-axis rotation speed ω _r and the γδ-axis rotation speed ω ₁ . θ _err is the relative phase between the γ-axis and the d-axis. The relative phase θ _err is equal to the phase difference between the γδ-axis phase θ ₁ and the dq-axis phase θ _r . The dq-axis phase _θr is equal to the actual rotor pole position.

The dq-axis rotation speed _ωr and the dq-axis phase _θr shown in FIG. _The γδ axis rotation speed ω1 and the γδ axis _phase θ1 can be arbitrarily selected. Therefore, by obtaining the relative speed ω _err and the relative phase θ _err , the dq-axis rotation speed ω _r and the dq-axis phase θ _r can be calculated.

The state equation of the high-frequency component generated when the γ-axis high-frequency alternating voltage v _γh is applied while the motor 1 is stopped can be expressed by Equation 3 if the armature resistance can be ignored. The γ-axis high-frequency alternating voltage v _γh is a high-frequency alternating voltage applied to windings provided in the motor 1 . In FIG. 1, illustration of the symbol “v _γh ” indicating the γ-axis high-frequency alternating voltage is omitted. The γ-axis high-frequency alternating voltage v _γh is a rectangular-wave voltage generated in a direction parallel to the γ-axis. The γ-axis high-frequency alternating voltage command value v _γh ^* may be simply referred to as v _γh ^* . In Equation 3, p is the differential operator, Ld is the _d -axis inductance, and Lq is the _q -axis inductance. L ₀ can be obtained by L ₀ =1/2(L _d +L _q ). L ₂ can be obtained by L ₂ =1/2(L _d −L _q ). The relative phase θ _err can be obtained by θ _err =θ _r −θ ₁ .

The γδ-axis high-frequency current amplitudes I _γh and I _δh , which are the amplitudes of the γδ-axis high-frequency currents flowing at this time, can be obtained by Equation (4). As shown in Equation 4, the γδ-axis high-frequency current amplitudes I _γh and I _δh can be obtained by integrating the state equation shown in Equation 3 over the period Th of v _γh ^* . The γδ-axis high-frequency current amplitudes I _γh and I _δh are equal to the amplitudes of the γδ-axis current detection values i _γ and i _δ . The γ-axis high-frequency current amplitude I _γh represents the amplitude of the high-frequency current on the γ-axis in the γδ orthogonal rotating coordinate system, and the δ-axis high-frequency current amplitude I _δh represents the amplitude of the high-frequency current on the δ-axis in the γδ orthogonal rotating coordinate system. show. The period Th is equal to one period of v _γh ^* and equal to one period of the high frequency alternating voltage v _γh . γ-axis current detection value i _γ is the current on the γ-axis. The δ-axis current detection value i _δ is the current on the δ-axis orthogonal to the γ-axis.

From Equation 4, it can be seen that when the high-frequency alternating voltage _vγh is applied in the direction parallel to the γ-axis, the δ-axis high-frequency current amplitude _Iδh changes at a cycle twice the relative phase _θerr . The same applies to the γ-axis high-frequency current amplitude _Iγh .

3 to 5, the relationship among the high-frequency alternating voltage v _γh applied to the windings provided in the motor 1, the γδ-axis current detection values i _γ and i _δ , and the γδ-axis high-frequency current amplitudes I _γh and I _δh Explain gender.

FIG. 3 shows the high frequency alternating voltage v _γh , the γδ axis current detection values i _γ and i _δ , and the γδ axis high frequency current amplitudes I _γh and I _δh with respect to the relative phase θ _err . The horizontal axis of FIG. 3 represents the relative phase θ _err . FIG. 3 shows, from top to bottom, the γ-axis high-frequency alternating voltage _vγh , the γδ-axis current detection values _iγ and _iδ , and the γδ-axis high-frequency current amplitudes _Iγh and _Iδh . The γ-axis current detection value iγ is equal to the high-frequency current on the _γ -axis estimated when the γ-axis high-frequency alternating voltage _vγh is applied. The δ-axis current detection value i _δ is equal to the high-frequency current on the δ-axis estimated when the δ-axis high-frequency alternating voltage v _δh is applied. Vγ shown in FIG. 3 is the amplitude of the _γ -axis high-frequency alternating voltage _vγh . The δ-axis high-frequency current amplitude I _δh shown in FIG. 3 can be calculated based on the deviation of the δ-axis high-frequency current i _δ corresponding to the timing at which the polarity of the high-frequency alternating voltage v _γh changes. A method for calculating the δ-axis high-frequency current amplitude I _δh will be specifically described with reference to FIGS. 4 and 5. FIG.

FIG. 4 is an enlarged first diagram showing the high-frequency alternating voltage _vγh and the _δ -axis current detection value iδ shown in FIG. FIG. 5 is a second diagram showing an enlarged view of the high-frequency alternating voltage _vγh and the _δ -axis current detection value iδ shown in FIG.

Here, the polarity of the δ-axis high-frequency current amplitude I _δh shown in Equation 4 changes positively or negatively as shown in FIG. When the polarity of the δ-axis high-frequency current amplitude I _δh is positive, as shown in FIG. 4, the polarity of the high-frequency alternating voltage v _γh and the polarity of the change rate of the δ-axis current detection value i _δ are the same. Specifically, as shown in FIG. 4, the time point at which the value of the high-frequency alternating voltage _vγh changes from negative to positive is t1, and a time corresponding to half the cycle Th of the high-frequency alternating voltage _vγh has elapsed from t1. When the polarity of the δ-axis current detection value i _δ with respect to time t on the horizontal axis is positive during t1 to t2 when the polarity of the high-frequency alternating voltage v _γh is positive when the time point is t2, the δ-axis high-frequency The polarity of the current amplitude I _δh becomes positive.

On the other hand, when the polarity of the _δ -axis high-frequency current amplitude I _δh is negative, as shown in FIG _. . Specifically, as shown in FIG. 5, the time point at which the value of the high-frequency alternating voltage _vγh changes from negative to positive is defined as t1, and a time corresponding to half the cycle Th of the high-frequency alternating voltage _vγh has elapsed from t1. When the polarity of the δ-axis current detection value i _δ with respect to time t on the horizontal axis is negative during t1 to t2 when the polarity of the high-frequency alternating voltage v _γh is positive when the time is t2, the δ-axis high-frequency The polarity of the current amplitude I _δh becomes negative.

Note that the γ-axis high-frequency current amplitude I _γh shown in FIG. 3 is the deviation of the γ-axis current detection value i _γ corresponding to the timing at which the polarity of the high-frequency alternating voltage v _γh changes, similarly to the δ-axis high-frequency current amplitude I _δh . can be calculated based on

Based on the fact that the high-frequency alternating voltage v _γh , the γδ-axis current detection values i _γ and i _δ , and the γδ-axis high-frequency current amplitudes I _γh and I _δh have the relationships shown in Equations 3 and 4, Next, the functions of the control unit 100 shown in FIG. 1 will be specifically described.

The control unit 100 includes a current coordinate conversion unit 5, a notch filter 7, a rectangular wave generation unit 10, a γ-axis current control unit 9a, a δ-axis current control unit 9b, a bandpass filter 14, a relative velocity/phase calculation unit 15, voltage coordinates. It includes a conversion unit 12, a PWM (Pulse Width Modulation) signal generation unit 13, an integration unit 16a, a subtraction unit 8a, a subtraction unit 8b, an addition unit 11a, an addition unit 11b, and an addition unit 11c. Below, the relative velocity/phase calculator 15 may be simply referred to as the calculator 15 .

Based on the γ-axis high-frequency current amplitude I _γh and the δ-axis high-frequency current amplitude I _δh , the calculation unit 15, the integration unit 16a, the addition unit 11b, and the addition unit 11c calculate the γ-axis phase θ ₁ , the dq-axis rotation speed ωr, and the dq-axis rotation speed _ωr and dq. An estimator 40 for estimating the axial phase _θr is constructed.

Based on the γδ-axis phase θ ₁ , the current coordinate transformation unit 5 coordinates-transforms the U-phase current i _u and the W-phase current i _w into the γδ-axis current detection values i _γ and i _δ . A coordinate conversion method by the current coordinate conversion unit 5 is described in Japanese Patent No. 3860031, for example.

The notch filter 7 receives the γ-axis current detection value _iγ and the _δ -axis current detection value iδ. The notch filter 7 removes the high-frequency current component that flows when the high-frequency alternating voltage is applied to the motor 1 from the γ-axis current detection value i _γ and the δ-axis current detection value i _δ . Currents are output as γδ-axis fundamental wave currents i _γf and i _δf .

The γ-axis fundamental wave current i _γf output from the notch filter 7 and the γ-axis current command value i _γ ^* are input to the subtractor 8a. The γ-axis current command value i _γ ^* is a command value for the current that should flow in the γ-axis direction, and is output from, for example, a host controller. The host controller is a function setting tool such as a touch panel attached to the power converter 2, for example. In this embodiment, the γ-axis current command value i _γ ^* is set to zero. The subtractor 8a calculates the deviation between the γ-axis current command value i _γ ^* and the γ-axis fundamental wave current i _γf , and outputs information indicating the calculated deviation. The γ-axis current control unit 9a receives information indicating the deviation output from the subtraction unit 8a. The γ-axis current control unit 9a amplifies the deviation to calculate the γ-axis fundamental wave voltage command value v _γf ^* .

The δ-axis fundamental wave current i _δf output from the notch filter 7 and the δ-axis current command value i _δ ^* are input to the subtractor 8b. The δ-axis current command value i _δ ^* is a command value for the current that should flow in the δ-axis direction, and is output from, for example, a host controller. In this embodiment, the δ-axis current command value i _δ ^* is set to zero. The subtractor 8b calculates the deviation between the δ-axis fundamental wave current i _δf and the δ-axis current command value i _δ ^* , and outputs information indicating the calculated deviation. Information indicating the deviation output from the subtraction unit 8b is input to the δ-axis current control unit 9b. The δ-axis current control unit 9b amplifies the deviation to calculate the δ-axis fundamental wave voltage command value v _δf ^* .

The rectangular wave generator 10 calculates v _γh ^* based on the γ-axis high-frequency voltage amplitude command value V _γh ^* . The γ-axis high-frequency voltage amplitude command value V _γh ^* is a signal for setting the amplitude of v _γh ^* . v _γh ^* is a rectangular signal whose amplitude is equal to the amplitude of the γ-axis high-frequency voltage amplitude command value V _γh ^* and whose period is equal to Th. The γ-axis high-frequency voltage amplitude command value V _γh ^* is output from, for example, a host controller.

v _γh ^* calculated by the rectangular wave generator 10 is input to the adder 11a. In addition to v _γh ^* , the γ-axis fundamental wave voltage command value v _γF ^* output from the γ-axis current control unit 9a is also input to the adder 11a. The adder 11a obtains the γ-axis voltage command value v _γ ^* by superimposing v _γh ^* on the γ-axis fundamental wave voltage command value v _γF ^* .

The γ-axis voltage command value v _γ ^* obtained by the addition section 11 a is input to the voltage coordinate conversion section 12 . In addition to the γ-axis voltage command value v _γ ^* , the δ-axis voltage command value v _δ ^* and the γδ-axis phase θ ₁ are input to the voltage coordinate conversion unit 12 . The δ-axis voltage command value v _δ ^* is equal to the δ-axis fundamental wave voltage command value v _δf ^* output from the δ-axis current control section 9b. Details of the voltage coordinate conversion unit 12 and the PWM signal generation unit 13 will be described later.

The γ-axis current detection value i _γ and the δ-axis current detection value i _δ output from the current coordinate conversion unit 5 are input to the band-pass filter 14 . The band-pass filter 14 calculates the high-frequency component of the _γ -axis current detection value iγ having the same frequency as the high-frequency alternating voltage applied to the motor 1, and converts the high-frequency current of the calculated _γ -axis current detection value iγ to the γ-axis Output as high frequency current amplitude I _γh . The γ-axis high-frequency current amplitude I _γh is, for example, a δ-axis current detection value detected during the period from time t1 to time t2 when the polarity of the rectangular γ-axis high-frequency alternating voltage command value v _γh ^* shown in FIG. 4 changes. It is calculated based on the amplitude of i _δ .

Further, the band-pass filter 14 calculates a high-frequency component of the δ-axis current detection value i _δ having the same frequency as the high-frequency alternating voltage, and converts the high-frequency current of the calculated δ-axis current detection value i _δ into a δ-axis high-frequency current amplitude Output as I _δh . The positive or negative polarity of the δ-axis high-frequency current amplitude I _δh is given to the δ-axis high-frequency current amplitude I _δh . Thus, the bandpass filter 14 functions as a current amplitude calculator.

The γ-axis high-frequency current amplitude I _γh and the δ-axis high-frequency current amplitude I _δh are input to the calculation unit 15 . The computing unit 15 computes the γδ-axis rotation speed ω ₁ , the relative speed ω _err , and the relative phase θ _err based on the γ-axis high-frequency current amplitude I _γh and the δ-axis high-frequency current amplitude I _δh . The details of the configuration of the calculation unit 15 will be described later.

The γδ-axis rotation speed ω1 calculated by the calculation unit ₁₅ is input to the integration unit 16a, and the integration unit 16a calculates the γδ _- axis phase θ1 by integrating the γδ _- axis rotation speed ω1.

The γδ-axis rotation speed ω ₁ and the relative speed ω _err calculated by the calculation unit 15 are input to the addition unit 11b. The adder 11b calculates the dq-axis rotation speed _ωr by adding the relative speed _ωerr to the γδ-axis rotation speed ω1, as shown in Equation ₁ above.

The adder 11c receives the relative phase θ _err calculated by the calculator 15 and the γδ-axis phase θ ₁ calculated by the integrator 16a. The adder 11c calculates the dq-axis phase _θr by adding the relative phase _θerr to the γδ _- axis phase θ1 as shown in Equation 2 above.

Based on the γδ-axis phase θ ₁ , the voltage coordinate conversion unit 12 converts the γ, δ-axis voltage command values v _γ ^* , v _δ ^* into the three-phase voltage command values v _u ^* , v _v ^* , v _w ^*. Convert to The phase voltage command values v _u ^* , v _v ^* , v _w ^* converted by the voltage coordinate conversion section 12 are input to the PWM signal generation section 13 . PWM signal generator 13 generates PWM signal 13 a for controlling the ON/OFF operation of the semiconductor switching element in power converter 2 . The PWM signal 13a is a rectangular wave signal that takes a binary value of high level or low level. The PWM signal 13a is input to a drive signal generator in the power converter 2, for example. The drive signal generator amplifies the PWM signal 13 a to a voltage capable of driving the semiconductor switching elements in the power converter 2 . The amplified signal is input to the inverter circuit in power converter 2 as a drive signal. The inverter circuit is provided with, for example, six semiconductor switching elements. When the semiconductor switching element is a bipolar transistor, the drive signal output from the drive signal generator is input to the base terminal of the bipolar transistor. When the semiconductor switching element is a field effect transistor, the drive signal output from the drive signal generator is input to the gate terminal of the field effect transistor. AC power is supplied to the motor 1 and the motor 1 rotates by turning ON/OFF the semiconductor switching element in the power converter 2 according to the drive signal.

Next, the configuration and operation of the calculation unit 15 will be described in detail. First, a method of calculating the γδ _- axis rotation speed ω1 by the calculation unit 15 will be described with reference to FIGS. 6 to 8. FIG.

FIG. 6 is a time chart for explaining the operation of the relative velocity/phase calculator 15. As shown in FIG. FIG. 6 shows the speed search operation by the relative speed/phase calculator 15 when the rotor of the synchronous reluctance motor 1 is rotating. FIG. 7 is a diagram showing the state of the δ-axis high-frequency current amplitude _{I δh} when the γδ-axis rotation speed ω1 is changed while the rotor is stopped. FIG. 7 shows the characteristics of the δ-axis high-frequency current amplitude I _δh when the rotor of the synchronous reluctance motor 1 is stopped. When the maximum value of the dq-axis rotation speed ω _r is +ω _rMax and the minimum value of the dq-axis rotation speed ω _r is −ω _rMax , the γδ-axis rotation speed ω ₁ is changed from +ω _rMax to − The waveform of the δ-axis high-frequency current amplitude I _δh when changed to ω _rMax changes as shown in FIG. When the relative speed ω _err becomes zero (ω _err =0), the δ-axis high-frequency current amplitude I _δh becomes maximum. The timing at which the relative speed ω _err becomes zero is equal to the timing at which the dq-axis rotation speed ω _r coincides with the γδ-axis rotation speed ω ₁ . The relative velocity ω _err shown in FIG. 8 is calculated by functions provided in the calculation section 15 shown in FIGS. 10 and 11, which will be described later.

Using such a property, the calculation _unit 15 _measures the _absolute value | _{I δh} | Set the velocity search value ω _r0 . The absolute value |I _δh | _Max is the absolute value of the maximum value of the δ-axis high-frequency current amplitude I _δh . The search value ω _r0 is the δ-axis high-frequency current amplitude I measured when the γδ-axis rotation speed ω1 is _{changed from +ω rMax to} −ω _rMax while the rotor of the motor 1 is rotating, as shown in FIG. The absolute value of _δh corresponds to the γδ-axis rotation speed ω ₁ when the absolute value |I _δh | _Max .

Next, a method of controlling the γδ _- axis rotation speed ω1 using the search value _ωr0 will be described. Since a high-pass filter is applied to the γδ-axis high-frequency current amplitudes I _γh and I _δh in the calculation of the relative velocity ω _err which will be described later, the γδ-axis high-frequency current amplitudes I _γh and I _δh must be AC quantities. Furthermore, in the control of the γδ _- axis rotation speed ω1 using the search value _ωr0 , the relative speed _ωerr is set to a relative speed command indicating any relative speed value other than 0 in order to generate the relative speed _ωerr . It is necessary to control the γδ-axis rotational speed ω ₁ to be equal to the value ω _err ^* . The relative speed command value ω _err ^* is information output from the host controller.

Two control methods are conceivable for controlling the γδ _- axis rotation speed ω1 using the search value _ωr0 . First, a simple _first control method for the γδ-axis rotation speed ω1 using the search value _ωr0 will be described. The calculation unit 15 can calculate the γδ-axis rotational speed ω ₁ by calculating Equation 5, that is, by adding the relative speed command value ω _err ^* to the search value ω _r0 . The calculation unit 15 uses the γδ-axis rotation speed ω1 calculated in this way as _a set value of the γδ-axis rotation speed _{ω1 used in the calculation of the relative speed ωerr ,} which will be described later.

According to this _first control method of the γδ-axis rotation speed _ω1 , the γδ _- axis rotation speed ω1 can be easily controlled. The velocity ω _err can become zero.

Next, _a second method of controlling the γδ-axis rotation speed ω1 by feedback control will be described with reference to FIG. FIG. 8 is a diagram showing a configuration example of the relative velocity/phase calculator 15 when controlling the γδ _- axis rotation speed ω1 by feedback control. The calculation unit 15 shown in FIG. 8 includes a subtraction unit 8c, a relative speed control unit 17, and an addition unit 11d. The relative speed command value ω _err ^* and the relative speed ω _err are input to the subtraction unit 8c. The subtraction unit 8c calculates the difference between the relative speed command value ω _err ^* and the relative speed ω _err by subtracting the relative speed ω _err from the relative speed command value ω _err ^* . The calculated difference is input to the relative speed control section 17 .

The relative speed control unit 17 is, for example, a PI (Proportional Integral) controller, and performs PI control on the input difference. A control result by the relative speed control unit 17 is input to the addition unit 11d. The relative speed control unit 17 is not limited to a PI controller as long as it can stabilize the input feedforward compensation information, and may be a PID (Proportional Integral Differential) controller, for example.

In addition to the control result by the relative speed control unit 17, the search value _ωr0 is also input to the addition unit 11d. _The addition unit 11d adds the search value ω _r0 to the control result of the relative speed control unit 17 as feedforward compensation, and outputs the result as the γδ axis rotation speed ω1. That is, the γδ _- axis rotation speed ω1 is calculated by adding the search value _ωr0 to the output of the feedback control by the relative speed control unit 17 as feedforward compensation. In this manner, the calculation unit 15 shown in FIG. 8 sets the γδ _- axis rotation speed ω1 by feedback control. According to the calculation unit 15 shown in FIG. 8, the relative speed ω _err can be made to match the relative speed command value ω _err ^* even when the rotor rotational speed fluctuates, the search value ω _r0 is measured, or the like occurs. , it is possible to suppress the relative velocity ω _err from becoming zero.

Next, a method of calculating the relative velocity ω _err and the relative phase θ _err will be described. It is clear that the position information of the magnetic poles of the motor 1 can be obtained from the γδ-axis high-frequency current amplitudes I _γh and I _δh that change as shown in FIG. Therefore, the calculation unit 15 of the control device 200 passes the γδ-axis high-frequency current amplitudes _Iγh and _Iδh through a high-pass filter, and uses the resulting γδ-axis high-frequency current information Iγ _(HPF) and _Iδ(HPF) . to calculate the magnetic pole position of the rotor.

FIG. 9 is a diagram showing waveforms of γδ-axis high-frequency current information I _γ(HPF) and I _δ(HPF) obtained as a result of passing the γδ-axis high-frequency current amplitudes I _γh and I _δh shown in FIGS. 1 and 3 through a high-pass filter. is. The upper part of FIG. 9 shows the waveform of the γ-axis high-frequency current information _Iγ(HPF) . The lower part of FIG. 9 shows the waveform of the δ-axis high-frequency current information I _{δ (HPF)} . The horizontal axis of FIG. 9 represents the relative phase θ _err .

As shown in FIG. 9, considering the phase advance of the δ-axis high-frequency current information I _{δ (HPF)} with respect to the γ-axis high-frequency current information I _{γ (HPF)} obtained as a result of passing through a high-pass filter, the relative phase θ _err is , can be calculated by Equation 6. ω _c is the cut-off angular frequency of the high-pass filter.

FIG. 10 is a first configuration example of the relative velocity/phase calculator 15 that calculates the relative velocity ω _err and the relative phase θ _err using the γδ-axis high-frequency current information I _{γ (HPF)} and I _{δ (HPF)} shown in FIG. It is a figure which shows. The relative velocity/phase calculator 15 shown in FIG. 10 includes a high-pass filter 18 a, a relative phase calculator 19 a, and a differentiator 20 . The γδ-axis high-frequency current amplitudes I _γh and I _δh are input to the high-pass filter 18a, and the high-pass filter 18a outputs γδ-axis high-frequency current information I _γ(HPF) and I _δ(HPF) . The γδ-axis high-frequency current information I _γ(HPF) and I _δ(HPF) are input to the relative phase calculator 19a, and the relative phase calculator 19a calculates the relative phase θ _err based on Equation (6). The relative phase θ _err is output to the adder 11c shown in FIG.

The differentiator 20 calculates the relative velocity ω _err by differentiating the relative phase θ _err calculated by the relative phase calculator 19a. The relative velocity ω _err is output to the adder 11b shown in FIG. 1 and also output to the subtractor 8c shown in FIG. By differentiating the relative phase θ _err , it is possible to easily calculate the relative velocity ω _err using the relative phase θ _err .

FIG. 11 is a second configuration example of the relative velocity/phase calculator 15 that calculates the relative velocity ω _err and the relative phase θ _err using the γδ-axis high-frequency current information I _{γ (HPF)} and I _{δ (HPF)} shown in FIG. It is a figure which shows. 11 includes a high-pass filter 18b, a relative phase calculator 19b, and a phase locked loop 30 instead of the relative phase calculator 19a shown in FIG.

In the calculation unit 15 shown in FIG. 11, the γδ-axis high-frequency current information I _γ(HPF) and I _δ(HPF) output from the high-pass filter 18b are input to the relative phase calculation unit 19b. The relative phase calculator 19b calculates the relative phase θ _err based on Equation 6, and outputs the calculated relative phase θ _err as the first relative phase θ _err0 . The relative phase θ _err0 is input to the subtractor 8 d provided in the phase synchronization circuit 30 . In addition to the relative phase θ _err0 , the relative phase θ _err is input to the subtraction unit 8d. The relative phase θ _err is phase information calculated by the integrator 16 b included in the phase synchronization circuit 30 . The subtractor 8 d calculates the difference between the relative phase θ _err0 and the relative phase θ _err , and information on the calculated difference is input to the relative speed estimator 21 .

The relative speed estimator 21 is, for example, a PI controller, and performs PI control on the input difference information. The relative velocity ω _err , which is the result of control by the relative velocity estimator 21, is output to the adder 11b shown in FIG. 1 and input to the integrator 16b.

The integrator 16b integrates the relative velocity ω _err to calculate the second relative phase, which is feedback information synchronized with the relative phase θ _err0 , as the relative phase θ _err . The relative phase θ _err is output to the adder 11c shown in FIG. According to the calculation unit 15 shown in FIG. 11, it is possible to obtain the effect of improving resistance to noise (phase noise) as compared with the calculation unit 15 shown in FIG.

FIG. 12 is a diagram showing a hardware configuration example of the control unit 100 shown in FIG. As shown in FIG. 12, the control section 100 can be implemented by a microcomputer 110 shown in FIG. The microcomputer 110 includes a memory 121, a CPU (Central Processing Unit) 122, an AD (Analog to Digital) conversion section 123, a PWM module 124, and a communication section 125 for communicating with an upper controller. The CPU 122 is a processor that controls the control unit 100 . A processor is a processing unit, an arithmetic unit, a microprocessor, a microcomputer or a DSP (Digital Signal Processor). If the host controller described above is, for example, a function setting tool such as a touch panel attached to the inverter, the communication unit 125 receives the γ-axis high-frequency voltage amplitude command value V _γh ^* output from the host controller, the relative speed command A value ω _err ^* is entered.

In the configuration example shown in FIG. 12, the functions of the control unit 100 are implemented by software, the functions of the control unit 100 are implemented by firmware, or the functions of the control unit 100 are implemented by a combination of software and firmware. Software and firmware are written as programs, and the programs are stored in the memory 121 . The memory 121 may be volatile or non-volatile such as RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory) and EEPROM (Electrically Erasable Programmable Read Only Memory), or Read Only Memory. semiconductor memories, magnetic discs, flexible discs, optical discs, compact discs, mini discs, and DVDs (Digital Versatile Discs).

The functions of the control unit 100 are realized by the CPU 122 reading and executing the programs stored in the memory 121 . Functions of the control unit 100 include, for example, the current coordinate conversion unit 5, the notch filter 7, the rectangular wave generation unit 10, the γ-axis current control unit 9a, the δ-axis current control unit 9b, the bandpass filter 14, and the calculation unit shown in FIG. 15, a voltage coordinate conversion unit 12, a PWM signal generation unit 13, an integration unit 16a, a subtraction unit 8a, a subtraction unit 8b, an addition unit 11a, an addition unit 11b and an addition unit 11c.

As described above, the estimation unit 40 of the control device 200 according to the present embodiment uses the γ-axis high-frequency current amplitude _Iγh , the δ-axis high-frequency current amplitude _Iδh , and the γδ-axis rotation speed ω ₁ , the dq-axis rotation speed _ωr , which is the second rotation speed of the rotor in the dq-axis orthogonal rotation coordinate system, is estimated. According to the estimating unit 40, the γδ orthogonal rotation generated when a high-frequency rectangular-wave alternating voltage is applied from the γδ orthogonal rotation coordinates rotating at an arbitrary speed to the motor 1 having a rotating rotor. Based on the γ-axis high-frequency current amplitude I _γh and the δ-axis high-frequency current amplitude I _δh on the coordinates, the rotational speed and magnetic pole position of the rotating rotor can be estimated. This makes it possible to estimate the rotation speed and the magnetic pole position of the rotor even when the rotor having a salient pole ratio rotates at high speed. Therefore, the rotation speed of the rotor naturally decelerates, so that the motor 1 can be started without waiting for the rotation speed to become low, or the motor 1 can be started without waiting for the rotation of the rotor to stop. As a result, delay in starting the motor 1 can be suppressed. Note that starting is the timing at which application of voltage for generating torque to the motor 1 is started after the search operation described above is completed.

Further, the estimation unit 40 of the control device 200 calculates the relative phase θ _err indicating the phase difference between the γ-axis phase and the d-axis phase based on the δ-axis high-frequency current amplitude I _δh and the γ-axis high-frequency current amplitude I _γh. Based on the relative phase θ _err , a relative speed ω _err indicating the speed difference between the first rotation speed γδ axis rotation speed ω ₁ and the second rotation speed dq axis rotation speed ω _r is calculated. The γδ-axis rotation speed ω ₁ is controlled so that the speed ω _err becomes equal to the relative speed command value ω _err ^* indicating a relative speed value other than 0, and the γ-axis phase θ ₁ and the relative phase ω _err is used to estimate the dq-axis phase θr, which is the magnetic pole position, and the dq-axis rotation speed _ωr is estimated based on the first rotation speed and the relative speed _{ω err} . As a result, even when a rotor having a salient pole ratio rotates at high speed, the dq-axis phase _θr , which is the magnetic pole position of the rotor, and the dq-axis rotation speed _ωr , which is the rotation speed of the rotor, can be estimated. can do. Therefore, the speed of the motor 1 can be controlled using the dq-axis rotation speed _ωr and the dq-axis rotation speed _ωr without causing a start delay of the motor 1 .

The configuration shown in the above embodiment shows an example of the content of the present invention, and it is possible to combine it with another known technology, and one configuration can be used without departing from the scope of the present invention. It is also possible to omit or change the part.

1 synchronous reluctance motor 2 power converter 3 rectifier circuit 4 three-phase AC power supply 5 current coordinate converter 6u U-phase current detector 6w W-phase current detector 7 notch filter 8a, 8b, 8c, 8d subtraction unit, 9a γ-axis current control unit, 9b δ-axis current control unit, 10 rectangular wave generation unit, 11a, 11b, 11c, 11d addition unit, 12 voltage coordinate conversion unit, 13 PWM signal generation unit, 13a PWM signal, 14 band-pass filter 15 relative velocity/phase calculator 16a, 16b integrator 17 relative velocity controller 18a, 18b high-pass filter 19a, 19b relative phase calculator 20 differentiator 21 relative velocity estimator 30 Phase lock circuit 40 estimating unit 50 three-phase AC wiring 100 control unit 110 microcomputer 121 memory 122 CPU 123 conversion unit 124 PWM module 125 communication unit 200 synchronous reluctance motor control device.

Claims (3)

A synchronous reluctance motor is operated by using an alternating current supplied to the synchronous reluctance motor without a position sensor for detecting the magnetic pole position of the rotor of the synchronous reluctance motor and a speed sensor for detecting the rotational speed of the rotor. A control device for a synchronous reluctance motor comprising a control unit for controlling rotation,
The control unit
When a high-frequency voltage is applied to the synchronous reluctance motor, γδ orthogonal rotation on control different from the dq-axis orthogonal rotation coordinate system consisting of the d-axis in the magnetic flux axis direction of the rotor and the q-axis orthogonal to the d-axis γ-axis high-frequency current amplitude indicating the amplitude of the high-frequency current on the γ-axis in the coordinate system, δ-axis high-frequency current amplitude indicating the amplitude of the high-frequency current on the δ-axis orthogonal to the γ-axis, and an estimating unit for estimating a second rotational speed of the rotor in the dq-axis orthogonal rotating coordinate system based on the first rotational speeds of the γ-axis and the δ-axis ;
The estimation unit
calculating a relative phase indicating a phase difference between the γ-axis phase and the d-axis phase based on the δ-axis high-frequency current amplitude and the γ-axis high-frequency current amplitude;
calculating a relative speed indicating a speed difference between the first rotation speed and the second rotation speed based on the relative phase;
controlling the first rotation speed so that the relative speed is equal to a relative speed command value indicating a value other than 0;
estimating the magnetic pole position based on the phase of the γ-axis and the relative phase;
A control device for a synchronous reluctance motor, which estimates the second rotational speed based on the first rotational speed and the relative speed . 2. The control device for a synchronous reluctance motor according to claim 1 , wherein said estimator calculates said relative speed by differentiating said relative phase. The estimator calculates a first relative phase indicating a phase difference between the γ-axis phase and the d-axis phase based on the high-frequency current information of the γ-axis high-frequency current amplitude and the δ-axis high-frequency current amplitude. death,
2. The control device for a synchronous reluctance motor according to claim 1 , wherein a second relative phase, which is feedback information synchronized with said first relative phase, is estimated as said relative phase.

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