JP5165545B2 - Electric motor magnetic pole position estimation device - Google Patents

Description

  The present invention relates to a magnetic pole position estimation device for an electric motor.

Conventionally, for example, the rotational speed of the motor is calculated from the γ-axis current in the γδ coordinate system having a phase difference with respect to the dq coordinate system and the current deviation between the δ-axis current and the model current of each axis calculated based on the motor model. An estimation device is known (see, for example, Patent Document 1).
Japanese Patent Laid-Open No. 08-308286

By the way, in the device according to the above prior art, the γ-axis current and the δ-axis current in the γδ coordinate system are acquired based on the detection value output from the phase current sensor that detects each phase current of the motor. The detection timing by the sensor coincides with the timing based on the motor model.
However, when the phase current sensor is omitted in the apparatus according to the prior art and each phase current of the motor is estimated by the DC side current sensor that detects the DC side current of the inverter that drives the motor, Since the timing of the current is different from the timing based on the motor model, there is a problem that it is difficult to appropriately estimate the rotation speed of the motor.
The present invention has been made in view of the above circumstances, and even in the case where each phase current of a motor is estimated by a DC-side current sensor, an electric motor capable of appropriately estimating the rotational speed and magnetic pole position of the motor. An object of the present invention is to provide a magnetic pole position estimation device.

In order to solve the above-described problems and achieve the object, a magnetic pole position estimation device for an electric motor according to a first aspect of the present invention uses a three-phase AC electric motor (for example, the motor 11 in the embodiment) by a pulse width modulation signal. ) In turn (for example, the inverter 13 in the embodiment) and pulse width modulation signal generating means (for example, the PWM signal in the embodiment) for generating the pulse width modulation signal from a carrier wave signal. Generator 25), a DC-side current sensor (for example, DC-side current sensor 31 in the embodiment) that detects a DC-side current of the inverter, and the DC-side current detected by the DC-side current sensor. Phase current estimation means for estimating the phase current (for example, the phase current estimation unit 23 in the embodiment) and a γδ coordinate system having a phase difference with respect to the dq coordinate system are set, and the phase current estimation The actual current (for example, the γ-axis current Iγ and the δ-axis current Iδ in the embodiment) obtained by converting the phase current estimated by the fixing means into the γδ coordinate system, and the voltage of the predetermined model of the motor Phase difference calculating means (for example, in the embodiment, for calculating the phase difference based on the current deviation from the model current corresponding to the equation (for example, the γ-axis model current IγM and the δ-axis model current IδM in the embodiment) Magnetic pole position error estimating unit 46) and magnetic pole position calculating means for calculating the magnetic pole position of the motor based on the phase difference calculated by the phase difference calculating means (for example, rotational speed-magnetic pole position calculating unit in the embodiment) 47), wherein the phase current estimation means is arranged at each of the carrier vertices in a voltage pattern symmetric with respect to the maximum or minimum carrier vertices in one period of the carrier signal. The current value at each one point of the phase current is estimated at symmetrical timings, and the predetermined timing is based on the current values at the two point timings consisting of the one point estimated by the phase current estimation means. Current value calculating means for calculating the current value of the phase current at (for example, also serving as the phase current estimating unit 23 in the embodiment), and the current value calculating means includes the current value calculating means for each of the three phase currents. Synchronization means that synchronizes the timing of the variable to the predetermined timing by synchronizing the timing of the variable by calculating the variable at the predetermined timing based on a plurality of timings at which the variable of the voltage equation is obtained. The phase difference calculation means includes the real current at the predetermined timing synchronized by the current value calculation means and the voltage equation at a continuous time as the predetermined timing . And the current deviation is calculated by the model current obtained by synchronizing the timing of the variable of the voltage equation with the predetermined timing by the synchronization means .

  Furthermore, in the magnetic pole position estimation device for an electric motor according to the second aspect of the present invention, the current value calculation means is based on the current value at the timing of two points each consisting of the one point estimated by the phase current estimation means. An average value is calculated as a current value of the phase current at the carrier apex, and the phase difference calculating means calculates the phase difference based on the voltage equation at the carrier apex.

  According to the magnetic pole position estimating apparatus for an electric motor according to the first aspect of the present invention, the current value for each of the three-phase phase currents estimated by the phase current estimating means is synchronized with a predetermined timing. Since the voltage equation is based on a predetermined timing, the timing of the actual current corresponding to the phase current matches the timing of the model current corresponding to the voltage equation, and the phase difference and the magnetic pole position can be calculated appropriately.

  Furthermore, according to the magnetic pole position estimation device of the electric motor according to the second aspect of the present invention, since the predetermined timing is the carrier apex, the current value for each of the three-phase phase currents can be reduced while reducing the error due to noise. The estimation accuracy can be improved, and the magnetic pole position of the electric motor can be estimated with high accuracy.

Embodiments of a magnetic pole position estimating apparatus for an electric motor according to the present invention will be described below with reference to the accompanying drawings.
The magnetic pole position estimating device 10 (hereinafter simply referred to as the magnetic pole position estimating device 10) according to this embodiment includes a motor phase current estimating device 10a (hereinafter simply referred to as the phase current estimating device 10a). Yes. The phase current estimation device 10a estimates each phase current supplied to, for example, a three-phase AC brushless DC motor 11 (hereinafter simply referred to as the motor 11), and the motor 11 uses a permanent magnet used for a field. And a stator (not shown) that generates a rotating magnetic field for rotating the rotor. Then, the magnetic pole position estimation device 10 estimates the magnetic pole position of the motor 11 (that is, the rotation angle of the rotor magnetic pole from a predetermined reference rotation position).
For example, as shown in FIG. 1, the phase current estimation device 10 a includes an inverter 13 that uses a battery 12 as a DC power source and a motor control device 14.

The three-phase (for example, U-phase, V-phase, and W-phase) AC motor 11 is driven by the inverter 13 in response to a control command output from the motor control device 14.
The inverter 13 includes a bridge circuit 13a formed by bridge connection using a plurality of switching elements (for example, MOSFETs: Metal Oxide Semi-conductor Field Effect Transistors) and a smoothing capacitor C, and the bridge circuit 13a performs pulse width modulation ( It is driven by the PWM signal.

  In this bridge circuit 13a, for example, a high-side and low-side U-phase transistor UH, UL paired for each phase, a high-side and low-side V-phase transistor VH, VL, a high-side and low-side W-phase transistor WH, WL is bridge-connected. Each transistor UH, VH, WH has a drain connected to the positive terminal of the battery 12 to form a high side arm, and each transistor UL, VL, WL has a source connected to the grounded negative terminal of the battery 12. It constitutes the low side arm. For each phase, the sources of the high-side arm transistors UH, VH, WH are connected to the drains of the low-side arm transistors UL, VL, WL, and the transistors UH, UL, VH, VL, WH, WL. Each of the diodes DUH, DUL, DVH, DVL, DWH, DWL is connected between the drain and the source so as to be in the forward direction from the source to the drain.

  The inverter 13 is, for example, a gate signal (that is, a PWM signal) that is a switching command that is output from the motor control device 14 when driving the motor 11 and is input to the gates of the transistors UH, VH, WH, UL, VL, WL. ), The DC power supplied from the battery 12 is converted into the three-phase AC power by switching the on / off (cut-off) state of each pair of transistors for each phase. By sequentially commutating energization to the windings, AC U-phase current Iu, V-phase current Iv and W-phase current Iw are passed through the stator windings of each phase.

  As will be described later, the motor control device 14 performs feedback control (vector control) of current on the γ-δ coordinates forming the rotation orthogonal coordinates, and calculates the command γ-axis current Iγc and the command δ-axis current Iδc. The phase voltage commands Vu, Vv, and Vw are calculated based on the command γ-axis current Iγc and the command δ-axis current Iδc, and a PWM signal that is a gate signal for the inverter 13 is calculated according to the phase voltage commands Vu, Vv, and Vw. Output. Then, the γ-axis current Iγ and the δ-axis current Iδ obtained by converting the phase currents Iu, Iv, Iw actually supplied from the inverter 13 to the motor 11 on the γ-δ coordinates, the command γ-axis current Iγc, and Control is performed so that each deviation from the command δ-axis current Iδc becomes zero.

The motor control device 14 includes, for example, a current sensor I / F (interface) 21, an overcurrent protection device 22, a phase current estimation unit 23, a control device 24, and a PWM signal generation unit 25. .
The current sensor I / F (interface) 21 is connected to a DC side current sensor 31 that detects a DC side current Idc of the bridge circuit 13a of the inverter 13 between the bridge circuit 13a of the inverter 13 and the negative terminal of the battery 12. Then, the detection signal output from the DC side current sensor 31 is output to the overcurrent protection device 22 and the phase current estimation unit 23.
The direct current sensor 31 may be disposed between the bridge circuit 13a of the inverter 13 and the positive terminal of the battery 12.

The overcurrent protection device 22 performs a predetermined overcurrent protection operation according to the DC side current Idc detected by the DC side current sensor 31.
The phase current estimation unit 23 actually uses the inverter 13 based on the DC side current Idc detected by the DC side current sensor 31 at the detection timing according to the gate signal (that is, PWM signal) output from the PWM signal generation unit 25. The phase currents Iu, Iv, and Iw supplied to the motor 11 are estimated. Details of the operation of the phase current estimation unit 23 will be described later.

For example, as shown in FIG. 4, the control device 24 has a phase difference Δθe that is a difference between the actual rotation angle and the estimated or designated rotation angle with respect to the dq axes of the rotation orthogonal coordinates of the actual motor 11. Then, a γ-δ axis of rotation orthogonal coordinates having a rotation speed ωe is set, and current feedback control (vector control) is performed on the γ-δ coordinates.
The control device 24 generates a command γ-axis current Iγc and a command δ-axis current Iδc, calculates each phase voltage command Vu, Vv, Vw based on the command γ-axis current Iγc and the command δ-axis current Iδc, and generates a PWM signal. To the unit 25.
Further, the control device 24 converts the phase currents Iu, Iv, and Iw output from the phase current estimation unit 23 onto the γδ coordinates, the γ-axis current Iγ and the δ-axis current Iδ, and the command γ-axis current Iγc. Current feedback control (vector control) is performed so that each deviation from the command δ-axis current Iδc becomes zero.
Details of the operation of the control device 24 will be described later.

  The PWM signal generation unit 25 compares each phase voltage command Vu, Vv, Vw with a carrier signal such as a triangular wave in order to pass a sinusoidal current to the three-phase stator winding, and A gate signal (that is, a PWM signal) for driving the transistors UH, VH, WH, UL, VL, WL on / off is generated. Then, the inverter 13 converts the DC power supplied from the battery 12 into three-phase AC power by switching the on (conductive) / off (cut-off) state of each transistor that forms a pair for each of the three phases. By sequentially commutating energization to each stator winding of the three-phase motor 11, AC U-phase current Iu, V-phase current Iv, and W-phase current Iw are energized to each stator winding.

The gate signal input from the PWM signal generation unit 25 to the inverter 13 is, for example, according to the combination of the on / off states of the transistors UH, UL and VH, VL and WH, WL that are paired for each phase. 1 and FIGS. 2A to 2H, the PWM (pulse width) corresponding to each of the eight switching states S1 to S8 (that is, the states of the basic voltage vectors V0 to V7 having different phases by 60 degrees). Modulation) signal. In Table 1 below, the transistors that are turned on among the high-side (High) and low-side (Low) transistors are shown, and the transistors that are turned on in FIGS. Is shaded.
Then, phase currents Iu, Iv, Iw are intermittently generated on the DC side of the bridge circuit 13a of the inverter 13 according to the switching states S1 to S8, and the DC side current Idc detected by the DC side current sensor 31. Is one of the phase currents Iu, Iv, Iw, or one of the phase currents Iu, Iv, Iw inverted, or zero.

  The phase current estimation unit 23, for example, in each of the switching states S2 to S7 (that is, the states of the basic voltage vectors V1 to V6 having different phases by 60 degrees) in one period of a carrier signal such as a triangular wave, for example. The two-phase phase currents of the three-phase phase currents are acquired from the DC-side current Idc detected by the DC-side current sensor 31 in two predetermined sets of states. Then, based on these two-phase phase currents, the other one-phase phase current among the three-phase phase currents is estimated. Then, each estimated value of the three-phase current obtained by estimating from the DC-side current Idc detected by the DC-side current sensor 31 is output to the control device 24.

For example, as shown in FIG. 3, in the case of three-phase modulation using a triangular carrier signal, the carrier signal with a voltage pattern symmetrical to the peak (carrier vertex) on the valley side of the triangular carrier signal is obtained. In the period of one cycle Ts, the detected value of each phase current for two phases can be acquired twice.
In other words, the phase current estimation unit 23 is symmetric with respect to the carrier vertex at times tu1 and tu2 (that is, at the time tp of the carrier vertex on the valley side) in the state of the two basic voltage vectors V1 symmetrical with respect to the carrier vertex. On the other hand, the first U-phase current Iu1 and the second U-phase current Iu2 are acquired from the DC-side current Idc detected by the DC-side current sensor 31 at the time having the same time interval T1 and at the DC-side current detection timing). In addition, in the state of the two times basic voltage vector V3 that is symmetric with respect to the carrier vertex, times tw1 and tw2 that are symmetric with respect to the carrier vertex (that is, the same for the time tp of the carrier vertex on the valley side) From the DC side current Idc detected by the DC side current sensor 31 at the time having the time interval T2 and at the DC side current detection timing), the first W It acquires current Iw1 and the 2W-phase current Iw2.

Then, the phase current estimation unit 23 calculates an average value for each phase based on the phase currents Iu1, Iu2 and Iw1, Iw2, and calculates each average value at the time tp at the peak of the carrier on the valley side (that is, the estimated phase current). Current value at synchronization timing). Thereby, the timing of the current value of the two-phase phase current (that is, the U-phase current and the W-phase current) is changed from the DC-side current detection timing (that is, each time tu1, tu2, tw1, tw2) to the carrier peak on the valley side. At time tp (estimated phase current synchronization timing).
Then, the phase current estimation unit 23 uses the fact that the sum of the current values of the respective phase currents at the same timing is zero, so that the current values of the two-phase currents (for example, the U-phase current and the W-phase current) ( That is, the current value of the other one-phase phase current (for example, V-phase current) is calculated from the current value at the time tp at the carrier apex on the valley side. Thereby, the timing of the current value of the three-phase phase current is synchronized with the time tp of the carrier peak on the valley side.
The phase current estimation unit 23 calculates the average value based on the phase currents Iu1, Iu2 and Iw1, Iw2, and estimates the phase current of the other one phase from the two-phase phase current. However, the present invention is not limited to this. Instead, each phase current may be estimated by another estimation method.

  For example, as shown in FIG. 5, the control device 24 includes a speed control unit 41, a command current generation unit 42, a current control unit 43, a γδ-3 phase conversion unit 44, a three phase-γδ conversion unit 45, A magnetic pole position error estimation unit 46, a rotation speed-magnetic pole position calculation unit 47, and an electrical angle-mechanical angle conversion unit 48 are provided.

The speed control unit 41 is based on the rotational speed command value ωrc input from the outside, for example, by the closed loop control corresponding to the rotational speed ωr (mechanical angle) output from the electrical angle-mechanical angle conversion unit 48, and the torque command Tc. Is calculated. Then, a torque command Tc is output.
The control device 24 may include a torque control unit instead of the speed control unit 41, and execute torque control.

  The command current generator 42 calculates a command δ-axis current Iδc based on the torque command Tc output from the speed controller 41. Based on the command δ-axis current Iδc, the command γ-axis current Iγc is calculated, and the command δ-axis current Iδc and the command γ-axis current Iγc are output.

  The current control unit 43 calculates a deviation ΔIγ between the command γ-axis current Iγc output from the command current generation unit 42 and the γ-axis current Iγ output from the three-phase-γδ conversion unit 45, and the command current generation unit 42 A deviation ΔIδ between the output command δ-axis current Iδc and the δ-axis current Iδ output from the three-phase-γδ converter 45 is calculated. Then, for example, by PI (proportional / integral) operation, the deviation ΔIγ is controlled and amplified to calculate the γ-axis voltage command value Vγ, and the deviation ΔIδ is controlled and amplified to calculate the δ-axis voltage command value Vδ. Then, the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ are output.

The γδ-3 phase conversion unit 44 uses the estimated magnetic pole position value θe of the motor 11 output from the rotation speed-magnetic pole position calculation unit 47 to generate a γ-axis voltage command value Vγ and a δ-axis voltage command value on the γ-δ coordinates. Vδ is converted into a U-phase voltage command Vu, a V-phase voltage command Vv, and a W-phase voltage command Vw, which are voltage command values on a three-phase AC coordinate that is a stationary coordinate.
The three-phase-γδ conversion unit 45 estimates each phase current Iu, Iv, Iw output from the phase current estimation unit 23 based on the magnetic pole position estimated value θe of the motor 11 output from the rotation speed-magnetic pole position calculation unit 47. The value is converted into a γ-axis current Iγ and a δ-axis current Iδ on the γ-δ coordinates.

  The magnetic pole position error estimation unit 46, for example, the γ-axis voltage command value Vγ and δ-axis voltage command value Vδ output from the current control unit 43 and the γ-axis currents Iγ and δ output from the three-phase-γδ conversion unit 45. Based on the shaft current Iδ, the phase difference Δθe is estimated using the fact that the induced voltage generated by the motor 11 when the motor 11 rotates changes with the rotational speed.

  As shown in FIG. 6, for example, the magnetic pole position error estimation unit 46 includes a motor model current calculation unit 51, a γ-axis current deviation calculation unit 52a, a δ-axis current deviation calculation unit 52b, and a magnetic pole position error calculation unit 53. Configured.

The motor model current calculation unit 51 uses the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ output from the current control unit 43 to calculate the γ-axis model current IγM and δ-axis according to the voltage equation of a predetermined model of the motor 11. The model current IδM is calculated, and the model currents IγM and IδM are output.
First, the voltage equation of a predetermined model of the motor 11 will be described below.
The voltage equation of the motor 11 in the γδ coordinate system (that is, the voltage equation in continuous time) is, for example, the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ, the winding resistance R, and the differential operator p. , D-axis inductance Ld and q-axis inductance Lq, phase difference Δθe, rotational speed estimation value ωe, d-axis current Id and q-axis current Iq, and magnetic flux component (induced voltage constant) φ of the permanent magnet, for example, It is described as shown in the following mathematical formula (1).

  In the above formula (1), if terms relating to the magnetic pole position and the rotational speed are deleted as disturbances, a simple voltage equation of the motor 11 is described as shown in the following formula (2).

Incidentally, the γ-axis current Iγ and the δ-axis current Iδ output from the three-phase-γδ conversion unit 45 are obtained by coordinate conversion of the estimated values of the phase currents Iu, Iv, and Iw output from the phase current estimation unit 23. Therefore, it has a value equivalent to the estimated value of each phase current Iu, Iv, Iw, that is, a value at time tp (estimated phase current synchronization timing) of the carrier peak on the valley side shown in FIG.
On the other hand, the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ output from the current control unit 43 are timings corresponding to the control cycle of the current feedback control executed by the motor control device 14, that is, The value at the time ts at the peak of the carrier shown in FIG. 3 (for example, the timing at which each phase current is detected in the comparative example including the phase current sensor that directly detects each phase current and the actual phase current detection timing) have.
Therefore, the timing of the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ output from the current control unit 43 is set to the timing of the γ-axis current Iγ and δ-axis current Iδ output from the three-phase-γδ conversion unit 45. When synchronized, the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ at the time tp (estimated phase current synchronization timing) of the carrier peak on the valley side shown in FIG. Is described as follows. In the following mathematical formula (3), arbitrary natural numbers n and m are an index corresponding to a peak on the peak side and an index corresponding to a peak on the valley side in the same arbitrary control cycle.

  The above equation (2), which is a simple voltage equation in continuous time, is discretized at the time tp (estimated phase current synchronization timing) of the carrier peak on the valley side shown in FIG. 3 using, for example, Euler approximation, and further When the mathematical formula (3) is applied, the discrete time state equation is described by, for example, the following mathematical formula (4) by one period Ts of the carrier signal.

  The motor model current calculation unit 51 outputs the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ output from the current control unit 43, that is, the time ts (actual phase current detection timing) at the peak of the carrier shown in FIG. (N-1) -th and (n-2) -th γ-axis voltage command values Vγ [n−1], Vγ [n-2] and δ-axis voltage command values Vδ [n−1], Vδ [ n-2], and the voltage equation of the predetermined model of the motor 11 shown in the above equation (4) has a value at the time tp (estimated phase current synchronization timing) of the carrier peak on the valley side shown in FIG. (M) The γ-axis model current IγM [m] and the δ-axis model current IδM [m] are calculated, and the model currents IγM [m] and IδM [m] are output.

  The γ-axis current deviation calculation unit 52a and the δ-axis current deviation calculation unit 52b, for example, as shown in the following formula (5), at the time tp (estimated phase current synchronization timing) of the valley-side carrier apex shown in FIG. m) The γ-axis current Iγ [m] and δ-axis current Iδ [m] output from the three-phase-γδ converter 45 and the model currents IγM [m] output from the motor model current calculator 51 , IδM [m] and ΔIδM [m] and ΔIδM [m] are calculated, and the current deviations ΔIγM [m] and ΔIδM [m] are output.

For example, the magnetic pole position error calculation unit 53 outputs (m) -th current deviations ΔIγM [m] and ΔIδM [m] output from the γ-axis current deviation calculation unit 52a and the δ-axis current deviation calculation unit 52b, The (m−1) th rotational speed estimation value ωe [m−1], the (m−1) th γ-axis current Iγ [m−1] and the δ-axis output from the three-phase-γδ converter 45. Based on the following formula (6), the current Iδ [m−1] and the electrical circuit constant of the motor 11 that is known in advance (that is, each inductance Ld, Lq and one period Ts of the carrier signal) The first phase difference Δθe is calculated. Then, the phase difference Δθe is output.
The following formula (6) is a discrete time state equation obtained by discretizing the voltage equation of the motor 11 shown in the formula (1), and a voltage equation of a predetermined model of the motor 11 shown in the formula (4). It is calculated based on a discrete time state equation obtained by discretizing a simple voltage equation of the motor 11.

  The rotation speed-magnetic pole position calculation unit 47 performs phase synchronization processing by PLL (Phase-locked loop) based on the phase difference Δθe output from the magnetic pole position error estimation unit 46.

  As the phase synchronization process, the phase synchronization unit 56 performs, for example, a PI (proportional / integral) operation, and, for example, based on the transfer function Ge (s) described as shown in the following formula (7), the proportional gain Kp and the integral Based on the gain Ki, the rotational speed estimated value ωe is calculated from the phase difference Δθe, for example, as shown in the following formula (8). Then, the estimated rotational speed value ωe is output.

  The integration calculation unit 57 calculates the magnetic pole position estimation value θe by integrating the rotational speed estimation value ωe output from the phase synchronization unit 56, for example, as shown in the following formula (9). Then, the magnetic pole position estimated value θe is output.

  Note that the rotation speed-magnetic pole position calculation unit 47 is not limited to the PI operation, and, for example, a follow-up calculation by a one-dimensional observer using the phase difference Δθe as an input value as in the modification shown in FIG. 8 and the following formula (10). Processing may be executed to calculate the rotational speed estimated value ωe and the magnetic pole position estimated value θe.

  The electrical angle-mechanical angle conversion unit 48 converts the rotational speed estimated value ωe output from the rotational speed-magnetic pole position calculation unit 47 into a rotational speed ωr (mechanical angle) according to the number of pole pairs q of the motor 11 and rotates. The speed ωr (mechanical angle) is output.

As described above, according to the magnetic pole position estimation apparatus 10 of the motor according to the present embodiment, the current values of the three-phase currents Iu, Iv, Iw estimated by the phase current estimation unit 23 are synchronized with the same predetermined timing. Furthermore, since the voltage equation of the predetermined model of the motor 11 is based on the same predetermined timing as the current value of each phase current Iu, Iv, Iw, γ corresponding to each phase current Iu, Iv, Iw The timings of the shaft current Iγ and the δ-axis current Iδ and the model currents IγM and IδM corresponding to the voltage equation coincide with each other, and the phase difference Δθe and the magnetic pole position estimated value θe can be appropriately calculated.
In addition, since the predetermined timing to be synchronized is set as the carrier apex, each phase current Iu, Iv, Iw is estimated for each of the three-phase phase currents Iu, Iv, Iw while reducing errors caused by noise. The estimation accuracy can be improved. Thereby, the magnetic pole position estimated value θe of the motor 11 can be accurately estimated.

  In the above-described embodiment, the magnetic pole position error estimation unit 46 estimates the phase difference Δθe by utilizing the fact that the induced voltage generated by the motor 11 changes according to the rotation speed when the motor 11 rotates. For example, the phase difference Δθe is utilized by applying a harmonic voltage to the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ output from the current control unit 43 and changing the inductance depending on the magnetic pole position. May be estimated. Even in this case, the voltage equation of the predetermined model of the motor 11 only needs to be synchronized at the same timing as the γ-axis current Iγ and the δ-axis current Iδ output from the three-phase-γδ conversion unit 45.

  In the above-described embodiment, the phase current estimation unit 23 uses the detected values of the respective phase currents for two phases in the period of one cycle Ts of the carrier signal for three-phase modulation using a triangular wave carrier signal. An average value is calculated for each phase, and each average value is defined as a current value at the time Ts / 2 at the peak of the carrier on the valley side. However, the present invention is not limited to this. The average value may be calculated for each phase from the detected values of the phase currents for two phases in the period of one cycle Ts, and each average value may be used as the current value at the time Ts / 2 of the carrier peak on the valley side. The phase current estimation unit 23 calculates the average value for each phase from the detected values of the phase currents for two phases, and estimates the phase current of the other one phase from the phase currents of the two phases. However, the present invention is not limited to this, and each phase current may be estimated by another estimation method.

It is a block diagram of the magnetic pole position estimation apparatus of the electric motor which concerns on embodiment of this invention. It is a figure which shows each switching state S1-S8 of the inverter shown in FIG. It is a figure which shows the example of the detection timing of the on-off pattern and each phase current of the carrier wave and each transistor UH, UL and VH, VL and WH, WL which concern on embodiment of this invention. It is a figure which shows an example of the (gamma) -delta axis | shaft of a rotation orthogonal coordinate and dq axis which concerns on embodiment of this invention. It is a block diagram of the magnetic pole position estimation apparatus of the electric motor which concerns on embodiment of this invention. It is a block diagram of the magnetic pole position error estimation part shown in FIG. It is a block diagram of the rotational speed-magnetic pole position calculating part shown in FIG. It is a block diagram of the rotational speed-magnetic pole position calculating part which concerns on the modification of embodiment of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Motor magnetic pole position estimation apparatus 11 Motor 13 Inverter 23 Phase current estimation part (Phase current estimation means, Current value calculation means)
24 controller 25 PWM signal generator (pulse width modulation signal generator)
31 DC-side current sensor 46 Magnetic pole position error estimator (phase difference calculation means)
47 Rotational speed-magnetic pole position calculator (magnetic pole position calculator)

Claims (2)

An inverter for sequentially commutating energization of a three-phase AC motor by a pulse width modulation signal, pulse width modulation signal generation means for generating the pulse width modulation signal from a carrier wave signal, and a direct current for detecting a DC side current of the inverter A side current sensor, phase current estimation means for estimating a phase current based on the DC side current detected by the DC side current sensor,
Set the γδ coordinate system having a phase difference with respect to the dq coordinate system, the phase current and the actual current obtained by coordinate converting the phase current estimated in the γδ coordinate system by estimating means, a predetermined model of the motor A phase difference calculating means for calculating the phase difference based on a current deviation from a model current corresponding to the voltage equation of the magnetic field, and a magnetic pole position for calculating the magnetic pole position of the electric motor based on the phase difference calculated by the phase difference calculating means An arithmetic means,
The phase current estimation means is configured to detect the phase current at a timing symmetric with respect to the carrier vertex in each of the voltage patterns symmetric with respect to the maximum or minimum carrier vertex in one period of the carrier signal. Estimate the current value of each one point,
Current value calculating means for calculating a current value of the phase current at a predetermined timing based on the current value at the timing of two points consisting of the one point estimated by the phase current estimating means;
The current value calculation means synchronizes the predetermined timing for each phase current of three phases,
Synchronization means for synchronizing the timing of the variable to the predetermined timing by calculating the variable at the predetermined timing based on a plurality of timings at which the variable of the voltage equation is obtained;
The phase difference calculating means, the actual current at the predetermined timing synchronized by the current value calculating means ,
The voltage equation in continuous time is discretized at the predetermined timing, and the current of the model equation is obtained by synchronizing the timing of the variable of the voltage equation with the predetermined timing by the synchronization means. An apparatus for estimating a magnetic pole position of an electric motor, wherein a deviation is calculated . The current value calculating means calculates an average value of the current values at the timing of two points consisting of the one point estimated by the phase current estimating means as a current value of the phase current at the carrier apex. And
2. The motor magnetic pole position estimation device according to claim 1, wherein the phase difference calculating unit calculates the phase difference based on the voltage equation at the carrier apex. 3.

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