JP7163641B2 - Synchronous motor controller - Google Patents

Description

The present invention relates to tuning performed by a control device for a synchronous motor.

A method of optimizing a magnetic flux model considering magnetic saturation including interference between d and q axes by auto-tuning in an embedded permanent magnet type synchronous motor (IPMSM) is known. As an example, there is a method of tuning the parameters related to the magnetic saturation characteristics of the IPMSM by applying high-frequency sinusoidal alternating currents (hereinafter referred to as tuning currents) to the d-axis and q-axis while the rotor is stopped. (See Patent Document 1, for example).

The magnetic saturation characteristic is the characteristic that the linearity between the d- and q-axis magnetic fluxes and the respective axis currents corresponding to them is lost due to the magnetic saturation of the motor iron core as the current increases. A characteristic is that the self-axis magnetic flux changes due to the influence of the other-axis current.

JP 2015-144502 A

The conventional auto-tuning method assumes that the rotor of the synchronous motor is stopped while the tuning current is flowing. However, the magnetic pole position may deviate from the position at the start of tuning due to high-frequency vibration torque generated in the synchronous motor during tuning due to the tuning current. This positional deviation is particularly likely to occur in an electric motor having a small moment of inertia.

If the synchronous motor is equipped with a position detector or speed detector, it is easy to detect the magnetic pole misalignment. However, in the case of position/speed sensorless control that controls the synchronous motor without using these detectors, the magnetic pole position deviation cannot be detected directly, so the tuning current flows in a phase shifted from the d-axis or q-axis. , the correct tuning result may not be obtained.

Accordingly, the present disclosure provides a controller for a synchronous motor that allows tuning currents to flow in the correct phase.

As one aspect of the technology of the present disclosure,
A coordinate system defined separately from the d, q orthogonal rotating coordinate system defined by the d-axis parallel to the rotor magnetic pole direction of the synchronous motor and the q-axis orthogonal to the d-axis, and is defined by the γ-axis and the δ-axis A control device for controlling the current and voltage supplied to the synchronous motor by a power converter in a defined γ, δ orthogonal rotating coordinate system,
A computing device that performs computation based on a magnetic flux model that takes into account at least the magnetic saturation characteristics of an iron core of the motor,
The computing device is
controlling the DC component of the γ-axis current of the synchronous motor to a command value;
applying a sinusoidal alternating voltage to the γ-axis,
performing at least one of the calculation of the d-axis inductance and the calculation of the parameters of the magnetic flux model;
calculating the Fourier coefficient of the fundamental wave component of the γ-axis voltage of the synchronous motor;
calculating the Fourier coefficient of the fundamental wave component of the δ-axis current of the synchronous motor;
Using the Fourier coefficient of the fundamental wave component of the γ-axis voltage and the Fourier coefficient of the fundamental wave component of the δ-axis current, the γ, δ orthogonal rotating coordinate system is set such that the d-axis and the γ-axis are aligned. A control device for a synchronous motor is provided, characterized by controlling the phase.

Further, as one aspect of the technology of the present disclosure,
A coordinate system defined separately from the d, q orthogonal rotating coordinate system defined by the d-axis parallel to the rotor magnetic pole direction of the synchronous motor and the q-axis orthogonal to the d-axis, and is defined by the γ-axis and the δ-axis A control device for controlling the current and voltage supplied to the synchronous motor by a power converter in a defined γ, δ orthogonal rotating coordinate system,
A computing device that performs computation based on a magnetic flux model that takes into account at least the magnetic saturation characteristics of an iron core of the motor,
The computing device is
controlling the DC component of the γ-axis current of the synchronous motor to a command value;
Applying a sinusoidal alternating voltage to the δ axis,
performing at least one of the calculation of the q-axis inductance and the calculation of the parameters of the magnetic flux model;
calculating the Fourier coefficient of the fundamental wave component of the δ-axis voltage of the synchronous motor;
calculating the Fourier coefficient of the fundamental wave component of the γ-axis current of the synchronous motor;
Using the Fourier coefficient of the fundamental wave component of the δ-axis voltage and the Fourier coefficient of the fundamental wave component of the γ-axis current, the γ, δ orthogonal rotating coordinate system is set such that the d-axis and the γ-axis are aligned. A control device for a synchronous motor is provided, characterized by controlling the phase.

Further, as one aspect of the technology of the present disclosure,
A coordinate system defined separately from the d, q orthogonal rotating coordinate system defined by the d-axis parallel to the rotor magnetic pole direction of the synchronous motor and the q-axis orthogonal to the d-axis, and is defined by the γ-axis and the δ-axis Equipped with a computing device that controls the current and voltage supplied to the synchronous motor by the power converter in the determined γ, δ orthogonal rotating coordinate system,
The computing device is
Detecting an oscillating current generated in the other of the γ-axis and the δ-axis perpendicular to one of the axes through which the tuning current flows, and correcting the phase of the tuning current flow based on the detected value of the oscillating current; A controller for a synchronous motor is provided.

According to the technique of the present disclosure, it is possible to pass the tuning current in the correct phase.

1 is a block diagram illustrating the overall configuration of a control device for a synchronous motor according to the present disclosure; FIG. FIG. 2 is a block diagram showing a configuration example of an axis deviation compensator used in the first embodiment; FIG. FIG. 10 is a block diagram showing a configuration example of an axis deviation compensator used in the second embodiment; FIG. 4 is an explanatory diagram of γ-δ axes used in this embodiment; It is a figure which illustrates the hardware constitutions of the arithmetic unit with which a control apparatus is provided.

Hereinafter, embodiments of a synchronous motor control device according to the present disclosure will be described with reference to the drawings. The same reference numerals are given to the same components, and redundant explanations are omitted. It should be noted that the present invention is not limited to the following embodiments, and can be modified as appropriate without changing the gist of the invention.

In order to control the torque of a synchronous motor such as an IPMSM or a synchronous reluctance motor (hereinafter referred to as SynRM) with high accuracy, a magnetic flux model that takes into account at least the magnetic saturation characteristics of the motor core is obtained, and the current is controlled based on this magnetic flux model. It is desirable to In order to perform current control based on the magnetic flux model, it is necessary to calculate each parameter of the magnetic flux model. Moreover, when trying to control the rotation of the synchronous motor with high accuracy, it may be necessary to calculate the inductance of the synchronous motor.

The control device for a synchronous motor according to this embodiment applies an alternating voltage while the rotor of the synchronous motor is stationary, and based on the response of the current that flows when the alternating voltage is applied, determines the relationship between each parameter of the magnetic flux model and the inductance of the synchronous motor. Perform auto-tuning to calculate at least one of them. The control device of the present embodiment is designed to prevent high-frequency noise generated in the shaft opposite to the shaft through which the tuning current flows when the magnetic pole position is deviated when tuning is performed with the rotor of a synchronous motor such as IPMSM or SynRM stopped. Detect oscillating currents. For example, if one of the γ-axis and δ-axis through which the tuning current flows is the γ-axis, the other axis (opposite axis) orthogonal to the γ-axis is the δ-axis, and one of the axes through which the tuning current flows is the δ-axis. If it is the δ-axis, the other axis (opposite axis) perpendicular to the δ-axis is the γ-axis. Then, the control device of the present embodiment calculates the amount of deviation of the magnetic pole positions of the synchronous motor based on the detected value of the oscillating current, and corrects the phase of the tuning current based on the calculated value of the amount of deviation.

With such correction, the control device of the present embodiment can perform tuning even when a deviation of the magnetic pole position occurs when the rotor of a synchronous motor that does not have a position/speed detector is stopped. A tuning current can be supplied in the correct phase by compensating for the deviation of the magnetic pole position. As a result, it is possible to obtain a correct tuning result even if the control target is a motor that has a small moment of inertia and is likely to cause magnetic pole position deviation.

The control device of the present embodiment can control the rotation of the synchronous motor with high precision by using the magnetic flux model for which auto-tuning of each parameter has been completed for current control of the synchronous motor. For example, the control device uses a d-axis current command value i _d ^* and a q-axis current command value i _d ^* calculated based on a magnetic flux model in which each parameter after auto-tuning is reflected. It controls the current and voltage supplied to the synchronous motor. This makes it possible to control the rotation of the synchronous motor with high accuracy.

Next, the details of the control device of this embodiment will be described. First, the γ-δ axes used for control by the control device of this embodiment will be described.

<1. γ-δ axis>
FIG. 4 is a diagram showing the dq axis and the γ-δ axis. The dq axis is an orthogonal axis that defines a d, q orthogonal rotating coordinate system synchronized with the actual magnetic pole position of the synchronous motor, and the d axis is defined as the magnetic pole position direction. The magnetic pole (N pole) direction of the rotor of the motor is defined as the d-axis, and the direction leading 90° from the d-axis parallel to the magnetic pole direction of the rotor is defined as the q-axis. On the other hand, the γ-δ axes are orthogonal axes defining an arbitrary γ, δ orthogonal rotating coordinate system defined separately from the d, q orthogonal rotating coordinate system. The control device of this embodiment realizes position/speed sensorless control by causing the γ-δ axes to follow the dq-axes.

The control device of this embodiment controls the tuning current of the synchronous motor on the γ-δ _axes . A shift occurs between the delta axes and the tuning current is controlled to a different position than the dq axes. Next, a method for correcting the axis deviation θ _err during tuning so that the d-axis and the γ-axis are aligned will be described.

<2. First Embodiment>
(2.1) Current and voltage control method First, FIG. 1 is a block diagram showing a control device according to the present disclosure together with a main circuit. ) will be described together with the configuration of the control device. The control calculation of the voltage and current supplied to the synchronous motor by the power converter is performed on the γ, δ orthogonal rotating coordinate system controlled to match the d, q orthogonal rotating coordinate system.

The controller 100 of the first embodiment controls the DC component of the γ-axis current to a command value, and controls the magnetic saturation of the motor iron core based on the current and voltage when the sinusoidal alternating voltage is applied to the γ-axis. Auto-tuning is performed to calculate each parameter of the magnetic flux model considering at least the characteristics. Alternatively, the control device 100 controls the DC component of the γ-axis current to a command value, and based on the current and voltage when a sinusoidal alternating voltage is applied to the γ-axis, the magnetic pole direction of the synchronous motor 1 is parallel to the direction of the magnetic poles of the synchronous motor 1 . Auto-tuning is performed to calculate (measure) the d-axis inductance in the direction. The control device 100 may perform auto-tuning to calculate both the parameters of the magnetic flux model and the d-axis inductance.

In the first embodiment, for example, the γ-axis current DC component command value _Iγ(0) ^* is set to an arbitrary value including 0, and the amplitude _Iγ(b1) ^* of the γ-axis alternating current command value is set to An arbitrary value greater than 0 is set, and the amplitude I _δ(b1) ^* of the δ-axis alternating current command value is set to 0. _Iγ(0) ^* , _Iγ(b1) ^* , and _Iδ(b1) ^* are respectively the command value for the DC component of the γ-axis current, the command value for the amplitude of the alternating current flowing through the γ-axis, and the command value for the amplitude of the alternating current flowing through the δ-axis. Represents the command value for the amplitude of the alternating current.

In the first embodiment, the integrator 5a integrates the angular frequency _ωh of the alternating voltage to calculate the angle _θh of the d-axis alternating current command value. The γ-axis alternating current command calculator 6a calculates the AC component i _γh ^* of the γ-axis alternating current command value as shown in Equation (1).

The adder 7a adds the γ-axis current DC component command value _Iγ(0) ^* and _iγh ^* to calculate the _γ -axis current command value iγ ^* .

The current coordinate converter 8 converts the u- and w-phase current detection values i _u and i _w respectively detected by the u-phase current detector 9 u and the w-phase current detector 9 w to the synchronous motor 1 based on the magnetic pole position estimation value θ _{r_est} of , the coordinates are transformed into γ, δ axis current detection values i _γ , i _δ .

The low-pass filter 10a removes high-frequency components from the _γ -axis current detection value _iγ to calculate the γ-axis current detection value iγf.

The deviation between the γ-axis current command value i _γ ^* and the γ-axis current detection value i _γf is calculated by the subtractor 11a, and this deviation is amplified by the γ-axis current controller 12a to obtain the γ-axis voltage command value v _γACR (γ Calculate the shaft voltage feedback control value v _γACR ). The γ-axis current controller 12a operates so that the deviation between the γ-axis current command value i _γ ^* and the γ-axis current detection value i _γf becomes zero to calculate the γ-axis voltage feedback control value v _γACR .

The parameters of the flux model are estimated using harmonic currents that flow due to magnetic saturation of the motor core. Therefore, in order to prevent the γ-axis current adjuster 12a from acting on harmonic currents, the response frequency of the γ-axis current adjuster 12a and the cutoff frequency of the low-pass filter 10a are adjusted to the angular frequency ω _h of the alternating voltage. It is set smaller than twice.

The DC armature resistance compensator 13 calculates the γ-axis armature resistance feedforward compensation value v _γra by Equation (2).

The voltage compensation value calculator 14 uses the d-axis reactance estimated value X _dhest and the d-axis armature resistance estimated value R _dhest calculated by the impedance estimator 15 to calculate the γ-axis voltage feedforward compensation value v _{γhFF according} to Equation 3. do. Estimation of impedance such as the d-axis reactance estimated value X _dhest and the d-axis armature resistance estimated value R _dhest may be performed by the impedance estimator 15 using a known technique described in Patent Document 1, for example. It is possible.

Also, the calculation of the d-axis inductance can be performed by the impedance estimator 15 using known techniques. For example, the impedance estimator 15 calculates the d-axis inductance L _dest by dividing the d-axis reactance estimated value X _dhest by the angular frequency ω _h (L _dest =X _dhest /ω _h ).

The Fourier coefficient calculator 16 calculates Fourier coefficients of the γ-axis voltage and the _γ -axis current based on θh, _vγ ^* , and _iγ . The Fourier coefficient calculator 16 calculates the Fourier cosine coefficient _Vγ(a1) of the fundamental wave component of the γ-axis voltage, the Fourier sine coefficient _Vγ(b1) of the fundamental wave component of the γ-axis voltage, and the Fourier coefficient of the DC component of the γ-axis current. I _γ(0) , Fourier cosine coefficient I _γ(a1) of the fundamental wave component of γ-axis current, Fourier sine coefficient I _γ(b1) of the fundamental wave component of γ-axis current, Double harmonic component of γ-axis current Fourier cosine coefficient I _γ(a2) , Fourier sine coefficient I _γ(b2) of the second harmonic component of the γ-axis current, Fourier cosine coefficient I _γ(a3) of the third harmonic component of the γ-axis current, and γ Calculate the Fourier sine coefficient _Iγ(b3) of the triple harmonic component of the shaft current. Fourier coefficients can be calculated using a known technique described in Patent Document 1, for example.

The Fourier cosine coefficient _Iγ(a2) of the double harmonic component of the γ-axis current, the Fourier sine coefficient _Iγ(b2) of the double harmonic component of the γ-axis current, and the γ-axis calculated by the Fourier coefficient calculator 16 From the Fourier cosine coefficient I _γ(a3) of the third harmonic component of the current and the Fourier sine coefficient I _γ(b3) of the third harmonic component of the γ-axis current, the armature resistance compensator 17 can be calculated by Equation 4 as follows: A harmonic armature resistance compensation value _vγhra is calculated.

The magnetic flux model coefficient estimator 18 receives the Fourier coefficient I _γ(0) of the γ-axis current DC component, the Fourier cosine coefficient I _γ(a1) of the γ-axis current fundamental wave component, γ Fourier sine coefficient _Iγ(b1) of the fundamental wave component of the axis current, Fourier cosine coefficient Iγ(a2) of the double harmonic component of the γ-axis current, Fourier sine coefficient _Iγ(a2) of the double harmonic component of the _γ -axis current _b2) , the Fourier cosine coefficient _Iγ(a3) of the third harmonic component of the γ-axis current, and the Fourier sine coefficient _Iγ(b3) of the third harmonic component of the γ-axis current are input. The magnetic flux model coefficient estimator 18 calculates coefficients (parameters) of the d-axis magnetic flux model, such as Equation 5, for example.

In the first embodiment, the δ-axis current i _δ is 0, so the magnetic flux model coefficient estimator 18 calculates K _Ld and K _sd using the above Fourier coefficients, for example, algebraic calculations, optimization calculations, etc. is calculated by a known calculation method. For example, the magnetic flux model coefficient estimator 18 performs auto-tuning to calculate parameters of the magnetic flux model such as K _Ld and K _sd using the known technique described in Patent Document 1 and the like.

The γ-axis voltage command value v _γ ^* is obtained by adders 7b and 7d into a γ-axis voltage feedback control value v _γACR , a γ-axis armature resistance feedforward compensation value v _γra , a γ-axis voltage feedforward compensation value v _γhFF , a γ-axis It is calculated by adding the harmonic armature resistance compensation value v _γhra . On the other hand, the δ-axis voltage command value v _δ ^* is fixed at zero.

The γ, δ axis voltage command values _{v γ} ^* , v _δ ^* calculated as described above are converted by the voltage coordinate converter 20 into u, It is converted into phase voltage command values v _u ^* , v _v ^* , v _w ^* of the v and w phases.

The rectifier circuit 3 rectifies the three-phase AC voltage from the three-phase AC power supply 4, converts it into a DC voltage, and supplies this DC voltage to the power converter 2 such as an inverter.

PWM (Pulse Width Modulation) circuit 21 adjusts the output voltage of power converter 2 to phase voltage command values v _u ^* , v _v ^* , v based on phase voltage command values v _u ^* , v _v ^* , v _w ^* . Generate a plurality of gating signals to control _w ^* . Based on a plurality of gate signals from the PWM circuit 21, the power converter 2 controls a plurality of semiconductor switching elements inside the power converter 2 to change the terminal voltage of the synchronous motor 1 such as IPMSM or SynRM to a phase voltage. Control to command values _vu ^* , _vv ^* , _vw ^* .

(2.2) Configuration of Axial Deviation Compensator The configuration of the axial deviation compensator 19 of FIG. 1 in the first embodiment will be described. First, the calculation principle of the axis deviation amount θ _err will be explained using the γ-δ axis and the dq axis. Equation 6 is the amplitude of the dq-axis voltage when the axis shift occurs. As a result, a fundamental wave current having an amplitude shown in Equation 7 is generated on the dq axes. However, for simplicity, the armature resistance is ignored.

Also, the γ-δ axis current fundamental wave amplitude can be expressed by Expression 8 using the d-q axis current fundamental wave amplitude and the axis deviation amount θ _err . By substituting Equations 6 and 7 into Equation 8 and arranging them, the γ-axis current fundamental wave amplitude and the δ-axis current fundamental wave amplitude are obtained as shown in Equations 9 and 10. Assuming that the γ-axis voltage amplitude V _γ is positive, the γ-axis current fundamental wave amplitude I _γ is positive from Equation 9 regardless of the amount of axis deviation _{θ err} . has polarities depending on the d-axis reactance X _d , the q-axis reactance X _q and the axis deviation θ _err .

Transforming Equation 10, Equation 11 is obtained. Here, assuming that 2θ _err is sufficiently small and approximated by sin θ≈θ, the axis deviation amount θ _err can be calculated by Equations 12 and 13.

Based on the principle described above, the axis deviation compensator 19 for correcting the axis deviation amount θ _err will be described with reference to FIG. 2 .

The amplitude calculator 22a calculates a δ-axis fundamental current Fourier sine coefficient correction value I _δ(b1) ' corresponding to the δ-axis current fundamental wave amplitude I _δ (oscillating current) of Equation 12 using Equations 14 and 15. . As a result, the amplitude can be calculated in consideration of the aforementioned amplitude polarity.

The G _θ calculator 23 a calculates the δ-axis current-axis deviation amount conversion gain G _θ δ by Equation (13). At this time, if the d-axis reactance _Xd and the _q -axis reactance Xq are unknown, assumed values are used. Also, the γ-axis voltage amplitude V _γ is calculated by Equation (16).

I _δ(b1) ' and G _θδ calculated as described above are multiplied by the multiplier 24a as shown in Equation 12 to calculate the compensation amount θ _err for compensating for _misalignment . It is input to the speed estimator 25a. The speed estimator 25a is composed of, for example, a PI controller, and calculates the speed estimated value ω _{r_est} as shown in Equation (17).

The integrator 5b integrates the speed estimated value ω _{r_est} and outputs the magnetic pole position estimated value θ _{r_est} . By constructing the axis deviation compensator as described above, it is possible to flow the tuning current in an appropriate phase even when the magnetic pole position is deviated during tuning. In other words, the control device 100 can control the phase of the γ, δ orthogonal rotating coordinate system so that the d-axis and the γ-axis are aligned by the compensation by the axis deviation compensator 19 .

<3. Second Embodiment>
(3.1) Current and Voltage Control Method A method for controlling the voltage and current of the synchronous motor 1 according to the second embodiment will be described together with the configuration of the control device with reference to FIG. The control calculation of the voltage and current supplied to the synchronous motor by the power converter is performed on the γ, δ orthogonal rotating coordinate system controlled to match the d, q orthogonal rotating coordinate system.

The control device 100 of the second embodiment controls the DC component of the γ-axis current to a command value, and controls the magnetic saturation of the motor iron core based on the current and voltage when a sinusoidal alternating voltage is applied to the δ-axis. Auto-tuning is performed to calculate each parameter of the magnetic flux model considering at least the characteristics. Alternatively, the control device 100 controls the DC component of the γ-axis current to a command value and applies a sine-wave alternating voltage to the δ-axis, based on the current and the voltage, which is perpendicular to the direction of the magnetic poles of the synchronous motor 1 . Auto-tuning is performed to calculate (measure) the q-axis inductance in the direction of The control device 100 may perform auto-tuning to calculate both the parameters of the magnetic flux model and the q-axis inductance.

In the second embodiment, for example, the γ-axis current DC component command value _Iγ(0) ^* is set to an arbitrary value including 0, and the amplitude _Iγ(b1) ^* of the γ-axis alternating current command value is set to 0, and the amplitude I _δ(b1) ^* of the δ-axis alternating current command value is set to an arbitrary value greater than 0.

In the second embodiment, the integrator 5a integrates the angular frequency _ωh of the alternating voltage to calculate the angle _θh of the d-axis alternating current command value. The δ-axis alternating current command calculator 6b calculates the AC component i _δh ^* of the δ-axis alternating current command value as shown in Equation (18).

The current coordinate converter 8 converts the u- and w-phase current detection values i _u and i _w respectively detected by the u-phase current detector 9 u and the w-phase current detector 9 w to the synchronous motor 1 based on the magnetic pole position estimation value θ _{r_est} of , the coordinates are transformed into γ, δ axis current detection values i _γ , i _δ .

The low-pass filter 10a removes high-frequency components from the _γ -axis current detection value _iγ to calculate the γ-axis current detection value iγf. The low-pass filter 10b removes high-frequency components from the _δ -axis current detection value _iδ to calculate a δ-axis current detection value iδf.

The deviation between the γ-axis current command value i _γ ^* and the γ-axis current detection value i _γf is calculated by the subtractor 11a, and this deviation is amplified by the γ-axis current controller 12a to obtain the γ-axis voltage command value v _γACR (γ Calculate the shaft voltage feedback control value v _γACR ). The γ-axis current controller 12a operates so that the deviation between the γ-axis current command value i _γ ^* and the γ-axis current detection value i _γf becomes zero to calculate the γ-axis voltage feedback control value v _γACR . Further, the deviation between the δ-axis current command value i _δ ^* and the δ-axis current detection value i _δf is calculated by the subtractor 11b, and this deviation is amplified by the δ-axis current controller 12b to obtain the δ-axis voltage command value v _δACR. (δ-axis voltage feedback control value v _δACR ) is calculated. The δ-axis current controller 12b operates so that the deviation between the δ-axis current command value i _δ ^* and the δ-axis current detection value i _δf becomes zero to calculate the δ-axis voltage feedback control value v _δACR .

The parameters of the flux model are estimated using harmonic currents that flow due to magnetic saturation of the motor core. Therefore, in order to prevent the γ-axis current regulator 12a and the δ-axis current regulator 12b from acting on harmonic currents, the response frequencies of the regulators 12a and 12b and the cutoff frequencies of the low-pass filters 10a and 10b are It is set smaller than twice the angular frequency _ωh of the alternating voltage.

The DC armature resistance compensator 13 calculates the γ-axis armature resistance feedforward compensation value v _γra by Equation 2, as in the first embodiment.

The voltage compensation value calculator 14 uses the q-axis reactance estimated value X _qhest and the q-axis armature resistance estimated value R _qhest calculated by the impedance estimator 15 to calculate the δ-axis voltage feedforward compensation value v _{δhFF according} to Equation 19. do. Impedance estimation such as the q-axis reactance estimated value X _qhest and the q-axis armature resistance estimated value R _qhest may be performed by the impedance estimator 15 using the known technique described in Patent Document 1, for example. It is possible.

Also, the calculation of the q-axis inductance can be performed by the impedance estimator 15 using known techniques. For example, the impedance estimator 15 calculates the q-axis inductance L _qest by dividing the q-axis reactance estimated value X _qhest by the angular frequency ω _h (L _qest =X _qhest /ω _h ).

The Fourier coefficient calculator 16 calculates Fourier coefficients of the _δ -axis voltage and the γ-axis current based on _{θh, vδ *} ^and iδ. The Fourier coefficient calculator 16 calculates the Fourier cosine coefficient V _δ(a1) of the fundamental wave component of the δ-axis voltage, the Fourier sine coefficient V _δ(b1) of the fundamental wave component of the δ-axis voltage, and the Fourier coefficient of the fundamental wave component of the δ-axis current. Cosine coefficient I _δ(a1) , Fourier sine coefficient I _δ(b1) of the fundamental wave component of the δ-axis current, Fourier cosine coefficient I _δ(a3) of the triple harmonic component of the δ-axis current, and Compute the Fourier sine coefficient I _δ(b3) of the triple harmonic component. Fourier coefficients can be calculated using a known technique described in Patent Document 1, for example.

The Fourier cosine coefficient _Iγ(a2) of the double harmonic component of the γ-axis current, the Fourier sine coefficient _Iγ(b2) of the double harmonic component of the γ-axis current, and the δ-axis calculated by the Fourier coefficient calculator 16 From the Fourier cosine coefficient I _δ(a3) of the third harmonic component of the current and the Fourier sine coefficient I _δ(b3) of the third harmonic component of the δ-axis current, the armature resistance compensator 17 calculates the γ-axis A harmonic armature resistance compensation value v _γhra and a δ-axis harmonic armature resistance compensation value v _δhra are calculated.

The magnetic flux model coefficient estimator 18 receives the Fourier coefficient I _γ(0) of the γ-axis current DC component and the Fourier cosine coefficient I _{γ(a2 )} , Fourier sine coefficient _Iγ(b2) of the double harmonic component of the γ-axis current, Fourier cosine coefficient _Iδ(a1) of the fundamental wave component of the δ-axis current, Fourier sine coefficient _{Iδ( b1)} , the Fourier cosine coefficient I _δ(a3) of the triple harmonic component of the δ-axis current, and the Fourier sine coefficient I _δ(b3) of the triple harmonic component of the δ-axis current are input. The magnetic flux model coefficient estimator 18 calculates respective coefficients (parameters) of the d-axis magnetic flux model such as Equation 5 and the q-axis magnetic flux model such as Equation 12, for example.

In a second embodiment, flux model coefficient estimator 18 computes K _{Ld ,} K _sd , K _sdq , K _Lq , K _sq and K _sqd using the above Fourier coefficients, e.g. Calculated by a known calculation method such as For example, the magnetic flux model coefficient estimator 18 uses the known technology described in Patent Document 1 or the like to calculate parameters of the magnetic flux model such as K _{Ld ,} K _sd , K _sdq , K _Lq , K _sq and K _sqd . perform auto-tuning.

The γ-axis voltage command value v _γ ^* is obtained by adding the γ-axis voltage feedback control value v _γACR , the γ-axis armature resistance feedforward compensation value v _γra , and the γ-axis harmonic armature resistance compensation value v _γhra by adders 7b and 7d. Calculated by adding. On the other hand, the δ-axis voltage command value v _δ ^* is obtained by adding the δ-axis voltage feedback control value v _δACR , the δ-axis voltage feedforward compensation value v _δhFF , and the δ-axis harmonic armature resistance compensation value v _δhra by adders 7c and 7e. Calculated by adding.

The γ, δ axis voltage command values _{v γ} ^* , v _δ ^* calculated as described above are converted by the voltage coordinate converter 20 into u, It is converted into phase voltage command values v _u ^* , v _v ^* , v _w ^* of the v and w phases.

The rectifier circuit 3 rectifies the three-phase AC voltage from the three-phase AC power supply 4, converts it into a DC voltage, and supplies this DC voltage to the power converter 2 such as an inverter.

The PWM circuit 21 controls the output voltage of the power converter 2 to the phase voltage command values _{vu*, vv*, vw} ^* _based ^on _the ^phase voltage command values _vu ^* , _vv ^* , _vw ^* . Generate multiple gate signals for Based on a plurality of gate signals from the PWM circuit 21, the power converter 2 controls a plurality of semiconductor switching elements inside the power converter 2 to change the terminal voltage of the synchronous motor 1 such as IPMSM or SynRM to a phase voltage. Control to command values _vu ^* , _vv ^* , _vw ^* .

(3.2) Configuration of Shaft Offset Compensator The configuration of the shaft offset compensator 19 of FIG. 1 in the second embodiment will be described. First, the calculation principle of the axis deviation amount θ _err will be explained using the γ-δ axis and the dq axis. Equation 22 is the amplitude of the dq axis voltage when the axis shift occurs. As a result, a fundamental wave current having an amplitude shown in Equation 7 is generated on the dq axes. However, for simplicity, the armature resistance is ignored.

Also, the γ-δ axis current fundamental wave amplitude can be expressed by Expression 8 using the d-q axis current fundamental wave amplitude and the axis deviation amount θ _err . By substituting Equation 22 and Equation 7 into Equation 8 and arranging them, the δ-axis current fundamental wave amplitude and the γ-axis current fundamental wave amplitude are obtained as shown in Equations 23 and 24. Assuming that the δ-axis voltage amplitude V _δ is positive, according to Equation 23, the δ-axis current fundamental wave amplitude I _δ is positive regardless of the _axis deviation amount θ _err . has polarities depending on the d-axis reactance X _d , the q-axis reactance X _q and the axis deviation θ _err .

Transformation of Equation 24 yields Equation 25. Here, assuming that 2θ _err is sufficiently small and approximated by sin θ≈θ, the axis deviation amount θ _err can be calculated by Equations 26 and 27.

Based on the principle described above, the axis deviation compensator 19 for correcting the axis deviation amount θ _err will be described with reference to FIG. 3 .

The amplitude calculator 22b calculates the γ-axis fundamental current Fourier sine coefficient correction value I _γ(b1) ' corresponding to the γ-axis current fundamental wave amplitude I _γ (oscillating current) of Equation 26 using Equations 28 and 29. . As a result, the amplitude can be calculated in consideration of the aforementioned amplitude polarity.

The G _θ calculator 23b calculates the γ-axis current-axis deviation amount conversion gain G _θγ from Equation (27). Also, the δ-axis voltage amplitude V _δ is calculated by Equation (30).

I _γ(b1) ' and G _θγ calculated as described above are multiplied by the multiplier 24b as shown in Equation 26 to calculate the compensation amount θ _err for compensating for _misalignment . It is input to the speed estimator 25b. The speed estimator 25b is composed of, for example, a PI controller, and calculates the speed estimated value ω _{r_est} as shown in Equation (17).

The integrator 5c integrates the speed estimated value ω _{r_est} and outputs the magnetic pole position estimated value θ _{r_est} . By constructing the axis deviation compensator as described above, it is possible to flow the tuning current in an appropriate phase even when the magnetic pole position is deviated during tuning. In other words, the control device 100 can control the phase of the γ, δ orthogonal rotating coordinate system so that the d-axis and the γ-axis are aligned by the compensation by the axis deviation compensator 19 .

FIG. 5 is a diagram illustrating a hardware configuration of an arithmetic unit included in the control device; The control device includes a computing device that performs computation based on a magnetic flux model that takes into account at least the magnetic saturation characteristics of the motor core. FIG. 5 shows a microcomputer 110, which is an example of an arithmetic device. The microcomputer 110 includes a memory 121 , a CPU (Central Processing Unit) 122 , an AD (Analog to Digital) conversion section 123 , a PWM module 124 , a communication section 125 and a timer 126 . The CPU 122 is a processor that controls the control device. The communication unit 125 communicates with a host controller outside the microcomputer 110 . The timer 126 counts timer values. The memory 121 stores programs and the like. A program in the memory 121 causes the CPU 122 to operate. The function of each control block in FIG. 1 is implemented by the CPU 122 operating according to a program stored in the memory 121 in a readable manner.

Each control block in FIG. , DC armature resistance compensator 13, low-pass filters 10b and 10c, voltage compensation value calculator 14, armature resistance compensator 17, integrator 5a, Fourier coefficient calculator 16, impedance estimator 15, magnetic flux model coefficient estimator 18 and a misalignment compensator 19 .

The function of each control block in FIG. 1 can be provided by a program that causes a computer to implement each function. Also, the function of each control block can be provided by a computer-readable recording medium recording the above program, or a computer program product such as the above program. Examples of recording media that can be used include flexible disks, hard disks, optical disks, magneto-optical disks, CD-ROMs, magnetic tapes, nonvolatile memory cards, and ROMs.

Although the control device for the synchronous motor has been described above with reference to the embodiments, the present invention is not limited to the above embodiments. Various modifications and improvements such as combination or replacement with part or all of other embodiments are possible within the scope of the present invention.

1 synchronous motor 2 power converter 3 rectifier circuit 4 three-phase AC power supply 5a-5c integrators 6a, 6b alternating current command calculators 7a-7e adder 8 current coordinate converters 9u, 9w current detectors 10a, 10b low-pass filter 11a , 11b subtractors 12a, 12b current controller 13 DC armature resistance compensator 14 voltage compensation value calculator 15 impedance estimator 16 Fourier coefficient calculator 17 armature resistance compensator 18 magnetic flux model coefficient estimator 19 axis deviation compensator 20 Voltage coordinate converter 21 PWM circuit 100 Control device 110 Microcomputer

Claims (6)

A coordinate system defined separately from the d, q orthogonal rotating coordinate system defined by the d-axis parallel to the rotor magnetic pole direction of the synchronous motor and the q-axis orthogonal to the d-axis, and is defined by the γ-axis and the δ-axis A control device for controlling the current and voltage supplied to the synchronous motor by a power converter in a defined γ, δ orthogonal rotating coordinate system,
A computing device that performs computation based on a magnetic flux model that takes into account at least the magnetic saturation characteristics of an iron core of the motor,
The computing device is
controlling the DC component of the γ-axis current of the synchronous motor to a command value;
applying a sinusoidal alternating voltage to the γ-axis,
performing at least one of the calculation of the d-axis inductance and the calculation of the parameters of the magnetic flux model;
calculating the Fourier coefficient of the fundamental wave component of the γ-axis voltage of the synchronous motor;
calculating the Fourier coefficient of the fundamental wave component of the δ-axis current of the synchronous motor;
Using the Fourier coefficient of the fundamental wave component of the γ-axis voltage and the Fourier coefficient of the fundamental wave component of the δ-axis current, the γ, δ orthogonal rotating coordinate system is set such that the d-axis and the γ-axis are aligned. A control device for a synchronous motor, characterized by controlling a phase. The computing device is
calculating the Fourier coefficient of the fundamental wave component of the γ-axis current;
2. The synchronous motor according to claim 1, further using a Fourier coefficient of a fundamental wave component of said γ-axis current to control the phase of said γ, δ orthogonal rotating coordinate system so that said d-axis and said γ-axis are aligned. controller. The computing device is
detecting an oscillating current generated in the δ-axis using the Fourier coefficient of the fundamental wave component of the δ-axis current and the Fourier coefficient of the fundamental wave component of the γ-axis current;
3. The phase of the γ, δ orthogonal rotating coordinate system is controlled so that the d-axis and the γ-axis are aligned using the fundamental wave component of the γ-axis voltage and the detected value of the oscillating current. Control device for a synchronous motor as described. A coordinate system defined separately from the d, q orthogonal rotating coordinate system defined by the d-axis parallel to the rotor magnetic pole direction of the synchronous motor and the q-axis orthogonal to the d-axis, and is defined by the γ-axis and the δ-axis A control device for controlling the current and voltage supplied to the synchronous motor by a power converter in a defined γ, δ orthogonal rotating coordinate system,
A computing device that computes based on a magnetic flux model that takes into account at least the magnetic saturation characteristics of the motor core,
The computing device is
controlling the DC component of the γ-axis current of the synchronous motor to a command value;
applying a sinusoidal alternating voltage to the δ-axis,
performing at least one of the calculation of the q-axis inductance and the calculation of the parameters of the magnetic flux model;
calculating the Fourier coefficient of the fundamental wave component of the δ-axis voltage of the synchronous motor;
calculating the Fourier coefficient of the fundamental wave component of the γ-axis current of the synchronous motor;
Using the Fourier coefficient of the fundamental wave component of the δ-axis voltage and the Fourier coefficient of the fundamental wave component of the γ-axis current, the γ, δ orthogonal rotating coordinate system is set such that the d-axis and the γ-axis are aligned. A control device for a synchronous motor, characterized by controlling a phase. The computing device is
calculating the Fourier coefficient of the fundamental wave component of the δ-axis current of the synchronous motor;
5. The synchronous motor according to claim 4, further using a Fourier coefficient of a fundamental wave component of said δ-axis current to control the phase of said γ, δ orthogonal rotating coordinate system so that said d-axis and said γ-axis are aligned. controller. The computing device is
detecting an oscillating current generated in the γ-axis using the Fourier coefficient of the fundamental wave component of the γ-axis current and the Fourier coefficient of the fundamental wave component of the δ-axis current;
6. The phase of the γ, δ orthogonal rotating coordinate system is controlled so that the d-axis and the γ-axis are aligned using the fundamental wave component of the δ-axis voltage and the detected value of the oscillating current. Control device for a synchronous motor as described.

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