JP5862125B2 - Control device for power converter - Google Patents

Description

  The present invention relates to a control device for a power conversion device that drives a motor at a variable speed, and more particularly to a control device for a power conversion device that includes a DC-DC converter and an inverter.

  As a power source for electric vehicles, a permanent magnet synchronous motor with a wide variable speed range is expected. As a power source for driving the permanent magnet synchronous motor at a variable speed, an inverter that is a variable frequency power source is generally used. Here, when a permanent magnet synchronous motor with a wide variable speed range is connected to the inverter as a load, the induced voltage of the permanent magnet synchronous motor rises in proportion to the rotational speed of the motor, and the counter electromotive voltage induced in the motor In contrast, the output voltage of the inverter may be insufficient. If the output voltage of the inverter is insufficient in this way, a desired current cannot be passed from the inverter to the motor, so that there is a possibility that a necessary torque cannot be generated in the motor. Therefore, conventionally, in the region where the inverter is operated in the asynchronous PWM (Pulse Width Modulation) mode, the output of the inverter with respect to the counter electromotive voltage induced in the motor by the maximum torque / current control or the flux weakening control. Measures were taken to avoid the problem of insufficient voltage margin. Here, the asynchronous PWM mode is a gate which is a PWM pulse by pulse width modulation using a voltage command for instructing an AC voltage waveform to be supplied from the inverter to the motor and a carrier having a predetermined frequency asynchronous to the voltage command. This is a mode for generating a signal.

In the maximum torque / current control and the flux weakening control, the current flowing to the motor is controlled. Current flowing through the motor armature winding (stator winding) of a d-axis current i _d is a component along the d-axis oriented in the direction of the N pole of the rotor of the permanent magnets, perpendicular to the d-axis It can be decomposed into a q-axis current i _q which is a component along the q-axis. Here, the q-axis current i _q contributes to the generation of magnet torque in the motor, and the d-axis current i _d and the q-axis current i _q contribute to the generation of reluctance torque. In the maximum torque / current control and the flux-weakening control, a current vector having the d-axis current _id and the q-axis current _iq as a component is controlled.

In the maximum torque / current control, a current vector having the maximum torque is selected from a current vector group having the same absolute value, and the maximum torque is associated with a current vector from which the torque is obtained. When the required torque is given to obtain the current vector associated with that torque, for the flow of such a current which is a component of the vector d-axis current i _d and the q-axis current i _q to the motor Perform current phase control. By performing this maximum torque / current control, the current flowing through the motor is minimized to obtain the required torque, and the copper loss generated in the motor is minimized, so that the back electromotive voltage of the motor and the output voltage of the inverter It is possible to reduce the margin to be provided between the two.

The field weakening control reduces the counter electromotive voltage generated in the armature winding by rotating the rotor by passing a negative d-axis current _id through the motor armature winding, thereby increasing the q-axis current i _q . To increase the torque of the motor. The field weakening control and the above-described maximum torque / current control are disclosed in Non-Patent Document 1, for example.

  By performing field weakening control, the problem of insufficient torque in a region where the rotational speed of the motor is high can be solved to some extent. However, there is a limit to field weakening control, and if the rotational speed of the motor exceeds a certain limit, a problem that a desired torque cannot be obtained in the high speed rotation region occurs even if field weakening control is performed in the asynchronous PWM mode.

  Therefore, there is a case where control is performed such that the gate signal generation mode in the control device is switched from the asynchronous PWM mode to, for example, the one-pulse synchronous PWM mode. Here, in the synchronous PWM mode, a gate signal that is a PWM pulse is generated by pulse width modulation using a voltage command instructing an AC voltage waveform to be supplied from the inverter to the motor and a carrier synchronized with the voltage command. It is a mode to do. The 1-pulse synchronous PWM mode is a mode in which one PWM pulse is generated during one cycle of the voltage command. By switching to the synchronous PWM mode such as one pulse, a high fundamental wave voltage can be supplied from the inverter to the motor, so that the problem of insufficient torque in the high-speed rotation region can be solved.

JP 2004-208409 A

Takeda / Matsui / Morimoto / Honda, “Design and Control of Embedded Permanent Magnet Synchronous Motor”, published by Ohmsha, 2001.10

  By the way, in the synchronous PWM mode, for example, when the output voltage of the inverter is larger than the counter electromotive voltage of the motor, a positive d-axis current flows through the motor, the magnetic flux density in the motor increases, and the motor loss (that is, In some cases, the output of the inverter that does not contribute to torque generation) increases. Patent Document 1 relates to a vehicle power control apparatus in which a DC voltage is applied to an inverter from a step-up DC-DC converter and a motor is driven by the inverter. In this vehicle power control apparatus, the power consumption of the motor is described. Based on the above, the circuit loss is reduced by reducing the step-up ratio of the DC-DC converter when the motor load is low and the power is low. Therefore, in accordance with Patent Document 1, it is conceivable to adopt a method of reducing the step-up ratio of the DC-DC converter that supplies a DC voltage to the inverter when the motor load is light during operation of the inverter in the synchronous PWM mode. . However, when the inverter is operated in the synchronous PWM mode, the motor rotates at a high speed and a high counter electromotive voltage is generated. Therefore, if the step-up ratio of the DC-DC converter is lowered based only on the motor load, the inverter output voltage becomes insufficient with respect to the back electromotive voltage of the motor, causing a problem that necessary torque cannot be obtained.

  The present invention has been made in view of the circumstances described above, and avoids an increase in motor loss without causing a problem of insufficient torque when the inverter is operated in the synchronous PWM mode. It aims at providing the control apparatus of the power converter device which can suppress an efficiency fall.

  The present invention relates to a control device for a power converter having DC voltage generating means for outputting a DC voltage, and an inverter for generating an AC voltage for driving a motor based on the DC voltage output from the DC voltage generating means. A means for generating a gate signal for switching ON / OFF of the switching element of the inverter, and a voltage command for instructing an AC voltage waveform to be supplied from the inverter to the motor as the generation mode of the gate signal; Asynchronous PWM mode for generating the gate signal by pulse width modulation using a carrier having a predetermined frequency asynchronous to the voltage command, and by pulse width modulation using the voltage command and a carrier synchronized with the voltage command. An inverter control means having a synchronous PWM mode for generating a gate signal; When the generation mode of the gate signal of the control means is the synchronous PWM mode, it is a component corresponding to the direction of the N pole of the permanent magnet provided in the rotor of the motor among the current supplied from the inverter to the motor. DC power command value calculating means for calculating a command value indicating a DC voltage supplied from the DC voltage generating means to the inverter so that a certain d-axis current becomes 0 or negative. A control device for a conversion device is provided.

  According to the present invention, when the inverter is operated in the synchronous PWM mode, the DC voltage supplied from the DC voltage generating means to the inverter is controlled so that the d-axis current flowing to the motor becomes zero or negative. Therefore, an increase in motor loss can be avoided and a decrease in efficiency of the motor drive system can be suppressed without causing a problem of insufficient torque.

  Many control devices for power conversion devices include a processor and a memory that stores a program to be executed by the processor. Therefore, assuming various motors, a program for causing a computer to function as the control device may be created, and this program may be distributed to users of the control device of the power conversion device.

It is a block diagram which shows the structure of the motor drive system containing the control apparatus which is 1st Embodiment of this invention. It is a figure which shows the relationship between the load angle and torque in the same embodiment. FIG. 5 is a voltage vector diagram when the back electromotive voltage of the motor is lower than the output voltage of the inverter and the motor is unloaded in the same embodiment. FIG. 6 is a voltage vector diagram when the back electromotive voltage of the motor is lower than the output voltage of the inverter and a constant load is applied to the motor. It is a voltage vector figure in case the counter electromotive voltage of a motor is equal to the output voltage of an inverter. It is a vector diagram which shows the magnetic flux in a motor in case a total magnetic flux is larger than the armature linkage magnetic flux by the permanent magnet of a motor, and it is no load. It is a vector diagram which shows the magnetic flux in a motor in case a total magnetic flux is larger than the armature linkage magnetic flux by the permanent magnet of a motor, and is a light load. It is a vector diagram which shows the magnetic flux in a motor in case the armature interlinkage magnetic flux by a permanent magnet of a motor is equal to a total magnetic flux, and the load is given to the motor. In the same embodiment, when the total magnetic flux is larger than the armature interlinkage magnetic flux by the permanent magnet of the motor, the result of the motor loss analysis for each of the case where the armature interlinkage magnetic flux by the permanent magnet is equal to the total magnetic flux is shown. FIG. It is a block diagram which shows the structure of the control apparatus which is 2nd Embodiment of this invention. It is a vector diagram which shows the magnetic flux in a motor and the electric current which flows into a motor in case the armature interlinkage magnetic flux by a permanent magnet of a motor is equal to a total magnetic flux, and a heavy load is given to the motor. It is a vector diagram which shows the effect of the embodiment. It is a block diagram which shows the structure of the control apparatus which is 3rd Embodiment of this invention. It is a block diagram which shows the structure of the control apparatus which is 4th Embodiment of this invention.

  Embodiments of the present invention will be described below with reference to the drawings.

<First Embodiment>
FIG. 1 is a block diagram showing a configuration of a motor drive system including a control device according to a first embodiment of the present invention. This motor drive system includes a DC power source 10 and a DC-DC converter 20, which constitute DC voltage generating means, a capacitor 30, an inverter 40, a motor 50, and a control device 100 according to the present embodiment.

  In this example, the DC-DC converter 20 includes two IGBTs (Insulated Gate Bipolar Transistors) connected in series with each other, two flywheel diodes connected in antiparallel to these IGBTs, respectively. , A well-known DC-DC converter composed of a step-up reactor. Here, one of the two IGBTs, the boost reactor, and the DC power supply 10 are connected in series to form a closed loop.

In the DC-DC converter 20, only one of the IGBTs is turned on to pass a current through a closed loop including the boosting reactor, and one of the IGBTs is turned off and the other IGBT is turned on to turn on the boosting reactor. The operation of flowing the current flowing in the capacitor 30 to the capacitor 30 side through the flywheel diode connected in antiparallel with the other IGBT is alternately repeated. Thereby, the capacitor 30 is charged with the DC intermediate voltage e _dc obtained by boosting the output voltage of the DC power supply 10. The DC-DC converter 20 can control the DC intermediate voltage e _dc charged in the capacitor 30 by adjusting the time for turning on the two IGBTs.

The inverter 40 is means for converting a DC intermediate voltage e _dc that is a charging voltage of the capacitor 30 into a three-phase AC voltage for driving the motor 50. This inverter 40 is a bridge circuit configured by using six sets of IGBTs and flywheel diodes as in the case of known inverters. In this example, the motor 50 is a permanent magnet synchronous motor.

  The control device 100 according to the present embodiment includes a current detection unit 101, a rotation speed detection unit 102, a DC voltage detection unit 103, an inverter control unit 110, an asynchronous / synchronous determination unit 120, a converter control unit 130, a pole A number storage unit 141, a counter electromotive voltage storage unit 142, a DC voltage command calculation unit 143, and a comparison unit 144.

The current detection unit 101 is a device that detects an AC current of each phase of U, V, and W supplied from the inverter 40 to the armature winding of the motor 50. The rotation speed detection unit 102 is a device that detects the rotation speed of the rotor of the motor 50 per unit time, that is, the rotation speed n. The DC voltage detection unit 103 is a device that detects the DC intermediate voltage e _dc charged in the capacitor 30.

  The inverter control unit 110 is a device that generates a gate signal for performing ON / OFF switching of each IGBT of the inverter 40. More specifically, the inverter control unit 110 generates a voltage command for instructing an AC voltage waveform to be supplied to the motor 50 based on a torque command or the like given from the outside, and uses this voltage command and a carrier to generate a pulse. Width modulation is performed, and a PWM pulse obtained by the pulse width modulation is supplied to each IGBT of the inverter 40 as a gate signal.

  The inverter control unit 110 has an asynchronous PWM mode and a one-pulse synchronous PWM mode as gate signal generation modes. As described above, the asynchronous PWM mode is a generation mode in which a PWM pulse is generated by pulse width modulation using a voltage command and a carrier having a predetermined frequency asynchronous to the voltage command, and is output as a gate signal. The synchronous PWM mode is a generation mode in which a PWM pulse is generated by pulse width modulation using a voltage command and a carrier synchronized with the voltage command, and is output as a gate signal. Hereinafter, an outline of the operation of the inverter control unit 110 in each of these modes will be described.

First, the asynchronous PWM mode will be described. The torque T generated in the rotor of the motor 50 that is a permanent magnet synchronous motor is given by the equation (1).

In this formula (1), _{P n} is the number of pole pairs, [psi _m is generated by the rotor of the permanent magnet, the armature winding and the magnetic flux interlinking, _{i d} is the d-axis current, _{i q} is the q-axis current, _{L d} Is a d-axis inductance, and L _q is a q-axis inductance. In the formula (1), the first term is a magnet torque generated by a magnetic flux generated by a permanent magnet, and the second term is a reluctance torque.

In the asynchronous PWM mode, the inverter control unit 110 controls a gate signal to be supplied to the inverter 40 so that a current for obtaining a torque corresponding to the torque command value is supplied from the inverter 40 to the motor 50. At this time, when the output voltage of the inverter 40 has a margin with respect to the terminal voltage of the motor 50, the d-axis current _id and the q-axis current _iq are controlled so that the current value becomes the minimum. When the output voltage of the inverter 40 is lower than the voltage, field weakening control is performed.

  Next, the synchronous PWM mode will be described. Here, a one-pulse synchronous PWM mode will be described as an example.

上記式（２）および（３）において、Ｒ_ａはモータ５０の電機子巻線の巻線抵抗、ωはモータ５０の回転速度により決まる電気角速度である。 In a steady state, when the AC voltage applied to the armature winding of the motor 50 is decomposed into a d-axis voltage v _d that is a component in the d-axis direction and a q-axis voltage v _q that is a component in the q-axis direction, these d-axes The voltage v _d and the q-axis voltage v _q are given by the equations (2) and (3).

In the above formulas (2) and (3), R _a is the winding resistance of the armature winding of the motor 50, and ω is the electrical angular speed determined by the rotational speed of the motor 50.

Further, the relationship between the terminal voltage v _mt of the motor 50, the d-axis voltage v _d and the q-axis voltage v _q is as shown in the following equation.

Here, assuming that the winding resistance is sufficiently small (R _a ≈0), substituting v _d = −V _a · sin δ and v _q = V _a · cos δ into the equations (2) and (3), When Expressions (2) and (3) are solved for i _d and i _q and substituted into Expression (1), Expression (5) is obtained. However, V _a is the output voltage of the inverter 40, [delta] is the load angle, i.e., an angle formed between the total magnetic flux [psi ₀ of the same direction as the armature flux linkage [psi _m by the rotor of the permanent magnet generated in the motor 50 is is there.

In the 1-pulse synchronous PWM mode, the inverter control unit 110 causes the inverter 40 to output _a constant rectangular wave voltage Va having the same frequency as the voltage command. When the inverter DC voltage charged in the capacitor 30 and _{e dc,} the output voltage _{V a} of the inverter 40 is given by equation (6).

In the synchronous PWM mode 1 pulse, the voltage V _a in the above formula (5) is constant, the torque T generated in the motor 50 is dependent on the load angle [delta]. FIG. 2 shows the relationship between the load angle δ and the torque in equation (5). The region where the load angle δ is positive is a region where power running (operation as a motor) is performed in the motor 50. The region where the load angle δ is negative is a region where regeneration (operation as a generator) is performed in the motor 50.
The above is the outline of the operations in the asynchronous PWM mode and the one-pulse synchronous PWM mode.

  The inverter control unit 110 changes the gate signal generation mode from the asynchronous PWM mode to the one-pulse synchronous PWM mode or the one-pulse synchronous PWM mode based on, for example, the rotation speed of the motor 50 detected by the rotation speed detection unit 102. To asynchronous PWM mode. The asynchronous / synchronous determination unit 120 is a device that determines whether the generation mode of the gate signal of the inverter control unit 110 is an asynchronous PWM mode or a synchronous PWM mode.

Converter control unit 130 is a device that controls DC intermediate voltage e _dc supplied from DC-DC converter 20 to inverter 40. In the present embodiment, the efficiency of the motor drive system is reduced by appropriately controlling the DC intermediate voltage e _dc by the converter control unit 130 during the period in which the inverter control unit 110 generates the gate signal in the one-pulse synchronous PWM mode. Suppress. In the following, in order to facilitate understanding of the control of the DC intermediate voltage e _dc in the present embodiment, prior to the description of the control of the DC intermediate voltage e _dc , the inverter control unit 110 outputs the gate signal in the 1-pulse synchronous PWM mode. The efficiency of the motor drive system when it is generated will be described.

First, the current supplied from the inverter 40 to the motor 50 depends on the magnitude relationship between the counter electromotive voltage v _me of the motor 50 and the output voltage V _a of the inverter 40. 3, when the counter electromotive voltage v _me of the motor 50 is lower than the output voltage V _a of the inverter 40, the motor 50 is in an unloaded condition, i.e., the voltage vector diagram in the state load angle δ of the motor 50 is 0 Is shown. However, the voltage drop due to the winding resistance of the armature of the motor 50 is ignored here. As shown in FIG. 3, the armature of the motor 50, in accordance with the difference between the output voltage _{V a} of the counter electromotive voltage _{v me} the inverter 40 of the motor 50, it flows d-axis current _{i d.} The polarity of the d-axis current _{i d} when the counter electromotive voltage _{v me} of the motor 50 is lower than the output voltage _{V a} of the inverter 40 becomes positive.

Here, in the asynchronous PWM mode, the inverter control unit 110 controls the inverter 40 such that a current necessary for obtaining a desired torque is supplied from the inverter 40 to the motor 50, and at that time, the motor 50 current value to control the d-axis current i _d and the q-axis current i _q to be the minimum when there is a margin in the output voltage V _a of the inverter 40 to the terminal voltage v _mt. Accordingly, in the asynchronous PWM mode, if the motor 50 is unloaded and the required torque is zero, the current supplied from the inverter 40 to the motor 50 is almost zero, and no loss occurs in the inverter 40.

However, the flow at the time of operation in the synchronous PWM mode, as shown in FIG. 3, the counter electromotive voltage v _me the motor 50 from the current inverter 40 which does not contribute to the torque when the output voltage is lower than V _a of the inverter 40 of the motor 50 A loss occurs in the inverter 40. At the same time, since a current also flows through the motor 50, copper loss occurs in the armature winding of the motor 50. Also, when d-axis current i _d is positive, occurs Zo磁effect, as a result, the magnetic flux density of the armature is increased, there is a problem that the iron loss of the armature is increased.

4, when the counter electromotive voltage v _me of the motor 50 is lower than the output voltage V _a of the inverter 40 shows a voltage vector diagram in the case where constant load is applied to the motor 50. If the DC intermediate voltage e _dc inverter 40 As is clear from the supra formula (6) is constant, the output voltage V _a of the inverter 40 is also constant, the output voltage V _a is to remain depicts a constant radius arc. In the example shown in FIG. 4, although the q-axis current i _q by the load applied to the motor 50 through the motor 50, still the output voltage V _a of the inverter 40 with respect to the counter electromotive voltage v _me of the motor 50 is excessive For this reason, the d-axis current i _d remains positive. Even in this state, there is a problem that the iron loss of the armature increases because the above-described magnetizing effect occurs.

However, from the state shown in FIG. 3, via the state shown in FIG. 4, further load on the motor 50 increases and the load angle δ increases, eventually negatively via the d-axis current i _d is 0 of the motor 50 Change.

Figure 5 is the back electromotive voltage v _me of the motor 50 indicates a voltage vector diagram in a case equal to the inverter output voltage V _a. If the counter electromotive voltage v _me and the inverter output voltage V _a of the motor 50 are equal, the current hardly flows in the motor 50 of the no-load state, the load of the motor 50 increases when the d-axis current i _d and the q-axis current i _q Flows, and the d-axis current _id is always negative. Therefore, no magnetizing effect is generated in the motor 50, and an increase in the iron loss of the armature can be suppressed.

Therefore, in this embodiment, in order to prevent the iron loss of the motor 50 is increased in the synchronous PWM mode, the DC-DC converter 20 as the d-axis current i _d flowing through the motor 50 becomes zero or negative in the inverter 40 The converter controller 130 controls the DC intermediate voltage edc to be supplied. This respect d-axis current i _d is 0 or a method of calculating the command value of the DC intermediate voltage e _dc negative to become various methods can be considered, in the present embodiment, the output voltage V _a of the inverter 40 is opposite the motor 50 The DC voltage command value calculation unit 143 calculates a command value of the DC intermediate voltage e _dc of the inverter 40 that is equal to the electromotive voltage v _me .

Hereinafter, control of the DC intermediate voltage e _dc performed in the present embodiment will be described. First, in the above equation (6), when V _a = v _me and e _dc is solved, equation (7) is obtained.

In Figure 1, the pole number storage section 141 of the control device 100, the counter electromotive voltage storage unit 142 and the DC voltage command value calculating unit 143 obtains the current counter electromotive voltage _{v me} of the motor 50, the counter electromotive voltage _{v me} The means for calculating the command value of the DC intermediate voltage e _dc is configured according to the above equation (7). More specifically, the counter electromotive voltage storage unit 142 stores, for example, a base frequency f _base and a counter electromotive voltage v _emf of the motor 50 at the base frequency. Here, the base frequency f _base is obtained by converting the maximum value of the rotation speed of the motor 50 at which the motor 50 can operate without reducing the maximum torque into the frequency of the counter electromotive voltage of the motor 50. The pole number storage unit 141 stores the number of magnetic pole pairs P of the rotor in the motor 50.

The DC voltage command value calculation unit 143 first determines the counter electromotive voltage of the motor 50 based on the number P of magnetic pole pairs stored in the pole number storage unit 141 and the rotation speed n of the motor 50 detected by the rotation number detection unit 102. Is calculated according to equation (8).

Next, the DC voltage command value calculation unit 143 calculates the current counter electromotive voltage v _me of the motor 50 by the following equation (9).

Then, the DC voltage command value calculation unit 143 calculates the command value of the DC intermediate voltage e _dc by substituting the back electromotive voltage v _me calculated in this way into the above equation (7). That is, the DC voltage command value calculation unit 143 determines a command value for the DC intermediate voltage e _dc according to the following equation (10).

The comparison unit 144 compares the command value of the DC intermediate voltage e _dc calculated by the DC voltage command value calculation unit 143 with the voltage value of the DC intermediate voltage e _dc detected by the DC voltage detection unit 103, Is calculated and supplied to the converter control unit 130.

Then, converter control unit 130, when the voltage value of the DC intermediate voltage e _dc detected by the DC voltage detection unit 103 is larger than the command value calculated by the DC voltage command value computing section 143 DC link voltage e _dc When the voltage value of the DC intermediate voltage e _dc detected by the DC voltage detector 103 is smaller than the command value calculated by the DC voltage command value calculator 143, the DC intermediate voltage edc is increased. The DC-DC converter 20 is controlled.

In synchronous PWM mode, thus the DC intermediate voltage _{e dc} output voltage _{V a} of the inverter 40 is supplied from the DC-DC converter 20 to the inverter 40 so as to match the counter electromotive voltage _{v me} of the motor 50 is controlled . As a result, d-axis current _{i d} flowing through the motor 50 becomes zero or negative. Accordingly, it is possible to prevent loss in the motor 50 during operation in the synchronous PWM mode, and it is possible to suppress a decrease in efficiency of the motor drive system.

In the above description, by noting that flows through the motor 50 in the synchronous PWM mode d-axis current i _d is dependent on the magnitude relationship between the output voltage V _a of the counter electromotive voltage v _me the inverter 40 of the motor 50. Then, from the viewpoint of controlling the DC intermediate voltage e _dc so that the output voltage V _a coincides with the back electromotive voltage v _me , the command value of the DC intermediate voltage e _dc for setting the d-axis current _id to 0 or negative is set. Calculation formula (10) was derived. However, a calculation formula (10) for a command value of the DC intermediate voltage e _dc for setting the d-axis current _id to 0 or negative can be derived from a different viewpoint. The flows to the motor 50 in the synchronous PWM mode d-axis current i _d is dependent on the magnitude relation between total flux [psi ₀ by the output voltage V _a of the armature flux linkage [psi _m and the inverter 40 by the permanent magnet of the motor 50 or less Focus on that. Then, from the viewpoint of controlling the DC link voltage e _dc such that [psi _{m =} [psi _0, equation for calculating the command value of the DC intermediate voltage e _dc for the d-axis current i _d and 0 or negative (10) To derive.

First, the magnitude relation between the total flux [psi ₀ by the output voltage V _a of the armature flux linkage [psi _m and the inverter 40 by the permanent magnet of the motor 50, how current is described how flowing to the motor 50.

The armature interlinkage magnetic flux Ψ _m by the permanent magnet can be calculated from the base frequency f _base and the counter electromotive voltage v _{emf at the} base frequency by the equation (11). Here, since the frequency and the counter electromotive voltage are in a proportional relationship, the frequency to be stored and the counter electromotive voltage need not be those of the base frequency.

ここで、角速度ωは式（１３）により表される。

この式（１３）における周波数ｆは、前掲式（８）により算出可能である。 On the other hand, total flux [psi ₀ can be calculated by the output voltage V _a of the inverter 40 and angular velocity ω Tocharian formula (12).

Here, the angular velocity ω is expressed by Expression (13).

The frequency f in the equation (13) can be calculated by the above equation (8).

FIG. 6 shows a vector diagram of magnetic fluxes when Ψ _m <Ψ ₀ and no load, that is, a load angle δ = 0. As shown in FIG. 6, the flow d-axis current i _d under the conditions of Ψ _{m <Ψ 0} be unloaded, further d-axis current i _d is the flux strengthened so positive. FIG. 7 shows a vector diagram of magnetic fluxes when Ψ _m <Ψ ₀ and in a light load state. When the load angle δ increases, the q-axis current i _q begins to flow. However, the d-axis current i _d remains positive. FIG. 8 shows a vector diagram of magnetic fluxes when Ψ _m = Ψ ₀ and a load is applied to the motor 50. In this case, what loads the d-axis current i _d does not become positive even given to the motor 50. Needless to say, if the load is zero, that is, if the load angle δ = 0, theoretically no current flows through the motor 50.
As described above, if the DC intermediate voltage e _dc applied to the inverter 40 is controlled so that Ψ _m = Ψ ₀ , the loss of the motor 50 is not increased.

FIG. 9 shows the result of the loss analysis of the motor 50 under the conditions of Ψ _m <Ψ ₀ and Ψ _m = Ψ ₀ . However, in this loss analysis, the d-axis inductance L _d of the motor 50 was 0.88 mH, and the q-axis inductance L _q was 2.08 mH. Moreover, it was set as (psi) _m = 0.18Wb in both conditions. In the former condition, Ψ ₀ = 0.20 Wb, and in the latter condition, Ψ ₀ = 0.18 Wb. Under either condition, the motor load factor was 50%.

この表１にも示すように、Ψ_ｍ＜Ψ_０の条件ではｄ軸電流ｉ_ｄは正となり、Ψ_ｍ＝Ψ_０の条件ではｄ軸電流ｉ_ｄは負となる。 For each of the above two conditions, the load angle δ is calculated from the equation (5), and the d-axis current i _d and the q-axis current i _q are calculated from the equations (2) and (3). Show.

As also shown in this Table 1, d-axis current i _d is a condition Ψ _{m <Ψ 0} is positive, the d-axis current i _d is negative in terms of Ψ _{m =} Ψ _0.

As shown in FIG. 9, in the condition of Ψ ₀ = 0.20Wb, compared to the conditions of Ψ ₀ = 0.18Wb, is remarkable increase in loss of particular stator. In this, as shown in Table 1, [psi _{0 =} for 0.20Wb the d-axis current i _d is a positive stronger flow in the magnetic flux, increasing the magnetic flux density of the stator, because the loss was increased It is believed that there is. When Ψ ₀ = 0.18 Wb, the loss of the entire motor 50 was reduced by about 13% compared to when Ψ ₀ = 0.20 Wb.

Next, a method for determining the DC intermediate voltage e _dc of the inverter 40 will be described. First, in order to set Ψ _m = Ψ ₀ , the following expression must be established from the above expressions (11) and (12).

When this equation (14) is solved for V _a , equation (15) is obtained.

Solving equation (6) for e _dc and substituting V _a in equation (14) yields:

  Substituting ω obtained by substituting Equation (8) into Equation (12) into Equation (15) and rearranging it yields Equation (10) described above.

As described above, the control method of the DC intermediate voltage e _dc according to the present embodiment, at the same time when the output voltage V _a of the inverter 40 is a method of controlling the DC link voltage e _dc to match the counter electromotive voltage v _me It can be said that the DC intermediate voltage e _dc is controlled such that the armature interlinkage magnetic flux Ψ _m by the permanent magnet of the motor 50 becomes equal to the total magnetic flux Ψ ₀ by the output voltage V _a of the inverter 40. Then, according to the control method of the DC intermediate voltage e _dc according to the present embodiment, the d-axis current i _d flowing through the motor 50 is 0 or negative, to prevent an increase in loss in the motor 50 during operation in the synchronous PWM mode Can do.

Second Embodiment
FIG. 10 is a block diagram showing a configuration of an inverter control device 100A according to the second embodiment of the present invention. In the control device 100A in the present embodiment, the inverter control unit 110 and the DC voltage command value calculation unit 143 of the control device 100 (see FIG. 1) according to the first embodiment are replaced with the inverter control unit 110A and the DC voltage command value calculation unit 143A. Has been replaced. In FIG. 10, in order to prevent the drawing from becoming complicated, illustrations corresponding to the current detection unit 101, the DC voltage detection unit 103, and the asynchronous / synchronous determination unit 120 in FIG. 1 are omitted.

In the present embodiment, the inverter control unit 110 </ b> A includes a d-axis current q-axis current determination unit 111, an inductance storage unit 112, and a total magnetic flux calculation unit 113. In the present embodiment, the DC voltage command value calculation unit 143A calculates a command value of the DC intermediate voltage e _{dc in} cooperation with the total magnetic flux calculation unit 113 of the inverter control unit 110A.

As in the first embodiment, when the DC intermediate voltage e _dc is controlled so that the armature interlinkage magnetic flux Ψ _m of the motor 50 and the total magnetic flux Ψ ₀ by the output voltage V _a of the inverter 40 are equal, In this case, as shown in FIG. 11, as the load angle δ increases, the d-axis current _id becomes excessive, and the output current of the inverter 50 increases. This embodiment is an improvement of the first embodiment in this regard.

In the present embodiment, the total magnetic flux calculation unit 113 of the inverter control unit 110A, as illustrated in FIG. 12, performs the total magnetic flux of the motor 50 such that the d-axis current _id and the q-axis current _iq have appropriate current values. obtaining the [psi _0, the DC voltage command value calculating portion 143A a command value for the DC intermediate voltage e _dc inverter 40 to obtain the total magnetic flux [psi ₀ is calculated.

Further details are as follows. The d-axis current q-axis current determination unit 111 determines the d-axis current _id and the q-axis current i _q based on the torque command. At that time, the data may be stored in advance as a table, or the d-axis current _id and the q-axis current _iq may be always calculated.

In a preferred embodiment, the d-axis current q-axis current determination unit 111 determines a d-axis current i _d and the q-axis current i _q by the maximum torque / current control described in 23-24 pages in Non-Patent Document 1.

More specifically, in this embodiment, the various torque, current absolute value of the current vector of each component of d-axis current i _d and the q-axis current i _q that can generate the torque is minimized Find a vector. Then, a table defining the d-axis current i _d and the q-axis current i _q of the current vector with the minimum absolute value capable of generating the torque is created in association with each torque, and the d-axis current q-axis current is created. It is stored in the determination unit 111 in advance. Then, the d-axis current q-axis current determination unit 111 reads the d-axis current _id and the q-axis current i _q corresponding to the torque command from this table and outputs them to the total magnetic flux calculation unit 113.

The inductance storage unit 112 stores a d-axis inductance L _d and a q-axis inductance L _q of the motor 50. Further, the DC voltage command value calculation unit 143A, based on the base frequency f _base stored in the back electromotive voltage storage unit 142 and the back electromotive voltage v _emf of the motor 50 at the base frequency, according to the above equation (11), It calculates the armature flux linkage [psi _m of the permanent magnets in, and notifies the armature flux linkage [psi _m in total magnetic flux calculation unit 113.

Overall flux calculating unit 113, d-axis current q-axis and d-axis current _{i d} and the q-axis current _{i q} determined by the current determining unit 111, d-axis inductance is stored in the inductance storage unit 112 _{L d} and q-axis inductance The total magnetic flux Ψ ₀ is calculated according to the following equation using L _q and the armature linkage magnetic flux Ψ _m calculated by the DC voltage command value calculation unit 143A. Then, the total magnetic flux calculation unit 113 notifies the calculated total magnetic flux Ψ ₀ to the DC voltage command value calculation unit 143A. In addition, it is clear also from FIG. 8 that Formula (17) is materialized.

The DC voltage command value calculation unit 143A calculates the command value of the DC intermediate voltage e _dc obtained by the total magnetic flux Ψ ₀ calculated by the total magnetic flux calculation unit 113. Specifically, it is as follows. First, solving supra equation (12) for the output voltage _{V a} of the inverter 40, the following equation is obtained.

Next, when the above equation (6) is solved for e _dc and V _a of this equation (18) is substituted, the following equation is obtained.

Therefore, the DC voltage command value calculation unit 143A includes the total magnetic flux Ψ ₀ calculated by the total magnetic flux calculation unit 113, the rotation speed n detected by the rotation number detection unit 102, and the magnetic pole stored in the pole number storage unit 142. From the logarithm P, a command value for the DC intermediate voltage e _dc is calculated according to the above equation (19). Then, the converter control unit 130 controls the converter 20 for making the DC intermediate voltage e _dc supplied to the inverter 40 coincide with this command value.

By performing such control, during operation in the synchronous PWM mode, as illustrated in FIG. 12, an appropriate q-axis current i _q and an appropriate d-axis current i _d that is 0 or negative are generated by the motor. 50 flows. Therefore, according to this embodiment, while preventing the d-axis current i _d to a heavy load becomes excessively large, it is possible to prevent an increase in iron loss of the motor 50.

<Third Embodiment>
FIG. 13 is a block diagram showing a configuration of an inverter control device 100B according to the third embodiment of the present invention. In the control device 100B in the present embodiment, the inverter control unit 110A and the DC voltage command value calculation unit 143A of the control device 100A (see FIG. 10) according to the second embodiment are replaced with the inverter control unit 110B and the DC voltage command value calculation unit 143B. Has been replaced. In FIG. 13, as in FIG. 10, illustrations corresponding to the current detection unit 101, the DC voltage detection unit 103, and the asynchronous / synchronous determination unit 120 are omitted to prevent the drawing from becoming complicated.

In the inverter control unit 110B, the total magnetic flux calculation unit 113 in the inverter control unit 110A of the second embodiment is replaced with a terminal voltage calculation unit 114. The terminal voltage calculation unit 114, according to the above formula (17), calculates the total magnetic flux [psi _0, the comprehensive magnetic flux [psi ₀ by multiplying the angular speed ω of the motor 50, the motor 50 obtained the overall magnetic flux [psi ₀ pin The voltage v _mt is calculated. Here, the angular velocity ω is calculated from the above formulas (8) and (8) based on the number P of magnetic pole pairs of the rotor of the motor 50 stored in the pole number storage unit 141 and the rotation speed n of the motor 50 detected by the rotation number detection unit 102. 13).

Instead of such a calculation method, the terminal voltage v _mt of the motor 50 may be calculated according to the following method. That is, the terminal voltage v _mt is calculated according to the following equation as shown in the vector diagram of FIG. 5 using the counter electromotive voltage and angular velocity of the motor 50 instead of the armature linkage flux Ψ ₀ of the permanent magnet.

Here, since the counter electromotive voltage v _me is proportional to the rotational speed of the motor 50, the base frequency f _base and the counter electromotive voltage v _emf at the base frequency stored in the counter electromotive voltage storage unit 142, and the rotation speed detection unit 102 Can be calculated based on the current rotational speed n of the motor 50 detected by.

本実施形態においても上記第２実施形態と同様な効果が得られる。 DC voltage command value calculating portion 143B calculates the command value of the DC intermediate voltage e _dc for the terminal voltage v _mt calculated by the terminal voltage calculation unit 114 to the terminal voltage v _mt of motor 50 according to the following equation.

Also in this embodiment, the same effect as the second embodiment can be obtained.

<Fourth embodiment>
FIG. 14 is a block diagram showing a configuration of a motor drive system including a control device 100C according to the fourth embodiment of the present invention. In this figure, elements corresponding to those of the first embodiment (FIG. 1) are denoted by the same reference numerals, and description thereof is omitted.

The control device 100C in the present embodiment includes a current value calculation unit 151 and a reference current storage unit 152. Here, the current value calculation unit 151 is a device that outputs a current value obtained by averaging the magnitudes of the U-phase current, the V-phase current, and the W-phase current detected by the current detection unit 101, for example, the amplitudes of these currents. is there. The reference current storage unit 152 stores a table in which reference current values are defined in association with various torques. Reference current value corresponding to the respective torque in this table, for example, a current value for performing the maximum torque / current control described above, i.e., from d-axis current i _d and the q-axis current i _q that can generate the torque The absolute value of the current vector having the smallest absolute value among the current vectors.

When the current value calculated by the current value calculation unit 151 is equal to or less than the reference current value stored in the reference current storage unit 152, the DC voltage command value calculation unit 143C is similar to the inverter 40 in the first embodiment. Of the DC intermediate voltage e _dc for matching the output voltage V _a of the motor to the counter electromotive voltage vme or the total magnetic flux Ψ ₀ by the output voltage V _a of the inverter 40 is changed to the armature interlinkage magnetic flux Ψ _m by the permanent magnet of the motor 50. The DC intermediate voltage e _dc command value to be matched with is calculated and supplied to the comparison unit 144.

On the other hand, when the current value calculated by the current value calculation unit 151 exceeds the reference current value stored in the reference current storage unit 152, the following control is performed in this embodiment. First, according to FIG. 5, in a situation where the control to match the output voltage V _a of the inverter 40 described in the first embodiment in the counter electromotive voltage v _me is being performed, from the current output voltage V _a of the inverter 40 When even larger, it is understood that the d-axis current i _d is decreased. Further, according to FIG. 8, control is performed so that the total magnetic flux Ψ ₀ by the output voltage V _a of the inverter 40 described in the first embodiment matches the armature linkage magnetic flux Ψ _m by the permanent magnet of the motor 50. In the situation, it can be seen that if the total magnetic flux Ψ ₀ due to the output voltage V _a of the inverter 40 is made larger than the current state, the d-axis current _id is lowered. Therefore, the DC voltage command value calculation unit 143C according to the present embodiment calculates the calculated DC intermediate voltage e when the current value calculated by the current value calculation unit 151 exceeds the reference current value stored in the reference current storage unit 152. increasing the DC link voltage _{e dc} inverter 40 to supply a large instruction value than the command value of the _dc to the comparison unit 144, the output voltage _{V a} of the inverter 40 is larger than the back electromotive voltage _{v me} of the motor 50, or The total magnetic flux Ψ ₀ is made larger than the armature linkage magnetic flux Ψ _m by the permanent magnet to decrease the d-axis current _id, and the output current of the inverter 40 is decreased.

Therefore, according to this embodiment, when the current value calculated by the current value calculation unit 151 is equal to or less than the reference current value during operation in the synchronous PWM mode, the d-axis current _id is 0 or negative. Thus, the control of the DC intermediate voltage e _dc is performed, and an increase in the loss of the motor 50 is prevented. Also, when synchronizing operation in the PWM mode, if the current value calculated by the current value computing section 151 exceeds the reference current value, the control is to reduce the DC link voltage e _dc increase in d-axis current i _d As a result, excessive current can be prevented from flowing through the motor 50 at high loads.

In the present embodiment, the DC voltage command value calculation unit 143C stores the upper limit value of the DC intermediate voltage e _dc of the inverter 40. This upper limit value is determined based on the upper limit of the withstand voltage of the switching elements of the DC-DC converter 20 and the inverter 40 and the capacitor 30. Then, the DC voltage command value calculation unit 143C performs torque control of the motor 50 by setting the command value equal to the upper limit value when the command value of the DC intermediate voltage e _dc exceeds the upper limit value.

Therefore, according to the present embodiment, the DC intermediate voltage e _dc can be prevented from exceeding the upper limit value, and the DC-DC converter 20 and the switching elements of the inverter 40 and the capacitor 30 can be protected.

<Other embodiments>
Although the first to fourth embodiments of the present invention have been described above, other embodiments can be considered in addition to these. For example:

(1) A method for controlling the DC voltage applied to the inverter so that the output voltage of the inverter described in the first embodiment is equal to the counter electromotive voltage of the motor, and a desired current for the motor described in the second embodiment. A method of controlling the DC voltage applied to the inverter so as to flow, and a method of increasing the command value of the DC voltage applied to the inverter from the current state when the current flowing through the motor described in the fourth embodiment exceeds the reference current. You may use together. For example, it may be switched by which method the DC voltage applied to the inverter is controlled based on the rotational speed n of the motor, or may be switched by a torque command. When switching by torque command, for example, in a light load region, the DC voltage applied to the inverter is controlled so that the output voltage of the inverter becomes equal to the counter electromotive voltage of the motor. A method for controlling the DC voltage applied to the power source is conceivable.

(2) In each of the above embodiments, the DC voltage generating means for supplying the DC voltage to the inverter is configured by the DC power supply and the step-up DC-DC converter. However, when the output voltage of the DC power supply is sufficiently high, the voltage is reduced. A type DC-DC converter or a buck-boost converter may be used. Moreover, you may use the AC-DC converter (PWM rectifier) which converts alternating current into direct current as a DC voltage generation means.

(3) In each of the above embodiments, the torque command is given to the inverter control unit. However, the speed command may be given, and the torque command may be obtained from the deviation between the speed command value and the actual speed.

(4) In each of the above embodiments, the three-phase current supplied from the inverter to the motor is detected. However, it is not always necessary to detect all three phases, two phases are detected, and the remaining one phase is obtained by calculation. Also good.

(5) A rotation speed prediction unit may be provided instead of the rotation speed detection unit.

(6) In the first embodiment, by using the above equation (10), a counter electromotive voltage v _me that generated in the motor 50, the DC intermediate voltage e of the fundamental wave component of the output voltage V _a of the inverter 40 is equal The command value of _dc was calculated. However, the command value of the DC intermediate voltage e _dc may be calculated by other methods. In the first embodiment, by using the above equation (10), the armature flux linkage ascribable to the permanent magnet in the total magnetic flux [psi ₀ and motor 50 to produce the motor 50 by the output voltage V _a of the inverter 40 [psi _m The command value of the DC intermediate voltage e _dc that is equal to is calculated. However, the command value of the DC intermediate voltage e _dc may be calculated by other methods.

(7) In each of the above embodiments, the single-pulse synchronous PWM mode is adopted as the synchronous PWM mode. However, when the torque control is performed by controlling the load angle of the output voltage of the inverter, the synchronous PWM mode such as three pulses is used. It may be adopted.

(8) Many control devices of the power conversion device include a processor and a memory that stores a program to be executed by the processor. Therefore, assuming various motors, a program for causing a computer to function as a control device according to the present invention may be created, and this program may be distributed to users of inverter control devices. For example, in the first embodiment (FIG. 1), the realities of the inverter control unit 110, the asynchronous / synchronous determination unit 120, the converter control unit 130, the DC voltage command value calculation unit 143, and the comparison unit 144 are executed by a processor according to a program. Arithmetic processing. Therefore, this program is created assuming various motors 50 and installed in the memory of the control device. At this time, parameters stored in various storage units such as the pole number storage unit 141 may be included in the program itself, or those stored in a nonvolatile memory or the like may be read into the program. The same applies when each embodiment other than the first embodiment is realized as a program.

DESCRIPTION OF SYMBOLS 10 ... DC power supply, 20 ... DC-DC converter, 30 ... Capacitor, 40 ... Inverter, 50 ... Motor, 100, 100A, 100B, 100C ... Control device, 101 ... Current detection part, 102 ... ... rotational speed detector 103 ... DC voltage detector 110, 110A ... inverter controller 120 ... asynchronous / synchronous determination part 130 ... converter controller 141 ... pole number storage part 142 ... Back electromotive voltage storage unit, 143, 143A, 143B, 143C ... DC voltage command calculation unit, 144 ... comparison unit, 111 ... d-axis current q-axis current determination unit, 112 ... inductance storage unit, 113 ... general Magnetic flux calculator, 114... Terminal voltage calculator, 151... Current value calculator, 152.

Claims (6)

In a control device for a power converter having DC voltage generating means for outputting DC voltage, and an inverter for generating AC voltage for driving a motor based on the DC voltage output from the DC voltage generating means,
Back electromotive force storage means for storing the back electromotive voltage of the motor at a specific frequency;
Pole number storage means for storing the number of magnetic pole pairs of the motor;
A means for generating a gate signal for switching ON / OFF of the switching element of the inverter, and a voltage command for instructing an AC voltage waveform to be supplied from the inverter to the motor as the generation mode of the gate signal; Asynchronous PWM mode for generating the gate signal by pulse width modulation using a carrier having a predetermined frequency asynchronous to the voltage command, and by pulse width modulation using the voltage command and a carrier synchronized with the voltage command. An inverter control means having a synchronous PWM mode for generating a gate signal;
When the generation mode of the gate signal of the inverter control means is the synchronous PWM mode, the current supplied from the inverter to the motor corresponds to the direction of the N pole of the permanent magnet provided in the rotor of the motor. Means for calculating a command value for instructing a DC voltage supplied from the DC voltage generating means to the inverter so that a d-axis current as a component becomes 0 or negative, the rotation speed of the motor, and the pole The counter electromotive voltage generated in the motor and the output voltage of the inverter based on the number of magnetic pole pairs stored in the number storage means and the counter electromotive voltage of the motor at a specific frequency stored in the counter electromotive voltage storage means And a direct-current voltage command value calculating means for calculating a direct-current voltage command value that equalizes the fundamental wave component of the control device. In a control device for a power converter having DC voltage generating means for outputting DC voltage, and an inverter for generating AC voltage for driving a motor based on the DC voltage output from the DC voltage generating means,
Back electromotive force storage means for storing the back electromotive voltage of the motor at a specific frequency;
Pole number storage means for storing the number of magnetic pole pairs of the motor;
A means for generating a gate signal for switching ON / OFF of the switching element of the inverter, and a voltage command for instructing an AC voltage waveform to be supplied from the inverter to the motor as the generation mode of the gate signal; Asynchronous PWM mode for generating the gate signal by pulse width modulation using a carrier having a predetermined frequency asynchronous to the voltage command, and by pulse width modulation using the voltage command and a carrier synchronized with the voltage command. An inverter control means having a synchronous PWM mode for generating a gate signal;
When the generation mode of the gate signal of the inverter control means is the synchronous PWM mode, the current supplied from the inverter to the motor corresponds to the direction of the N pole of the permanent magnet provided in the rotor of the motor. Means for calculating a command value for instructing a DC voltage supplied from the DC voltage generating means to the inverter so that a d-axis current as a component becomes 0 or negative, the rotation speed of the motor, and the pole Based on the number of magnetic pole pairs stored in the number storage means and the counter electromotive voltage of the motor at a specific frequency stored in the counter electromotive voltage storage means, and the total magnetic flux generated in the motor by the output voltage of the inverter; DC voltage command value calculation means for calculating a command value of the DC voltage to equalize the armature flux linkage by the permanent magnet in the motor;
The control apparatus of the power converter device characterized by comprising. In a control device for a power converter having DC voltage generating means for outputting DC voltage, and an inverter for generating AC voltage for driving a motor based on the DC voltage output from the DC voltage generating means,
Back electromotive force storage means for storing the back electromotive voltage of the motor at a specific frequency;
Pole number storage means for storing the number of magnetic pole pairs of the motor;
A means for generating a gate signal for switching ON / OFF of the switching element of the inverter, and a voltage command for instructing an AC voltage waveform to be supplied from the inverter to the motor as the generation mode of the gate signal; Asynchronous PWM mode for generating the gate signal by pulse width modulation using a carrier having a predetermined frequency asynchronous to the voltage command, and by pulse width modulation using the voltage command and a carrier synchronized with the voltage command. An inverter control means having a synchronous PWM mode for generating a gate signal;
When the generation mode of the gate signal of the inverter control means is the synchronous PWM mode, the current supplied from the inverter to the motor corresponds to the direction of the N pole of the permanent magnet provided in the rotor of the motor. DC voltage command value calculating means for calculating a command value indicating a DC voltage supplied from the DC voltage generating means to the inverter so that the d-axis current as a component becomes 0 or negative,
The inverter control means includes
D-axis current q-axis current determining means for determining a d-axis current flowing through the motor and a q-axis current that is a component along the q-axis orthogonal to the d-axis based on a torque command;
Inductance storage means for storing d-axis inductance and q-axis inductance of the motor;
The d-axis current and the q-axis current determined by the d-axis current and the q-axis current determining means, the d-axis inductance and the q-axis inductance stored in the inductance storage means, and the specification stored in the back electromotive voltage storage means A total magnetic flux calculating means for calculating a total magnetic flux to be generated by the motor based on an armature linkage magnetic flux of a permanent magnet of the motor determined from a counter electromotive voltage of the motor at a frequency of
The control apparatus for a power converter, wherein the DC voltage command calculation means calculates a command value of the DC voltage for causing the motor to generate a total magnetic flux calculated by the total magnetic flux calculation means. In a control device for a power converter having DC voltage generating means for outputting DC voltage, and an inverter for generating AC voltage for driving a motor based on the DC voltage output from the DC voltage generating means,
Back electromotive force storage means for storing the back electromotive voltage of the motor at a specific frequency;
Pole number storage means for storing the number of magnetic pole pairs of the motor;
A means for generating a gate signal for switching ON / OFF of the switching element of the inverter, and a voltage command for instructing an AC voltage waveform to be supplied from the inverter to the motor as the generation mode of the gate signal; Asynchronous PWM mode for generating the gate signal by pulse width modulation using a carrier having a predetermined frequency asynchronous to the voltage command, and by pulse width modulation using the voltage command and a carrier synchronized with the voltage command. An inverter control means having a synchronous PWM mode for generating a gate signal;
When the generation mode of the gate signal of the inverter control means is the synchronous PWM mode, the current supplied from the inverter to the motor corresponds to the direction of the N pole of the permanent magnet provided in the rotor of the motor. DC voltage command value calculating means for calculating a command value indicating a DC voltage supplied from the DC voltage generating means to the inverter so that the d-axis current as a component becomes 0 or negative,
The inverter control means includes
D-axis current q-axis current determining means for determining a d-axis current flowing through the motor and a q-axis current that is a component along the q-axis orthogonal to the d-axis based on a torque command;
Inductance storage means for storing d-axis inductance and q-axis inductance of the motor;
The d-axis current and the q-axis current determined by the d-axis current and the q-axis current determining means, the d-axis inductance and the q-axis inductance stored in the inductance storage means, and the specification stored in the back electromotive voltage storage means Terminal voltage calculation means for calculating a terminal voltage to be applied to the motor based on the armature linkage magnetic flux of the permanent magnet of the motor determined from the counter electromotive voltage of the motor at a frequency of
The DC voltage command calculating means calculates a command value of the DC voltage for giving the terminal voltage calculated by the terminal voltage calculating means to the motor. The d-axis current q-axis current determining means is a d-axis current and a q-axis current from which a torque corresponding to the torque command is obtained, and a d-axis current having a minimum absolute value of a current vector having both components as a component and The control device for a power converter according to claim 3 or 4, wherein a q-axis current is determined. 6. The DC voltage command value calculating means makes the command value equal to the upper limit value when the calculated DC voltage command value exceeds a predetermined upper limit value. The control apparatus of the power converter device according to claim 1.

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