JP5332305B2 - Control device for permanent magnet type synchronous motor - Google Patents

Description

  The present invention relates to a control device for a permanent magnet type synchronous motor (hereinafter also referred to as PMSM) that does not have a magnetic pole position detector. Specifically, the magnetic pole position and speed of a rotor are determined based on a current detection value of a motor, a terminal voltage detection value, and the like. The present invention relates to a control device that stably controls a permanent magnet type synchronous motor by calculating with high accuracy from the above and controlling current based on these.

  In order to reduce the cost of the PMSM control device, so-called sensorless control, which is operated without using a magnetic pole position detector, has been put into practical use. In sensorless control, stable torque control and speed control are realized by calculating the magnetic pole position and speed of the rotor from information on the terminal voltage and current of the electric motor, and performing current control based on the calculated position.

  For example, in Patent Document 1, an expansion induced voltage generated in a direction orthogonal to the magnetic pole direction of a rotor is calculated, a calculation error of the magnetic pole position is detected from the angle of the expansion induced voltage, and this is used to The magnetic pole position and speed are calculated. This conventional technique is characterized in that the magnetic pole position and speed of an embedded magnet structure permanent magnet type synchronous motor (hereinafter also referred to as IPMSM) having saliency in the rotor can be accurately calculated.

  In Patent Document 2 and Non-Patent Document 1, current and magnetic flux are estimated using a magnetic flux observer, and the magnetic pole position and speed of the synchronous motor are calculated using these. According to these conventional techniques, the magnetic pole position and speed can be calculated with high accuracy even at low speed.

Among these, in the prior art described in Patent Document 2, the magnetic pole position and speed are calculated using a magnetic flux observer derived from the voltage equation in the rotating coordinate system.
In the prior art described in Non-Patent Document 1, the product of the current (q-axis current) in the direction orthogonal to the magnetic pole direction of the rotor is equal to the torque and is generated in the magnetic pole direction (d-axis direction) of the rotor. A new magnetic flux (hereinafter referred to as an expanded magnetic flux) is defined, and the magnetic pole position is calculated from the angle of the expanded magnetic flux. Here, the expanded magnetic flux is estimated using a magnetic flux observer derived from a voltage equation of a fixed coordinate system. On the other hand, the speed is calculated by amplifying the vector product of the current estimation error and the expanded magnetic flux estimated value based on the theory of the model reference adaptive system using the technique described in Non-Patent Document 2. This prior art has a feature that the magnetic pole position and speed of the IPMSM having saliency in the rotor can be accurately calculated even at a low speed.

Japanese Patent No. 3411878 (paragraphs [0132] to [0141], FIG. 8, FIG. 9, etc.) Re-published Patent No. 2002-91558 (page 7, line 49 to page 8, line 21, FIG. 1, FIG. 2, etc.) Hideaki Hatta, Masaru Hasegawa, Kazuki Matsui, “Adaptive Magnetic Flux Observer on Stator Coordinates for IPMSM Sensorless Control”, 2005 Annual Conference of the Institute of Electrical Engineers of Japan, 4th volume, p. 244-245, 2005 Takashi Tsuji, Ritsuko Tomioka, Toru Nakano, Tokai Kim, “Position Sensorless Control of Brushless DC Motor Using Adaptive Observer”, IEEJ Transactions D, Vol. 113, No. 5, p. 579-586, 1993

The amplitude of the expansion induced voltage used in Patent Document 1 is proportional to the speed of the rotor. For this reason, the expansion induced voltage cannot be accurately calculated at a low speed, and the calculation accuracy tends to decrease.
On the other hand, the magnetic flux observer shown in Patent Document 2 is configured based on a state equation when the calculation error of the magnetic pole position is near zero. For this reason, there is a problem that the calculation accuracy decreases when the calculation error of the magnetic pole position becomes large, and this problem becomes apparent particularly when an IPMSM having saliency in the rotor is operated.

In addition, the magnetic flux observer shown in Non-Patent Document 2 can stably operate an IPMSM having a saliency in the rotor. However, since the control calculation is performed in a fixed coordinate system, the current to be estimated The maximum speed of the electric motor is limited due to the fact that the expansion magnetic flux is an alternating current amount.
To summarize the above, each conventional technique has advantages and disadvantages, and there is a problem that a permanent magnet type synchronous motor such as IPMSM cannot be stably operated in a wide speed range.

  Therefore, the problem to be solved by the present invention is that a permanent magnet synchronous motor without a magnetic pole position detector can stably operate the permanent magnet synchronous motor by calculating the magnetic pole position and speed with high accuracy over a wide speed range. An object of the present invention is to provide a control device.

In order to solve the above-described problem, the invention according to claim 1 is directed to a control device for a permanent magnet synchronous motor that does not have a magnetic pole position detector.
Taking the current, voltage and magnetic flux of the motor as vectors,
A magnetic flux observer for estimating an expanded magnetic flux generated in the direction of the magnetic pole of the rotor from the current detection value of the electric motor, the terminal voltage equivalent value and the speed estimation value;
Angle calculating means for calculating the angle of the expanded magnetic flux estimated value estimated by the magnetic flux observer;
Speed estimating means for amplifying the angle of the expanded magnetic flux estimated value and calculating the speed estimated value;
Electrical angle calculation means for amplifying the speed estimated value and calculating the magnetic pole position estimated value.

The invention according to claim 2 is the control device for the permanent magnet type synchronous motor according to claim 1,
The magnetic flux observer is
Means for calculating an armature resistance voltage drop proportional to an estimated current value of the motor;
Means for calculating an armature reaction voltage drop proportional to a product of a vector obtained by rotating the current estimation value by 90 ° and the speed estimation value;
Means for calculating an expansion induced voltage from a product of a vector obtained by rotating the expanded magnetic flux estimated value by 90 ° and the speed estimated value;
Means for subtracting the armature resistance voltage drop, the armature reaction voltage drop and the expansion induced voltage from the terminal voltage equivalent value to calculate a transient voltage;
Means for calculating a current differential value proportional to the transient voltage;
Means for amplifying a deviation between the estimated current value and the detected current value to calculate a current correction value;
Means for amplifying a sum of the current differential value and the current correction value to calculate the current estimated value;
And means for amplifying a deviation between the estimated current value and the detected current value to calculate the expanded magnetic flux estimated value.

  The invention according to claim 3 is for speeding up the response of the speed estimation value. In the control device according to claim 1 or 2, the means for calculating the speed feedforward compensation value from the speed command value; And a means for compensating the speed estimation value by the speed feedforward compensation value.

According to the present invention, in a control device for a permanent magnet type synchronous motor having no magnetic pole position detector, the magnetic pole position and speed can be calculated with high accuracy in a wide speed range based on the expanded magnetic flux estimated by the magnetic flux observer. it can. In particular, in the present invention, since the control calculation in the rotating coordinate system is possible, the maximum speed of the electric motor to be controlled can be increased.
Further, the present invention can accurately calculate the expanded magnetic flux even when the magnetic pole position calculation error (estimation error) is large, and is effective for stable operation of the IPMSM having a saliency in the rotor.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
First, PMSM can realize highly accurate torque control by performing current control according to the d-axis of the rotor (the magnetic pole direction of the rotor) and the q-axis advanced 90 degrees from the d-axis. However, if the magnetic pole position detector is not provided, the d and q axes cannot be directly detected, and therefore the γ and δ axes of the orthogonal rotation coordinate system that rotates at the estimated speed value ω ₁ corresponding to the d and q axes are controlled. The control calculation is performed by estimating to the side. Here, the angles of the d and q axes (magnetic pole positions) are defined as θ _r , and the angles of the γ and δ axes (magnetic pole position estimated values) are defined as θ ₁ .
The definition of the γ and δ axes is shown in FIG. In FIG. 2, ω _r is the rotational angular velocity of the d and q axes, and θ _err is the angular difference (magnetic pole position estimation error) between the d and q axes and the γ and δ axes.

FIG. 1 is a control block diagram of a first embodiment of the present invention corresponding to claim 1. As described above, a permanent magnet type synchronous motor is operated by a power converter 70 using current and voltage on the γ and δ axes. The structure for driving 80 is shown.
First, the main circuit for driving the permanent magnet type synchronous motor 80 having no magnetic pole position detector will be described. 50 is a three-phase AC power source, and the rectifier circuit 60 rectifies the three-phase AC voltage of the power source 50 to generate a DC voltage. Convert to This DC voltage is supplied to a power converter 70 composed of a PWM inverter, and is converted into a predetermined three-phase AC voltage for driving the electric motor 80.

Next, a method for controlling the speed of the permanent magnet synchronous motor 80 using the magnetic pole position estimated value θ ₁ and the speed estimated value ω ₁ will be described together with the configuration of the control device.
The deviation between the speed command value ω ^* and the estimated speed value ω ₁ is calculated by the subtractor 16, and the deviation is amplified by the speed regulator 17 to calculate the torque command value τ ^* . The current command calculator 18 calculates γ and δ-axis current command values i _γ ^* and i _δ ^* that output a desired torque from the torque command value τ ^* .
The current coordinate converter 14 detects the phase current detection values i _u and i _w detected by the u-phase current detector 11u and the w-phase current detector 11w, respectively, and detects the γ and δ-axis currents using the magnetic pole position estimated value θ _1. Coordinates are converted to values i _γ and i _δ .

The subtractor 19a calculates the deviation between the γ-axis current command value i _γ ^* and the detected γ-axis current value i _γ, and the difference is amplified by the γ-axis current regulator 20a to calculate the γ-axis voltage command value v _γ ^* . To do. On the other hand, the subtractor 19b calculates a deviation between the δ-axis current command value i _δ ^* and the δ-axis current detection value i _δ, and the δ-axis current regulator 20b amplifies the deviation to obtain the δ-axis voltage command value v _δ ^*. Is calculated.
These γ and δ-axis voltage command values v _γ ^* and v _δ ^* are converted into phase voltage command values v _u ^* , v _v ^* and v _w ^* by the voltage coordinate converter 15.

The PWM circuit 13 generates a gate signal from the phase voltage command values v _u ^* , v _v ^* , v _w ^* and the DC voltage detection value E _dc detected by the voltage detector 12. The power converter 70 controls the internal semiconductor switching element based on the gate signal, thereby controlling the terminal voltage of the permanent magnet type synchronous motor 80 to the phase voltage command values v _u ^* , v _v ^* , v _w ^* . To do.

Next, the calculation principle of the speed and magnetic pole position using the expanded magnetic flux will be described.
First, the amplitude Ψ _{ex of the} expanded magnetic flux is defined by Equation 1.

When the direction of the expanded magnetic flux Ψ _exγδ is defined as the magnetic pole direction (d-axis direction) of the rotor, the relationship of Mathematical Formula 2 is established between the γ and δ-axis expanded magnetic fluxes Ψ _exγ and Ψ _exδ and the angle difference θ _err .

  From Equations 1 and 2, the γ and δ-axis voltage equations of the permanent magnet type synchronous motor 80 can be derived as Equation 3 using the extended magnetic flux.

In Equation 3, the first term on the right side is the voltage drop due to the armature resistance, the second term on the right side is a transient voltage that is parallel to the current differential value and proportional to the d-axis inductance, and the third term on the right side is the voltage drop due to the armature reaction. Since the matrix J is a 90-degree rotation coordinate transformation, the voltage drop due to the armature reaction is equal to the product of the vector obtained by advancing the current by 90 degrees, the q-axis inductance, and the angular velocity ω ₁ of the γ and δ axes. The fourth term on the right side of Equation 3 is the expansion induced voltage, which is equal to the product of the vector obtained by advancing the expansion magnetic flux by 90 degrees and the angular velocity ω _r of the d and q axes.

FIG. 3 is a vector diagram showing the relationship between the γ- and δ-axis voltage equations of Equation 3. The vector diagram of FIG. 3 shows a case where the current differential value is zero and the speed estimated value ω ₁ matches the actual speed value ω _r .
Since the expanded magnetic flux Ψ _exγδ is generated in the d-axis direction, the angle difference θ _err can be detected from the angle δ _Ψex of the expanded magnetic flux Ψ _exγδ observed on the γ and δ axes from FIG. Here, the expanded magnetic flux Ψ _exγδ is _obtained from γ, δ-axis voltage command values v _γ ^* , v _δ ^* , γ, δ-axis current detection values i _γ , i _δ, and the estimated speed value ω ₁ as shown in FIG. It is possible to calculate using the vector relationship.

Next, a magnetic flux observer for estimating the γ and δ axis expanded magnetic flux will be described. The configuration and operation of the magnetic flux observer correspond to the invention according to claim 2.
From Equation 3, the state equations of γ and δ-axis currents i _γ and i _δ including the expanded magnetic flux are Equation 4 and Equation 5, respectively.

In Equations 4 and 5, d and q-axis current differential values pi _d and pi _q are approximated to zero, and the estimated speed value ω ₁ is approximated to the actual velocity value ω _r , and γ, By approximating that the δ-axis terminal voltages v _γ and v _δ can be controlled to γ and δ-axis voltage command values v _γ ^* and v _δ ^* , respectively, the magnetic flux observer is configured by Equations 6 and 7.

From the relational expression of Expression 3, the first line on the right side of Expression 6 physically corresponds to the γ and δ axis current differential values, and among these, the calculation result in parentheses corresponds to the transient voltage. That is, the first line on the right side of Equation 6 specifically includes an armature resistance voltage drop, an armature reaction voltage drop, and an armature reaction voltage drop from a γ and δ axis voltage command value (here, equal to γ and δ axis voltage detection values). extended electromotive force by subtracting the calculated transient voltage, the transient voltage obtained by multiplying the inverse of d-axis inductance L _d gamma, and calculates the δ-axis current differential value.
The second line on the right side of Equation 6 corresponds to a current correction value calculated by amplifying the deviation between the current estimated value i _γδest and the detected current value i _γδ, and γ on the first line on the right side of Equation 6 The current estimated value i _γδest is calculated based on the addition result of the δ-axis current differential value and the current correction value, and the deviation between the current estimated value i _γδest and the detected current value i _γδ is amplified by Equation 7. This indicates that the expanded magnetic flux estimated value Ψ _exγδest is calculated.

The above-described prior art related to Patent Document 2 also estimates the rotor magnetic flux using a magnetic flux observer and estimates the speed based on this magnetic flux. However, the major difference between this embodiment and the prior art according to Patent Document 2 is that in this embodiment, as shown in Equation 6, the armature reaction voltage drop is advanced by 90 degrees in the current estimated value i _γδest. Further, the point calculated from the product of the vector, the q-axis inductance L _d and the speed estimated value ω ₁ , and the γ and δ-axis current differential values are calculated from the product of the transient voltage calculated value and the inverse of the d-axis inductance L _d. Is a point.
The prior art according to Patent Document 2 is based on the condition that the magnetic pole position calculation error is near zero. However, in the magnetic flux observer according to the present embodiment shown in Equations 6 and 7, the magnetic pole in the process of deriving the expanded magnetic flux Since no approximation is performed on the position estimation error θ _err , the magnetic flux position estimation error θ _err can be accurately calculated even when the magnetic pole position estimation error θ _err is large.

Next, calculation of the speed estimated value ω ₁ and the magnetic pole position estimated value θ ₁ using the magnetic flux observer will be described with reference to the block diagram of FIG.
In FIG. 1, the magnetic flux observer 31 calculates γ and δ-axis expanded magnetic flux estimated values ψ _exγest and ψ _exδest using the magnetic flux observer shown in Equations 6 and 7. Angle calculator 32, gamma, [delta] axis extended flux estimate _Ψ exγest, the angle [delta] _Pusaiexest of [psi _Exderutaest be calculated by Equation 8.

The speed estimator 33 calculates the speed estimated value ω ₁ by amplifying the angle δ _Ψexest of the expanded magnetic flux estimated value. The speed estimator 33 is composed of a PI controller, and obtains a speed estimated value ω ₁ by Equation 9.

Electrical angle calculator 34 amplifies the speed estimation value omega ₁ calculates the magnetic pole position estimation value theta _1. The electrical angle calculator 34 is composed of an integrator, and the magnetic pole position estimated value θ ₁ is obtained by Equation 10.

As shown in FIG. 3, since the position estimation error θ _err can be detected by the expanded magnetic flux angle δ _Ψex observed from the γ and δ axes, the PLL circuit that controls the position estimation error θ _err to zero using Expression 7 and Expression 8 Thus, the speed estimated value ω ₁ and the magnetic pole position estimated value θ ₁ can be accurately calculated.

Next, a second embodiment of the present invention corresponding to claim 3 will be described based on the block diagram of FIG.
In this embodiment, the response of the speed estimated value is improved, and the block diagram of FIG. 4 corresponds to an improved method of calculating the speed estimated value ω ₁ in FIG. Below, it demonstrates centering on the difference with FIG. 1 in FIG.

The speed feedforward compensator 41 calculates a speed feedforward compensation value ω _1FF from the first order lag filter output of the speed command value ω ^* .
The speed estimator 33 calculates the speed correction value ω _1comp by amplifying the angle δ _Ψexest of the expanded magnetic flux estimated value. Here, the speed estimator 33 is composed of a PI controller, and calculates a speed correction value ω _{1comp according} to Equation 11.

The adder 42 adds the speed feedforward compensation value ω _1FF and the speed correction value ω _1comp to calculate the speed estimated value ω ₁ , and this speed estimated value ω ₁ is used to calculate the magnetic pole position estimated value θ _1. .
According to this embodiment, by performing speed feedforward compensation, the response of the speed estimated value ω ₁ can be made faster, and the transient characteristics can be improved.

1 is a block diagram showing a first embodiment of the present invention. It is a figure which shows the definition of d, q axis | shaft and (gamma), (delta) axis. It is a vector diagram which shows the relationship between the electric current of a permanent magnet type synchronous motor, a terminal voltage, and an extended magnetic flux. It is a block diagram which shows 2nd Embodiment of this invention.

Explanation of symbols

11u u-phase current detector 11w w-phase current detector 12 voltage detector 13 PWM circuit 14 current coordinate converter 15 voltage coordinate converter 16 subtractor 17 speed controller 18 current command calculator 19a subtractor 19b subtractor 20a γ-axis Current regulator 20b δ-axis current regulator 31 Magnetic flux observer 32 Angle calculator 33 Speed estimator 34 Electrical angle calculator 41 Speed feedforward compensator 42 Adder 50 Three-phase AC power supply 60 Rectifier circuit 70 Power converter 80 Permanent magnet type Synchronous motor

Claims (3)

In a control device for a permanent magnet type synchronous motor having no magnetic pole position detector,
Taking the current, voltage and magnetic flux of the motor as vectors,
A magnetic flux observer for estimating an expanded magnetic flux generated in the direction of the magnetic pole of the rotor from the current detection value of the electric motor, the terminal voltage equivalent value and the speed estimation value;
Angle calculating means for calculating the angle of the expanded magnetic flux estimated value estimated by the magnetic flux observer;
Speed estimating means for amplifying the angle of the expanded magnetic flux estimated value and calculating the speed estimated value;
An electrical angle calculating means for amplifying the speed estimated value to calculate a magnetic pole position estimated value;
A control device for a permanent magnet type synchronous motor, comprising: In the control device for the permanent magnet type synchronous motor according to claim 1,
The magnetic flux observer is
Means for calculating an armature resistance voltage drop proportional to an estimated current value of the motor;
Means for calculating an armature reaction voltage drop proportional to a product of a vector obtained by rotating the current estimation value by 90 ° and the speed estimation value;
Means for calculating an expansion induced voltage from a product of a vector obtained by rotating the expanded magnetic flux estimated value by 90 ° and the speed estimated value;
Means for subtracting the armature resistance voltage drop, the armature reaction voltage drop and the expansion induced voltage from the terminal voltage equivalent value to calculate a transient voltage;
Means for calculating a current differential value proportional to the transient voltage;
Means for amplifying a deviation between the estimated current value and the detected current value to calculate a current correction value;
Means for amplifying a sum of the current differential value and the current correction value to calculate the current estimated value;
Means for amplifying a deviation between the current estimated value and the current detected value to calculate the expanded magnetic flux estimated value;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to claim 1 or 2,
Means for calculating a speed feedforward compensation value from the speed command value;
Means for compensating the speed estimate by the speed feedforward compensation value;
A control device for a permanent magnet type synchronous motor.

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