JP6573213B2 - Control device for permanent magnet type synchronous motor - Google Patents

Description

  The present invention relates to a control device for a permanent magnet type synchronous motor having no magnetic pole position detector, and more particularly to a technique for improving the stability of control.

  In order to reduce the cost of a control device for a permanent magnet type synchronous motor, so-called sensorless vector control that operates without using a magnetic pole position detector for detecting the magnetic pole position of a rotor has been put into practical use. In the sensorless vector control, the magnetic pole position and speed of the rotor are calculated from information on the terminal voltage and current of the electric motor, and current control is performed based on these to realize torque control and speed control.

For example, in Non-Patent Document 1, in position sensorless vector control of a synchronous motor having an embedded magnet structure, an expansion induced voltage generated in a direction orthogonal to the magnetic pole direction of the rotor is calculated, and the magnetic pole is calculated from the expansion induced voltage calculation value. It is described that a position estimation error is detected and the magnetic pole position and velocity are calculated using this magnetic pole position estimation error.
However, the sensorless vector control described in Non-Patent Document 1 has a problem in stability at low speed. For this reason, in the low speed range, as described in, for example, Japanese Patent Application Laid-Open No. 2001-190093, the current angular velocity is controlled to a speed command value, and the rotor of the permanent magnet type synchronous motor is operated. A technique of operating by drawing in an electric current may be applied. Hereinafter, such an operation method is referred to as “current drawing control”.

As described above, stable control is possible if the control method is selected according to the speed of the motor. However, if a shock occurs during switching of the control method, smooth acceleration / deceleration operation becomes difficult. It is desirable to switch with less.
From such a viewpoint, for example, in the control device described in Japanese Patent No. 5277724, the expansion induced voltage is calculated from the current of the motor, the terminal voltage, and the angular velocity of the current command value when switching from the current pull-in control to the sensorless vector control. Then, the magnetic pole position estimation error is calculated from the angle of the expansion induced voltage to initialize the magnetic pole position estimation value, the speed estimation value is initialized using the angular velocity of the current command value, and the magnetic pole position estimation error is used. These command values are initialized so that the terminal voltage and current of the motor do not change suddenly, and the torque command value is initialized using the torque calculated from the motor current, the expansion induced voltage and the angular velocity of the current command value. The switch is possible.

Koji Tanaka and Ichiro Miki, “Position Sensorless Control of Embedded Magnet Synchronous Motor Using Extended Inductive Voltage”, IEEJ Transactions D, Vol.125, No.9, p.833-p.838, 2005

In order to increase the maximum torque of the motor in the current pull-in control, in principle, it is necessary to perform control so as to increase the d-axis current that is a current parallel to the magnetic pole direction of the rotor.
On the other hand, in order to stably perform the sensorless vector control according to Non-Patent Document 1, the following extended induced voltage E _ex described as the equation (5) in this document needs to be sufficiently large.
E _ex = (L _d −L _q ) (ω i _d −pi _q ) + ωK _E
In this equation, L _d is a d-axis inductance, L _q is a q-axis inductance, ω is a rotor angular velocity, i _d is a d-axis current, i _q is a q-axis current, p is a differential operator, and _KE is an induced voltage. It is a constant.
In a synchronous motor having an embedded magnet structure, the q-axis inductance L _q is larger than the d-axis inductance L _d, and according to the above formula, when the d-axis current _id is large in the positive direction, the expansion induced voltage E _ex becomes small, and as a result, the stability of the sensorless vector control is lowered.

Therefore, in the technique described in Non-Patent Document 1, there is a problem in stability immediately after switching from current pull-in control to sensorless vector control, and it is required to solve this.
Therefore, a problem to be solved by the present invention is to provide a control device for a permanent magnet type synchronous motor capable of stably realizing sensorless vector control even when the d-axis current is large.

In order to solve the above-mentioned problem, the invention according to claim 1 estimates the rotor speed and magnetic pole position of a permanent magnet synchronous motor by calculation without using a magnetic pole position detector, and based on these estimated values, In a control device that controls the torque and speed of the motor by controlling the armature current of the motor,
Taking the current and terminal voltage of the motor as vectors,
Position estimation error calculation means for calculating the magnetic pole position estimation error of the motor, speed estimation means for estimating the speed of the electric motor from the magnetic pole position estimation error, and integration means for estimating the magnetic pole position by integrating a speed estimated value And having
The position estimation error calculation means includes
Expansion induced voltage calculation means for calculating a first expansion induced voltage using the current of the motor, the terminal voltage equivalent value, and the estimated speed value; a d-axis current that is a current parallel to the magnetic pole of the rotor; An extension induced voltage compensation means for calculating a second extension induced voltage using the estimated speed value and the first extension induced voltage, and an angle for calculating the magnetic pole position estimation error using the second extension induced voltage. Difference calculating means,
The extended induced voltage compensation means includes
When the d-axis current is equal to or smaller than the first threshold, the second expansion induced voltage value is calculated to be equal to the first expansion induced voltage calculation value, and the d-axis current is calculated from the first threshold value. The second expansion induced voltage is calculated so as to be equal to the induced voltage of the electric motor when it is equal to or larger than the larger second threshold value.
Thereby, even when the d-axis current is large, sensorless vector control can be stably realized.

  According to a second aspect of the present invention, in the control device for the permanent magnet type synchronous motor according to the first aspect, the position estimation error calculating means becomes zero when the d-axis current is equal to or less than the first threshold, means for calculating a weighting factor that becomes 1 when the d-axis current is equal to or greater than the second threshold, and the weighting factor, the speed estimation value, the d-axis current, and the first expansion induced voltage calculation value And calculating the second expansion induced voltage calculation value.

  According to a third aspect of the present invention, in the control device for a permanent magnet synchronous motor according to the second aspect, the speed estimation means includes a proportional regulator, and an upper limit value of the proportional gain in the proportional regulator is set to the weight coefficient. The speed estimation means as a function includes a proportional regulator, and an upper limit value of the proportional gain in the proportional regulator is a function of the weight coefficient.

  According to the present invention, it is possible to improve stability when shifting to sensorless vector control while a large d-axis current is flowing, such as when switching from current pull-in control to sensorless vector control.

It is a control block diagram which shows embodiment of this invention. It is a vector diagram showing the definition of d, q axis and γ, δ axis. It is explanatory drawing of the compensation function for calculating the weighting coefficient in embodiment of this invention.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a control block diagram showing an embodiment of the present invention.
A permanent magnet type synchronous motor (hereinafter also referred to as PMSM) can realize high-performance torque control and speed control by performing control on the d and q axes constituting orthogonal rotation coordinates synchronized with the rotor. it can. Here, for the d and q axes, the direction parallel to the N pole of the rotor is defined as the d axis, and the direction advanced by 90 ° from the d axis is defined as the q axis. However, in the case of sensorless control in which the PMSM is operated without using the magnetic pole position detector, the positions of the d and q axes cannot be directly detected. Are controlled internally, and current and voltage control calculations are performed on the γ and δ axes.

FIG. 2 is a vector diagram showing the definition of the d, q axis and the γ, δ axis. As shown in FIG. 2, the angular velocities of the d and q axes are defined as ω _r (rotor speed), and the angular velocities (speed estimated values) of the γ and δ axes are defined as ω ₁ .
Also, the angle difference (position estimation) between the angle of the γ-axis (magnetic pole position estimated value) θ ₁ with respect to the u-phase winding of PMSM and the angle of the d-axis (magnetic pole position) θ _r with reference to the u-phase winding. Error) θ _err is defined by Equation 1.

Next, the configuration and operation of the control device will be described based on the control block diagram of FIG.
First, parts relating to speed control, current control, and voltage control of the PMSM 80 will be described.
In FIG. 1, a subtractor 16 obtains a deviation between the speed command value ω _r ^* and the speed estimated value ω _1, and a speed regulator 17 calculates a torque command value τ ^* so that the deviation becomes zero. The current command calculator 18 calculates a d-axis current command value i _d ^* and a q-axis current command value i _q ^* for generating a desired torque based on the torque command value τ ^*. The current command value i _γ ^* and the δ-axis current command value i _δ ^* are used for control.

On the other hand, the u-phase current detection value i _u and the w-phase current detection value i _w detected by the u-phase current detector 11 _u and the w-phase current detector 11 _w are input to the coordinate converter 14, and the magnetic pole position estimated value θ ₁ is obtained. The coordinate is converted into a detected γ-axis current value i _γ and a detected δ-axis current value i _δ .
The subtractor 19a obtains the deviation between the γ-axis current command value i _γ ^* and the detected γ-axis current value i _γ, and the γ-axis current regulator 20a calculates the γ-axis voltage command value v _γ ^* so that this deviation becomes zero. Generate. Further, the subtractor 19b obtains a deviation between the δ-axis current command value i _δ ^* and the δ-axis current detection value i _δ, and the δ-axis current regulator 20b causes the δ-axis voltage command value v _δ to be zero. ^{* Is} generated.

The coordinate converter 15 converts the γ-axis voltage command value v _γ ^* and the δ-axis voltage command value v _δ ^* into the phase voltage command values v _u ^* , v _v ^* , and v _w ^* using the magnetic pole position estimated value θ _1. Convert coordinates. The PWM circuit 13 generates a gate signal to be supplied to the semiconductor switching element of the power converter 70 such as an inverter based on the phase voltage command values v _u ^* , v _v ^* , v _w ^* .

The rectifier circuit 60 rectifies the three-phase AC voltage of the three-phase AC power supply 50 to convert it into a DC voltage, and supplies the DC voltage to the power converter 70.
The power converter 70 controls the semiconductor switching element based on the gate signal output from the PWM circuit 13 and controls the terminal voltage of the PMSM 80 to the phase voltage command values v _u ^* , v _v ^* , v _w ^* .
With the above control, the speed of the PMSM 80 can be controlled to the speed command value ω _r ^* .

Next, the PMSM 80 speed and magnetic pole position estimation technique will be described.
First, the expansion induced voltage calculator 31 calculates the first γ-axis and δ-axis expansion induced voltages e _exγest and e _{exδest according} to Equation 2.

In Equation 2, the γ-axis voltage command value v _γ ^* and the δ-axis voltage command value v _δ ^* are used as the γ-axis voltage v _γ and the δ-axis voltage v _δ, which are the terminal voltage equivalent values of the PMSM 80, respectively.
Here, Equation 2 measures the phase voltage or line voltage of the PMSM 80 using a voltage detection circuit (not shown), and calculates the γ-axis voltage v _γ calculated from these measured values and the estimated magnetic pole position θ ₁ . The calculation may be performed using the δ-axis voltage v _δ . Also, gamma-axis current i _gamma, gamma-axis current command value instead of the [delta] -axis current i _δ i _{γ ^*,} may be used [delta] -axis current value i _[delta] ^*, a speed command instead of the estimated speed value omega ₁ The value ω _r ^* may be used.

The weighting factor calculator 41 calculates the weighting factor K _EexEmf according to the compensation function shown in FIG. This weight coefficient K _EexEmf is _zero when the d-axis current command value i _d ^* is in the range of 0 to the first threshold value i _dthEex , and is 1 when i _d ^* is in the range greater than the second threshold value i _dthEmf. , I _d ^* is a function that increases in proportion to i _d ^* when it is in the range of i _{dthEex to} i _dthEmf .
Note that the calculation of the weight coefficient K _EexEmf may be performed using the γ-axis current i _γ instead of the d-axis current command value i _d ^* .

The extended induced voltage compensator 42 in FIG. 1 calculates the second γ-axis and δ-axis extended induced voltages e _exω1γ and e _{exω1δ according} to Equation 3.

Second extended electromotive force calculation value _{e exω1γ, e exω1δ} is extended induced voltage calculation value _{e Exganmaest} when the weighting factor _{K EexEmf} is zero, respectively equal to _{e Exderutaest,} the induced voltage when the weighting factor _{K EexEmf} is 1 It becomes equal to the calculated values e _mfγest and e _mfδest (the reason why it becomes equal to e _mfγest and e _mfδest when the weight coefficient K _EexEmf is 1 will be described later).
The calculation of Equation 3 may be performed using the d-axis current command value i _d ^* instead of the γ-axis current i _γ .

Next, a method of calculating the speed and the magnetic pole position using the second extended induced voltage calculation values e _exω1γ and e _exω1δ will be described.
The angle difference calculator 32 _obtains a position estimation error calculation value θ _errest from the second extended induced voltage calculation values e _exω1γ and e _exω1δ using Equation 4.

Further, the proportional / integral controller constituting the speed estimator 33 obtains a speed estimated value ω ₁ by performing proportional / integral calculations on the position estimation error calculation value θ _{errest according} to Equation 5.

The integrator 34 integrates the speed estimated value ω ₁ to calculate the magnetic pole position estimated value θ ₁ .
By these calculations, the speed estimated value ω ₁ and the magnetic pole position estimated value θ ₁ are calculated so that the position estimation error θ _err becomes zero, and these values can be converged to true values.
In the above configuration, the extended induced voltage calculator 31, the weighting coefficient calculator 41, the extended induced voltage compensator 42, and the angle difference calculator 32 constitute position estimation error calculation means in the claims.

Here, in the calculation of the speed estimation value ω ₁ according to Equation 5, the upper limit value of the proportional gain K _Pω1 in the proportional regulator of the speed estimator and the integral regulator of the speed estimator are _derived from the stability of the speed / position estimation system. The upper limit value of the reciprocal of the integration time constant T _Iω1 at is limited by Equation 6.

As mentioned earlier, the second extended electromotive force calculation value _{e exω1γ, e exω1δ is, i} ^{d *} is small, expand when the weighting factor _{K EexEmf} are zero induced voltage calculation value _{e Exganmaest,} respectively _{e Exderutaest} When i _d ^* is large and the weighting coefficient K _EexEmf is 1, they are equal to the induced voltage calculation values e _mfγest and e _mfδest .
Therefore, the position estimation error calculated value θ _errest is equal to the angle of the extended induced voltage calculated values e _exγest and e _exδest when the weight coefficient K _EexEmf is zero, and the induced voltage when the weight coefficient K _EexEmf is 1. It becomes equal to the angle of the calculated values e _mfγest and e _mfδest . As a result, the responsiveness of the speed / position estimation system _{varies depending on} the value of the weight coefficient K _EexEmf .

Therefore, as shown in Equation 6, the coefficient ω _dθerrKP is _{used as} a function of the weighting coefficient K _EexEmf, and the upper limit of the speed estimator proportional gain K _Pω1 is calculated according to the weighting coefficient K _EexEmf to stabilize the system. .
The calculation of Equation 6 may be performed using γ, δ-axis currents i _δ , i _γ instead of d, q-axis current command values i _d ^* , i _q ^* .

Next, the _principle that the second extended induced voltage calculation values e _exω1γ and e _exω1δ can be made equal to the induced voltage calculation values e _mfγest and e _mfδest by the above-described Equation 3 by _setting the weight coefficient K _EexEmf to 1 will be described.
The voltage equation on the γ and δ axes of the PMSM 80 can be approximated as Equation 7 using the induced voltage E _mf when the position estimation error θ _err is small.

As a result, the γ and δ-axis induced voltage calculation values e _mfγest and e _mfδest can be obtained as in Expression 8.

ここで、数式９の右辺の演算は、数式３において重み係数Ｋ_{ＥｅｘＥｍｆ}を１としたときの演算と同じである。 From Equations 2 and 8, when the current differential term is approximated to zero, the γ and δ-axis induced voltage calculation values _emfγest and _emfδest are obtained using the first γ and δ-axis expansion induced voltage calculation values e _exγest and e _exδest. It can be obtained by Equation 9.

Here, the calculation on the right side of Expression 9 is the same as the calculation when the weighting coefficient K _EexEmf is _set to 1 in Expression 3.

From the above, the weighting coefficient calculator 41 controls the weighting coefficient K _EexEmf from zero to 1 according to the magnitude of the d-axis current command value i _d ^* , so that the second γ, δ axis extended electromotive force calculation value _{e exω1γ, e exω1δ} the first gamma, [delta] axis extended electromotive force calculation value _{e exγest,} from _{e exδest} γ, δ-axis induced voltage calculation value _{e mfγest,} clear that can migrate smoothly to the _{e Mfderutaest} It is.

Then, even if the d-axis current _{i d} is large, gamma, [delta] axis induced voltage calculation value _{e Mfganmaest,} can calculates the position estimation error theta _err from _{e Mfderutaest,} the sensorless vector control will be described can be realized stably.
From Equations 7 and 8, there is a relationship of Equation 10 between the γ and δ-axis induced voltage calculation values e _mfγest and e _mfδest and the position estimation error θ _err .

From Equation 10, the position estimation error θ _err can be calculated from the angles of the γ and δ-axis induced voltage calculation values e _mfγest and e _mfδest . Also, from Equation 7, the induced voltage _{E mf} is not dependent on the d-axis current _{i d.} Therefore, even when the d-axis current _{i d} is large, gamma, [delta] axis induced voltage calculation value _{e Mfganmaest,} is capable calculating the position estimation error theta _err from _{e Mfderutaest,} the sensorless vector control can be realized stably it can.

  As described above, according to the present embodiment, when the d-axis current is small, the speed estimated value and the position estimated value are calculated using the angle of the extended induced voltage, and when the d-axis current is large, the angle of the induced voltage is calculated. By calculating the estimated speed value and the estimated position value using, sensorless vector control can be stably realized.

11u: u-phase current detector 11w: w-phase current detector 13: PWM circuit 14, 15: coordinate converters 16, 19a, 19b: subtractor 17: speed controller 18: current command calculator 20a: γ-axis current controller 20b: δ-axis current regulator 31: extended induced voltage calculator 32: angle difference calculator 33: speed estimator 34: integrator 41: weighting factor calculator 42: extended induced voltage compensator 50: three-phase AC power supply 60 : Rectifier circuit 70: Power converter 80: Permanent magnet synchronous motor (PMSM)

Claims (3)

The rotor speed and magnetic pole position of the permanent magnet type synchronous motor are estimated by calculation without using the magnetic pole position detector, and the torque of the motor is controlled by controlling the armature current of the motor based on these estimated values. In the control device that controls the speed,
Taking the current and terminal voltage of the motor as vectors,
Position estimation error calculation means for calculating the magnetic pole position estimation error of the motor, speed estimation means for estimating the speed of the electric motor from the magnetic pole position estimation error, and integration means for estimating the magnetic pole position by integrating a speed estimated value And having
The position estimation error calculation means includes
Expansion induced voltage calculation means for calculating a first expansion induced voltage using the current of the motor, the terminal voltage equivalent value, and the estimated speed value; a d-axis current that is a current parallel to the magnetic pole of the rotor; An extension induced voltage compensation means for calculating a second extension induced voltage using the estimated speed value and the first extension induced voltage, and an angle for calculating the magnetic pole position estimation error using the second extension induced voltage. Difference calculating means,
The extended induced voltage compensation means includes
When the d-axis current is equal to or smaller than the first threshold, the second expansion induced voltage value is calculated to be equal to the first expansion induced voltage calculation value, and the d-axis current is calculated from the first threshold value. A control device for a permanent magnet synchronous motor, wherein the second expansion induced voltage is calculated so as to be equal to the induced voltage of the electric motor when it is equal to or larger than a larger second threshold value. In the control device for the permanent magnet type synchronous motor according to claim 1,
The position estimation error calculation means includes
Means for calculating a weighting factor that becomes zero when the d-axis current is less than or equal to the first threshold and becomes 1 when the d-axis current is greater than or equal to the second threshold; A control apparatus for a permanent magnet type synchronous motor, wherein the second expansion induced voltage calculation value is calculated using a value, the d-axis current, and the first expansion induced voltage calculation value. In the control device for the permanent magnet type synchronous motor according to claim 2,
The speed estimator includes a proportional regulator, and an upper limit value of a proportional gain in the proportional regulator is a function of the weighting coefficient.

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