JP5200927B2 - Analog circuits and electronic equipment - Google Patents

Description

  The present invention relates to an analog circuit and an electronic device.

  The noise of the transistors constituting the analog circuit includes thermal noise and flicker noise. Of these, thermal noise is dominant noise in a high frequency band, and is proportional to absolute temperature. On the other hand, flicker noise is dominant noise in a low frequency region, and the noise level increases as the signal frequency decreases.

  On the other hand, in an analog circuit such as a radio clock receiver or a gyro sensor detector, frequency conversion is performed by a mixer or the like. Specifically, for example, conversion from the frequency of the carrier signal to the frequency of the desired signal is performed. For this reason, a high signal frequency and a low signal frequency are mixed as signal frequencies to be subjected to small amplitude amplification of the operational amplifier constituting the analog circuit.

However, in analog circuits such as radio clock receivers and gyro sensor detectors so far, optimal sizing design of operational amplifiers in consideration of such difference in signal frequency has not been performed. Moreover, no consideration has been given to the compatibility between low noise and low power consumption.
Japanese Patent Laid-Open No. 4-214054

  The present invention has been made in view of the above technical problems, and an object of the present invention is to provide an analog circuit and an electronic apparatus that can achieve both low noise and low power consumption.

  The present invention includes a first circuit having a first-type operational amplifier in which a frequency of a signal to be amplified is a first frequency, and a second frequency in which the frequency of the signal to be amplified is lower than the first frequency. A second circuit having a two-type operational amplifier, a channel width of a differential stage transistor of a differential portion of the first-type operational amplifier is W1a, a channel length is L1a, and the difference between the first-type operational amplifiers The bias current flowing in the moving part is Ia, the channel width of the differential stage transistor in the differential part of the second-type operational amplifier is W1b, the channel length is L1b, and flows in the differential part of the second-type operational amplifier. When the bias current is Ib, it relates to an analog circuit in which W1b × L1b> W1a × L1a and Ia> Ib.

  According to the present invention, the first circuit of the analog circuit includes a first-type operational amplifier, and the second circuit includes a second-type operational amplifier. Between the WL product W1a × L1a of the differential stage transistor of the first type operational amplifier and the WL product W1b × L1b of the differential stage transistor of the second type operational amplifier, there is a relationship of W1b × L1b> W1a × L1a. It holds. Therefore, since the WL product W1b × L1b of the second-type operational amplifier can be increased, the flicker noise of the second-type operational amplifier can be reduced, and the SNR (Signal-to-Nose Ratio) of the analog circuit can be improved. On the other hand, since the WL product W1a × L1a of the first-type operational amplifier can be reduced, a situation in which the circuit area of the first-type operational amplifier becomes unnecessarily large can be prevented, and the analog circuit can be reduced in size.

  Further, a relationship of Ia> Ib is established between the bias current Ia of the differential section of the first-type operational amplifier and the bias current Ib of the differential section of the second-type operational amplifier. Therefore, since the bias current Ia of the first-type operational amplifier can be increased, the thermal noise of the first-type operational amplifier can be reduced and the SNR of the analog circuit can be improved. On the other hand, since the bias current Ib of the second-type operational amplifier can be reduced, it is possible to prevent the consumption current of the second-type operational amplifier from becoming unnecessarily large and to reduce the power consumption of the analog circuit.

  According to the present invention, when the first frequency is f1, the second frequency is f2, and the corner frequency of flicker noise and thermal noise in the frequency-noise characteristic is fcr, the first-type operational amplifier F1-fcr <fcr-f2, and in the second-type operational amplifier, fcr-f2 <f1-fcr may be satisfied.

  Thus, if the relationship of f1−fcr <fcr−f2 is established, the corner frequency fcr can be brought closer to the frequency f1 side, and both noise reduction and power consumption reduction of the operational amplifier can be achieved. If the relationship of fcr−f2 <f1−fcr is established, the corner frequency fcr can be made closer to the frequency f2 side, and both the noise reduction and the area reduction of the operational amplifier can be achieved.

  The present invention further includes a mixer that mixes a signal having a given frequency with a signal including a desired signal, wherein the first circuit is a circuit provided on the front side of the mixer, and the second circuit May be a circuit provided on the subsequent stage side of the mixer.

  When such a mixer is provided, the signal frequency in the first circuit on the front stage side of the mixer is different from the signal frequency in the second circuit on the rear stage side of the mixer. Also in this case, according to the present invention, both the noise reduction and the power consumption reduction can be achieved by properly using the first-type operational amplifier and the second-type operational amplifier on the front stage side and the rear stage side of the mixer.

  In the present invention, the first circuit may be a first amplifier circuit that amplifies an input signal, and the first amplifier circuit may be constituted by the first-type operational amplifier.

  In this way, thermal noise in the first amplifier circuit can be reduced, and noise in the analog circuit can be reduced.

  In the present invention, the second circuit is a second amplifier circuit that amplifies a signal after mixing from the mixer or a filter unit that performs filtering processing of the signal after mixing, and the second amplifier circuit or The filter unit may be configured by the second-type operational amplifier.

  In this way, flicker noise in the second amplifier circuit or filter unit can be reduced, and noise in the analog circuit can be reduced.

  In the present invention, when the channel length of the active load stage transistor of the differential section of the second-type operational amplifier is L3b, L1b <L3b may be satisfied.

  In this way, flicker noise in the second-type operational amplifier can be further reduced.

  In the present invention, when the WL ratio of the differential stage transistor of the first-type operational amplifier is RT1a and the WL ratio of the active load stage transistor of the first-type operational amplifier is RT3a, RT1a> RT3a. May be.

  In this way, thermal noise in the first-type operational amplifier can be further reduced.

In the present invention, the effective gate voltage of the differential stage transistor of the second-type operational amplifier is Veff, the current flowing between the drain and the source is Ids, the mobility is μ, and the gate capacitance per unit area is Cox. Where the WL ratio is RT1b, the Boltzmann constant is k, the absolute temperature is T, the electronic charge is q, and the process variation parameter is P (P> 1), P × (k × T / q )> Veff = {2 × Ids / (μ × Cox × RT1b)} ^{1/2 The} ratio WL1 RT1b may be set to a ratio that satisfies the relationship of k × T / q.

  In this way, the differential stage transistor of the second-type operational amplifier can be prevented from operating at the weak inversion region or the boundary between the weak inversion region and the strong inversion region, so flicker noise caused by making the WL ratio RT1b too large. The increase of can be minimized.

  In the present invention, the area of the arrangement region of the differential stage transistor among the elements constituting the second type operational amplifier is Sdf, and the elements other than the differential stage transistor among the elements constituting the second type operational amplifier. When the area of the element arrangement region is Sre, Sdf> Sre may be satisfied.

  In this way, the differential stage transistor having a large WL product can be arranged in the arrangement region having the area of Sdf, and flicker noise can be reduced.

  In the present invention, the differential stage transistor of the second-type operational amplifier is composed of J (J> 2) transistors connected in parallel, and the arrangement region of the differential stage transistors is connected in parallel. The J transistors may be arranged.

  In this way, the WL ratio can be increased while increasing the WL product of the differential stage transistor, and flicker noise can be reduced efficiently.

  In the present invention, the active load stage transistor of the second-type operational amplifier is composed of I (J> I> 2) transistors connected in parallel, and the J number of transistors constituting the differential load stage transistor. The transistors are arranged side by side along the X direction, and the I transistors constituting the active load stage transistor are arranged side by side along the X direction on the Y direction side of the J transistors. Also good.

  In this way, the J transistors constituting the differential stage transistors and the I transistors constituting the active load stage transistors can be efficiently arranged symmetrically, so that the layout efficiency can be improved.

  In the present invention, the mixer includes a third-type operational amplifier, the channel width of the differential stage transistor of the differential section of the third-type operational amplifier is W1c, the channel length is L1c, and the third-type operational amplifier When the bias current flowing in the differential section of the operational amplifier is Ic, W1c × L1c> W1a × L1a and Ic> Ib may be satisfied.

  If the relationship of W1c × L1c> W1a × L1a is established in this way, the WL product W1c × L1c of the third-type operational amplifier can be increased, and flicker noise of the third-type operational amplifier can be reduced. If the relationship of Ic> Ib is established, the bias current Ic of the third-type operational amplifier can be increased, so that the thermal noise of the third-type operational amplifier can be reduced. As a result, the SNR of the analog circuit can be improved.

  The present invention further includes a reference voltage supply circuit for supplying an analog reference voltage, the reference voltage supply circuit having a first-type operational amplifier for a reference voltage, and supplying the analog reference voltage to the first circuit. A second supply circuit for supplying an analog reference voltage to the second circuit, the first supply circuit for supplying the reference voltage, and a second type operational amplifier for the reference voltage; The channel width of the differential stage transistor of the differential section of the type 1 operational amplifier is W1d, the channel length is L1d, the bias current flowing in the differential section of the first type operational amplifier for the reference voltage is Id, and the reference The channel width of the differential stage transistor of the differential section of the voltage-type second-type operational amplifier is W1e, the channel length is L1e, and the bias current flowing in the differential section of the second-type operational amplifier for reference voltage is I When a, W1e × L1e> W1d × L1d, it may be Id> Ie.

  Thus, if the relationship of Id> Ie is established, the thermal noise superimposed on the analog reference voltage supplied from the first supply circuit can be minimized, and the thermal noise in the first circuit of the analog circuit can be suppressed. Can be prevented. Further, if the relationship of W1e × L1e> W1d × L1d is established, flicker noise superimposed on the analog reference voltage supplied from the second supply circuit can be minimized, and the analog circuit in the second circuit can be suppressed. An increase in flicker noise can be prevented.

  In the present invention, the first frequency may be a frequency of a carrier signal, and the second frequency may be a frequency of a desired signal carried by the carrier signal.

  The present invention also relates to an electronic device including any one of the analog circuits described above and a processing unit that performs processing based on information from the analog circuit.

The present invention also includes a differential section and an output section connected to an output node of the differential section, and the differential section includes a differential stage transistor and an active load stage transistor to constitute an operational amplifier. Of the elements, the area of the arrangement region of the differential stage transistor is Sdf, among the elements constituting the operational amplifier, the area of the arrangement area of elements other than the differential stage transistor is Sre, and the channel length of the differential stage transistor is L 1, the effective gate voltage V eff, the drain-source current Ids, the mobility μ, the gate capacitance per unit area Cox, the WL ratio RT 1, the channel of the active load stage transistor When the length is L3, the Boltzmann constant is k, the absolute temperature is T, the electronic charge is q, and the process variation parameter is P (P> 1) In a Sdf> Sre, L1 <with a L3, P × (k × T / q)> Veff = {2 × Ids / (μ × Cox × RT1)} 1/2 of> k × T / q It relates to an operational amplifier in which the WL ratio RT1 is set to a ratio that satisfies the relationship.

  In the present invention, since Sdf> Sre holds, a differential stage transistor having a large WL product can be arranged in an arrangement region having an area of Sdf, and flicker noise can be reduced. Moreover, flicker noise can be further reduced by setting L1 <L3. In order to satisfy the relationship of Sdf> Sre and L1 <L3, it is efficient to increase the WL ratio RT1. However, if the WL ratio RT1 is increased too much, the differential stage transistor operates in the weak inversion region, which may increase flicker noise. In this regard, in the present invention, the WL ratio RT1 is set so that the effective gate voltage is P × (k × T / q) Veff> k × T / q. In this way, the differential stage transistor of the second-type operational amplifier can be prevented from operating at the weak inversion region or the boundary between the weak inversion region and the strong inversion region, and thus flicker noise caused by excessively increasing the WL ratio RT1. The increase of can be minimized.

  Hereinafter, preferred embodiments of the present invention will be described in detail. The present embodiment described below does not unduly limit the contents of the present invention described in the claims, and all the configurations described in the present embodiment are indispensable as means for solving the present invention. Not necessarily. For example, in the following description, a radio timepiece receiver or a gyro sensor detector will be described as an analog circuit to which the present invention is applied, but the present invention is not limited to this.

1. Configuration of Analog Circuit FIG. 1 shows a configuration example of an analog circuit (analog front end circuit) 300 according to this embodiment. The analog circuit 300 includes a first circuit 310 and a second circuit 320.

  Here, the first circuit 310 includes a first-type operational amplifier OP1 in which the frequency of the amplification target signal (the signal to be subjected to the small signal amplification of the operational amplifier) is the first frequency. That is, the first circuit 310 includes at least one first-type operational amplifier OP1.

  Further, the second circuit 320 includes a second-type operational amplifier OP2 in which the frequency of the signal to be amplified is a second frequency lower than the first frequency. That is, the second circuit 320 includes at least one second-type operational amplifier OP2.

  For example, a first frequency input signal (received signal, sensor signal) from an antenna, a sensor, or the like is input to the first circuit 310. The first circuit 310 includes, for example, an amplifier circuit, and the amplifier circuit amplifies the input signal by the first-type operational amplifier OP1. In this case, the input signal (carrier signal) that is the signal to be amplified is a signal having a high first frequency.

  A mixer (frequency conversion circuit) 322 can be provided between the first circuit 310 and the second circuit 320. That is, the first circuit 310 can be a circuit provided on the upstream side of the mixer 322, and the second circuit 320 can be a circuit provided on the downstream side of the mixer 322. Then, the mixer 322 mixes a signal having a given frequency (for example, a local frequency signal and a synchronization signal) with a signal including a desired signal.

  The signal converted to the second frequency by mixing (frequency conversion) by the mixer 322 is input to the second circuit 320. The amplifier circuit, the filter unit, or the output circuit of the second circuit 320 performs small amplitude amplification on the signal of the second frequency that is the signal to be amplified.

  The first-type operational amplifier OP1 is an operational amplifier with lower thermal noise at a first frequency (for example, carrier frequency, modulation frequency, resonance frequency) than, for example, the second-type operational amplifier OP2. On the other hand, the second-type operational amplifier OP2 has a flicker noise at a second frequency (for example, the frequency of the desired signal carried by the carrier signal, the maximum frequency in the frequency band of the desired signal) as compared with the first-type operational amplifier OP1. It is a low operational amplifier.

  Specifically, a first frequency (for example, several tens of KHz to several hundred KHz) that is a signal frequency of the first-type operational amplifier OP1 is set to f1, and a second frequency that is a signal frequency of the second-type operational amplifier OP2 ( For example, let f2 be several Hz to several hundred Hz, and let fcr be the corner frequency of flicker noise and thermal noise. In this case, in the first-type operational amplifier OP1, for example, a relationship of f1-fcr <fcr-f2 is established. That is, the first-type operational amplifier OP1 is sized so that f1 is set to a frequency in the vicinity of the corner frequency fcr. Further, in the second-type operational amplifier OP2, for example, a relationship of fcr−f2 <f1−fcr is established. That is, the sizing of the second-type operational amplifier OP2 is performed so that f2 is set to a frequency near the corner frequency fcr.

  Further, the channel width of the differential stage transistor in the differential section of the first-type operational amplifier OP1 is W1a, the channel length is L1a, and the bias current (current value) flowing through the differential section is Ia. Further, it is assumed that the channel width of the differential stage transistor in the differential section of the second-type operational amplifier OP2 is W1b, the channel length is L1b, and the bias current flowing in the differential section is Ib. In this case, for example, the relationship of W1b × L1b> W1a × L1a and Ia> Ib is established.

  When a mixer 322 (frequency conversion circuit) is provided between the first circuit 310 and the second circuit 320, the mixer 322 can include, for example, a third-type operational amplifier OP3. The third-type operational amplifier OP3 has lower thermal noise at the frequency f1 than the second-type operational amplifier OP2 (or OP1), and flicker noise at the frequency f2 compared to the first-type operational amplifier OP1 (or OP2). Is a low operational amplifier. For example, when the channel width of the differential stage transistor in the differential section of the third-type operational amplifier OP3 is W1c, the channel length is L1c, and the bias current flowing through the differential section is Ic, W1c × L1c> W1a × L1a , Ic> Ib. Alternatively, the relationship of W1c × L1c> W1b × L1b and Ic> Ia may be established.

  2 and 3 show configuration examples of a radio clock receiver as an example of the analog circuit 300. FIG. The receiving apparatus is not limited to the configuration shown in FIGS. 2 and 3, and various modifications may be made such as omitting some of the components or adding other components.

  The receiving apparatus in FIG. 2 is an example of a direct conversion method. Currently, in each country (Japan, Germany, UK, etc.), a long wave standard radio wave including a time code as time data is transmitted. In Japan, 40 KHz and 60 KHz long-wave standard radio waves modulated with time codes are transmitted from two transmitting stations. That is, time data is transmitted as binary data with 60 seconds as one cycle. The radio timepiece receives the radio wave of such a time code and corrects the time data of the clock circuit. For this reason, the receiving device of the radio timepiece extracts a signal of a desired frequency by detecting and demodulating the received radio wave, and acquires a time code.

  Specifically, in FIG. 2, an antenna 330 constituted by a bar antenna or the like receives a long wave standard radio wave, and the received radio wave is converted into an electric signal and output. The amplifier circuit (RF amplifier circuit) 332 amplifies the signal from the antenna 330 and outputs it. The filter unit 334 has a frequency characteristic of a high-pass filter, for example, and performs a filter process on the signal from the amplifier circuit 332.

  The mixer (frequency conversion circuit) 336 directly converts the signal from the filter unit 334 into a baseband signal (several Hz) by mixing the signal of the local oscillation frequency from the local oscillation circuit 344. I do.

  The amplifier circuit 338 amplifies the signal from the mixer 336. The filter unit 340 has a frequency characteristic of a low-pass filter, for example, and performs a filter process on the signal from the amplifier circuit 338. The demodulation unit 342 demodulates the signal from the filter unit 340 and outputs a time code obtained by the demodulation. In the radio timepiece, the time data is corrected by this time code.

  The receiving apparatus in FIG. 3 is an example of a superheterodyne system. The amplifier circuit 352 amplifies the signal from the antenna 350. The filter unit 354 has a frequency characteristic of, for example, a band-pass filter, and performs a filter process on the signal from the amplifier circuit 352.

  The mixer 356 performs super-heterodyne conversion that generates a signal having an intermediate frequency (several hundreds of Hz) by mixing the signal having the local oscillation frequency from the local oscillation circuit 364 with the signal from the filter unit 354. .

  The filter unit 358 has a frequency characteristic of, for example, a bandpass filter, and allows a signal from the mixer 356 to pass a frequency component in a predetermined range centered on the intermediate frequency and cut off a frequency component outside the range. The detection circuit 360 detects a signal from the filter unit 358 (for example, envelope detection). The demodulator 362 demodulates the detection signal from the detection circuit 360 and outputs a time code obtained by the demodulation.

  In FIG. 2, for example, the amplifier circuit 332 (first amplifier circuit) and the filter unit 334 become the first circuit 310 of FIG. 1, and the amplifier circuit 332 and the filter unit 334 are configured by the first-type operational amplifier OP1. be able to.

  In FIG. 2, for example, the amplifier circuit 338 (second amplifier circuit) and the filter unit 340 become the second circuit 320 of FIG. 1, and the amplifier circuit 338 and the filter unit 340 are configured by the second-type operational amplifier OP2. can do.

  In FIG. 3, for example, the amplifier circuit 352 and the filter unit 354 become the first circuit 310 of FIG. 1, and the amplifier circuit 352 and the filter unit 354 can be configured by the first-type operational amplifier OP1.

  In FIG. 3, for example, the filter unit 358 and the detection circuit 360 become the second circuit 320 of FIG. 1, and the filter unit 358 and the detection circuit 360 can be configured by the second-type operational amplifier OP2.

  Note that the analog circuit 300 to which the method of the present embodiment can be applied is not limited to the radio clock receiver as shown in FIGS. For example, the present invention may be applied to a receiving device in an infrared remote controller or a detection device of various sensors such as a gyro sensor described later.

2. 2. Noise Reduction Technique 2.1 Noise Analysis FIG. 4A shows a configuration example of an operational amplifier used in this embodiment. This operational amplifier includes a differential unit 200 and an output unit 210.

  The differential unit 200 includes differential stage transistors M1 and M2 and active load stage transistors M3 and M4. A bias stage transistor M5 is also included. The differential stage transistors M1 and M2 are provided between the node N1 and the nodes N2 and N3, and differential input signals IM and IP are input to gates thereof. The active load stages M3 and M4 are provided between the nodes N2 and N3 and AGND (first power supply), and the node N2 is connected to the gate thereof. The bias stage transistor M5 is provided between VDD (second power supply) and the node N1. The bias node N4 of the bias circuit 212 constituted by the transistor M8 and the current source IS is connected to the gate of the bias stage transistor M5. As a result, a bias current IBD corresponding to the bias current IBS of the bias circuit 212 flows through the differential unit 200.

  The output unit 210 includes a drive stage transistor M6 and a bias stage transistor M7 provided between VDD and AGND. The output node N3 of the differential unit 200 is connected to the gate of the drive stage transistor M6, and the bias node N4 is connected to the gate of the bias stage transistor M7. A phase compensation capacitor CF and a resistor RF are provided between the nodes N3 and N5.

  Note that the configuration of the operational amplifier according to this embodiment is not limited to that shown in FIG. For example, FIG. 4A shows an example in which the differential stage transistors M1 and M2 and the bias stage transistor M5 are P-type transistors and the active load stage transistors M3 and M4 are N-type transistors. M2 and M5 may be N-type transistors, and M3 and M4 may be P-type transistors. Also, modifications such as omitting a part of the elements (transistors and capacitors) in FIG. 4A and adding other elements are possible.

Next, noise analysis of the operational amplifier in FIG. FIG. 4B shows an equivalent circuit of a small signal amplitude of a single transistor. Noise because it is handled in units of ^{V 2 (V} 2 / Hz), it will be handled in units of equivalent circuits ^{V 2} in FIG. 4 (B). In the equivalent circuit of FIG. 4B, in order to calculate the input conversion noise (gate conversion noise) of the transistor, a voltage source of noise Svg = Vn ² is provided at the gate of the transistor. Between the drain and the source, a current source of gm ² (Vgs ² + Vn ² ) and a resistance of 1 / gds ² are provided.

  As shown in FIG. 5A, noise includes flicker noise (1 / f noise) and thermal noise. Flicker noise is noise that occurs when electrons are captured and emitted from dangling bonds at the interface between the gate oxide film and the silicon substrate, and increases as the frequency decreases. On the other hand, thermal noise is noise generated by random movement of electrons when the channel region of a transistor is regarded as a resistance, and is proportional to absolute temperature.

  In the equivalent circuit of the input conversion noise in FIG. 4B, flicker noise and thermal noise are obtained as shown in the following equations (1) and (2), respectively.

  In the above equation (1), Cox is the gate capacitance per unit area of the transistor, W is the channel width, L is the channel length, f is the frequency, and K is a flicker noise constant depending on the manufacturing process. In the above equation (2), gm is a trans (mutual) conductance, k is a Boltzmann constant, and T is an absolute temperature.

  In this embodiment, the transfer function is calculated by replacing the operational amplifier circuit of FIG. 4A with the equivalent circuit of FIG. At that time, assuming that the size (W, L), noise level, and drain-source current of all the transistors of the operational amplifier are the same, only the upper term that accounts for 99% of the whole is derived by numerical analysis. Then, the noise Svg (noise spectrum) of the operational amplifier is obtained as in the following equation (3).

  In the above equation (3), Vn1, Vn2, Vn3, and Vn4 are gate noise voltages of the transistors M1, M2, M3, and M4 in FIG. 4A, and gm1, gm2, gm3, and gm4 are the transistors M1, This is the transconductance of M2, M3, and M4.

  As is apparent from the numerical analysis result of the above equation (3), the noise Svg of the operational amplifier is the noise of the differential stage transistors M1 and M2 and the active load stage transistors M3 and M4 of the differential section 200 in FIG. It is understood that most of the whole is occupied. Therefore, when sizing the operational amplifier, the channel width W and channel length L of these transistors M1, M2, M3, and M4 may be optimized.

  First, flicker noise is analyzed. The gate noise voltages Vn1, Vn2, Vn3, and Vn4 for flicker noise are obtained from the above equation (1) as in the following equations (4), (5), (6), and (7).

  In the above equations (4) to (7), W1 and L1 are the channel width and channel length of the differential stage transistors M1 and M2, and W3 and L3 are the channel width and channel length of the active load stage transistors M3 and M4. . Here, the channel width W2 and channel length L2 of M2 are the same as W1 and L1 of M1, and the channel width W4 and channel length L4 of M4 are the same as W3 and L3 of M3. Kp and Kn are process-dependent constants of the P-type transistor and the N-type transistor.

  Further, the transconductance gm1 (= gm2) of the differential stage transistors M1 and M2 and the transconductance gm3 (= gm4) of the active load stage transistors M3 and M4 are obtained by the following equations (8) and (9).

  In the above equations (8) and (9), μp and μn are the mobility of the P-type transistor and the N-type transistor, Ids is the drain-source current of the transistor, and Ids is the same in all the transistors M1 to M4. There is.

  By substituting the above equations (4) to (9) into the above equation (3), the flicker noise SvgFlc of the operational amplifier is obtained as the following equation (10).

  Next, thermal noise is analyzed. The gate noise voltages Vn1, Vn2, Vn3, and Vn4 for thermal noise are obtained from the above equation (2) as in the following equations (11), (12), (13), and (14).

  When the above equations (11) to (14) are substituted into the above equation (3), the thermal noise SvgThm of the operational amplifier is represented by the following equation (15).

  By substituting the above equations (8) and (9) into the above equation (15), the thermal noise SvgThm of the operational amplifier is obtained as the following equation (16).

When constants determined in the natural world and constants that depend only on the process are excluded from the above expression (10) regarding the flicker noise SvgFlc and the above expression (16) regarding the thermal noise SvgThm, W1, W3, L1, L3, and Ids are It becomes a variable that can be managed by design. Therefore, from the above equation (10), in order to reduce the flicker noise SvgFlc, the following holds.
(P1) The WL product W1 × L1 (gate area) of the differential stage transistors M1 and M2 is increased as much as possible. As a result, the first term of the above equation (10) becomes small, and SvgFlc becomes small.
(P2) The ratio of L1 / L3 is made as small as possible. That is, for example, L1 <L3. As a result, the second term of the above equation (10) is reduced, and SvgFlc is reduced. As a result, the WL product W3 × L3 of the active load stage transistors M3 and M4 also increases.
(P3) Flicker noise SvgFlc does not depend on Ids. Therefore, if only flicker noise is taken into account, the power consumption can be reduced by reducing Ids.

  From the above, in order to achieve both reduction of flicker noise SvgFlc and low power consumption, it is necessary to increase the WL product W1 × L1 of the differential stage transistor and reduce the bias current IBD (Ids) flowing in the differential section. I know it ’s good.

Further, from the above equation (16), the following holds in order to reduce the thermal noise SvgThm. (Q1) Increase the current Ids (IBD) as much as possible. As a result, the first term and the second term of the above equation (16) are reduced, and SvgThm is reduced.
(Q2) The WL ratio RT1 = W1 / L1 of the differential stage transistors M1 and M2 is made as large as possible, and the WL ratio RT3 = W3 / L3 of the active load stage transistors M3 and M4 is made as small as possible. That is, for example, RT1> RT3.
(Q3) The thermal noise SvgThm does not depend on the WL product W1 × L1 or W3 × L3. Therefore, if only thermal noise is considered, the area of the operational amplifier can be reduced by reducing W1 × L1 and W3 × L3.

  As described above, in order to achieve both the reduction of the thermal noise SvgThm and the reduction of the area of the operational amplifier, the operational amplifier whose size is reduced by increasing the bias current IBD flowing through the differential unit and reducing W1 × L1 and W3 × L3. It can be seen that it should be designed.

2.2 Use of operational amplifier As is clear from the above (P1) and (Q1), in order to reduce the noise of the operational amplifier, the WL product W1 × L1 of the differential stage transistor is increased to reduce the flicker noise. The thermal noise can be reduced by increasing the bias current IBD of the differential section.

  However, when the WL product W1 × L1 is increased, the layout area of the operational amplifier is increased, and the circuit becomes larger. On the other hand, when the bias current IBD is increased, the consumption current of the operational amplifier increases, which hinders low power consumption.

  Therefore, in this embodiment, in order to achieve both low noise, small circuit area, and low power consumption, first-type and second-type operational amplifiers OP1 and OP2 are prepared, and these operational amplifiers are used separately. Is adopted.

  For example, as shown in FIG. 5A, flicker noise is dominant in the low frequency region and thermal noise is dominant in the high frequency region.

  On the other hand, as shown in FIG. 5A, the frequency f1 (first frequency) of the amplification target signal of the first circuit 310 in FIG. The frequency f2 (second frequency) of the amplification target signal of the circuit 320 is low.

  Specifically, the frequency f1 corresponds to the frequency of the carrier signal, and is, for example, a frequency in a band (AC band) of several tens KHz to several hundreds KHz.

  On the other hand, the frequency f2 corresponds to the frequency of the desired signal carried by the carrier signal (the maximum frequency of the frequency band of the desired signal), and is, for example, a frequency in the band (DC band) of several Hz to several hundred Hz. .

  Therefore, in the present embodiment, in the first circuit 310 on the front stage side of the mixer 322 that extracts a desired signal from the carrier signal, the first-type operational amplifier OP1 that is an operational amplifier that is more important to reduce thermal noise than the second-type operational amplifier OP2 is used. use. Specifically, an operational amplifier having lower thermal noise at the frequency f1 of the carrier signal than the second-type operational amplifier OP2 is used.

  On the other hand, in the circuit on the rear stage side of the mixer 322, the second-type operational amplifier OP2, which is an operational amplifier in which flicker noise reduction is more important than the first-type operational amplifier OP1, is used. Specifically, an operational amplifier having lower flicker noise at the frequency f2 of the desired signal than the first-type operational amplifier OP1 is used.

  For example, in FIG. 5B, the WL product of the differential stage transistors of the first-type operational amplifier OP1 is expressed as W1 × L1 = W1a × L1a, and the bias current flowing through the differential unit is expressed as IBD = Ia. The frequency (operating frequency) of the signal to be amplified by the first-type operational amplifier OP1 is expressed as fp = f1. The WL ratio of the differential stage transistor is expressed as RT1 = RT1a = W1a / L1a, and the WL ratio of the active load stage transistor is expressed as RT3 = RT3a = W3a / L3a.

  On the other hand, the WL product of the differential stage transistors of the second-type operational amplifier OP2 is expressed as W1 × L1 = W1b × L1b, and the bias current flowing through the differential unit is expressed as IBD = Ib. The frequency of the amplification target signal of the second-type operational amplifier OP2 is expressed as fp = f2. The ratio of the channel length of the differential stage transistor and the active load stage transistor is expressed as L1 / L3 = L1b / L3b. The WL product of the third-type operational amplifier OP3 is expressed as W1 × L1 = W1c × L1c, and the bias current flowing through the differential unit is expressed as IBD = Ic.

  In this case, in the present embodiment, as shown in FIG. 5C, the relationship between W1b × L1b> W1a × L1a, Ia> Ib, and f1> f2 between the first-type and second-type operational amplifiers OP1 and OP2. Holds. Further, the relationship RT1a> RT3a holds for the first-type operational amplifier OP1, and the relationship L1b <L3b holds for the second-type operational amplifier OP2. For the third-type operational amplifier OP3, the relationship of W1c × L1c> W1a × L1a and Ic> Ib is established. Note that the relationship of W1c × L1c> W1b × L1b and Ic> Ia may be established.

  For example, as shown in FIG. 6A and the above-described (P1), flicker noise can be reduced by increasing the WL product W1 × L1 of the differential stage transistor, and in FIG. 6A and the above-described (Q1). As shown, the thermal noise can be reduced by increasing the bias current IBD of the differential section. On the other hand, the signal frequency f1 of the first-type operational amplifier OP1 is high, and the signal frequency f2 of the second-type operational amplifier OP2 is low.

  Therefore, in the present embodiment, as shown in FIG. 6B, in the first-type operational amplifier OP1 having a high signal frequency f1, by increasing the bias current IBD = Ia, the dominant thermal noise is obtained at the high frequency f1. To reduce the noise of the entire system. Specifically, the bias current Ia of the first-type operational amplifier OP1 is set to, for example, about 2 to 10 times, more preferably about 4 to 7 times the bias current Ib of the second-type operational amplifier OP2. On the other hand, since the influence of flicker noise is small at the high frequency f1, even if the WL product W1 × L1 = W1a × L1a of the first-type operational amplifier OP1 is increased, the layout area of the operational amplifier is only increased unnecessarily. Does not contribute to overall noise reduction. In this regard, in the present embodiment, W1a × L1a is smaller than W1b × L1b, so that a situation in which the layout area becomes unnecessarily large can be prevented.

  Further, in the present embodiment, as shown in FIG. 6C, in the second-type operational amplifier OP2 having a low signal frequency f2, the WL product W1 × L1 = W1b × L1b is increased so that it is dominant at the low frequency f2. Flicker noise is effectively reduced, and noise of the entire system is reduced. Specifically, the WL product W1b × L1b of the second-type operational amplifier OP2 is set to, for example, about 10 to 100 times the WL product W1a × L1a of the first-type operational amplifier OP1, more preferably 30 to 60 times. Set to degree. On the other hand, since the influence of thermal noise is small at the low frequency f2, even if the bias current IBD = Ib is increased, the current consumption of the operational amplifier only increases unnecessarily, and does not contribute much to the low power consumption of the entire system. . In this regard, in the present embodiment, since Ib is smaller than Ia, it is possible to prevent a situation in which the current consumption becomes unnecessarily large.

  Further, in the first-type operational amplifier OP1 that emphasizes thermal noise reduction, as is clear from the above-described equations (16) and (Q2), the WL ratio RT1a = W1a / L1a is increased as much as possible, while the WL ratio RT3a = By making W3a / L3a as small as possible, thermal noise can be reduced. Therefore, in the first-type operational amplifier OP1, transistors are sized so that the relationship of RT1a> RT3a is established. Specifically, RT1a is set to, for example, about 2 to 8 times, more preferably about 3 to 6 times that of RT3a. Thereby, the noise of the whole system can be further reduced.

  On the other hand, in the second-type operational amplifier OP2 in which reduction of flicker noise is important, flicker noise can be reduced by making the ratio of L1b / L3b as small as possible, as is clear from the aforementioned equations (10) and (P2).

  Therefore, in the second-type operational amplifier OP2, transistors are sized so that the relationship L1b <L3b is established. Specifically, L1b is set to, for example, about 0.4 to 0.8 times L3b. Thereby, the noise of the whole system can be further reduced.

2.3 Corner Frequency In the present embodiment, the first-type and second-type operational amplifiers OP1 are set so that the frequencies f1 and f2 and the corner frequency fcr have a relationship as shown in FIGS. 6B and 6C, for example. IP2 transistor sizing.

  That is, in the first-type operational amplifier OP1, as shown in FIG. 6B, the relationship of f1-fcr <fcr-f2 is established. On the other hand, in the second-type operational amplifier OP2, as shown in FIG. 6C, the relationship of fcr−f2 <f1−fcr is established.

  The corner frequency fcr is a frequency corresponding to the intersection of the flicker noise characteristic line and the thermal noise characteristic line in the frequency-noise characteristics of FIGS. 6 (A) to 6 (C).

  For example, in the first-type operational amplifier OP1 of FIG. 7A, when the corner frequency fcr is set at a position indicated by E1, noise at the frequency f1 increases, and thermal noise cannot be effectively reduced. On the other hand, when the corner frequency fcr is set to the position indicated by E2, the bias current IBD becomes unnecessarily large, which hinders low power consumption.

  For this reason, in the present embodiment, for the first-type operational amplifier OP1, the transistors of the operational amplifier are sized so that the corner frequency fcr is set in the vicinity of the frequency f1. In this case, ideally, the thermal noise and power consumption of the operational amplifier can be reduced optimally by setting fcr = f1 as indicated by E3 in FIG. However, if fcr = f1 is set, the thermal noise level may become larger than the desired noise level when there is a process variation.

  Therefore, in FIG. 6B, the relationship of f1−fcr <fcr−f2 is established, and the corner frequency fcr is set as close as possible to the frequency f1 side. As a result, it is possible to achieve both noise reduction and low power consumption of the operational amplifier while taking into account process variations.

  Further, in the second-type operational amplifier OP2 of FIG. 7B, when the corner frequency fcr is set at a position indicated by E4, noise at the frequency f2 increases, and flicker noise cannot be effectively reduced. On the other hand, if the corner frequency fcr is set to the position indicated by E5, the WL product W1 × L1 becomes unnecessarily large, which hinders the reduction of the circuit area.

  Therefore, in the present embodiment, for the second-type operational amplifier OP2, the transistors of the operational amplifier are sized so that the corner frequency fcr is set in the vicinity of the frequency f2. In this case, ideally, by setting fcr = f2 as indicated by E6 in FIG. 7B, the flicker noise and layout area of the operational amplifier can be reduced optimally. However, if fcr = f2 is set, the flicker noise level may become larger than the desired noise level when there is a process variation.

  Therefore, in FIG. 6C, the relationship of fcr−f2 <f1−fcr is established, and the corner frequency fcr is made as close as possible to the frequency f2. As a result, it is possible to achieve both noise reduction and area reduction of the operational amplifier while taking into account process variations.

2.4 Effective Gate Voltage As shown in FIG. 5C and the above equation (10), flicker is achieved by increasing the WL product W1b × L1b of the differential stage transistor as much as possible and reducing the channel length L1b as much as possible. Noise can be reduced. Therefore, it is considered that flicker noise can be efficiently reduced by performing sizing such that the WL ratio W1b / L1b increases.

  However, it has been found that if the WL ratio W1b / L1b is increased too much, the effective gate voltage Veff decreases and flicker noise increases. The effective gate voltage Veff is expressed by the following equation (17).

  Here, Vgs is the gate-source voltage of the transistor, Vth is the threshold voltage, Ids is the drain-source current, μ is the mobility, Cox is the gate capacitance per unit area, and RT1b is the WL of the differential stage transistor. The ratio is RT1b = W1b / L1b.

  For example, FIGS. 8A and 8B show the measurement results regarding the relationship between the effective gate voltage Veff and noise (noise level Svg). FIG. 8A shows an example of an N-type transistor, and FIG. 8B shows an example of a P-type transistor. 8A and 8B, the effective gate voltage Veff is changed by changing Vgs. The drain-source voltage Vds is equal to the gate-source voltage Vgs.

  As indicated by E7 in FIG. 8A and E8 in FIG. 8B, when the effective gate voltage Veff decreases, the noise increases sharply. This is because when the effective gate voltage Veff, which is a reference voltage having a small signal amplitude, decreases, the transistor operates in the weak inversion region, and flicker noise increases rapidly. For example, in FIGS. 8A and 8B, the flicker noise increases rapidly from Veff = 10 mV to 100 mV. Therefore, in order to suppress an increase in flicker noise resulting from the operation in the weak inversion region, it is desirable that the effective gate voltage Veff be larger than 10 mV to 100 mV.

  FIG. 9A is a simulation result showing the relationship between the area of the operational amplifier, the current consumption, and the noise when the effective gate voltage Veff is changed. The X axis is the area (the length of one side when the operational amplifier is square). ), The Y axis represents the current consumption, and the Z axis represents the noise level.

  In FIG. 9A, the effective gate voltage Veff is changed by changing the WL ratio RT1b = W1b / L1b under the condition that the WL product W1b × L1b is constant. For example, the directions indicated by arrows F1 and F2 in FIG. 9A are directions in which the effective gate voltage Veff decreases, and the effective gate voltage Veff is decreased by increasing the WL ratio RT1b = W1b / L1b. .

  As indicated by F1 in FIG. 9A, when the effective gate voltage Veff is reduced, the area of the operational amplifier is reduced and the current consumption is reduced, but the noise is hardly changed.

  However, the noise increases rapidly at F3 in FIG. That is, when the WL ratio RT1b = W1b / L1b becomes larger than a predetermined value, the flicker noise increases abruptly because the transistor operates in the weak inversion region as described with reference to FIGS. 8A and 8B. The noise of the operational amplifier also increases rapidly.

  In this case, at the point F4 in FIG. 9A, although the noise is small, the area and current consumption of the operational amplifier are large, so that the area and current are wasted.

  Therefore, in the present embodiment, the WL ratio RT1b = W1b / L1b of the differential stage transistor is determined (the value is narrowed down) so as to be set to the point F3 in FIG. 9A. For example, the WL ratio RT1b can be a value in the range of 50 to 200. Then, under the condition of RT1b determined in this way, the transistors of the operational amplifier are sized so that W1b × L1b becomes larger and L1b / L3b becomes smaller as shown in FIG. 5C.

  Specifically, in order to prevent the transistor from operating in the weak inversion region, the following equation (18) may be established.

Here, k is a Boltzmann constant, T is an absolute temperature, q (= 1.602 × 10 ⁻¹⁹ coulomb) is an electronic charge amount, and k × T / q = 25.7 mV at room temperature (25 degrees). .

  However, as shown in FIGS. 8A and 8B, flicker noise also rises at the boundary between the weak inversion region and the strong inversion region, and it is necessary to consider process variation. Therefore, when the process variation parameter (process-dependent parameter) is P (P> 1), the following equation (19) is established.

  Here, the process variation parameter can be, for example, P = 3.0, and more preferably, P can be a value in the range of 1.5 to 2.0.

  In this embodiment, the WL ratio RT1b = W1b / L1b of the differential stage transistor is determined so that the effective gate voltage Veff is in a range satisfying the above equation (19). Then, the transistors of the operational amplifier are sized so that W1b × L1b becomes larger and L1b / L3b becomes smaller under the determined RT1b range. It should be noted that after W1b / L1b is determined and W1b × L1b or L1b / L3b is determined, RT1b is changed again to perform fine adjustment.

  FIG. 9B shows an operational amplifier when L1b / L3b (in other words, L3b / L1b), which is the ratio of the gate length L1b of the differential stage transistor and the gate length L3b of the active load stage transistor, is changed. It is a simulation result which shows the relationship of an area, current consumption, and noise.

  When L1b / L3b is increased as indicated by F5 in FIG. 9B (L3b / L1b is decreased), the noise increases rapidly from the point indicated by F6 as indicated by F7. In this case, as L1b / L3b is increased, the area of the operational amplifier is reduced, but the consumption current hardly changes.

  From the result of FIG. 9B, it can be seen that in order to optimally set L1b / L3b, it is only necessary to set L1b / L3b so as to be the point indicated by F6. Specifically, the ratio of L1b / L3b can be set to a value within the range of 0.4 to 0.8, for example.

  In the present embodiment, by sizing the transistors of the operational amplifier as described above, it is possible to achieve both noise reduction, circuit scale reduction, and low power consumption.

  For example, in FIG. 10, the noise level of each circuit when the transistor of the operational amplifier is sized by the method of the present embodiment and the noise level of each circuit in the comparative example that is not sized by the method of the present embodiment, Indicated by a bar graph.

  FIG. 10 shows the results when the method of the present embodiment is applied to a gyro sensor detection device to be described later. QVAMP, DIFF, PGA, SYNCD, and FLT are a Q / V conversion circuit, a differential amplification circuit, The noise level of a sensitivity adjustment circuit, a synchronous detection circuit, and a filter part is shown.

  As shown in FIG. 10, according to the present embodiment, the op amp is used properly for each of the circuit handling the AC signal and the circuit handling the DC signal, and the optimum low noise design is performed. As a result, the entire system is efficiently reduced in noise. Have succeeded in doing.

2.5 Layout Next, a layout method of the first-type and second-type operational amplifiers OP1 and OP2 will be described. FIG. 11A is a layout example of the first-type operational amplifier OP1, and FIG. 11B is a layout example of the second-type operational amplifier OP2.

  In FIG. 11B, H2 indicates an arrangement region of the differential stage transistors M1 and M2 among the elements (transistors, capacitors, resistors, etc.) constituting the second-type operational amplifier OP2. The area of the arrangement region of H2 is Sdf, and the area of the arrangement region of elements other than the differential stage transistors M1 and M2 among the elements constituting the second-type operational amplifier is Sre. The elements other than the differential stage transistors M1 and M2 are, for example, transistors M3, M4, M5, M6, and M7 other than M1 and M2 in FIG. 4A, a capacitor CF, and a resistor RF. In this case, in FIG. 11B, a relationship of Sdf> Sre is established. If such a relationship is established, the WL product W1b × L1b can be increased, and flicker noise can be reduced.

  On the other hand, in the first-type operational amplifier OP1 in FIG. 11A, since the arrangement region of the differential stage transistors M1 and M2 is small as indicated by H1, the relationship of Sdf> Sre does not hold. Instead, thermal noise is reduced by sizing the transistor so that the bias current IBD = Ia is increased.

  In FIG. 11B, the differential stage transistor M1 of the second-type operational amplifier OP2 includes a plurality (J) of transistors TR11, TR12, TR13, and TR14 connected in parallel. Similarly, the differential stage transistor M2 of OP2 includes a plurality of transistors TR21, TR22, TR23, and TR24 connected in parallel. A plurality of transistors TR11 to TR14 and TR21 to TR24 connected in parallel are arranged in the arrangement region of the differential stage transistors M1 and M2 indicated by H2.

  In FIG. 11B, the X direction is the channel length direction, and the Y direction is the channel width direction. The channel lengths of the transistors TR11 to TR14 and TR21 to TR24 connected in parallel are L1b. On the other hand, if the number of transistors TR11 to TR14 and TR21 to TR24 connected in parallel is J (J = 4 in FIG. 11B), the channel width of each transistor of TR11 to TR14 and TR21 to TR24 is W11b = W1b / J.

  If the transistors TR11 to TR14 and TR21 to TR24 connected in parallel as shown in FIG. 11B are arranged, the WL ratio W1b / L1b can be increased while the WL product W1b × L1b is increased, and the flicker is increased. Noise can be reduced efficiently.

  In FIG. 11B, the active load stage transistors M3 and M4 are also composed of a plurality (I pieces) of transistors TR31, TR32, TR41, and TR42 connected in parallel. The channel length of each transistor of TR31, TR32, TR41, and TR42 connected in parallel is L3b. On the other hand, if the number of transistors TR31, TR32, TR41, TR42 connected in parallel is I (I = 2 in FIG. 11B), the channel width of each transistor of TR31, TR32, TR41, TR42 is W33b = W3b. / I.

  In FIG. 11B, J (for example, J = 4) transistors TR11 to TR14 and TR21 to TR24 are arranged in the X direction in the arrangement region of the differential stage transistors M1 and M2, while the active load stage. In the arrangement region of the transistors M3 and M4, I (I <J, for example, I = 2) transistors TR31, TR32, TR41, and TR42 are arranged in the X direction. In this way, while satisfying the relationship of L1b <L3b described in FIG. 5C, the transistors TR11 to TR14, TR21 to TR24 that constitute M1 and M2, and the transistors TR31 to TR32 that constitute M3 and M4, TR41 to TR42 can be efficiently arranged symmetrically in a rectangular arrangement region. Thereby, layout efficiency can be improved.

  FIG. 12 shows a layout example of the third-type operational amplifier OP3. In FIG. 12, H3 indicates an arrangement region of the differential stage transistors M1 and M2. In FIG. 12, the relationship of Sdf> Sre is established as in FIG. 11B, and the WL product W1c × L1c can be increased, so that flicker noise can be reduced.

  Also in FIG. 12, the differential stage transistors M1 and M2 are constituted by a plurality of transistors TS11 to TS14 and TS21 to TS24 connected in parallel. A plurality (J) of the transistors TS11 to TS14 and TS21 to TS24 connected in parallel are arranged in the arrangement region of the differential stage transistors M1 and M2 indicated by H3.

  The channel length of the transistors TS11 to TS14 and TS21 to TS24 connected in parallel is L1c, and the channel width is W11c = W1c / J.

  As apparent from a comparison between FIG. 12 and FIG. 11B, in the third-type operational amplifier OP3 in FIG. 12, the area of the arrangement region of the differential stage transistors M1 and M2 indicated by H3 is larger than that in FIG. Is even larger. Thus, an operational amplifier that can further reduce flicker noise than that in FIG. 11B can be realized.

3. Modification FIG. 13 shows a modification of the present embodiment. In FIG. 13, the configuration of the reference voltage supply circuit 20 is further added to the configuration of FIG.

  The reference voltage supply circuit 20 supplies AGND (analog reference voltage in a broad sense) to the first circuit 310. This AGND (analog ground) is a voltage serving as a reference of the analog circuit, and the signal amplification of the operational amplifier is performed with reference to this AGND. Therefore, the reference voltage supply circuit 20 needs to supply AGND having a stable potential to the first circuit 310.

  The reference voltage supply circuit 20 includes a first supply circuit 21 and a second supply circuit 22. The first supply circuit 21 supplies AGND (analog reference voltage) to the first circuit 310. The second supply circuit 22 supplies AGND to the second circuit 320. The first supply circuit 21 has a first-type operational amplifier OP1 for reference voltage, and the second supply circuit 22 has a second-type operational amplifier OP2 for reference voltage.

  Here, it is assumed that the channel width of the differential stage transistor in the differential portion of the first-type operational amplifier OP1 for the reference voltage is W1d, the channel length is L1d, and the bias current flowing in the differential portion is Id. Further, it is assumed that the channel width of the differential stage transistor in the differential portion of the second-type operational amplifier OP2 for the reference voltage is W1e, the channel length is L1e, and the bias current flowing in the differential portion is Ie. In this case, the relationship of W1e × L1e> W1d × L1d and Id> Ie is established.

  That is, in the first-type operational amplifier OP1 for the reference voltage, transistor sizing similar to that of the first-type operational amplifier included in the first circuit 310 described with reference to FIGS. 4A to 12 is performed. In the second-type operational amplifier OP2 for the reference voltage, the same transistor sizing as that of the second-type operational amplifier included in the second circuit 320 described with reference to FIGS. Specifically, for example, regarding the sizing of the differential stage transistor of the first-type operational amplifier OP1 for the reference voltage, W1d = W1a and L1d = L1a can be set. The sizing of the differential stage transistor of the second-type operational amplifier OP2 for the reference voltage can also be set to W1e = W1b and L1e = L1b. The same applies to sizing of the active load stage transistor.

  For example, in the first circuit 310 that operates by supplying AGND from the first supply circuit 21, the frequency f1 of the signal to be amplified is a high frequency as shown in FIG. 5A, and at this frequency f1, As mentioned above, thermal noise is dominant. Therefore, even if the first circuit 310 is configured with a first-type operational amplifier, if thermal noise is superimposed on AGND, the thermal noise of the first circuit 310 eventually increases.

  In this regard, in FIG. 13, the first supply circuit 21 has a first-type operational amplifier OP1 for a reference voltage, and supplies AGND using this OP1. The first-type operational amplifier OP1 for the reference voltage is an operational amplifier with an emphasis on thermal noise reduction as described with reference to FIG. Therefore, thermal noise superimposed on the AGND line AGL1 can be minimized, and increase in thermal noise in the first circuit 310 can be prevented.

  Further, in the second circuit 320 that operates by being supplied with AGND from the second supply circuit 22, the frequency f2 of the signal to be amplified is a low frequency as shown in FIG. 5A, and at this frequency f2, As described above, flicker noise is dominant. Therefore, even if the second circuit 320 is constituted by a second-type operational amplifier, if flicker noise is superimposed on AGND, the flicker noise of the second circuit 320 eventually increases.

  In this regard, in FIG. 13, the second supply circuit 22 has a second-type operational amplifier OP2 for a reference voltage, and supplies AGND using this OP2. The second-type operational amplifier OP2 for the reference voltage is an operational amplifier with an emphasis on reducing flicker noise as described with reference to FIG. Accordingly, flicker noise superimposed on the AGND line AGL2 can be minimized, and increase in flicker noise in the second circuit 320 can be prevented.

  The AGND from the first supply circuit 21 is supplied via the AGND line AGL1 (first analog reference voltage line in a broad sense). On the other hand, AGND from the second supply circuit 22 is supplied via an AGND line AGL2 (second analog reference voltage line in a broad sense). In this case, the AGND lines AGL 1 and AGL 2 are separated and wired from the reference voltage supply circuit 20 to the first and second circuits 310 and 320. That is, the two AGND lines AGL 1 and AGL 2 are separated in terms of layout and connected to the first circuit 310 and the second circuit 320. By doing so, it is possible to prevent the noise from the AGND line AGL1 from being transmitted to the AGND line AGL2, and the noise from the AGND line AGL2 from being transmitted to the AGND line AGL1.

  For example, if thermal noise from the AGND line AGL2 is transmitted to the AGND line AGL1, even if the first-type operational amplifier OP1 that emphasizes thermal noise reduction is used in the first supply circuit 21, the thermal noise from the AGL2 1 is transmitted to the first circuit 310, and the SNR deteriorates. Similarly, if flicker noise from the AGND line AGL1 is transmitted to the AGND line AGL2, the flicker noise from the AGL1 is obtained even if the second type operational amplifier OP2 that is important for reducing flicker noise is used in the second supply circuit 22. Is transmitted to the second circuit 320, and the SNR deteriorates.

  In this regard, if the AGND lines AGL1 and AGL2 are separated and wired in the layout as shown in FIG. 13, the above situation can be prevented and the SNR of the entire system can be improved.

  FIG. 14 shows a detailed configuration example of the reference voltage supply circuit 20. The reference voltage supply circuit 20 is not limited to the configuration shown in FIG. 14, and various modifications such as omitting some of the components or adding other components are possible.

  The reference voltage supply circuit 20 includes first, second, and third supply circuits 21, 22, and 23. A reference voltage generation circuit 26 is also included.

  The first supply circuit 21 (first impedance conversion circuit) performs voltage impedance conversion using, for example, a first-type operational amplifier OP1 for reference voltage. That is, the first-type operational amplifier OP1 included in the first supply circuit 21 is a voltage follower-connected operational amplifier in which an inverting input terminal (second input terminal in a broad sense) is connected to an output terminal. Is connected to the AGND line AGL1.

  The second supply circuit 22 (second impedance conversion circuit) performs voltage impedance conversion using, for example, a second-type operational amplifier OP2 for reference voltage. That is, the second-type operational amplifier OP2 included in the second supply circuit 22 is a voltage follower-connected operational amplifier whose inverting input terminal (second input terminal) is connected to the output terminal, and the output terminal of OP2 Are connected to the AGND line AGL2.

  The third supply circuit 23 is provided in front of the first and second supply circuits 21 and 22 and supplies the voltage V3Q to the first and second supply circuits 21 and 22. The first and second supply circuits 21 and 22 perform impedance conversion of the output voltage V3Q from the third supply circuit 23 and output AGND.

  The third supply circuit 23 includes the third-type operational amplifier OP3 described with reference to FIGS. In addition, a voltage dividing circuit including resistors RJ1, RJ2, and RJ3 can be included.

  The reference voltage generation circuit 26 generates a reference voltage VR for generating AGND. As the reference voltage generation circuit 26, for example, a circuit that generates the reference voltage VR by a band gap can be employed.

  For example, in FIG. 14, the voltage at the node NJ1 becomes equal to the reference voltage VR due to an imaginary short of the third-type operational amplifier OP3. Therefore, when the resistance values of the resistors RJ2 and RJ3 are R2 and R3, the output voltage of the third supply circuit 23 is V3Q = VR × {(R2 + R3) / R3}. The first and second supply circuits 21 and 22 perform impedance conversion of the output voltage V3Q = AGND = VR × {(R2 + R3) / R3}. This stabilizes the potential of AGND.

  Now, assume that the channel width of the differential stage transistor in the differential section of the third-type operational amplifier OP3 for the reference voltage is W1f, the channel length is L1f, and the bias current flowing in the differential section is If. In this case, the relationship of W1f × L1f> W1d × L1d and If> Ie is established.

  That is, in the third-type operational amplifier OP3 for the reference voltage, the same transistor sizing as the third-type operational amplifier included in the mixer 322 described with reference to FIGS. 4A to 12 is performed. Specifically, for example, the sizing of the differential stage transistor of the third-type operational amplifier OP3 for the reference voltage can be set to W1f = W1c and L1f = L1c. The same applies to sizing of the active load stage transistor.

  For example, the first supply circuit 21 receives the output voltage V3Q from the third supply circuit 23 and supplies AGND to the first circuit 310. Therefore, when thermal noise is superimposed on the output voltage V3Q of the third supply circuit 23, even if the first-type operational amplifier OP1 that emphasizes thermal noise reduction is used as the first supply circuit 21, it is superimposed on V3Q. Thermal noise is transmitted to the first circuit 310. As a result, the SNR of the entire system deteriorates.

  The second supply circuit 22 receives the output voltage V3Q from the third supply circuit 23 and supplies AGND to the second circuit 320. Therefore, if flicker noise is superimposed on the output voltage V3Q of the third supply circuit 23, it is superimposed on V3Q even if the second supply circuit 22 uses the second-type operational amplifier OP2 that emphasizes the reduction of flicker noise. Flicker noise is transmitted to the second circuit 320. As a result, the SNR of the entire system deteriorates.

  In this regard, in the third-type operational amplifier OP3 of the third supply circuit 23 in FIG. 14, the relationship of W1f × L1f> W1d × L1d and If> Ie (W1c × L1c> W1a × L1a, Ic> Ib) is established. More preferably, the relationship of W1f × L1f> W1e × L1e and If> Id (W1c × L1c> W1b × L1b, Ic> Ia) is established. Therefore, the third-type operational amplifier OP3 is an operational amplifier with low thermal noise and flicker noise. For example, the third-type operational amplifier OP3 has a large WL product and a large bias current as shown in the layout example of FIG. Therefore, the operational amplifier is very low noise in both thermal noise and flicker noise.

  As described above, when the third-type operational amplifier OP3 having low noise in both thermal noise and flicker noise is used as the third supply circuit 23, the thermal noise and flicker noise of the output voltage V3Q of the third supply circuit 23 are minimized. To the limit. Accordingly, thermal noise of the output voltage V3Q is transmitted to the first circuit 310 via the first supply circuit 21, and flicker noise of the output voltage V3Q is transmitted to the second circuit 320 via the second supply circuit 22. The situation of transmission can be minimized, and the SNR of the entire system can be greatly improved.

4). Next, a case where the method of the present embodiment is applied to a gyro sensor detection device will be described. FIG. 15 shows a configuration example of the detection device 30. The detection device 30 includes a drive circuit 40 and a detection circuit 60. The detection device 30 is not limited to the configuration of FIG. 15, and various modifications such as omitting some of the components or adding other components are possible. For example, when the synchronization signal can be extracted based on the detection signal from the vibrator 10, the configuration of the drive circuit 40 may be omitted.

  The vibrator 10 (vibration gyro), which is a physical quantity transducer, is a piezoelectric vibrator formed of a piezoelectric material such as quartz. FIG. 16A shows a tuning fork type piezoelectric vibrator as an example of the vibrator 10. The vibrator 10 includes drive vibrators 11 and 12 and detection vibrators 16 and 17. The drive vibrators 11 and 12 are provided with drive terminals 2 and 4, and the detection vibrators 16 and 17 are provided with detection terminals 6 and 8. FIG. 16A shows an example in which the vibrator 10 is a tuning fork type, but the vibrator 10 of the present embodiment is not limited to such a structure. For example, it may be T-shaped or double T-shaped. The piezoelectric material of the vibrator 10 may be other than quartz. Further, the vibrator 10 that is a physical quantity transducer may be an electrostatic MEMS (Micro Electro Mechanical Systems) that similarly performs a drive / detection operation by electrostatic capacitance. A physical quantity transducer is an element for converting a physical quantity (a quantity representing the degree of the property of an object, the unit of which is defined) into another physical quantity. As physical quantities to be converted, in addition to Coriolis force, force such as gravity, acceleration, and mass can be considered. In addition to the current (charge), the physical quantity obtained by the conversion may be a voltage or the like.

  The drive circuit 40 outputs a drive signal (drive voltage) VD to drive the vibrator 10 (physical quantity transducer in a broad sense) and receives a feedback signal VF from the vibrator 10. Thereby, the vibrator 10 is excited. The detection circuit 60 receives the detection signals (detection current, charge) ISP and ISM from the vibrator 10 driven by the drive signal VD, and detects (extracts) a desired signal (Coriolis force signal) from the detection signal.

  Specifically, an alternating drive signal (drive voltage) VD from the drive circuit 40 is applied to the drive terminal 2 of the drive vibrator 11 in FIG. Then, the driving vibrator 11 starts to vibrate due to the reverse voltage effect, and the driving vibrator 12 also starts to vibrate due to the tuning fork vibration. At this time, a current (charge) generated by the piezoelectric effect of the drive vibrator 12 is fed back from the drive terminal 4 to the drive circuit 40 as a feedback signal VF. As a result, an oscillation loop including the vibrator 10 is formed.

  When the driving vibrators 11 and 12 vibrate, the detection vibrators 16 and 17 vibrate in the direction shown in FIG. Then, currents (charges) generated by the piezoelectric effect of the detection vibrators 16 and 17 are output from the detection terminals 6 and 8 as detection signals ISP and ISM. The detection circuit 60 receives these detection signals ISP and ISM and detects a desired signal (desired wave) that is a signal corresponding to the Coriolis force.

  That is, when the vibrator 10 (gyro sensor) rotates around the detection axis 19 in FIG. 16A, a Coriolis force Fc is generated in a direction orthogonal to the vibration direction of the vibration speed v. For example, FIG. 16B schematically shows a view of the detection shaft 19 of FIG. In FIG. 16B, when the angular velocity when rotating around the detection axis 19 is ω, the mass of the vibrator is m, and the vibration speed of the vibrator is v, the Coriolis force is Fc = 2m · v · ω. It is expressed. Therefore, when the detection circuit 60 detects (extracts) a desired signal that is a signal corresponding to the Coriolis force, the rotational angular velocity ω of the gyro sensor (vibrator) can be obtained.

  The vibrator 10 has a drive side resonance frequency fd and a detection side resonance frequency fs. Specifically, the natural resonance frequency of drive vibrators 11 and 12 (the natural resonance frequency of drive vibration mode) is fd, and the natural resonance frequency of detection vibrators 16 and 17 (the natural resonance frequency of detection vibration mode). Is fs. In this case, the drive vibrators 11 and 12 and the detection vibrators 16 and 17 can perform the detection operation and have an appropriate inter-mode coupling that does not cause unnecessary resonance coupling. Has a certain frequency difference. The detuning frequency Δf = | fd−fs |, which is this frequency difference, is set to a frequency that is sufficiently smaller than fd and fs.

The drive circuit (oscillation circuit) 40 includes an amplification circuit 42, an AGC (Automatic Gain Control) circuit 44 that performs automatic gain control, and a binarization circuit (comparator) 46. In the drive circuit 40, in order to keep the sensitivity of the gyro sensor constant, it is necessary to keep the amplitude of the drive voltage supplied to the vibrator 10 (drive vibrator) constant. Therefore, an AGC circuit 44 for automatically adjusting the gain is provided in the oscillation loop of the drive vibration system. Specifically, the AGC circuit 44 automatically adjusts the gain variably so that the amplitude (vibration speed v of the vibrator) of the feedback signal FD becomes constant. The phase is adjusted so that the phase shift in the oscillation loop is 0 degree (0 deg). At the time of oscillation startup, the gain of the oscillation loop is set to a gain larger than 1 in order to enable high-speed oscillation startup.

  The amplifier circuit 42 amplifies the feedback signal FD from the vibrator 10. Specifically, the I / V conversion circuit included in the amplifier circuit 42 converts the current (charge) that is the feedback signal FD from the transducer 10 into a voltage, amplifies it, and outputs it as the drive side amplified signal VD2.

  The AGC circuit 44 monitors the drive side amplified signal VD2 that is a signal after being amplified by the drive side amplifier circuit 42, and controls the gain of the oscillation loop. The AGC circuit 44 includes a gain control amplifier (GCA) for controlling the oscillation amplitude in the oscillation loop and a gain control circuit for outputting a control voltage for adjusting the gain of the gain control amplifier in accordance with the oscillation amplitude. be able to. In addition, this gain control circuit corresponds to a rectifier circuit (full-wave rectifier circuit) that converts the AC signal VD2 from the amplifier circuit 42 into a DC signal, or the difference between the voltage of the DC signal from the rectifier circuit and the reference voltage. A circuit for outputting a control voltage can be included.

  The binarization circuit 46 performs binarization processing of the drive side amplified signal VD2 that is a sine wave, and the synchronization signal (reference signal) CLK obtained by the binarization processing is supplied to the synchronization detection circuit 100 of the detection circuit 60. Output. The binarization circuit 46 can be realized by a comparator that receives the sine wave (alternating current) signal VD2 from the amplifier circuit 42 and outputs a rectangular wave synchronization signal CLK. Another circuit may be provided between the amplifier circuit 42 and the binarization circuit 46 or between the binarization circuit 46 and the synchronous detection circuit 100. For example, a high-pass filter or a phase shift circuit (phase shifter) may be provided.

  The detection circuit 60 includes an amplifier circuit 70, a synchronous detection circuit 100, and a filter unit 110. Note that some of these components may be omitted, or other components may be added.

  The amplifier circuit 70 amplifies the detection signals ISP and ISM from the vibrator 10. Specifically, a Q / V conversion circuit (I / V conversion circuit) included in the amplifier circuit 70 receives signals ISP and ISM from the vibrator 10 and converts electric charges (current) generated in the vibrator 10 into a voltage. Amplify.

  A synchronous detection circuit (detection circuit, detector) 100 performs synchronous detection based on a synchronous signal (synchronous clock, reference signal) CLK. This synchronous detection makes it possible to eliminate unnecessary signals for mechanical vibration leakage.

  The filter unit 110 provided on the subsequent stage side of the synchronous detection circuit 100 performs a filtering process on the signal VS6 after the synchronous detection. Specifically, low-pass filter processing for removing high frequency components is performed.

  The detection signal (sensor signal) from the vibrator 10 includes a desired signal (desired wave) and an unnecessary signal (unnecessary wave). Since the amplitude of the unnecessary signal is generally about 100 to 500 times the amplitude of the desired signal, the required performance for the detection device 30 is increased. This unnecessary signal may be caused by mechanical vibration leakage, electrostatic coupling leakage, detuning frequency Δf, 2fd (2ωd), DC offset, or the like. The unnecessary signal of mechanical vibration leakage is generated due to an imbalance of the shape of the vibrator 10 or the like. Further, the unnecessary signal of electrostatic coupling leakage is generated when the drive signal VD of FIG. 15 leaks to the input terminals of the ISP and ISM through the parasitic capacitors CP and CM.

  FIGS. 17A to 17C are frequency spectra for explaining the removal of unnecessary signals. FIG. 17A shows a frequency spectrum before synchronous detection. As shown in FIG. 17A, in the detection signal before synchronous detection, a DC offset unnecessary signal exists in the DC frequency band. Further, an unnecessary signal and a desired signal of mechanical vibration leakage exist in the frequency band of fd.

  FIG. 17B shows a frequency spectrum after synchronous detection. The desired signal in the fd frequency band in FIG. 17A appears in the DC and 2fd frequency bands after synchronous detection, as shown in FIG. 17B. Further, an unnecessary signal (DC offset) in the DC frequency band in FIG. 17A appears in the fd frequency band after synchronous detection, as shown in FIG. 17B. Further, an unnecessary signal (mechanical vibration leakage) in the fd frequency band in FIG. 17A appears in the 2fd frequency band after synchronous detection, as shown in FIG. 17B.

  FIG. 17C shows the frequency spectrum after the filter processing. By smoothing (LPF) the signal after synchronous detection by the filter unit 110, it is possible to remove frequency components of unnecessary signals in frequency bands such as fd and 2fd.

  In the present embodiment, the amplifier circuit 70 includes the first-type operational amplifier OP1, and the filter unit 110 includes the second-type operational amplifier OP2.

  Here, the first-type operational amplifier OP1 is an operational amplifier having lower thermal noise at the frequency of the carrier signal (for example, the resonance frequency of the vibrator and the drive-side resonance frequency) than that of the second-type operational amplifier OP2, for example. On the other hand, the second-type operational amplifier OP2 is an operational amplifier with lower flicker noise at a desired signal frequency (for example, the maximum frequency in the frequency band of the desired signal) than the first-type operational amplifier OP1.

  As described above, the relationship of f1−fcr <fcr−f2 is established in the first type operational amplifier OP1, and the relationship of fcr−f2 <f1−fcr is established in the second type operational amplifier OP2. Furthermore, the relationship of W1b × L1b> W1a × L1a and Ia> Ib is established. The synchronous detection circuit 100 can include, for example, a third-type operational amplifier OP3. In this case, the relationship of W1c × L1c> W1a × L1a and Ic> Ib is established. Alternatively, the relationship of W1c × L1c> W1b × L1b and Ic> Ia may be established.

  FIG. 18 shows a detailed configuration example of the detection circuit 60. 18 includes an amplifier circuit 70, a sensitivity adjustment circuit 80, a synchronous detection circuit 100, and a filter unit 110.

  The amplifier circuit 70 includes Q / V conversion circuits 72 and 74 and a differential amplifier circuit 76. The Q / V (I / V) conversion circuits 72 and 74 convert the electric charge (current) generated in the vibrator 10 into a voltage. The differential amplifier circuit 76 performs differential amplification of the output signals VS1P and VS1M from the Q / V conversion circuits 72 and 74.

  The Q / V conversion circuit 72 includes a capacitor CA1 and a resistor RA1 provided between the nodes NA1 and NA2, and an operational amplifier (operational amplifier) OPA1, and has a frequency characteristic of a low-pass filter. The operational amplifier OPA1 has an inverting input terminal (first input terminal) connected to the input node NA1, and a non-inverting input terminal (second input terminal) connected to AGND (reference power supply voltage). The Q / V conversion circuit 74 has the same configuration.

  The differential amplifier circuit 76 includes resistors RB1, RB2, RB3, RB4 and an operational amplifier OPB. By making the resistance ratio of RB1 and RB2 equal to the resistance ratio of RB3 and RB4, the differential amplifier circuit 76 performs differential amplification that amplifies the difference between the input signals VS1P and VS1M, which are opposite phase signals. By this differential amplification, unnecessary signals such as common mode noise and electrostatic coupling leakage input from the vibrator 10 to the Q / V conversion circuits 72 and 74 can be removed.

  In this embodiment, the Q / V conversion circuits 72 and 74 (the first and second charge / voltage conversion circuits or the first and second current / voltage conversion circuits) and the differential amplifier circuit 76 are shown in FIG. C) and the first-type operational amplifier OP1 described in FIG. That is, the first-type operational amplifier OP1 is used as the operational amplifiers OPA1, OPA2, and OPB in FIG.

  In this way, as the operational amplifiers OPA1, OPA2, and OPB of the Q / V conversion circuits 72 and 74 and the differential amplifier circuit 76 that amplify signals in a high frequency band (drive side resonance frequency band) in which thermal noise is dominant. Thus, the first-type operational amplifier OP1 that emphasizes thermal noise reduction is used. As a result, the thermal noise of the Q / V conversion circuits 72 and 74 and the differential amplifier circuit 76 can be minimized, and the SNR of the entire system can be improved.

  The sensitivity adjustment circuit 80 adjusts the sensitivity (change amount per unit angular velocity of the output voltage) by variably controlling the gain.

  In FIG. 18, the sensitivity adjustment circuit 80 is provided on the upstream side of the synchronous detection circuit 100. If the sensitivity adjustment circuit 80 (Programmable Gain Amp) is provided on the upstream side of the synchronous detection circuit 100 in this way, sensitivity adjustment is performed in the state of the signal of the frequency fd instead of the DC signal. Therefore, the adverse effect of flicker noise that becomes smaller as the frequency is higher can be minimized. In addition, since the number of circuit blocks on the front stage side of the sensitivity adjustment circuit 80 is reduced as compared with the method of providing the sensitivity adjustment circuit on the rear stage side of the filter unit 110, the sensitivity adjustment circuit 80 amplifies the noise of these circuit blocks. SNR degradation can be minimized.

  The sensitivity adjustment circuit 80 in FIG. 18 is an example of a non-inverting amplification type. Note that an inversion amplification type circuit may be used as the sensitivity adjustment circuit 80.

  In the sensitivity adjustment circuit 80, the resistance value of the variable resistor RD2 between the output node ND3 and the output tap QT and the resistance value of the variable resistor RD1 between the output tap QT and the node of AGND are based on the sensitivity adjustment data DPGA. Variablely controlled. Thereby, the gain of the sensitivity adjustment circuit 80 is adjusted, and sensitivity adjustment is performed. For example, assuming that the resistance values of the variable resistors RD1 and RD2 are R1 and R2, the gain of the sensitivity adjustment circuit 80 which is a PGA is G = (R1 + R2) / R1.

  Further, the sensitivity adjustment circuit 80 in FIG. 18 operates as a variable gain amplifier and also operates as, for example, a high-pass filter. Specifically, a high-pass active filter is configured by the capacitor CD1, the resistor RD3, and the operational amplifier OPD. That is, the operational amplifier OPD functions as a buffer for a high-pass filter including the capacitor CD1 and the resistor RD3. The variable resistors RD1 and RD2 and the operational amplifier OPD constitute a variable gain amplifier. That is, the operational amplifier OPD is shared by the high-pass active filter and the variable gain amplifier.

  If the sensitivity adjustment circuit 80 is operated as a high-pass filter, the DC component can be cut, and the situation where the DC signal is amplified by the variable gain amplifier (PGA) can be prevented. Therefore, it is possible to prevent a situation in which the variable gain amplifier of the sensitivity adjustment circuit 80 and the operational amplifier on the rear stage side (for example, the operational amplifier of the synchronous detection circuit) are saturated due to excessive input and the output overflows. In addition, DC noise can be removed by this high-pass filter, and the SNR can be improved.

  In the sensitivity adjustment circuit 80, the operational amplifier OPD is shared by the high-pass active filter and the variable gain amplifier. Therefore, the number of operational amplifiers can be reduced as compared with the case where an operational amplifier for an active filter and an operational amplifier for a variable gain amplifier are provided separately. Therefore, it is possible to reduce the circuit scale and reduce the number of circuit blocks serving as noise sources, thereby improving the SNR.

  In this embodiment, the sensitivity adjustment circuit 80 of FIG. 18 is configured by the first-type operational amplifier OP1. That is, the operational amplifier OPD shared by the high-pass filter and the variable gain amplifier in the sensitivity adjustment circuit 80 is configured by the first-type operational amplifier OP1.

  In this way, the first-type operational amplifier OP1 that emphasizes thermal noise reduction is used as the operational amplifier OPD of the sensitivity adjustment circuit 80 that handles signals in a high frequency band (driving side resonance frequency band) where thermal noise is dominant. It becomes like this. As a result, the thermal noise of the sensitivity adjustment circuit 80 can be minimized, and the SNR of the entire system can be improved.

  The synchronous detection circuit 100 performs synchronous detection based on the synchronous signal CLK from the drive circuit 40. The synchronous detection circuit 100 includes a switching element SE1 that is on / off controlled by a synchronization signal CLK and a switching element SE2 that is on / off controlled by an inverted synchronization signal CLKN, and performs synchronous detection by a single balance mixer system. A signal VS5 is input to the switching element SE1, and an inverted signal VS5N of the signal VS5 is input to the switching element SE2.

  FIG. 19 shows an example of a signal waveform for explaining the synchronous detection. As shown in FIG. 19, in the first period T1 in which the synchronization signal CLK is at the H level, the input signal VS5 is output to the output terminal as the signal VS6, and in the second period T2 in which the synchronization signal CLK is at the L level, An inverted signal VS5N of the input signal VS5 is output to the output terminal as the signal VS6. By this synchronous detection, a gyro output signal that is a desired signal can be detected and extracted. Note that a double balance mixer system may be employed as the synchronous detection circuit 100.

  In FIG. 18, the synchronous detection circuit 100 includes an offset adjustment circuit 90 (0-point adjustment circuit). The offset adjustment circuit 90 performs adjustment to remove the initial offset voltage (offset voltage) of the output signal VSQ of the detection device 30. For example, when the typical temperature is 25 ° C., offset adjustment processing is performed so that the voltage of the output signal VSQ matches the reference output voltage.

  The offset adjustment circuit 90 includes a D / A conversion circuit 92 and an addition circuit (addition / subtraction circuit) 94. The D / A conversion circuit 92 converts the initial offset adjustment data DDA into an analog initial offset adjustment voltage VA.

  The adder circuit 94 adds the adjustment voltage VA from the D / A conversion circuit 92 to the voltage of the input signal VS5. The adder circuit 94 includes resistors RE1, RE2, and RE3 provided between the nodes NE5 and NE6, NE1, and NE2, respectively. Also included is an operational amplifier OPE whose node NE5 is connected to its inverting input terminal and whose node AGND is connected to its non-inverting input terminal.

  In this embodiment, the synchronous detection circuit 100 is configured by the third-type operational amplifier OP3 described with reference to FIGS. 5B, 5C, and 12. That is, the third-type operational amplifier OP3 is used as the operational amplifier OPE of the offset adjustment circuit 90 of the synchronous detection circuit 100.

  For example, the synchronous detection circuit 100 functions as a mixer that is a frequency conversion circuit. Therefore, in the synchronous detection circuit 100, a low frequency signal and a high frequency signal are mixed. For example, the signals VS5 and VS5N are high frequency AC signals, while the adjustment voltage VA of the offset adjustment circuit 90 is a DC signal. Therefore, the operational amplifier OPE of the synchronous detection circuit 100 handles both a low frequency signal and a high frequency signal as amplification target signals. For this reason, it is necessary to consider both thermal noise and flicker noise in the operational amplifier OPE.

  In this regard, in the present embodiment, as the operational amplifier OPE, the third-type operational amplifier OP3 that has low noise in both thermal noise and flicker noise is used. Therefore, even when a low-frequency signal and a high-frequency signal are mixed as signals to be amplified, low-noise signal amplification can be realized and the SNR of the entire system can be improved.

  In FIG. 18, a modification in which the offset adjustment circuit 90 is not provided in the synchronous detection circuit 100 is also possible. In this case, the third-type operational amplifier OP3 may be used as an inverting amplifier for generating the inverted signal VS5N of the input signal VS5.

  Of the unnecessary signals described in FIGS. 17A to 17C, the unnecessary signal caused by the detuning frequency Δf = | fd−fs | is the signal of the detection-side resonance frequency fs as the gyro detection signal. Is generated when this detection signal is synchronously detected by the synchronous detection circuit 100. For example, in order to improve the response of the gyro sensor, the detection vibrator may be oscillated at a natural resonance frequency fs with a minute amplitude in an idling manner. Alternatively, an external vibration from outside the gyro sensor may be applied to the vibrator, so that the detection vibrator may vibrate at the natural resonance frequency fs. When the detection vibrator vibrates at the frequency fs in this way, a signal of the frequency fs is mixed into the signal VS5 input to the synchronous detection circuit 100. Since the synchronous detection circuit 100 performs synchronous detection based on the synchronous signal CLK having the frequency fd, an unnecessary signal having a detuning frequency Δf = | fd−fs | corresponding to the difference between the frequencies fd and fs is generated.

  Here, the detuning frequency Δf = | fd−fs | is sufficiently smaller than fd and fs. Accordingly, a steep attenuation characteristic as shown in FIG. 20 is necessary to remove the unnecessary signal of the component of the detuning frequency Δf. Therefore, there is a problem that it is difficult to remove such an unnecessary signal of the component of the detuning frequency Δf only with the continuous-time low-pass filter.

  In order to solve such a problem, in FIG. 18, an SCF (switched capacitor filter) 114 that is a discrete time filter is provided in the filter unit 110. The SCF 114 removes the component of the detuning frequency Δf = | fd−fs | corresponding to the difference between the drive-side resonance frequency fd and the detection-side resonance frequency fs of the vibrator, and removes the frequency component (DC component) of the desired signal. Has frequency characteristics to pass. The filter unit 110 includes a pre-filter (pre-filter) 112 provided on the front side of the SCF 114 and an output circuit 116 provided on the rear side of the SCF 114 and functioning as an output buffer and a post filter (post-filter). . These pre-filter 112 and output circuit 116 are continuous-time filters.

  As shown in FIG. 18, if the filter unit 110 is provided with an SCF 114 (discrete time filter in a broad sense), it is easy to realize a steep attenuation characteristic as shown in FIG. Therefore, even when the detuning frequency Δf is extremely smaller than the frequency fd, the unnecessary signal component in the frequency band of the detuning frequency Δf is reliably and easily removed without adversely affecting the desired signal in the passband. it can.

  As shown in FIG. 18, the SCF 114 includes switched capacitor circuits 210, 212, and 214, capacitors CG4, CG5, CG6, and CG7, and operational amplifiers OPG1 and OPG2. Note that the configuration of the SCF 114 is not limited to that shown in FIG. 18, and various known configurations can be used.

  When the SCF 114 is provided in the filter unit 110 as shown in FIG. 18, since the signal is sampled in discrete time in the SCF 114, aliasing that is a frequency folding phenomenon due to sampling occurs.

  In order to prevent such an adverse effect of aliasing, in FIG. 18, an anti-aliasing prefilter 112 (continuous time type filter in a broad sense) is provided on the front side of the SCF 114. That is, when the sampling frequency is fsp (= fd), the prefilter 112 has anti-aliasing frequency characteristics for removing frequency components equal to or higher than fsp / 2 (= fd / 2).

  In this case, the frequency band of the desired signal is, for example, fa0 or less as shown in FIG. 20, and the frequency is low. On the other hand, the sampling frequency fsp of the SCF 114 is about 50 to 500 times, for example, fa0, and the frequency is high. Therefore, if it is a general anti-aliasing prefilter, the attenuation characteristic is not so steep.

  However, it has been found that a sensor that handles minute signals, such as a gyro sensor, cannot remove unnecessary signals with a general anti-aliasing attenuation characteristic. That is, in the detection signal of the gyro sensor, the amplitude of the unnecessary signal is, for example, about 100 to 500 times the amplitude of the desired signal. Therefore, in the general anti-aliasing attenuation characteristic, the amplitude of the unnecessary signal becomes larger than the amplitude of the desired signal (DC component), and the SNR deteriorates due to the return to the DC component by sampling of the SCF 114. .

Therefore, the amplitude of the unnecessary signal that appears in the frequency band of frequency k × fd (k is a natural number) by the synchronous detection by the synchronous detection circuit 100 is set to be equal to the desired signal (minimum resolution). It is desirable to have frequency characteristics (filter characteristics, attenuation characteristics) that attenuate below the amplitude. The amplitude of the desired signal is an amplitude corresponding to the minimum resolution of the desired signal, and is an amplitude corresponding to dps (degree per second). The amplitude of the desired signal is the amplitude of the desired signal in the DC frequency region.

  In this way, even when an unnecessary signal having an amplitude of, for example, about 100 to 500 times the desired signal appears at the frequency k × fd, the frequency component of the unnecessary signal can be reliably removed by the pre-filter 112. .

  In the present embodiment, the filter unit 110 in FIG. 18 is configured by the second-type operational amplifier OP2 described in FIGS. 5B, 5C, and 11B. Specifically, the second-type operational amplifier OP2 is used as the operational amplifiers OPH, OPG1, OPG2, and OPI of the filter unit 110.

  That is, in FIG. 18, the SCF 114 having a steep attenuation characteristic is used to remove unnecessary signals due to the detuning frequency, and the operational amplifiers OPG1 and OPG2 are necessary to realize the SCF 114.

  In FIG. 18, the anti-aliasing prefilter 112 is also used as a filter for removing unnecessary signals appearing at the frequency k × fd by synchronous detection. For this reason, for example, a secondary active low-pass filter is used as the pre-filter 112, and an operational amplifier OPH is required to realize this active low-pass filter.

  Further, in FIG. 18, the output circuit 116 functions as an output buffer that performs impedance conversion of the output signal VSQ and also functions as a post filter of the SCF 114. In order to realize such an output buffer function and a post filter function, an operational amplifier OPI is required.

  As described above, the filter unit 110 is provided with a large number of operational amplifiers OPH, OPG1, OPG2, and OPI. When the noise level of these operational amplifiers is high, the SNR of the entire system is greatly deteriorated.

  In this regard, in the present embodiment, the second-type operational amplifier OP2 with low flicker noise is used as the operational amplifiers OPH, OPG1, OPG2, and OPI. As a result, even when such a large number of operational amplifiers are used, the flicker noise of the filter unit 110 can be minimized and the SNR of the entire system can be improved.

  In the second-type operational amplifier OP2, the bias current Ib flowing through the differential unit is smaller than the bias current Ia of the first-type operational amplifier OP1 (Ib <Ia). Therefore, even when a large number of operational amplifiers OPH, OPG1, OPG2, and OPI are provided in the filter unit 110 as shown in FIG. 18, an increase in power consumption of the entire system can be minimized.

5. Electronic Device FIG. 21 shows a configuration example of a gyro sensor 510 (sensor in a broad sense) including the detection device 30 of the present embodiment and an electronic device 500 including the gyro sensor 510. Note that the electronic device 500 and the gyro sensor 510 are not limited to the configuration in FIG. 21, and various modifications such as omitting some of the components or adding other components are possible. In addition, as the electronic device 500 of the present embodiment, various devices such as a digital camera, a video camera, a mobile phone, a car navigation system, a robot, a game machine, and a portable information terminal can be considered.

  Electronic device 500 includes a gyro sensor 510 and a processing unit 520. Further, a memory 530, an operation unit 540, and a display unit 550 can be included. A processing unit (CPU, MPU, etc.) 520 performs control of the gyro sensor 510 and the like and overall control of the electronic device 500. The processing unit 520 performs processing based on information (angular velocity information, physical quantity) detected by the gyro sensor 510. For example, processing for camera shake correction, posture control, GPS autonomous navigation, and the like is performed based on the detected angular velocity information. A memory (ROM, RAM, etc.) 530 stores a control program and various data, and functions as a work area and a data storage area. The operation unit 540 is for the user to operate the electronic device 500, and the display unit 550 displays various information to the user. According to the detection device 30 of the present embodiment, a small sensor can be adopted as the gyro sensor 510 incorporated in the electronic apparatus 500. Thereby, the electronic device 500 can be reduced in size and cost.

  Although the present embodiment has been described in detail as described above, it will be easily understood by those skilled in the art that many modifications can be made without departing from the novel matters and effects of the present invention. Accordingly, all such modifications are intended to be included in the scope of the present invention. For example, in the specification or drawings, terms (vibrators, gyro sensors, AGND, SCF, etc.) described at least once together with different terms (physical quantity transducers, sensors, analog reference voltages, discrete time filters, etc.) in a broader sense or the same meaning ) May be replaced by the different terms anywhere in the specification or drawings. Further, the configurations of the analog circuit, the reception device, the detection device, and the electronic device are not limited to those described in this embodiment, and various modifications can be made.

2 is a configuration example of an analog circuit of the present embodiment. 2 is a configuration example of a radio clock receiver. The other structural example of the receiver of a radio timepiece. 4A and 4B are explanatory diagrams of noise analysis of the operational amplifier. FIG. 5A to FIG. 5C are explanatory diagrams of the noise reduction method of this embodiment. 6A to 6C are explanatory diagrams of the relationship between the frequencies f1 and f2 and the corner frequency fcr. FIGS. 7A and 7B are explanatory diagrams of a method for setting the corner frequency fcr in the first-type and second-type operational amplifiers. 8A and 8B are explanatory diagrams of the relationship between the effective gate voltage and noise. FIG. 9A and FIG. 9B are explanatory diagrams of simulation results of this embodiment. The bar graph figure for comparing the noise level of this embodiment and a comparative example. 11A and 11B show layout examples of first-type and second-type operational amplifiers. A layout example of a third-type operational amplifier. The modification of this embodiment. 3 shows a detailed configuration example of a reference voltage supply circuit. 2 is a configuration example of a gyro sensor detection device. 16A and 16B are explanatory diagrams of the vibrator. FIGS. 17A to 17C are explanatory diagrams of a frequency spectrum. Detailed configuration of the detection circuit. The signal waveform example for demonstrating synchronous detection. Explanatory drawing of a detuning frequency. Configuration examples of electronic devices and gyro sensors.

Explanation of symbols

20 reference voltage supply circuit, 21 first supply circuit, 22 second supply circuit,
23 third supply circuit, 26 reference voltage generation circuit,
300 analog circuit, 310 first circuit, 320 second circuit, 322 mixer,
330 antenna, 332 amplifier circuit, 334 filter unit, 336 mixer,
338 amplifier circuit, 340 filter unit, 342 demodulator unit, 344 local oscillator circuit,
350 antenna, 352 amplifier circuit, 354 filter section, 356 mixer,
358 filter unit, 360 detection circuit, 362 demodulation unit, 364 local oscillation circuit,
500 electronic equipment, 510 gyro sensor, 520 processing unit, 530 memory,
540 operation unit, 550 display unit

Claims (12)

A first circuit having a first-type operational amplifier in which the frequency of the signal to be amplified is the first frequency;
And a second circuit having a second-type operational amplifier whose frequency to be amplified is a second frequency lower than the first frequency,
The channel width of the differential stage transistor in the differential section of the first-type operational amplifier is W1a, the channel length is L1a, the bias current flowing in the differential section of the first-type operational amplifier is Ia, and the second-type W1b × L1b> W1a when the channel width of the differential stage transistor of the differential section of the operational amplifier is W1b, the channel length is L1b, and the bias current flowing through the differential section of the second-type operational amplifier is Ib. × L1a, Ia> Ib,
When the first frequency is f1, the second frequency is f2, and the corner frequency of flicker noise and thermal noise in the frequency-noise characteristic is fcr, in the first-type operational amplifier, f1−fcr < fcr−f2, and in the second-type operational amplifier, fcr−f2 <f1−fcr ,
An analog circuit characterized in that the bias current Ia is set to 2 to 10 times the bias current Ib, and the WL product W1b × L1b is set to 10 to 100 times the WL product W1a × L1a . In claim 1,
Including a mixer for mixing a signal of a given frequency with a signal including a desired signal;
The first circuit is a circuit provided on the front side of the mixer,
2. The analog circuit according to claim 1, wherein the second circuit is a circuit provided on a subsequent stage side of the mixer. In claim 2,
The first circuit includes:
A first amplifier circuit for amplifying an input signal;
The analog circuit, wherein the first amplifier circuit is constituted by the first-type operational amplifier. In claim 2 or 3,
The second circuit includes:
A second amplifying circuit that amplifies the signal after mixing from the mixer or a filter unit that performs filtering of the signal after mixing;
The analog circuit, wherein the second amplifier circuit or the filter unit is constituted by the second-type operational amplifier. In any one of Claims 1 thru | or 4,
An analog circuit characterized in that L1b <L3b, where L3b is the channel length of the active load stage transistor in the differential section of the second-type operational amplifier. In any one of Claims 1 thru | or 5,
When the WL ratio of the differential stage transistor of the first-type operational amplifier is RT1a and the WL ratio of the active load stage transistor of the first-type operational amplifier is RT3a, RT1a> RT3a. Analog circuit. In any one of Claims 1 thru | or 6.
Of the elements constituting the second type operational amplifier, the area of the arrangement region of the differential stage transistor is Sdf, and among the elements constituting the second type operational amplifier, the arrangement area of the elements other than the differential stage transistor An analog circuit, wherein Sdf> Sre when the area is Sre. In claim 7,
The differential stage transistors of the second-type operational amplifier are configured by J (J> 2) transistors connected in parallel, and the arrangement region of the differential stage transistors includes the J transistors connected in parallel. An analog circuit in which a transistor is arranged. In claim 8,
The active load stage transistor of the second-type operational amplifier is composed of I (J>I> 2) transistors connected in parallel,
The J transistors constituting the differential load stage transistor are arranged side by side along the X direction, and the I pieces constituting the active load stage transistor are arranged on the Y direction side of the J transistors. The analog circuit is characterized by being arranged side by side along the X direction. In any of claims 2 to 4,
The mixer includes a third-type operational amplifier,
When the channel width of the differential stage transistor in the differential section of the third-type operational amplifier is W1c, the channel length is L1c, and the bias current flowing in the differential section of the third-type operational amplifier is Ic, W1c An analog circuit characterized in that × L1c> W1a × L1a and Ic> Ib. In any one of Claims 1 thru | or 10.
The analog circuit, wherein the first frequency is a frequency of a carrier signal, and the second frequency is a frequency of a desired signal carried by the carrier signal. An analog circuit according to any one of claims 1 to 11 ,
A processing unit for performing processing based on information from the analog circuit;
An electronic device comprising:

Images