JP5261915B2 - Detection device, gyro sensor, electronic device, and detection device adjustment method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a detector or the like for materializing offset adjustment while preventing the scale of a circuit from enlarging. <P>SOLUTION: This detector comprises a drive circuit and a detection circuit while the detection circuit comprises an amplification circuit amplifying an output signal from a vibrator. The amplification circuit comprises a differential amplification circuit 76 performing differential amplification on a differential signal corresponding to the output signal, and an offset adjustment circuit 78 performing offset adjustment by variably controlling the resistance value of at least one resistance the amplification circuit 76 has. The offset adjustment is performed by amplifying at least one of first and second unnecessary signals superposed, respectively, on first and second signals constituting the differential signal. <P>COPYRIGHT: (C)2008,JPO&amp;INPIT

Description

  The present invention relates to a detection device, a gyro sensor, an electronic device, and a method for adjusting the detection device.

  Electronic devices such as a digital camera, a video camera, a mobile phone, and a car navigation system incorporate a gyro sensor (physical quantity transducer) for detecting a physical quantity that changes due to an external factor. Such a gyro sensor detects a physical quantity such as an angular velocity and is used for so-called camera shake correction, attitude control, GPS autonomous navigation, and the like.

  In recent years, the gyro sensor is required to be lighter and smaller and to have high detection accuracy, and a piezoelectric vibration gyro sensor has attracted attention as one of the gyro sensors. Among them, a quartz piezoelectric vibration gyro sensor using quartz as a piezoelectric material is expected as an optimum sensor for incorporation into many devices. This vibration gyro sensor detects a physical quantity corresponding to the Coriolis force generated by the rotation.

  In such a vibration gyro sensor, the output signal (output current) from the vibrator becomes a very weak signal as the vibrator becomes lighter and smaller. Therefore, the detection device that detects a desired signal (a signal corresponding to a physical quantity such as Coriolis force) based on such a weak output signal has a performance capable of detecting the desired signal with as much gain as possible without distortion and low noise. Is required.

  In the vibration gyro sensor, an offset adjustment for removing an offset voltage generated due to various causes is performed.

However, if a dedicated offset adjustment circuit for adjusting such an offset is provided, the scale of the circuit increases. In addition, in the dedicated offset adjustment circuit, an accurate minute voltage must be generated using a D / A conversion circuit or the like, which complicates the circuit. Furthermore, due to the dedicated offset adjustment circuit itself, there is a problem that noise increases and the S / N ratio deteriorates.
JP 2005-127978 A

  The present invention has been made in view of the technical problems as described above. The object of the present invention is to provide a detection device, a gyro sensor, an electronic device capable of realizing offset adjustment while minimizing the scale of the circuit. An object of the present invention is to provide a method for adjusting a device and a detection device.

  The present invention includes a drive circuit that drives a vibrator to excite the vibrator, and a detection circuit that receives an output signal from the vibrator and detects a desired signal, and the detection circuit outputs the output from the vibrator. An amplifier circuit for amplifying a signal, the amplifier circuit comprising: a differential amplifier circuit for performing differential amplification of a differential signal corresponding to the output signal; and a resistance value of at least one resistor included in the differential amplifier circuit. The present invention relates to a detection apparatus including an offset adjustment circuit that performs offset adjustment by variably controlling.

  According to the present invention, the offset adjustment is performed by the offset adjustment circuit variably controlling the resistance value of at least one resistor included in the differential amplifier circuit. In this way, for example, offset adjustment is realized by breaking the balance of the resistance values of the differential amplifier circuit, so that it is not necessary to provide a dedicated offset adjustment circuit. Therefore, the circuit can be reduced in size. Further, since the number of circuit blocks serving as noise sources is reduced, the S / N ratio can be improved.

  According to the present invention, the differential amplifier circuit includes a first resistor provided between a first input node to which a first signal constituting the differential signal is input and the first node, and the first resistor. A second resistor provided between the first node and the output node; and a third resistor provided between the second input node to which the second signal constituting the differential signal is input and the second node. , A fourth resistor provided between the second node and the first power supply voltage node, its inverting input terminal connected to the first node, and its non-inverting input terminal connected to the first node The offset adjustment circuit variably controls the resistance value of at least one of the first and third resistors, and is connected to the second node, the output terminal of which is connected to the output node. Offset adjustment may be performed.

  In this way, the gain on the first signal side and the gain on the second signal side can be controlled independently, and the offset adjustment can be simplified.

  In the present invention, the offset adjustment circuit performs offset adjustment by amplifying at least one of the first and second unnecessary signals superimposed on the first and second signals constituting the differential signal, respectively. You may make it perform.

  In this way, the first and second unnecessary signals superimposed on the first and second signals, respectively, can be effectively used to realize the offset adjustment, thereby reducing the circuit scale and the S / N ratio. Can be improved.

  The present invention also includes a sensitivity adjustment circuit that variably controls the gain to adjust the sensitivity, and the sensitivity adjustment circuit adjusts the gain of the detection device set by the offset adjustment by the offset adjustment circuit, thereby adjusting the sensitivity. Adjustments may be made.

  In this way, it is possible to realize sensitivity adjustment in which the gain changed by the offset adjustment by the offset adjustment circuit is reset by sensitivity adjustment and the sensitivity is matched with the reference value.

  In the present invention, a synchronous detection circuit that performs synchronous detection based on a reference signal may be included, and the sensitivity adjustment circuit may be provided on the upstream side of the synchronous detection circuit.

  In this way, the sensitivity adjustment is performed not in the DC signal but in the state of the signal of a given frequency, so that flicker noise can be reduced. Further, since the number of circuit blocks on the front stage side of the sensitivity adjustment circuit is reduced, it is possible to minimize the deterioration of the S / N ratio due to the sensitivity adjustment circuit amplifying the noise of these circuit blocks.

  In the present invention, the sensitivity adjustment circuit may operate as a variable gain amplifier and as a high-pass filter.

  In this way, the DC component can be cut by the high-pass filter, and the situation where the DC signal is amplified by the sensitivity adjustment circuit can be prevented. Therefore, it is possible to prevent a situation where the variable gain amplifier of the sensitivity adjustment circuit, the operational amplifier on the rear stage, and the like are in a saturated operation state. Further, since the number of circuit blocks can be reduced, the S / N ratio can be improved.

  In the present invention, an operational amplifier may be shared by the high-pass filter that is an active filter and the variable gain amplifier.

  In this way, since the number of operational amplifiers serving as noise sources can be reduced, the circuit can be reduced in size and the S / N ratio can be improved.

  Further, the present invention includes a synchronous detection circuit that performs synchronous detection based on a reference signal, and a filter unit that performs filtering of the signal after synchronous detection, and the offset adjustment circuit calculates an initial offset voltage of the output signal of the detection device. The offset adjustment is performed so that the filter unit includes a discrete-time filter and a continuous-time filter provided in front of the discrete-time filter, and the continuous-time filter is unnecessary due to environmental changes. You may have a frequency characteristic which removes the offset variation part of a signal.

  In this way, even when an environmental change such as a temperature fluctuation occurs after the offset adjustment, the offset fluctuation of the unnecessary signal due to the environmental change is removed by the continuous time filter. Therefore, even if a special circuit for removing the offset fluctuation is not provided, the offset fluctuation can be removed by effectively using the continuous time filter on the upstream side of the discrete time filter.

  In the present invention, the continuous-time filter attenuates an offset variation of an unnecessary signal that appears in a frequency band of frequency k × fd (k is a natural number) by the synchronous detection by the synchronous detection circuit below the amplitude of the desired signal. It may have frequency characteristics.

  In this way, the offset fluctuation that appears in the frequency band of the frequency k × fd by synchronous detection can be effectively removed by using the continuous time filter. Accordingly, it is possible to efficiently remove the offset fluctuation.

  In the present invention, the continuous-time filter is a first-order low-pass filter, wherein the amplitude of the desired signal is A0, the offset fluctuation of the unnecessary signal appearing at the frequency k × fd is ΔAk, and the filter at the frequency fd When the attenuation factor is a, the continuous-time filter may have a frequency characteristic that attenuates an offset variation of an unnecessary signal so that ΔAk × (a / k) ≦ A0.

  In this way, even when a first-order low-pass filter is used as the continuous-time filter, it is possible to easily realize a filter that can remove the offset variation of the frequency k × fd.

In the present invention, the continuous-time filter is a second-order low-pass filter, wherein the amplitude of the desired signal is A0, the offset fluctuation of the unnecessary signal appearing at the frequency k × fd is ΔAk, and the filter at the frequency fd When the attenuation factor is a, the continuous-time filter may have a frequency characteristic that attenuates an offset variation of an unnecessary signal so that ΔAk × (a / k ² ) ≦ A0.

  In this way, even when a secondary low-pass filter is used as the continuous-time filter, it is possible to easily realize a filter that can remove the offset variation of the frequency k × fd.

  In the present invention, the discrete-time filter removes a component of the detuning frequency Δf = | fd−fs | corresponding to the difference between the drive-side resonance frequency fd and the detection-side resonance frequency fs of the vibrator, thereby obtaining a desired signal. It may have a frequency characteristic that allows the frequency component to pass through.

  In this way, even when the detuning frequency Δf is sufficiently small with respect to the frequency fd, the unnecessary signal component of the detuning frequency Δf can be reliably and easily removed. Further, while the unnecessary signal having the detuning frequency Δf is removed by the discrete time filter, the offset fluctuation of the unnecessary signal appearing in the frequency band of k × fd can also be removed by the continuous time filter. Therefore, an unnecessary signal and its offset fluctuation can be efficiently removed.

  The present invention also relates to a gyro sensor including any one of the detection devices described above and the vibrator.

  The present invention also relates to an electronic device including the gyro sensor described above and a processing unit that performs processing based on angular velocity information detected by the gyro sensor.

  According to another aspect of the present invention, there is provided a method for adjusting a detection device according to any one of the above, wherein the output signal of the detection device is monitored and a resistance value of at least one of the resistors included in the differential amplifier circuit is controlled. The offset adjustment of the detection device is performed, the output signal of the detection device is monitored after the offset adjustment, and the sensitivity of the detection device matches the reference value based on the output signal of the detection device monitored after the offset adjustment. This relates to the adjustment method for adjusting the sensitivity.

  In this way, the gain changed by the offset adjustment can be set again by the sensitivity adjustment, and the sensitivity can be matched with the reference value.

  Hereinafter, preferred embodiments of the present invention will be described in detail. The present embodiment described below does not unduly limit the contents of the present invention described in the claims, and all the configurations described in the present embodiment are indispensable as means for solving the present invention. Not necessarily.

1. FIG. 1 shows a configuration example of a gyro sensor 510 including a detection device 30 of the present embodiment and an electronic apparatus 500 including a gyro sensor 510. The electronic device 500 and the gyro sensor 510 are not limited to the configuration shown in FIG. 1, and various modifications such as omitting some of the components or adding other components are possible. In addition, as the electronic device 500 of the present embodiment, various devices such as a digital camera, a video camera, a mobile phone, a car navigation system, a robot, a game machine, and a portable information terminal can be considered.

  Electronic device 500 includes a gyro sensor 510 and a processing unit 520. Further, a memory 530, an operation unit 540, and a display unit 550 can be included. A processing unit (CPU, MPU, etc.) 520 performs control of the gyro sensor 510 and the like and overall control of the electronic device 500. The processing unit 520 performs processing based on angular velocity information (physical quantity) detected by the gyro sensor (physical quantity transducer) 510. For example, processing for camera shake correction, posture control, GPS autonomous navigation, and the like is performed based on the angular velocity information. A memory (ROM, RAM, etc.) 530 stores a control program and various data, and functions as a work area and a data storage area. The operation unit 540 is for the user to operate the electronic device 500, and the display unit 550 displays various information to the user.

  The gyro sensor 510 includes the vibrator 10 and the detection device 30. The vibrator 10 in FIG. 1 is a tuning fork type piezoelectric vibrator formed from a thin plate of a piezoelectric material such as quartz, and includes driving vibrators 11 and 12 and detection vibrators 16 and 17. The drive vibrators 11 and 12 are provided with drive terminals 2 and 4, and the detection vibrators 16 and 17 are provided with detection terminals 6 and 8.

  The drive circuit 40 included in the detection device 30 outputs a drive signal (drive voltage) to drive the vibrator 10 (physical quantity transducer in a broad sense) and receives a feedback signal from the vibrator 10. Thereby, the vibrator 10 is excited. The detection circuit 60 receives a detection signal (detection current, electric charge) from the vibrator 10 driven by the drive signal, and detects (extracts) a desired signal (Coriolis force signal) from the detection signal.

  Specifically, an alternating drive signal (drive voltage) from the drive circuit 40 is applied to the drive terminal 2 of the drive vibrator 11. Then, the driving vibrator 11 starts to vibrate due to the reverse voltage effect, and the driving vibrator 12 also starts to vibrate due to the tuning fork vibration. At this time, a current (charge) generated by the piezoelectric effect of the drive vibrator 12 is fed back from the drive terminal 4 to the drive circuit 40 as a feedback signal. As a result, an oscillation loop including the vibrator 10 is formed.

  When the drive vibrators 11 and 12 vibrate, the detection vibrators 16 and 17 vibrate at a vibration speed v in the direction shown in FIG. Then, a current (charge) generated by the piezoelectric effect of the detection vibrators 16 and 17 is output from the detection terminals 6 and 8 as a detection signal. Then, the detection circuit 60 receives the detection signal from the vibrator 10 and detects a desired signal (desired wave) that is a signal corresponding to the Coriolis force. That is, when the vibrator 10 (gyro sensor) rotates around the detection shaft 19, a Coriolis force Fc is generated in a direction orthogonal to the vibration direction of the vibration speed v. For example, when the angular velocity when rotating around the detection axis 19 is ω, the mass of the vibrator is m, and the vibration speed of the vibrator is v, the Coriolis force is expressed as Fc = 2 m · v · ω. Accordingly, the detection circuit 60 detects (extracts) a desired signal (sensor signal) that is a signal corresponding to the Coriolis force, whereby the rotational angular velocity ω of the gyro sensor (vibrator) can be obtained. By using the obtained angular velocity ω, the processing unit 520 can perform various processes for camera shake correction, posture control, GPS autonomous navigation, and the like.

  The vibrator 10 has a drive side resonance frequency fd and a detection side resonance frequency fs. Specifically, the natural resonance frequency of drive vibrators 11 and 12 (the natural resonance frequency of drive vibration mode) is fd, and the natural resonance frequency of detection vibrators 16 and 17 (the natural resonance frequency of detection vibration mode). Is fs. In this case, a constant frequency difference is provided between fd and fs so that the drive vibrators 11 and 12 and the detection vibrators 16 and 17 do not cause unnecessary resonance coupling. The detuning frequency Δf = | fd−fs |, which is this frequency difference, is set to a frequency that is sufficiently smaller than fd and fs.

  Although FIG. 1 shows an example in which the vibrator 10 is a tuning fork type, the vibrator 10 of the present embodiment is not limited to such a structure. For example, it may be T-shaped or double T-shaped. The piezoelectric material of the vibrator 10 may be other than quartz.

2. Detection Device FIG. 2 shows a configuration example of the detection device 30 of the present embodiment. The detection device 30 is not limited to the configuration in FIG. 2, and various modifications such as omitting some of the components or adding other components are possible.

  The detection device 30 includes a drive circuit 40 that drives the vibrator 10 to excite the vibrator, and a detection circuit 60 that receives an output signal (charge, current) from the vibrator 10 and detects a desired signal (desired wave). Including.

  The drive circuit (oscillation circuit) 40 includes an I / V conversion circuit 42 that converts current into voltage, an AGC (Automatic Gain Control) circuit 44 that performs automatic gain control, and a binarization circuit (comparator) 46. In the drive circuit 40, in order to keep the sensitivity of the gyro sensor constant, it is necessary to keep the amplitude of the drive voltage supplied to the vibrator 10 (drive vibrator) constant. Therefore, an AGC circuit 44 for automatically adjusting the gain is provided in the oscillation loop of the drive vibration system. Specifically, the AGC circuit 44 automatically adjusts the gain variably so that the amplitude of the input signal ID (vibration vibration speed v) becomes constant. The phase is adjusted so that the phase shift in the oscillation loop is 0 degree. At the time of oscillation startup, the gain of the oscillation loop is set to a gain larger than 1 in order to enable high-speed oscillation startup.

  The I / V conversion circuit 42 converts a current (charge) that is a signal ID from the vibrator 10 into a voltage, and outputs the voltage as a drive signal VD1. The I / V conversion circuit 42 can be realized by a capacitor, a resistor, and an operational amplifier.

  The AGC circuit 44 monitors the drive signal VD1 and controls the gain of the oscillation loop. The AGC circuit 44 includes a gain control amplifier (GCA) for controlling the oscillation amplitude in the oscillation loop and a gain control circuit for outputting a control voltage for adjusting the gain of the gain control amplifier in accordance with the oscillation amplitude. be able to. The gain control circuit also includes a rectifier circuit (full-wave rectifier circuit) that converts the AC drive signal VD1 from the I / V converter circuit 42 into a DC signal, and the voltage of the DC signal from the rectifier circuit and the reference voltage. A circuit that outputs a control voltage corresponding to the difference can be included.

  The binarization circuit 46 binarizes the drive signal VD1 that is a sine wave, and outputs a reference signal (synchronization signal) RS obtained by the binarization process to the synchronous detection circuit 100. The reference signal RS is also output to the filter unit 110 (SCF 114). The binarization circuit 46 can be realized by a comparator that receives the sine wave (alternating current) signal VD1 from the I / V conversion circuit 42 and outputs a rectangular wave reference signal RS. Another circuit may be provided between the I / V conversion circuit 42 and the binarization circuit 46 or between the binarization circuit 46 and the synchronous detection circuit 100. For example, a high-pass filter or a phase shift circuit (phase shifter) may be provided.

  The detection circuit 60 includes an amplifier circuit 70, a sensitivity adjustment circuit 80, a synchronous detection circuit 100, and a filter unit 110. Note that some of these may be omitted.

  The amplifier circuit 70 amplifies the output signals ISP and ISM from the vibrator 10. The amplifier circuit 70 includes Q / V conversion circuits 72 and 74, a differential amplifier circuit 76, and an offset adjustment circuit 78.

  The Q / V conversion circuits 72 and 74 receive the signals ISP and ISM from the vibrator 10 and convert the charge (current) generated in the vibrator 10 into a voltage. The differential amplifier circuit 76 performs differential amplification of the signals VS1P and VS1M from the Q / V conversion circuits 72 and 74.

  FIG. 3A shows a configuration example of the Q / V (I / V) conversion circuits 72 and 74. The Q / V conversion circuits 72 and 74 include a feedback capacitor CA1 and a feedback resistor RA1 provided between nodes NA1 and NA2, and an operational amplifier (operational amplifier) OPA, and have frequency characteristics of a low-pass filter. An input node NA1 is connected to the inverting input terminal (−) of the operational amplifier OPA, and a node of the reference power supply voltage AGND (analog ground) is connected to the non-inverting input terminal (+).

  When the circuit of FIG. 3A is made to function as a Q / V conversion circuit, the capacitance value of CA1 and the resistance value of RA1 so that the cut-off frequency fc = 1 / 2πCR is sufficiently smaller than the resonance frequency fd. Set. As a result, the phase changes by about −90 degrees at the resonance frequency fd. On the other hand, when the circuit of FIG. 3A is made to function as an I / V conversion circuit, the capacitance value of CA1 and the value of RA1 are set so that the cutoff frequency fc = 1 / 2πCR is sufficiently higher than the resonance frequency fd. Set the resistance value. In this case, since the phase hardly changes, a phase shift circuit for changing the phase of the reference signal RS by +90 degrees or -90 degrees is necessary.

  The differential amplifier circuit 76 performs differential amplification of the differential signal corresponding to the output signals ISP and ISM from the vibrator 10. This differential signal is composed of a first signal (positive phase input signal) VS1P and a second signal (negative phase input signal) VS1M. By the differential amplification of the differential amplifier circuit 76, the electrostatic coupling leakage signal that is an unnecessary signal (interference signal) in phase with the sensor signal (desired signal) can be removed.

  The offset adjustment circuit 78 performs offset adjustment processing. Specifically, adjustment is performed to remove the initial offset voltage of the output signal VSQ of the detection device 30. For example, when the typical temperature is 25 ° C., offset adjustment processing is performed so that the voltage of the output signal VSQ matches the reference output voltage.

  More specifically, the offset adjustment circuit 78 performs offset adjustment by variably controlling the resistance value of at least one resistor included in the differential amplifier circuit 76. For example, offset adjustment is performed by variably controlling the resistance value of at least one of the first and third resistors which are resistors on the input side of the differential amplifier circuit 76 (resistors connected to the input node). In other words, at least one of the first and second unnecessary signals (unnecessary signals for electrostatic coupling leakage) superimposed on the first and second signals VS1P and VS1M constituting the differential signal is amplified. By doing so, offset adjustment is performed.

  The sensitivity adjustment circuit 80 performs sensitivity adjustment processing. Specifically, the sensitivity is adjusted by variably controlling the gain. In this case, in the present embodiment, the sensitivity adjustment circuit 80 performs sensitivity adjustment by adjusting (changing) the gain of the detection device 30 set by the offset adjustment by the offset adjustment circuit 78. The sensitivity adjustment circuit 80 amplifies a signal with a variable resistor whose resistance value is variably controlled based on sensitivity adjustment data, or a gain (amplification factor) determined by the resistance value (resistance ratio) of the variable resistor, for example. Operational amplifiers and the like. The sensitivity adjustment circuit 80 preferably operates as a variable gain amplifier and preferably operates as a high-pass filter. It is more desirable that the high-pass filter that is an active filter and the variable gain amplifier share an operational amplifier.

  The synchronous detection circuit (detection circuit, detector) 100 performs synchronous detection on the amplified signal VS5 based on the reference signal (reference clock) RS. By this synchronous detection, a mechanical vibration leakage signal that is an unnecessary signal having a phase difference of 90 degrees with respect to the sensor signal can be removed.

  FIG. 3B shows a configuration example of the synchronous detection circuit 100. The synchronous detection circuit 100 includes an inverting amplifier 102, a non-inverting amplifier 104, an inverter 106, and switching elements SE1 and SE2. One end of the switching element SE1 and one end of the switching element SE2 are connected to the node NE4 of the output signal Q (VS6). The input signal IN (VS5) is inverted and amplified by the inverting amplifier 102, and the signal IN1 having a phase opposite to that of the input signal IN is input to the other end of the switching element SE1. The input signal IN is amplified by the non-inverting amplifier 104, and a signal IN2 having the same phase as the input signal IN is input to the other end of the switch SE2. The switching element SE1 composed of MOS transistors is on / off controlled by an inverted signal RSX of the reference signal RS, and the switching element SE2 is on / off controlled by a reference signal RS. That is, synchronous detection is performed by alternately switching on the switching elements SE1 and SE2. Note that the function of the inverting amplifiers 102 and 104 may be replaced by an operational amplifier included in a circuit on the preceding stage of the synchronous detection circuit 100.

  The filter unit 110 performs a filtering process on the signal VS6 after synchronous detection. Specifically, low-pass filter processing for removing high frequency components is performed.

  FIG. 4 shows an example of a signal waveform for explaining the operation of the detection device 30. The drive signal VD1 is a sine wave whose frequency is the drive-side natural frequency fd. The drive signal VD1 is binarized by the binarization circuit 46, whereby a rectangular wave reference signal RS is obtained. The signal VS5 (sensor signal) input to the synchronous detection circuit 100 is amplitude-modulated (AM-modulated) according to the magnitude (angular velocity) of the Coriolis force. The signal VS5 is synchronously detected by the reference signal RS, and the obtained signal VS6 is smoothed by the filter unit 110, so that the DC component of the desired signal is output as the signal VSQ. That is, the voltage level of the signal VSQ becomes a voltage level corresponding to the magnitude of the Coriolis force, and the rotational angular velocity of the gyro sensor can be obtained by obtaining this voltage level.

3. Unnecessary signal The sensor signal includes a desired signal (desired wave) and an unnecessary signal (unnecessary wave). In the present embodiment, a combination of a desired signal and an unnecessary signal is referred to as a sensor signal.

  Since the amplitude of the unnecessary signal is generally much larger than the amplitude of the desired signal, the required performance for the detection device 30 is high. This unnecessary signal may be caused by mechanical vibration leakage, electrostatic coupling leakage, detuning frequency Δf, 2fd (2ωd), DC offset, or the like.

  The unnecessary signal of mechanical vibration leakage is generated due to an imbalance of the shape of the vibrator 10 or the like. Since the unnecessary signal of mechanical vibration leakage superimposed on the signal ISP and the unnecessary signal of mechanical vibration leakage superimposed on the signal ISM are in opposite phases, they cannot be removed by the differential amplifier circuit 76. However, the unnecessary signal of mechanical vibration leakage superimposed on the signal VS5 has a 90-degree phase difference from the desired signal in the signal VS5, and therefore can be removed by the synchronous detection circuit 100.

  The unnecessary signal of the electrostatic coupling leakage is generated when the drive signal VD2 in FIG. 2 leaks to the input terminals of the ISP and the ISM through the parasitic capacitances CP and CM. That is, capacitive coupling is formed by the parasitic capacitances CP and CM, and a signal whose phase is advanced by about 90 degrees (π / 2) from the drive signal VD2 near the drive signal frequency is an ISP as an unnecessary signal for electrostatic coupling leakage. , Superimposed on both IM. Since the unnecessary signal superimposed on the signal ISP and the unnecessary signal superimposed on the signal ISM are in phase with each other, the differential amplifier circuit 76 can remove them. However, since CP = CM is not always satisfied, when CP and CM are not equal, an unnecessary signal for electrostatic coupling leakage superimposed on the signal ISP and an unnecessary electrostatic coupling leakage superimposed on the signal ISM are unnecessary. The signal will vary in magnitude. Therefore, in this case, the unnecessary signal of the electrostatic coupling leakage remains without being removed.

  For example, a desired signal in ISP and ISM and an unnecessary signal of electrostatic coupling leakage have a phase relationship as shown in FIG. That is, the desired signal S1 (t) in the ISP and the desired signal S2 (t) in the ISM are in reverse phase (phase difference 180 degrees). On the other hand, the unnecessary signal N1 (t) for electrostatic coupling leakage in the ISP and the unnecessary signal N2 (t) for electrostatic coupling leakage in the ISM are in phase (in phase).

ISP = S1 (t) + N1 (t) (1)
ISM = S2 (t) + N2 (t) (2)
S1 (t) = − S2 (t) (3)
N1 (t) = N2 (t) (4)
Therefore, the relationship of the above equation (3) is established between S1 (t) and S2 (t), and the relationship of the above equation (4) is established between N1 (t) and N2 (t). As long as the ISP and ISM are input to the differential amplifier circuit 76 that functions as a subtractor, the output of the differential amplifier circuit 76 is
ISP-ISM = 2S1 (t) (5)
It becomes. Therefore, ideally, only the desired signal (2S1 (t)) whose amplitude is doubled remains at the output of the differential amplifier circuit 76, and unnecessary signals are removed.

However, in practice,
| S2 (t) | = | S1 (t) | + ΔS (t) (6)
| N2 (t) | = | N1 (t) | + ΔN (t) (7)
It has become. Here, | S1 (t) | and | S2 (t) | represented by absolute value symbols represent the amplitudes of the desired signals S1 (t) and S2 (t). Similarly, | N1 (t) | and | N2 (t) | represent the amplitudes of the unnecessary signals N1 (t) and N2 (t). ΔS (t) represents an unbalance component of the desired signal (amplitude difference of the desired signal in ISP and ISM), and ΔN (t) represents an unbalance component of the unnecessary signal (amplitude difference of the unwanted signal in ISP and ISM). Represents. Therefore, the output of the differential amplifier circuit 76 is
ISP-ISM = 2S1 (t) + ΔS (t) + ΔN (t) (8)
It becomes. Therefore, not only a desired signal but also an unnecessary signal (an unbalance component ΔN (t)) remains at the output of the differential amplifier circuit 76.

  The 2fd unnecessary signal is generated when the vibrator vibrates at a harmonic frequency of 2fd for some reason. DC offset unnecessary signals are caused by input leakage, electrostatic coupling leakage imbalance, phase shift between the desired signal and the reference signal, duty shift of the reference signal, DC offset of the circuit block, etc. Occur.

  Next, the removal of unnecessary signals will be described in detail using the frequency spectrum of FIGS. 6 (A) to 6 (C). FIG. 6A shows a frequency spectrum before synchronous detection. As shown in FIG. 6A, in the sensor signal before synchronous detection, there is a DC offset unnecessary signal in the DC frequency band. Further, an unnecessary signal and a desired signal of mechanical vibration leakage exist in the frequency band of fd.

  FIG. 6B shows a frequency spectrum after synchronous detection. The desired signal in the fd frequency band in FIG. 6A appears in the DC and 2fd frequency bands after synchronous detection, as shown in FIG. 6B. Further, an unnecessary signal (DC offset) in the DC frequency band in FIG. 6A appears in the fd frequency band after synchronous detection, as shown in FIG. 6B. Further, an unnecessary signal (mechanical vibration leakage) in the fd frequency band in FIG. 6A appears in the 2fd frequency band after synchronous detection, as shown in FIG. 6B. In FIG. 6A, when an unnecessary signal exists in the 2fd frequency band, it appears in the 3fd and fd frequency bands after synchronous detection. The mixed noise after detection is noise generated by a circuit subsequent to the synchronous detection circuit 100.

  FIG. 6C shows the frequency spectrum after the filter processing. By smoothing (LPF) the signal after the synchronous detection by the filter unit 110, the frequency components of the unnecessary signals in the frequency bands such as fd and 2fd are removed.

  Since the desired signal in VS5 in FIG. 2 is amplitude-modulated as described in FIG. 4, it can be expressed as A (t) sin (ωd × t). Further, an unnecessary signal (interference wave) of mechanical leakage vibration in VS5 is 90 degrees out of phase with the desired signal, and therefore can be expressed as Bsin (ωd × t + π / 2). Further, since the sensor signal at VS5 is the sum of the desired signal and the unnecessary signal at VS5, it can be expressed as A (t) sin (ωd × t) + Bsin (ωd × t + π / 2). The drive signal can be expressed as Csin (ωd × t). A (t), B, and C are amplitudes, and ωd = 2πfd.

The synchronous detection can be regarded as multiplication of the sensor signal (desired signal + unnecessary signal) and the drive signal (reference signal) at VS5. Therefore, for the desired signal among the sensor signals in VS5,
A (t) sin (ωd × t) × Csin (ωd × t)
= {(A (t) × C) / 2} × {1-cos (2ωd × t)}
It becomes. Therefore, as shown in FIG. 6B, the desired signal appears in the DC and 2fd frequency bands after synchronous detection.

On the other hand, regarding the unnecessary signal of mechanical vibration leakage among the sensor signals in VS5,
Bsin (ωd × t + π / 2) × Csin (ωd × t)
= {− (B × C) / 2} × cos (2ωd × t + π / 2)
It becomes. Therefore, as shown in FIG. 6B, the unnecessary signal of mechanical vibration leakage appears in the frequency band of 2fd + (2ωd) after synchronous detection and does not appear in DC.

  Next, synchronous detection will be described with reference to the schematic diagrams of FIGS. In practice, the amplitude B of the unnecessary signal (mechanical leakage vibration) is much larger than the amplitude A (t) of the desired signal, but for the convenience of the drawing, the amplitude A (t) is equal to B.

  As shown in FIG. 7A, when the phase of the desired signal at VS5 and the phase of the reference signal (drive signal) are completely aligned, the desired signal and the unnecessary signal at VS6 after synchronous detection are shown in FIG. It becomes like (B). That is, the desired signal has a complete full-wave rectified waveform, and the unnecessary signal has a waveform in which the areas of the positive part and the negative part are equal. Therefore, smoothing by the filter unit 110 causes the DC component of the desired signal to be output as the signal VSQ, and the component of the unnecessary signal does not appear as the signal VSQ.

  On the other hand, when the phase of the desired signal at VS5 and the phase of the reference signal (drive signal) are shifted by γ as shown in FIG. 7C, the desired signal and unnecessary signal at VS6 after synchronous detection are shown in FIG. 7 (D). That is, the desired signal is not a complete full-wave rectified waveform but includes a negative component. In addition, the area of the positive portion and the negative portion of the unnecessary signal is not equal. Therefore, in the signal VSQ obtained by the smoothing in the filter unit 110, the DC component of the desired signal is smaller than in the case of FIG. 7B, and the unnecessary signal component appears as the signal VSQ.

4). FIG. 8 shows examples of signal waveforms of the input signals (desired signals) VS1P and VS1M and the output signal VS4 of the differential amplifier circuit 76. The signals VS1P and VS1M constitute a differential signal. That is, since the signals ISP and ISM from the transducers 16 and 17 for detection are opposite in phase, the signals VS1P and VS1M from the Q / V conversion circuits 72 and 74 are also opposite in phase from each other by 180 degrees. To form a differential signal.

  As described above, the unnecessary signal of the electrostatic coupling leakage is generated by capacitive coupling by the parasitic capacitances CP and CM between the drive signal VD2 of FIG. 2 and the input terminals of the ISP and ISM. That is, the vibration component of the drive signal VD2 having the frequency fd is transmitted to the input terminals of the ISP and ISM via the parasitic capacitors CP and CM, and appears as an unnecessary signal of electrostatic coupling leakage as shown in FIG. For this reason, the unnecessary signal for electrostatic coupling leakage superimposed on the signal VS1P and the unnecessary signal for electrostatic coupling leakage superimposed on the signal VS1P are in-phase signals as shown in FIG. Therefore, when the signals VS1P and VS1M have the same amplitude of the electrostatic coupling leakage unnecessary signal, the differential amplification (subtraction) by the differential amplifier circuit 76 causes the output signal VS4 to have a static voltage as shown in FIG. Unnecessary signal of electric coupling leakage will not appear.

  As described above, the differential amplifier circuit 76 is generally used for differential amplification of the signals VS1P and VS1M and removal of unnecessary signals in phase such as electrostatic coupling leakage. Conventionally, offset adjustment for removing the initial offset voltage is performed by, for example, a dedicated offset adjustment circuit provided on the rear stage side of the filter unit 110.

  However, if such a dedicated offset adjustment circuit is provided, the circuit scale increases. Further, if an offset adjustment circuit is provided on the rear stage side of the filter unit 110, noise itself is amplified by the gain of the offset adjustment circuit. In addition, the number of circuit blocks serving as noise sources increases, and the S / N ratio deteriorates.

  Therefore, in the present embodiment, the differential amplifier circuit 76 has not only a function of differential amplification but also a function of offset adjustment, and the resistance value of the resistor of the differential amplifier circuit 76 is variably controlled. Offset adjustment is performed.

  For example, in FIG. 9A, when the environmental temperature is 25 ° C. (typical temperature) and the gyro sensor is stationary, the output voltage VQ of the detection device 30 matches the reference output voltage VR (for example, VDD / 2). To adjust the offset. That is, in FIG. 9A, the output voltage VQ does not match the reference output voltage VR. In this case, as shown in FIG. 9B, the offset adjustment is performed so that the initial offset voltage | VQ−VR | is removed and becomes zero. Specifically, after the gyro sensor is manufactured, the gyro sensor is brought into a stationary state, and the output voltage VQ of the detection device 30 is monitored. Then, offset adjustment data for making the output voltage VQ coincide with the reference output voltage VR is written in a non-illustrated nonvolatile memory or the like. Then, the offset adjustment is performed based on the adjustment data so that the output voltage VQ of the detection device 30 matches VR.

  FIG. 10 shows a configuration example of the differential amplifier circuit 76 and the offset adjustment circuit 78 of the present embodiment. The differential amplifier circuit 76 includes a first resistor RB1 provided between the first input node NB1 to which the first signal VS1P constituting the differential signal is input and the first node NB3, a node NB3, and an output. A second resistor RB2 provided between nodes NB5 is included. Further, the third resistor RB3 provided between the second input node NB2 and the second node NB4 to which the second signal VS1M constituting the differential signal is input, the node NB4 and the reference voltage AGND (in a broad sense). Includes a fourth resistor RB4 provided between the first power supply voltage nodes. Furthermore, an inverting input terminal (−) is connected to the node NB3, a non-inverting input terminal (+) is connected to the node NB4, and an output terminal is connected to the output node NB5.

  The offset adjustment circuit 78 variably controls the resistance value of at least one of the resistors RB1, RB2, RB3, and RB4 included in the differential amplifier circuit 76, so that the offset adjustment circuit 78 in FIGS. 9A and 9B is performed. Perform the offset adjustment described. Specifically, the offset adjustment circuit 78 receives offset adjustment data DDA [m: 0]. Based on the adjustment data DDA [m: 0], the offset adjustment is performed by variably controlling the resistance values of the resistors RB1 and RB3.

  In FIG. 10, the resistance values of both the resistors RB1 and RB3 are controlled, but only one resistance value of the resistors RB1 and RB3 may be controlled. Further, the configurations of the differential amplifier circuit 76 and the offset adjustment circuit 78 are not limited to those in FIG. 10, and modifications such as omitting some of the components in FIG. 10 or adding other components are possible.

In FIG. 10, the voltages of the signals VS1P, VS1M, and VS4 are V1, V2, and V3, respectively, and the resistance values of the resistors RB1, RB2, RB3, and RB4 are R1, R2, R3, and R4, respectively. Then, V3 is expressed as the following equation.
V3 = {R4 × (R1 + R2) × V2-R2 × (R3 + R4) × V1} / {R1 × (R3 + R4)}
= (K1 * V2-K2 * V1) / K3 (9)
here,
K1 = R4 × (R1 + R2) (10)
K2 = R2 × (R3 + R4) (11)
K3 = R1 × (R3 + R4) (12)
It is.

  In the conventional example, only the differential amplification function is provided to the differential amplifier circuit 76, and R1 = R3 and R2 = R4 are set. When R1 = R3 and R2 = R4 are set, K1 = K2 = R2 × (R1 + R2) and K3 = R1 × (R1 + R2). In the conventional example, V3 is expressed by the following equation.

V3 = K1 × (V2-V1) / K3
= (R2 / R1) x (V2-V1) (13)
When the gyro sensor is stationary, the voltages V1 and V2 of the desired signals VS1P and VS1M proportional to the angular velocity of the gyro sensor are both zero. Therefore, as shown in FIG. 8, when the amplitude of the unnecessary signal of the electrostatic coupling leakage on the VS1P side and the VS1M side is the same (when the unbalance component is 0), the differential amplifier circuit 76 when the gyro sensor is stationary. The output voltage V3 becomes zero.

  However, as described above, the unnecessary signal of electrostatic coupling leakage is generated by capacitive coupling due to parasitic capacitance. Therefore, as shown in FIG. 11 and FIG. 12, there is an unbalance component (ΔN (t)) in the unnecessary signal of the electrostatic coupling leakage between the VS1P side and the VS1M side, and the amplitude of the unnecessary signal on the VS1P side and the VS1M side is It will not be the same. If such an unbalanced component exists, even if differential amplification is performed by the differential amplifier circuit 76, an unnecessary signal of electrostatic coupling leakage remains in the output signal VS4 as shown in FIGS.

  In the present embodiment, the offset adjustment is realized by using the unbalanced component of the in-phase unnecessary signal in reverse.

  That is, as shown in FIG. 11 and FIG. 12, when an unnecessary signal of electrostatic coupling leakage remains in the output signal VS4, the unnecessary signal is a signal in phase with the desired signal or 180 degrees out of phase. It is not removed by detection. Since the unnecessary signal is in the same frequency fd band as the desired signal before synchronous detection (see FIG. 6A), after the synchronous detection, it appears in the DC frequency band in the same manner as the desired signal (see FIG. 6A). (See FIG. 6B). Therefore, the unnecessary signal is not removed even by the low-pass filter processing by the filter unit 110 (see FIG. 6C), and appears as an offset voltage (initial offset voltage) of the output signal VSQ.

  Here, the offset voltage includes, in addition to those caused by the electrostatic coupling leakage as described above, an input leakage, a phase shift existing between the desired signal and the reference signal, a duty shift of the reference signal, and a circuit block. Some of them are caused by DC offset.

  Therefore, in the present embodiment, the offset voltage generated by the unnecessary signal (input leakage, phase shift, etc.) other than the electrostatic coupling leakage is canceled by the offset voltage generated by the unnecessary signal of electrostatic coupling leakage. That is, as shown in FIGS. 11 and 12, offset adjustment is performed by amplifying at least one of unnecessary signals (first and second unnecessary signals in a broad sense) of electrostatic coupling leakage superimposed on the signals VS1P and VS1M, respectively. Is realized.

  For example, the unnecessary signal of electrostatic coupling leakage superimposed on the signal VS4 in FIG. 11 is in phase with the desired signal of the signal VS4 in FIG. Therefore, if the unnecessary signal (second unnecessary signal) for electrostatic coupling leakage superimposed on the signal VS1M in FIG. 11 is amplified, the unnecessary signal for electrostatic coupling leakage in the signal VS4 in FIG. 11 is amplified. The offset voltage of the output signal VSQ of the device 30 can be changed in the positive direction.

  On the other hand, the unnecessary signal of the electrostatic coupling leakage superimposed on the signal VS4 in FIG. 12 is in reverse phase (shifted by 180 degrees) from the desired signal of the signal VS4 in FIG. Accordingly, if the unnecessary signal (first unnecessary signal) of the electrostatic coupling leakage superimposed on the signal VS1P of FIG. 8 is amplified, the unnecessary signal of the electrostatic coupling leakage of the signal VS4 of FIG. 12 is amplified. The offset voltage of the output signal VSQ of the device 30 can be changed in the negative direction.

  Therefore, even when the offset voltage (initial offset voltage) in FIGS. 9A and 9B is positive or negative, the offset voltage generated by the unnecessary signal of electrostatic coupling leakage is effectively used. Thus, it is possible to realize offset adjustment by removing the offset voltage to zero.

  Next, the offset adjustment method of the present embodiment will be described in more detail with reference to FIGS. 13 (A) and 13 (B).

  V3 in FIG. 13A is expressed as V3 = (K1 × V2−K2 × V1) / K3 as in the above equation (9). When V3 is synchronously detected by the reference signal RS and further passed through the LPF of the filter unit 110, V4 is obtained. This V4 is a DC signal, and the sign of V4 is positive if the phase of V3 and the phase of the reference signal RS are in phase, and negative if it is in reverse phase (shifted 180 degrees). The magnitude of V4 is proportional to the amplitude of V3.

  For example, in case 1 of FIG. 13B, the unnecessary signals of the electrostatic coupling leakage of V1 and V2 are in phase, and the amplitude of V1 is smaller than the amplitude of V2. In this case, a positive DC voltage is output as V4. Therefore, when there is no unnecessary signal other than electrostatic coupling leakage, the offset voltage becomes positive.

  In case 2 of FIG. 13B, the unnecessary signals of the electrostatic coupling leakage of V1 and V2 are in phase, and the amplitude of V1 is larger than the amplitude of V2. In this case, a negative DC voltage is output as V4. Accordingly, when there is no unnecessary signal other than the electrostatic coupling leakage, the offset voltage becomes negative.

  Since the angular velocity = 0 in the stationary state, V1 and V2 do not include a desired signal (Coriolis force signal). In this state, by adjusting K1 and K2 so that V4 becomes 0V, the entire gyro module can be tuned so that the output voltage at rest becomes 0V.

  At this time, mechanical vibration leakage may be applied. If tuning is performed in a state where mechanical vibration leakage is applied, DC generated due to the phase of the reference signals RS and V3 being shifted from 90 degrees in the synchronous detection described with reference to FIGS. It is also possible to cancel the offset.

  When an angular velocity, that is, a Coriolis force signal that is a desired signal is applied to the gyro sensor adjusted in this way, the output signal corresponding to the angular velocity is correctly zero-adjusted according to the principle of superposition, and the output of the filter unit 110 As will be obtained.

  In this embodiment, offset adjustment is realized by controlling at least one of R1 and R3 among the resistance values R1 to R4 of the resistors RB1 to RB4.

  For example, as shown in the above equations (10) and (11), when K1 and K2 are controlled, the resistance values R1 and R3 can be regarded as independent variables. That is, when the resistance value R1 is changed, K1 = R4 × (R1 + R2) changes, while K2 = R2 × (R3 + R4) does not change. When the resistance value R3 is changed, K2 = R2 × (R3 + R4) is changed, while K1 = R4 × (R1 + R2) is not changed.

  It is assumed that the unnecessary signal of the electrostatic coupling leakage is in a state as shown in FIG. 11 and the offset voltage is negative. In this case, the resistance value R1 of the resistor RB1 in FIG. 10 is increased to increase K1 = R4 × (R1 + R2) of V3 = (K1 × V2−K2 × V1) / K3. This increases the amplitude of the unnecessary signal of the electrostatic coupling leakage of the output signal VS4 of FIG. 11 and increases the positive offset voltage component due to the electrostatic coupling leakage, so that the negative offset voltage can be changed in the positive direction. it can. This makes it possible to perform offset adjustment by removing the offset voltage to zero. Note that the resistance value R3 of the resistor RB3 may be decreased to decrease K2 = R2 × (R3 + R4) of V3 = (K1 × V2−K2 × V1) / K3. Even in this case, since the amplitude of the unnecessary signal of the electrostatic coupling leakage of the signal VS4 of FIG. 11 is increased, the negative offset voltage can be changed in the positive direction and removed.

  Further, it is assumed that the unnecessary signal of electrostatic coupling leakage is in a state as shown in FIG. 12, and the offset voltage is positive. In this case, the resistance value R3 of the resistor RB3 in FIG. 10 is increased, and K2 = R2 × (R3 + R4) of V3 = (K1 × V2−K2 × V1) / K3 is increased. By doing so, the amplitude of the unnecessary signal of the electrostatic coupling leakage of the output signal VS4 of FIG. 12 increases, and the negative offset voltage component due to the electrostatic coupling leakage increases, so that the positive offset voltage can be changed in the negative direction. it can. This makes it possible to perform offset adjustment by removing the offset voltage to zero. Note that the resistance value R1 of the resistor RB1 may be reduced to reduce K1 = R4 × (R1 + R2) of V3 = (K1 × V2−K2 × V1) / K3. Even in this case, since the amplitude of the unnecessary signal of the electrostatic coupling leakage of the signal VS4 in FIG. 12 is increased, the positive offset voltage can be changed in the negative direction and removed.

  FIG. 14 shows a configuration example of the offset adjustment circuit 78. FIG. 14 shows a configuration example for realizing variable control of the resistance value of the resistor RB1 of FIG. 10, but variable control of the resistance value of the resistor RB3 can also be realized by the same configuration. The offset adjustment circuit 78 of FIG. 14 includes a decoder 79 and transistors (for example, N-type transistors) TB0 to TBn. The decoder 79 receives the offset adjustment data DDA [m: 0] and decodes the DDA [m: 0]. Then, selection signals SL0 to SLn are output. The resistor RB1 is configured by resistors RBR and DR0 to DRn connected in series.

  Selection signals SL0 to SLn from the decoder 79 are input to the gates of the transistors TB0 to TBn, whereby the TB0 to TBn are on / off controlled. For example, when the selection signal SL0 becomes active and the transistor TB0 is turned on, the output tap TR0 of the resistor RB1 and the node NB3 are connected. Therefore, when the resistance value of the resistor RBR is RR and the resistance values of the resistors DR0 to DRn are ΔR, the resistance value of the resistor RB1 is set to R1 = RR.

  When the selection signal SL1 becomes active and the transistor TB1 is turned on, the output tap TR1 and the node NB3 are connected. Therefore, the resistance value of the resistor RB1 is set to R1 = RR + ΔR. Similarly, when the selection signal SL2 becomes active and the transistor TB2 is turned on, the output tap TR2 and the node NB3 are connected. Therefore, the resistance value of the resistor RB1 is set to R1 = RR + 2 × ΔR. As described above, according to the configuration of FIG. 14, the resistance value R1 of the resistor RB1 can be variably controlled based on the offset adjustment data DDA [m: 0].

5. Sensitivity Adjustment In the gyro sensor 510, sensitivity adjustment is performed to adjust the overall gain of the detection device 30 so that the sensitivity matches a given reference value. This sensitivity (V / degree / sec) is the amount of change per unit angular velocity of the output voltage VQ (VSQ voltage) in FIG. 15, and corresponds to the slope of the straight line of the output voltage VQ.

  Until now, from the viewpoint of ease of sensitivity adjustment and ease of understanding, a circuit for adjusting such sensitivity has been provided on the rear side of the filter unit 110 in FIG. That is, the sensitivity adjustment is generally performed after removing unnecessary signals and the like to obtain a DC (direct current) signal.

  However, it has been found that if the sensitivity adjustment circuit is provided on the rear side of the filter unit 110, system noise increases. That is, when the sensitivity adjustment is performed after the low-pass filter processing of the filter unit 110 becomes a DC signal, noise generated by the sensitivity adjustment circuit itself appears in the output voltage VQ. In addition, flicker noise (1 / f noise) increases as the frequency decreases. Therefore, if sensitivity adjustment is performed in a DC signal state, the adverse effect of flicker noise also increases. Further, if a sensitivity adjustment circuit for performing gain adjustment is provided on the rear stage side, not only the desired signal but also noise generated by the circuit on the front stage of the sensitivity adjustment circuit is amplified, and the S / N ratio deteriorates.

  Therefore, in FIG. 2, the sensitivity adjustment circuit 80 is provided on the upstream side of the synchronous detection circuit (detector) 100. Specifically, a sensitivity adjustment circuit 80 is provided between the amplifier circuit 70 and the synchronous detection circuit 100 (offset adjustment circuit).

  If the sensitivity adjustment circuit 80 is provided on the upstream side of the synchronous detection circuit 100 in this way, sensitivity adjustment is performed in the state of a signal of frequency fd instead of a DC signal. Therefore, the adverse effect of flicker noise (1 / f noise) that decreases as the frequency increases can be minimized. Noise generated in the sensitivity adjustment circuit 80 itself appears in the frequency band fd by synchronous detection as shown in FIG. 6B and can be removed by the filter unit 110. Therefore, the adverse effect of noise generated in the sensitivity adjustment circuit 80 itself can be reduced. Furthermore, since the number of circuit blocks on the front stage side of the sensitivity adjustment circuit 80 is reduced compared to the case where the sensitivity adjustment circuit is provided on the rear stage side of the filter unit 110, the sensitivity adjustment circuit 80 amplifies the noise of these circuit blocks. Degradation of the S / N ratio due to can be minimized.

6). Configuration of Sensitivity Adjustment Circuit FIGS. 16A and 16B show a configuration example of the sensitivity adjustment circuit 80 (Programmable Gain Amp). In FIGS. 16A and 16B, the sensitivity adjustment circuit 80 operates as a variable gain amplifier (PGA) and as a high-pass filter. An operational amplifier is shared by the high-pass filter, which is an active filter, and the variable gain amplifier.

  FIG. 16A shows a non-inverting amplification type example. A sensitivity adjustment circuit 80 in FIG. 16A includes a capacitor CD1 provided between an input node ND8 and a node ND9 (first node), and a node of the node ND9 and a reference power supply voltage AGND (first power supply voltage). Including a resistor RD5. Variable resistors RD6 and RD7 provided between output node ND11 and the node of AGND are also included. Further, the node ND9 is connected to the non-inverting input terminal, the output tap QT (node ND10) of the variable resistors RD6 and RD7 is connected to the inverting input terminal, and the operational amplifier OPD3 whose output node is connected to the output node ND11 is connected. Including.

  In FIG. 16A, the resistance value of the variable resistor RD7 between the output node ND11 and the output tap QT and the resistance value of the variable resistor RD6 between the node of the output tap QT and AGND are the sensitivity adjustment data DPGA [m : 0]. Thereby, the gain of the sensitivity adjustment circuit 80 is adjusted, and sensitivity adjustment is performed.

  For example, when the resistance values of the variable resistors RD6 and RD7 are R6 and R7, the gain of the sensitivity adjustment circuit 80 which is PGA is G = (R6 + R7) / R6. Specifically, a plurality of output taps are provided for the variable resistor, and the variable resistor RD6 is selected by selecting the output tap QT corresponding to the adjustment data DPGA [m: 0] from the plurality of output taps. , The resistance values R6 and R7 of RD7 are determined, and the gain G = (R6 + R7) / R6 is determined.

  FIG. 16B shows an example of an inverting amplification type. The sensitivity adjustment circuit 80 in FIG. 16B is provided between the resistor RD8 provided between the input node ND12 and the node ND13 (first node), and between the node ND13 and the node ND14 (second node). The capacitor CD2 is included. Further, it includes a variable resistor RD9 which is provided between the output node ND15 and the node ND14 and whose resistance value is variably controlled based on the adjustment data DPGA [m: 0]. Further, it includes an operational amplifier OPD4 whose node ND14 is connected to its inverting input terminal and whose node of the power supply voltage AGND is connected to its non-inverting input terminal.

  In FIG. 16B, when the resistance values of the variable resistors RD8 and RD9 are R8 and R9, the gain of the sensitivity adjustment circuit 80 is G = −R9 / R8. Note that it is possible to modify the components shown in FIGS. 16A and 16B or add other components. For example, the capacitors CD1 and CD2 may be omitted, and the sensitivity adjustment circuit 80 may not have a high-pass filter function.

  The sensitivity adjustment using the sensitivity adjustment circuit 80 of FIGS. 16A and 16B is specifically realized as follows. First, after the manufacture of the gyro sensor, the output voltage VQ of the detection device 30 is monitored. Then, for example, the gyro sensor is rotated from a stationary state at a given rotational angular velocity, and the sensitivity that is the amount of change in the output voltage VQ at that time (straight line in FIG. 15) is obtained. Then, adjustment data DPGA [m: 0] for making the obtained sensitivity coincide with the reference value (reference sensitivity) is written in a non-illustrated nonvolatile memory or the like. Then, the sensitivity adjustment circuit 80 adjusts the gain of the operational amplifier based on the adjustment data DPGA [m: 0] so that the sensitivity of the detection device 30 matches the reference value.

  Here, the sensitivity adjustment circuit 80 in FIGS. 16A and 16B operates as a variable gain amplifier and as a high-pass filter. If the sensitivity adjustment circuit 80 is operated as a high-pass filter, the DC component can be cut, and the situation where the DC signal is amplified by the variable gain amplifier (PGA) can be prevented. Therefore, it is possible to prevent the variable gain amplifier of the sensitivity adjustment circuit 80 and the operational amplifier on the rear stage (for example, the offset adjustment circuit and the synchronous detection circuit operational amplifier) from being saturated due to excessive input and causing the output to overflow. . Further, DC noise can be removed by this high-pass filter, and the S / N ratio can be improved.

  In FIGS. 16A and 16B, an operational amplifier is shared by the high-pass active filter and the variable gain amplifier. Therefore, the number of operational amplifiers can be reduced as compared with the case where an operational amplifier for an active filter and an operational amplifier for a variable gain amplifier are provided separately. Therefore, the circuit can be reduced in size and the number of circuit blocks that become noise sources is reduced, so that the S / N ratio can be improved.

  The circuit of FIG. 16A has an advantage that the cutoff frequency and the gain (resistance value of the variable resistor) can be independently controlled as compared with FIG. 16B. For example, in a high-pass filter, in order to cut a DC component and pass a desired signal, it is desirable to make the cut-off frequency fc as low as possible. Here, when the capacitance value is C and the resistance value is R, the cutoff frequency is expressed as fc = 1 / 2πCR. When the capacitance value C2 of CD2 in FIG. 16B is increased, the circuit area is increased, so that the resistance value R8 of the resistor RD8 needs to be increased in order to reduce the cut-off frequency fc.

  However, when the resistance value R8 is increased in this way, noise (thermal noise or the like) generated at the resistor RD8 increases, and the S / N ratio decreases.

  On the other hand, in FIG. 16A, the cut-off frequency fc is determined by the capacitance value C1 of the capacitor CD1 and the resistance value R5 of the resistor RD5. On the other hand, the gain of the variable gain amplifier is G = (R6 + R7) / R6, and is determined by the resistance values R6 and R7 of the variable resistors RD6 and RD7. That is, the cutoff frequency fc and the gain can be adjusted separately. Therefore, even when the resistance value R5 of RD5 is increased in order to lower the cut-off frequency fc, it is not necessary to increase the resistance values R6 and R7 of RD6 and RD7. Accordingly, since the resistance values R6 and R7 of RD6 and RD7 can be made sufficiently smaller than the resistance value R9 of RD9 in FIG. 16B, the noise generated by these variable resistors RD6 and RD7 can be reduced, and S / The N ratio can be improved.

7). Method for Adjusting Detection Device Next, a method for adjusting the detection device 30 of this embodiment will be described with reference to the flowchart of FIG.

  In FIG. 17, offset adjustment processing is first performed (step S1). That is, the output voltage VQ of the detection device 30 is monitored (step S2), and it is determined whether VQ is equal to or higher than the reference output voltage VR (step S3).

  When the output voltage VQ is equal to or higher than the reference output voltage VR and the offset voltage is positive, the resistance value R1 of RB1 or the resistance value R3 of RB3 in FIG. 10 is set by offset adjustment data so that VQ changes to a small value. Adjust (step S4). For example, in the state shown in FIG. 12, the resistance value R3 is increased, and K2 = R2 × (R3 + R4) of V3 = (K1 × V2−K2 × V1) / K3 is increased. Thereby, the positive offset voltage can be changed in the negative direction. Then, it is determined whether or not the output voltage VQ matches the reference output voltage VR (step S5). If not, the process returns to step S4 and adjustment is performed again.

  On the other hand, when the output voltage VQ is smaller than the reference output voltage VR and the offset voltage is negative, the resistance value R1 or R3 is adjusted by the offset adjustment data so that VQ changes to a large value (step S6). For example, in the state as shown in FIG. 11, the resistance value R1 is increased, and K1 = R4 × (R1 + R2) of V3 = (K1 × V2−K2 × V1) / K3 is increased. Thereby, the negative offset voltage can be changed in the positive direction. Then, it is determined whether or not the output voltage VQ matches the reference output voltage VR (step S7). If not, the process returns to step S6 and adjustment is performed again.

  Then, the finally determined offset adjustment data is written into the nonvolatile memory (step S8), and the offset adjustment process is terminated.

  Next, sensitivity adjustment processing is performed (step S9). That is, the output voltage VQ of the detection device 30 is monitored again (step S10). Then, the sensitivity adjustment data is adjusted so that the sensitivity matches the reference value (step S11). Then, the sensitivity adjustment data finally determined is written in the nonvolatile memory (step S12), and the sensitivity adjustment process is terminated.

  As described above, in this embodiment, the output signal of the detection device 30 is monitored, the resistance value of at least one resistor included in the differential amplifier circuit 76 is controlled, and the offset adjustment of the detection device 30 is performed (steps S1 to S8). ). Then, after the offset adjustment, the output signal of the detection device 30 is monitored again, and based on the output signal of the detection device 30 monitored after the offset adjustment, sensitivity adjustment is performed so that the sensitivity of the detection device 30 matches the reference value. (Steps S9 to S12).

  That is, as described above, the output voltage of the differential amplifier circuit 76 is expressed as V3 = (K1 × V2−K2 × V1) / K3. In this embodiment, by controlling the resistance values R1 and R3, the offset adjustment is realized by changing K1 = R4 × (R1 + R2) and K2 = R2 × (R3 + R4).

  However, when the resistance values R1 and R3 are changed, K3 = R1 × (R3 + R4) also changes, so that the gain of the differential amplifier circuit 76 also changes, and the overall gain of the detection device 30 also changes.

  Therefore, in this embodiment, as shown in FIG. 17, first, offset adjustment is performed (steps S1 to S8). Next, sensitivity adjustment is performed to adjust the gain of the detection device 30 set (changed) by this offset adjustment again (steps S9 to S12). Thus, even when the gain of the detection device 30 changes due to the offset adjustment, the gain can be adjusted again by the sensitivity adjustment, and the sensitivity of the detection device 30 can be set to an appropriate reference value. . Note that the adjustment method of the present embodiment is not limited to FIG. For example, sensitivity adjustment may be performed first and then offset adjustment may be performed.

8). As shown in FIG. 9A, when offset adjustment is performed so that the initial offset voltage becomes 0, at least the environmental temperature is 25 ° C., and the gyro sensor is detected at rest when the angular velocity is 0. The output voltage (zero point voltage) VQ of the device 30 becomes equal to the reference output voltage VR.

  However, as shown in FIG. 9B, the offset voltage has a positive or negative temperature characteristic (temperature drift). Therefore, even if the initial offset voltage is removed to zero, there is a problem that the offset fluctuation due to temperature change (environmental change in a broad sense) does not become zero.

  For example, even if the initial offset voltage is removed, the DC offset of FIG. 6A and the offset fluctuation amount of the mechanical leakage vibration appear in the frequency bands of fd and 2fd after synchronous detection. If there is a temperature variation in the phase shift described in FIGS. 7C and 7D, the offset variation of the phase shift unnecessary signal appears in the DC frequency band. Therefore, in order to solve such a problem, the following method is adopted in the present embodiment.

  FIG. 18 shows a configuration example of the filter unit 110 of the present embodiment. The filter unit 110 includes an SCF (switched capacitor filter) 114 that is a discrete-time filter. The filter unit 110 includes a pre-filter (pre-filter) 112 provided on the front stage side of the SCF 114 (discrete time filter in a broad sense) and a post filter (post-filter) 116 provided on the rear stage side of the SCF 114. Including. These pre-filter 112 and post-filter 116 are continuous-time filters.

  In the present embodiment, the prefilter 112 (continuous time type filter in a broad sense) has a frequency characteristic (attenuating below the amplitude of the desired signal) that eliminates offset fluctuations of unnecessary signals due to environmental changes such as temperature changes and voltage changes. Frequency characteristics).

  That is, when the SCF 114 is provided in the filter unit 110 as shown in FIG. 18, since the signal is sampled at discrete times in the SCF 114, aliasing that is a frequency folding phenomenon due to sampling occurs. For example, when the sampling frequency is fsp, a harmonic frequency signal of fsp / 2 (= fd / 2) is folded back into the DC frequency region and the S / N ratio is deteriorated.

  In order to prevent such an adverse effect of aliasing, in FIG. 18, an anti-aliasing prefilter 112 is provided in front of the SCF 114. That is, when the sampling frequency is fsp (= fd), the prefilter 112 has anti-aliasing frequency characteristics for removing frequency components equal to or higher than fsp / 2 (= fd / 2).

  In this embodiment, paying attention to the presence of such an anti-aliasing pre-filter 112, the pre-filter 112 is effectively utilized. That is, the anti-aliasing pre-filter 112 is also used as a filter for removing an offset variation of an unnecessary signal due to an environmental change. In this way, it is not necessary to provide a separate filter for removing the offset variation of the unnecessary signal. Therefore, the circuit can be reduced in size and the number of circuit blocks that become noise sources is reduced, so that the S / N ratio can be improved.

  FIG. 19A to FIG. 19C show examples of frequency spectra after the initial offset voltage is removed as shown in FIG. 9B. If the offset voltage has a positive or negative temperature characteristic as shown in FIG. 9B, even if the initial offset voltage is removed, the offset fluctuation of the DC offset unnecessary signal as shown in FIG. Minutes appear in the DC frequency band. Further, the offset fluctuation of the unnecessary signal of mechanical vibration leakage appears in the fd frequency band.

  After synchronous detection, as shown in FIG. 19B, the offset fluctuation of the DC offset unnecessary signal appears in the fd frequency band, and the offset fluctuation of the mechanical vibration leakage unnecessary signal appears in the 2fd frequency band.

  That is, when the environmental temperature is 25 ° C., the offset adjustment by the offset adjustment circuit 90 removes unnecessary signal components in the DC and fd frequency bands of FIG. 19A so that they do not appear.

  However, when the environmental temperature is deviated from 25 ° C., offset fluctuations of these unnecessary signals due to temperature changes appear in the frequency bands of DC and fd. For example, when the amplitude of an unnecessary signal of mechanical vibration leakage increases due to temperature change, the increase appears in the fd frequency band before synchronous detection, and appears in the 2fd frequency band after synchronous detection. Such an offset variation of the unnecessary signal cannot be removed in the adjustment process after manufacturing the gyro sensor, but needs to be adjusted and removed dynamically during the operation of the gyro sensor.

  Further, when such an offset fluctuation appears in the DC frequency band after synchronous detection or is turned back to the DC frequency band by discrete sampling of the SCF 114, the sensitivity adjusted by the sensitivity adjustment circuit 80 also shifts. That is, it appears as a temperature variation of sensitivity.

  In this case, a method of incorporating a special temperature compensation circuit for removing such an offset variation in the detection device 30 is also conceivable.

  However, according to this method, the scale of the circuit is increased by the provision of the temperature compensation circuit. Further, since the number of circuit blocks serving as noise sources increases, the S / N ratio deteriorates.

  In this regard, in this embodiment, the pre-filter 112 necessary for anti-aliasing is effectively used from the beginning, and the offset fluctuations of these unnecessary signals are removed as shown in FIG. 19C. Therefore, the circuit can be reduced in size, and the number of circuit blocks serving as noise sources is reduced, so that the S / N ratio can be improved.

That is, the purpose of the normal anti-aliasing prefilter is as follows (A1).
(A1) Unnecessary signals such as random noise and pulse noise generated in the circuit are prevented from returning to the SCF passband.

On the other hand, in this embodiment, in addition to the role (A1), the prefilter 112 has the following role (A2).
(A2) The quality of the desired signal (Coriolis force signal) existing in DC is generated by synchronous detection, and the offset fluctuation of the unnecessary signal such as fd, 2fd, etc. always present in k × fd is turned back to DC by sampling at SCF114. Deteriorating (S / N ratio) is prevented.

The above (A2) is caused by the following conditions (B1) to (B3) peculiar to the gyro sensor.
(B1) The gyro sensor performs synchronous detection.
(B2) A strong spectrum corresponding to the offset fluctuation of the unnecessary signal appears at fd and 2fd by synchronous detection.
(B3) Since the sampling frequency of the SCF is fsp = fd, offset fluctuations of unnecessary signals such as fd and 2fd are turned back to DC where the desired signal exists.

  That is, the offset fluctuation (see FIG. 19B) of the unnecessary signal appearing at fd and 2fd after synchronous detection is larger than the amplitude of the desired signal.

  Also, as will be described later, in order to simplify the system configuration, it is desirable that the sampling frequency of the SCF 114 be fsp = fd. When fsp = fd, the offset variation of the unnecessary signals of fd and 2fd is folded back to DC exactly by sampling at the SCF 114.

  On the other hand, the desired signal (see FIG. 19A) present in fd before synchronous detection appears in DC by synchronous detection (see FIG. 19B). Therefore, unless any countermeasure is taken, the quality of the desired DC signal is extremely deteriorated due to aliasing of the offset fluctuation of the unnecessary signal existing in fd and 2fd. Specifically, when the offset fluctuation amount of the unnecessary signals of fd and 2fd is turned back and the offset fluctuation amount larger than the minimum resolution of the desired signal is superimposed on the DC, even if the gyro sensor is in a stationary state, it is as if the gyro sensor is stationary. It gives false information as if the sensor is rotating at a constant angular velocity.

  In order to solve such a problem, the present embodiment pays attention to the presence of the pre-filter 112 in the preceding stage of the SCF 114, and gives this pre-filter 112 not only the above (A1) but also the role (A2). ing.

9. Frequency characteristics of continuous-time filter In a sensor that handles a minute signal such as a gyro sensor, the amplitude of an unnecessary signal is much larger than the amplitude of a desired signal. Therefore, when the amplitude of an unnecessary signal such as mechanical vibration leakage fluctuates due to temperature variation, the offset variation that is the amplitude variation of the unnecessary signal also becomes very large compared to the amplitude (DC component) of the desired signal. Therefore, if the attenuation of the pre-filter 112 is not set appropriately, the S / N ratio may be deteriorated due to the return to the DC component of the offset fluctuation due to the sampling of the SCF 114.

  Therefore, in the present embodiment, the offset fluctuation amount of the unnecessary signal that appears in the frequency band of frequency k × fd (k is a natural number) by the synchronous detection by the synchronous detection circuit 100 is desired for the prefilter 112 that is a continuous-time filter. A frequency characteristic (filter characteristic, attenuation characteristic) that attenuates below the amplitude of the signal (minimum resolution) is provided. For example, it has a frequency characteristic that attenuates an offset variation (see FIG. 19B) of an unnecessary signal that appears in the frequency bands of the frequencies fd, 2fd, and 3fd to be equal to or less than the amplitude of the desired signal. Here, the amplitude of the desired signal is an amplitude corresponding to the minimum resolution of the desired signal, and is an amplitude value corresponding to an angular velocity (unit: dps: degree per second) that is a detection target of the detection apparatus of the present embodiment. The amplitude of the desired signal is the amplitude of the desired signal in the DC frequency region.

  In this way, even when a very large offset variation (amplitude variation of the unnecessary signal) appears in the frequency k × fd compared to the desired signal, the offset variation can be reliably removed by the pre-filter 112. . Therefore, it is possible to prevent the S / N ratio from deteriorating due to the return to the DC component of the offset fluctuation due to the sampling in the SCF 114. Therefore, it is possible to provide a detection device that is optimal for a gyro sensor that handles minute signals.

  FIG. 20 schematically shows the frequency characteristics of the prefilter 112. As indicated by D1 in FIG. 20, the prefilter 112 has a sufficient attenuation characteristic at fsp / 2 (= fd / 2). Therefore, it is possible to prevent the S / N ratio from deteriorating due to the return of random noise (thermal noise, 1 / f noise, etc.) due to sampling in the SCF 114, and the prefilter 112 serves as a normal anti-aliasing filter. Can be given.

  When the pre-filter 112 is a primary low-pass filter, the attenuation gradient is −20 dB / dec. The amplitude (minimum resolution) of the desired signal (DC component) is A0, the offset fluctuation of the unnecessary signal appearing at the frequency k × fd (k is a natural number) is ΔAk, and the filter attenuation rate at the frequency fd is a. And In this case, the prefilter 112 may have a frequency characteristic that attenuates the offset fluctuation of the unnecessary signal so that ΔAk × (a / k) ≦ A0.

  For example, in D2 of FIG. 20, the offset fluctuation amount of the unnecessary signal appearing at the frequency fd is ΔA1, and the attenuation factor (attenuation degree) of the filter at the frequency fd is a. Therefore, ΔA1 × a ≦ A0 is satisfied.

  Further, in D3 of FIG. 20, the offset fluctuation amount of the unnecessary signal appearing at the frequency 2fd is ΔA2, and the attenuation factor of the filter at the frequency 2fd is a / k = a / 2 because the prefilter 112 is first-order. is there. Therefore, ΔA2 × (a / 2) ≦ A0 is satisfied.

  Although not shown in FIG. 20, the offset fluctuation amount of the unnecessary signal appearing at the frequency 3fd is ΔA3, and the attenuation factor of the filter at the frequency 3fd is a / k = a because the prefilter 112 is first order. / 3. Therefore, ΔA3 × (a / 3) ≦ A0 is established.

  If the above conditions are satisfied, when the pre-filter 112 is a first-order low-pass filter, the offset fluctuation amount of the unnecessary signal that appears in the frequency bands of the frequencies fd, 2fd, and 3fd is less than the amplitude of the desired signal. Can be attenuated.

Note that a secondary low-pass filter may be used as the prefilter 112. When the pre-filter 112 is a secondary low-pass filter, the attenuation gradient is −40 dB / dec. Therefore, in this case, the prefilter 112 may have a frequency characteristic that attenuates the offset fluctuation of the unnecessary signal so that ΔAk × (a / k ² ) ≦ A0.

  For example, the offset fluctuation amount of the unnecessary signal appearing at the frequency fd is ΔA1, and the attenuation factor of the filter at the frequency fd is a, so that ΔA1 × a ≦ A0 is satisfied.

The offset fluctuation amount of the unnecessary signal appearing at the frequency 2fd is ΔA2, and the attenuation factor of the filter at the frequency 2fd is a / k ² = a / 4 because the prefilter 112 is second order. Therefore, ΔA2 × (a / 4) ≦ A0 is satisfied.

The offset fluctuation amount of the unnecessary signal appearing at the frequency 3fd is ΔA3, and the attenuation factor of the filter at the frequency 3fd is a / k ² = a / 9 because the prefilter 112 is second order. Therefore, ΔA3 × (a / 9) ≦ A0 is satisfied.

  If the above conditions are satisfied, when the pre-filter 112 is a second-order low-pass filter, the offset fluctuation amount of the unnecessary signal that appears in the frequency bands of the frequencies fd, 2fd, and 3fd is less than the amplitude of the desired signal. Can be attenuated.

  FIG. 21A shows a configuration example of the prefilter 112. FIG. 21A shows an example of a first-order low-pass filter. Prefilter 112 includes a resistor RI1 and a capacitor CI1 provided between nodes NI2 and NI3, and a resistor RI2 provided between nodes NI1 and NI2. Also included is an operational amplifier OPI having its inverting input terminal connected to the node NI2 and its non-inverting input terminal connected to the AGND node. The circuit of FIG. 21A can also be used as the post filter 116 of FIG.

  Note that a secondary low-pass filter may be used as the prefilter 112. FIG. 21B is an example of a secondary low-pass filter. The low-pass filter in FIG. 21B includes resistors RH1, RH2, and RH3 provided between the node NH2 and nodes NH4, NH3, and NH1, a capacitor CH1 provided between the nodes NH3 and NH4, and a node NH2. And a capacitor CH2 provided between the nodes of AGND. Also included is an operational amplifier OPH having its inverting input terminal connected to the node NH3 and its non-inverting input terminal connected to the AGND node.

  In the first-order low-pass filter of FIG. 21A, the cutoff frequency is fc = 1 / (2π × C1 × R1), where C1 is the capacitance value of the capacitor CI1 and R1 is the resistance value of the resistor RI1. The attenuation gradient is −20 dB / dec. Therefore, when a first-order low-pass filter is used, the attenuation at the frequencies fd and 2fd is reduced as shown by G2 and G3 by sufficiently reducing the cutoff frequency fc as shown by G1 in FIG. can do. Therefore, the offset fluctuation amount of the unnecessary signal as shown in G7 can also be attenuated, and even when this offset fluctuation amount is turned back to DC, it becomes possible to make it less than the minimum resolution of the desired signal.

In the secondary low-pass filter of FIG. 21B, when the capacitance values of the capacitors CH1 and CH2 are C1 and C2, and the resistance values of the resistors RH1 and RH2 are R1 and R2, the cutoff frequency is fc = 1 / {2π × (C1 × C2 × R1 × R2) ^1/2 }. The attenuation gradient is −40 dB / dec. As described above, the secondary low-pass filter has a large attenuation gradient although the number of elements increases. Therefore, sufficient attenuation can be obtained at the frequencies fd and 2fd as indicated by G5 and G6 without making the cut-off frequency fc as low as indicated by G4 in FIG. Therefore, the circuit can be made smaller than the first-order low-pass filter of FIG. Further, the offset fluctuation of the unnecessary signal can be sufficiently attenuated, and the offset fluctuation can be reliably removed.

10. Among the unnecessary signals of the detuning frequency, the unnecessary signal caused by the detuning frequency Δf = | fd−fs | is mixed with the signal of the detection side resonance frequency fs in the sensor signal, and this sensor signal is synchronously detected by the synchronous detection circuit 100. Is generated. For example, in order to improve the response of the gyro sensor, the detection vibrator may be oscillated at a natural resonance frequency fs with a minute amplitude in an idling manner. Alternatively, an external vibration from outside the gyro sensor may be applied to the vibrator, so that the detection vibrator may vibrate at the natural resonance frequency fs. When the detection vibrator vibrates at the frequency fs in this way, a signal of the frequency fs is mixed into the signal VS5 input to the synchronous detection circuit 100. Since the synchronous detection circuit 100 performs synchronous detection based on the reference signal RS having the frequency fd, an unnecessary signal having a detuning frequency Δf = | fd−fs | corresponding to the difference between the frequencies fd and fs is generated.

For example, the mixed frequency fs signal can be expressed as Dsin (ωs × t). Note that ωs = 2πfs. And since synchronous detection can be regarded as multiplication of a sensor signal and a drive signal (reference signal), about a signal of frequency fs among sensor signals,
Csin (ωd × t) × Dsin (ωs × t)
= {− (D × C) / 2} × [cos {(ωd + ωs) t} −cos {(ωd−ωs) t}]
It becomes. As is clear from the above equation, when a signal with a frequency fs is mixed, an unnecessary signal with a detuning frequency Δf = | fd−fs | is generated after synchronous detection.

  Here, the detuning frequency Δf = | fd−fs | is sufficiently smaller than fd and fs. Therefore, a steep attenuation characteristic as shown in FIG. 23 is required to remove the unnecessary signal of the component of the detuning frequency Δf. Accordingly, there is a problem that it is difficult to remove unnecessary signals of such a component of the detuning frequency Δf only with a conventional continuous-time low-pass filter.

  In order to solve such a problem, in FIG. 18, the filter unit 110 is provided with an SCF 114 that is a discrete time filter. The SCF 114 removes the component of the detuning frequency Δf = | fd−fs | corresponding to the difference between the drive-side resonance frequency fd and the detection-side resonance frequency fs of the vibrator, and removes the frequency component (DC component) of the desired signal. Has frequency characteristics to pass.

  As shown in FIG. 18, if the filter unit 110 is provided with an SCF 114 (discrete time filter in a broad sense), it is easy to realize a steep attenuation characteristic as shown in FIG. Therefore, even when the detuning frequency Δf is extremely smaller than the frequency fd, the unnecessary signal component in the frequency band of the detuning frequency Δf is reliably and easily removed without adversely affecting the desired signal in the passband. it can.

  In addition, the continuous-time filter has a disadvantage that if the capacitance value C of the capacitor and the resistance value R of the resistor vary, the frequency characteristic of the filter also varies, making it difficult to obtain a stable frequency characteristic. For example, the absolute values of C and R vary by about ± 20%, and the cut-off frequency of the continuous-time filter (RC filter) is determined by C × R. When the cutoff frequency varies, the amplitude of the desired signal in the pass band is attenuated or the phase is changed, thereby degrading the signal quality.

  On the other hand, in the SCF 114, the filter characteristics can be determined by the capacitance ratio and the sampling frequency (clock frequency). For example, since the accuracy of the capacitance ratio is 0.1% or less, there is little variation in the cut-off frequency. Therefore, according to the SCF 114, it is possible to easily realize a steep attenuation characteristic that reliably removes an unnecessary signal having the detuning frequency Δf while passing a desired signal in the pass band.

  In the present embodiment, the SCF 114 is operated based on a clock corresponding to the reference signal RS (the reference signal itself or a clock generated by the reference signal; a clock having the same frequency as the reference signal). Specifically, for example, a non-overlapping two-phase clock (sampling clock) is generated by the reference signal RS. Based on the generated clock, the switching elements included in the SCF 114 are turned on / off to operate the SCF 114.

  In this way, since the operation clock of the SCF 114 can be generated by effectively using the reference signal RS, the circuit can be reduced in scale. In addition, since the frequency fd of the reference signal RS and the sampling frequency (clock frequency) of the SCF 114 can be matched, the design of the frequency characteristics of the filter can be facilitated. In addition, when fd, which is the drive-side resonance frequency, varies due to an environmental change (temperature change) or a change with time, the sampling frequency of the SCF 114 also varies according to this variation. Therefore, the cutoff frequency of the SCF 114 can be changed and adjusted according to the change in the frequency fd. Therefore, even when an environmental change or a change with time occurs, an unnecessary signal having the detuning frequency fd can be reliably removed.

  As described above, in the present embodiment, unnecessary signals having the detuning frequency fd are removed by the SCF 114, and offset fluctuations of unnecessary signals appearing in fd, 2fd, 3fd, and the like by synchronous detection are used for anti-aliasing of the SCF 114. Each filter has a different role sharing such as removal by the pre-filter 112 provided in the filter. That is, an unnecessary signal that requires a steep attenuation characteristic such as an unnecessary signal due to a detuning frequency is removed by the SCF 114, and an offset variation of the unnecessary signal is removed by the pre-filter 112. By clarifying the division of roles in this way, unnecessary signals can be efficiently removed with a small circuit.

  Although the present embodiment has been described in detail as described above, it will be easily understood by those skilled in the art that many modifications can be made without departing from the novel matters and effects of the present invention. Accordingly, all such modifications are intended to be included in the scope of the present invention. For example, in the specification or drawings, terms (SCF, prefilter, temperature change, etc.) described at least once together with different terms (discrete time filter, continuous time filter, environmental change, etc.) having a broader meaning or the same meaning are The different terms can be used anywhere in the specification or drawings. Further, the structure of the vibrator, the configuration of the detection device, the gyro sensor, and the electronic device are not limited to those described in this embodiment, and various modifications can be made. It is also possible to realize the discrete time filter with a filter other than the SCF (for example, a digital filter). In addition, the detection circuit may not be provided with the synchronous detection circuit or the filter unit, and the detection processing of the signal from the vibrator may be realized by another configuration.

Configuration examples of electronic devices and gyro sensors. The structural example of a detection apparatus. 3A and 3B are configuration examples of a Q / V (I / V) conversion circuit and a synchronous detection circuit. The signal waveform example for demonstrating operation | movement of a detection apparatus. Explanatory drawing about electrostatic coupling leakage. 6A to 6C show examples of frequency spectra. 7A to 7D are explanatory diagrams of synchronous detection. The signal waveform example for demonstrating operation | movement of a differential amplifier circuit. 9A and 9B are explanatory diagrams of offset adjustment. 2 is a configuration example of a differential amplifier circuit and an offset adjustment circuit. Explanatory drawing of the offset adjustment method of this embodiment. Explanatory drawing of the offset adjustment method of this embodiment. 13A and 13B are explanatory diagrams of the offset adjustment method of the present embodiment. 2 is a configuration example of an offset adjustment circuit. Explanatory drawing of sensitivity adjustment. FIG. 16A and FIG. 16B are configuration examples of the sensitivity adjustment circuit. The flowchart for demonstrating the adjustment method of a detection apparatus. The structural example of a filter part. 19A to 19C show examples of frequency spectra. Explanatory drawing of the frequency characteristic of a pre filter. 21A and 21B show configuration examples of the prefilter. Explanatory drawing of the frequency characteristic at the time of using a primary and secondary pre-filter. Explanatory drawing of a detuning frequency.

Explanation of symbols

10 vibrator, 30 detector, 40 drive circuit, 42 I / V conversion circuit,
44 AGC circuit, 46 binarization circuit, 60 detection circuit, 70 amplification circuit,
72, 74 Q / V (I / V) conversion circuit, 76 differential amplifier circuit,
78 offset adjustment circuit, 79 decoder, 80 sensitivity adjustment circuit,
100 synchronous detection circuit, 110 filter unit, 112 pre-filter,
114 SCF, 116 post filter, 500 electronic device, 510 gyro sensor,
520 processing unit, 530 memory, 540 operation unit, 550 display unit

Claims (10)

A drive circuit for driving the vibrator to excite the vibrator;
A detection circuit that receives an output signal from the vibrator and detects a desired signal;
The detection circuit includes:
An amplification circuit for amplifying the output signal from the vibrator ;
A synchronous detection circuit that performs synchronous detection based on a reference signal;
Including a filter unit for filtering the signal after synchronous detection;
The amplifier circuit is
A differential amplifier circuit for performing differential amplification of a differential signal corresponding to the output signal;
The differential amplifier circuit that variably controlling the resistance value at least one resistance possessed, see contains an offset adjustment circuit for performing offset adjustment,
The offset adjustment circuit includes:
Perform offset adjustment to remove the initial offset voltage of the output signal of the detector,
The filter unit is
A discrete time filter;
A continuous-time filter provided on the front side of the discrete-time filter,
The discrete time filter is:
A frequency characteristic that removes the component of the detuning frequency Δf = | fd−fs | corresponding to the difference between the drive-side resonance frequency fd and the detection-side resonance frequency fs of the vibrator and allows the frequency component of the desired signal to pass;
The continuous time filter is:
It has a frequency characteristic that attenuates an offset variation of an unnecessary signal due to an environmental change, which appears in a frequency band of frequency k × fd (k is a natural number) by synchronous detection by the synchronous detection circuit, below the amplitude of the desired signal. Detection device. In claim 1 ,
Includes a sensitivity adjustment circuit that variably controls gain and adjusts sensitivity,
The sensitivity adjustment circuit includes:
A detection apparatus, wherein sensitivity adjustment is performed by adjusting a gain of the detection apparatus set by offset adjustment by the offset adjustment circuit. In claim 2 ,
The sensitivity adjustment circuit includes:
A detection apparatus provided on a preceding stage of the synchronous detection circuit. In claim 3 ,
The sensitivity adjustment circuit includes:
A detection apparatus that operates as a variable gain amplifier and operates as a high-pass filter. In claim 4 ,
An operational amplifier is shared by the high-pass filter that is an active filter and the variable gain amplifier. In any one of Claims 1 thru | or 5 ,
The continuous-time filter is a first-order low-pass filter;
When the amplitude of the desired signal is A0, the offset fluctuation amount of the unnecessary signal appearing at the frequency k × fd is ΔAk, and the attenuation factor of the filter at the frequency fd is a,
The continuous time filter is:
A detection apparatus having frequency characteristics for attenuating offset fluctuation of an unnecessary signal so that ΔAk × (a / k) ≦ A0 is satisfied. In any one of Claims 1 thru | or 5 ,
The continuous-time filter is a second-order low-pass filter;
When the amplitude of the desired signal is A0, the offset fluctuation amount of the unnecessary signal appearing at the frequency k × fd is ΔAk, and the attenuation factor of the filter at the frequency fd is a,
The continuous time filter is:
A detection apparatus having a frequency characteristic for attenuating an offset fluctuation amount of an unnecessary signal so that ΔAk × (a / k ² ) ≦ A0 is satisfied. A detection device according to any one of claims 1 to 7 ,
The vibrator;
A gyro sensor comprising: A gyro sensor according to claim 8 ,
A processing unit for performing processing based on angular velocity information detected by the gyro sensor;
An electronic device comprising: A method for adjusting a detection device according to any one of claims 1 to 7 ,
Monitoring the output signal of the detection device;
Controlling the resistance value of at least one of the resistors included in the differential amplifier circuit, and performing offset adjustment of the detection device;
Monitor the output signal of the detector after the offset adjustment;
An adjustment method comprising: performing sensitivity adjustment for matching the sensitivity of the detection device with a reference value based on an output signal of the detection device monitored after offset adjustment.

Images