JP2008172894A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply circuit in which the number of components is reduced to reduce ripple components and spike components contained in a DC output voltage. <P>SOLUTION: A primary side circuit includes a switching element Q1 with its switching frequency controlled by a control circuit 1 and an oscillation drive circuit 2, and a primary side parallel resonance circuit consisting of a leakage inductor L1 and a primary side parallel resonance capacitor C1. A secondary side circuit includes a secondary side parallel resonance circuit and a secondary side series resonance circuit consisting of a leakage inductor L2, a secondary side parallel resonance capacitor C3, and a secondary side series resonance capacitor C4, wherein the secondary side high speed diode Do1, an inductor Lo and a secondary side smoothing capacitor Co function as a step-down converter. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

  As an example of the step-up / step-down converter that is a converter for stepping up and stepping down a DC voltage, an inductor L11, an inductor L12, a capacitor C11, a capacitor Co11, a high-speed diode Do11, and a switching element Q11 as shown in FIG. A Cuk converter is known that converts the voltage to supply a voltage Eo to a load R. The Chuuk converter is a so-called zero ripple DC-DC converter that not only has a function as a step-up / step-down converter, but also makes the ripple voltage included in the DC output voltage zero. The outline of the operation principle of the Chuuk converter will be described below with reference to the drawings.

  The Chuk converter shown in FIG. 22 includes an inductor L11, a capacitor Co11, a high speed diode Do12, and a switching element Q12 as shown in FIG. 23, and boosts the input voltage Ei and supplies it to the load R, and FIG. As shown, the inductor L12, the capacitor Co11, the high-speed diode Do13, and the switching element Q13 can be considered to be combined with a step-down converter that steps down the input voltage Ei and supplies it to the load R. That is, it can be considered that the step-up converter and the step-down converter are coupled by the capacitor C11. Here, each of the step-up converter shown in FIG. 23 and the step-down converter shown in FIG. 24 is a circuit technology as a well-known converter.

  Further, the Chuk converter shown in FIG. 22 can be modified to a converter having a transformer which is an equivalent circuit. The circuit shown in FIG. 25 is obtained by rewriting the circuit shown in FIG. 22 in order to finally derive a converter having a transformer which is an equivalent circuit. The capacitor C11 is a series connection circuit of a capacitor C21 and a capacitor C22. To replace. Further, the circuit shown in FIG. 26 fixes the floating potential at the connection point between the capacitor C21 and the capacitor C22 in the circuit shown in FIG. 25 by the inductor L13. In the circuit shown in FIG. 27, the inductor L13 in the circuit shown in FIG. 26 is configured by a transformer having a primary winding N1 and a secondary winding N2. The circuit shown in FIG. 28 is a circuit equivalent to FIG. 27 in which the functions of the primary winding N1 and the secondary winding N2 in the circuit shown in FIG. 27 are separated into a voltage conversion function and an inductance function. In the circuit shown in FIG. 28, the inductance of the leakage inductor L1 corresponds to the inductance generated in the primary winding N1, and the inductance of the leakage inductor L2 corresponds to the inductance generated in the secondary winding N2. Moreover, the transformer formed by winding the primary winding N1 and the secondary winding N2 in the circuit shown in FIG. 28 is an ideal transformer having only a winding ratio.

  In the circuit shown in FIG. 28, the zero ripple condition in which the ripple voltage generated at both ends of the load R is zero is a turns ratio that is a ratio of the number of turns of the primary winding N1 / the number of turns of the secondary winding N2. The relationship between n and the magnetic coupling coefficient k between the primary winding N1 and the secondary winding N2 of the transformer is to have the relationship of the turns ratio n = magnetic coupling coefficient k. FIG. 29 shows a current waveform in the switching cycle of the primary-side current i1 (see FIG. 28) when this condition is satisfied. Here, a region where the current i1 increases corresponds to a period during which the switching element Q11 is turned on (conducted), and a region where the current i1 decreases corresponds to a period during which the switching element Q11 is turned off (disconnected). FIG. 30 shows the secondary current i2 (see FIG. 28). As shown, the current i2 is a direct current, and no ripple is generated in the voltage applied across the load R. However, the design and manufacture of a composite transformer (integrated magnetex transformer) formed so as to have a turn ratio n satisfying such a condition and having a leakage inductor L1 and a leakage inductor L2 satisfying such condition. It was difficult. As a result, the range of operation with zero ripple with respect to changes in the DC voltage Ei and the load R was narrow. Further, both the switching element Q11 and the high-speed diode Do11 are performing a so-called hard switching operation and have a high noise generation level. Thus, practical application of the circuit shown in FIG. 28 has been considered difficult due to the narrow range of operation with zero ripple and the difficulty of noise suppression. In recent years, Dr. Chuuk has succeeded in developing a Tesla converter (not shown) improved from this Chuuk converter (see, for example, Patent Document 1). In this Tesla converter, an active clamp circuit is added in parallel with the primary winding N1 in the circuit shown in FIG. 27, and the high-speed diode Do11 is a synchronous rectifier circuit using a MOS-FET to perform a soft switching operation.

  As a circuit similar to the circuit shown in FIG. 28, there is a switching power supply circuit shown in FIG. This switching power supply circuit includes, on the primary side, a choke coil PCC that functions as an inductor Lp1, a switching element Q1 (which includes a body diode DD1), a primary side parallel resonant capacitor C1, a primary side series resonant capacitor C2, and a converter. The transformer PIT includes a primary winding N1 and is configured as a class E switching amplifier that forms a voltage resonance circuit and a current resonance circuit. The secondary side includes a secondary winding N2 of the converter transformer and a secondary side series resonance. The capacitor C4, the high-speed diode Do1, the high-speed diode Do2, the first secondary-side smoothing capacitor Co1, the second secondary-side smoothing capacitor Co2, and the inductor Lo form a current resonance circuit and a voltage doubler half-wave rectifier circuit Is a multi-resonance converter. Here, the primary winding N1 and the secondary winding N2 are magnetically loosely coupled, and the primary winding N1 functions as a leakage inductor L1, and the secondary winding N2 functions as a leakage inductor L2. Further, the control circuit 1 and the oscillation / drive circuit 2 function as control means, and make the voltage value of the secondary side voltage Eo2 constant.

  FIG. 32 shows the operation of the main part of the circuit shown in FIG. 31 by showing voltages and currents. From the top of the drawing, current IQ1 (see FIG. 31), current ID2 (see FIG. 31), current ID1 (see FIG. 31), voltage Eo1 (see FIG. 31), and voltage Eo2 (see FIG. 31). Each is shown.

A converter having a voltage resonance converter on the primary side and a step-down converter on the secondary side is also known (see Patent Document 2).
US Pat. No. 6,462,962 specification JP-A-6-169568

  However, in the Chuk converter shown in FIG. 28, as described above, it is difficult to design and manufacture a composite transformer having good characteristics. Further, in the Tesla converter improved in this, three MOS-FETs and high-speed diodes are required, the number of components increases, and it is still difficult to obtain a composite transformer. It has not been solved. In the converter circuit using the class E switching amplifier shown in FIG. 31, the voltage Eo1 is turned off when the high speed diode is turned off due to the reverse recovery time (trr) of the high speed diode Do1 and the high speed diode Do2 forming the voltage doubler half-wave rectifier circuit. In addition, a spike voltage having a magnitude of about five times the magnitude of the ripple voltage generated in the switching period is generated. As a countermeasure for suppressing this spike voltage, a low-pass filter formed by the inductor Lo and the secondary side smoothing capacitor Co2 is required, but the inductance of the inductor Lo and the secondary side smoothing capacitor are not adversely affected on the load side. It is difficult to select the capacitance of Co2. In the circuit configuration described in Patent Document 1, since two converter circuits are provided on the primary side and the secondary side, the number of parts is large and the circuit configuration is complicated.

  This is a switching power supply circuit that solves the above-mentioned problems, reduces the number of components, and reduces the ripple component and spike component contained in the DC output voltage without using a complex transformer that is difficult to configure. It is to provide.

  The switching power supply circuit according to the present invention includes a primary circuit for generating AC power by inputting DC power, and a primary winding and a secondary winding for transmitting the AC power to the secondary circuit at the core. A converter transformer that is wound and formed by magnetically loosely coupling the primary winding and the secondary winding, and a secondary side DC output voltage supplied to the load from the secondary side circuit Constant voltage control means for controlling the primary side circuit so as to be constant voltage, the primary side circuit comprising: a switching element whose switching frequency is controlled by the constant voltage control means; The primary side parallel resonant frequency is governed by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side parallel resonant capacitor connected in parallel to the switching element. A primary side parallel resonant circuit formed in this manner, and the secondary side circuit includes a leakage inductance component generated in the secondary winding of the converter transformer and a direct or alternating current in the secondary winding. And a secondary side parallel resonance circuit formed such that the secondary side parallel resonance frequency is dominated by the capacitance of the secondary side parallel resonance capacitor connected in parallel with each other, and leakage occurring in the secondary winding of the converter transformer The secondary side series resonance formed such that the secondary side series resonance frequency is governed by the inductance component and the capacitance of the secondary side series resonance capacitor connected in series with the secondary winding directly or AC. A secondary side DC output voltage generating means for rectifying the AC power obtained in the secondary winding, and the secondary side DC output voltage generating means includes the secondary side series resonant capacitor. A diode connected in parallel to a series connection circuit of the second winding and the secondary winding, an inductor connected at one end to the diode, and a secondary-side smoothing capacitor connected to the other end of the inductor. Then, the voltage generated in the secondary winding is stepped down.

  In this switching power supply circuit, a primary side parallel resonant circuit is provided on the primary side, a secondary side parallel resonant circuit, a secondary side series resonant circuit, a secondary side DC output voltage generating means, and a load are supplied to the secondary side. And a constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage becomes a constant voltage. The secondary side DC output voltage generating means is formed to step down the voltage generated in the secondary winding, thereby reducing the magnitude of the ripple voltage and spike voltage included in the secondary side DC output voltage. To do.

  Another switching power supply circuit according to the present invention includes a primary side circuit for generating AC power by inputting DC power, and a primary winding and a secondary winding for transmitting the AC power to the secondary side circuit. A converter transformer wound around a core and formed by magnetically loosely coupling the primary winding and the secondary winding, and a secondary side DC output supplied to the load from the secondary side circuit And a constant voltage control means for controlling the primary side circuit so that the voltage becomes a constant voltage, wherein the primary side circuit is a switching whose switching frequency is controlled by the constant voltage control means. The primary side parallel resonant frequency is governed by the element, the leakage inductance component generated in the primary winding of the converter transformer, and the capacitance of the primary side parallel resonant capacitor connected in parallel to the switching element. A primary side parallel resonant circuit formed as described above, and the secondary side circuit includes a leakage inductance component generated in the secondary winding of the converter transformer and a direct or alternating current in the secondary winding. A secondary side series resonance circuit formed such that the secondary side series resonance frequency is dominated by the capacitance of the secondary side series resonance capacitors connected in series, and the secondary winding is directly or alternatingly connected A snubber circuit formed as a series connection circuit of a capacitor and a resistor connected in parallel to each other, and secondary side DC output voltage generating means for rectifying AC power obtained in the secondary winding, The side DC output voltage generating means includes a diode connected in parallel to a series connection circuit of the secondary side series resonant capacitor and the secondary winding, an inductor having one end connected to the diode, and the inductor. Has a secondary side smoothing capacitor connected to the other end of the inductor, and is formed so as to step down the voltage generated in said secondary winding.

  In this switching power supply circuit, a primary side parallel resonance circuit is provided on the primary side, a secondary side series resonance circuit on the secondary side, a snubber circuit formed as a series connection circuit of a capacitor and a resistor, and a secondary side DC output Voltage generating means and constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied to the load becomes a constant voltage. The secondary side DC output voltage generating means is formed to step down the voltage generated in the secondary winding, thereby reducing the magnitude of the ripple voltage included in the secondary side DC output voltage. Prevents the occurrence of spike voltage.

  ADVANTAGE OF THE INVENTION According to this invention, the switching power supply circuit which reduces the ripple component of DC output voltage can be provided, without reducing the number of components and using the composite transformer which is difficult to configure.

“First Embodiment”
The switching power supply circuit according to the first embodiment shown in FIG. 1 includes a primary circuit that generates high-frequency AC power by inputting DC power obtained by rectifying and smoothing AC power from an AC power supply, and this AC power. A primary winding N1 and a secondary winding N2 for transmission to the secondary circuit are wound around the core, and the converter transformer PIT is supplied to the load (not shown) from the secondary circuit. The switching power supply circuit includes a control circuit 1 and an oscillation / drive circuit 2 that function as constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage Eo becomes a constant voltage. Here, the primary side circuit and the secondary side circuit are separated by the primary winding N1 and the secondary winding N2 of the converter transformer, and are arranged on the AC power supply AC side including the primary winding N1. The circuit unit is a primary circuit, and the circuit unit arranged on the secondary smoothing capacitor Co side including the secondary winding N2 is a secondary circuit.

  The primary side circuit includes a switching element Q1 whose switching frequency fs is controlled by the oscillation / drive circuit 2, and a leakage inductor L1 and a switching element Q1 that are leakage inductance components generated in the primary winding N1 of the converter transformer PIT. A primary side parallel resonance circuit formed such that the primary side parallel resonance frequency fpo1 is dominated by the capacitance of the primary side parallel resonance capacitor C1 connected in parallel, and the operation of the primary side circuit is a voltage resonance type. It has. Here, the fact that the primary side parallel resonance frequency is dominated means that the leakage inductor L1 and the primary side parallel resonance capacitor C1 are the main factors that determine the primary side parallel resonance frequency fpo1. For example, the primary side smoothing capacitor Ci also affects the primary side parallel resonance frequency, but the capacitance value of the primary side smoothing capacitor Ci is, for example, 1 as compared with the capacitance value of the primary side parallel resonance capacitor C1. Since it is 10,000 times larger, the primary side smoothing capacitor Ci can be regarded as a short circuit in terms of alternating current, and does not dominate the primary side parallel resonance frequency.

  The secondary side circuit includes a leakage parallel inductor L2 that is a leakage inductor generated in the secondary winding N2 and a secondary parallel resonance capacitor C3 connected to both ends of the secondary winding N2, and a secondary parallel resonance frequency fpo2. The secondary side parallel resonance circuit formed so that is controlled is provided. The secondary side circuit is formed so that the secondary side series resonance frequency fso2 is dominated by the leakage inductor L2 and the secondary side series resonance capacitor C4 connected in series to the secondary winding N2. A secondary series resonant circuit is provided. Here, the fact that the secondary side parallel resonance frequency is dominated means that the leakage inductor L2 and the secondary side parallel resonance capacitor C3 are the main factors that determine the secondary side parallel resonance frequency fpo2. The fact that the secondary side series resonance frequency is dominated means that the leakage inductor L2 and the secondary side series resonance capacitor C4 are the main factors that determine the secondary side series resonance frequency fso2. For example, the secondary side smoothing capacitor Co also affects the secondary side parallel resonance frequency and the secondary side series resonance frequency, but the capacitance value of the secondary side smoothing capacitor Co is the capacitance value of the secondary side parallel resonance capacitor C3. For example, the secondary smoothing capacitor Co can be regarded as a short circuit in terms of alternating current, and the secondary parallel resonant frequency and 2 are larger than the capacitance value of the secondary series resonant capacitor C4. It does not dominate any of the secondary series resonance frequencies.

  Further, the secondary side circuit includes a high speed diode Do1, an inductor Lo, and a secondary side smoothing capacitor Co for functioning as a step-down converter.

  Here, the principle correspondence between the circuit shown in FIG. 1 and the circuit shown in FIG. 23 is that the secondary side series resonant capacitor C4 in the circuit shown in FIG. 1 is replaced with a capacitor Co11 in the boost converter circuit shown in FIG. Can be considered.

  With respect to the switching power supply circuit shown in FIG. 1, the power path from the input side to the output side will be described in further detail in order. First, for a commercial AC power supply AC, a common mode noise filter is formed by two filter capacitors CL and a common mode choke coil CMC.

  A full-wave rectifier circuit including a primary-side rectifying element Di and a primary-side smoothing capacitor Ci that are bridge-connected is connected to the output side of the common mode noise filter. When this full-wave rectifier circuit receives AC power from a commercial AC power supply AC and performs a full-wave rectification operation, a rectified and smoothed DC voltage is obtained at both ends of the primary side smoothing capacitor Ci. DC power is supplied from both ends of the side smoothing capacitor Ci. In this case, the value of the DC voltage Ei is at a level corresponding to the AC input voltage VAC.

  One end of the primary side smoothing capacitor Ci and one end of the primary winding N1 of the converter transformer PIT are connected, and the switching element Q1 is connected to the other end of the primary winding N1, and the DC voltage Ei is 1 It is supplied to the switching element Q1 connected to the other end of the primary winding N1 via the secondary winding N1. A body diode DD1 is connected in parallel between the drain and source of the switching element Q1. The anode and cathode of the body diode DD1 are connected to the source and drain of the switching element Q1. The body diode DD1 is a body diode included in the switching element Q1, and is an element that is incidentally formed in the MOS-FET manufacturing process.

  In addition, an oscillation / drive circuit 2 is provided for switching the switching element Q1. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. For example, a general-purpose IC can be used in this case. The oscillation circuit of the oscillation / drive circuit 2 generates an oscillation signal having a required frequency, and the drive circuit uses, for example, the gate voltage for switching driving the MOS-FET as the switching element Q1 using the oscillation signal. A switching drive signal is generated and applied to the gate. As a result, the switching element Q1 continuously performs a switching operation at the period of the switching drive signal (reciprocal of the switching frequency fs).

  The control circuit 1 is provided to stabilize the secondary side DC output voltage Eo by the switching frequency control method. In this case, the control circuit 1 generates a detection output corresponding to the level change of the error voltage obtained as a difference between the secondary side DC output voltage Eo as a detection input and a reference voltage to be stabilized. To supply. The oscillation / drive circuit 2 drives the switching element Q1 such that the switching frequency fs is varied according to the input detection output of the control circuit 1. For this purpose, the frequency of the oscillation signal generated by the internal oscillation circuit is varied. By changing the switching frequency fs of the switching element Q1, the resonance impedance of the primary side parallel resonance circuit changes, and the amount of power transmitted from the primary winding N1 of the converter transformer PIT to the secondary winding N2 side is changed. Although it changes, this operates so as to stabilize the level of the secondary side DC output voltage Eo.

  The converter transformer PIT is provided in order to transmit the switching output converted to AC power again to the secondary side by switching DC power with the switching element Q1. Here, the frequency of the switching output, which is again AC power, is selected to be 1000 times or more the frequency of the commercial AC power supply AC, for example, so that the size of the converter transformer PIT can be made small. This converter transformer PIT has a structure shown in a sectional view in FIG. The converter transformer PIT is an E-shaped core that is a ferrite core (a core in which both the cross-sectional direction shown in FIG. 2 and the cross-section orthogonal thereto are rectangular, or the cross-sectional direction shown in FIG. 2 is rectangular. The cross section perpendicular to this has a circular core). As described above, the converter transformer PIT is an EE type core that combines the core CR1 and the core CR2 that are E-shaped cores made of a ferrite material (a core that is a combination of the above-described cores each having a rectangular core cross section) or an EER type A core (a core in which one of the core cross sections is combined with the above-described core) is provided, and the winding portion is divided on the primary side and the secondary side, and then the primary winding N1 and the secondary winding N2 is wound on the bobbin B that covers the central magnetic leg of the EE type core or the EER type core. Here, the primary winding N1 and the secondary winding N2 are magnetically loosely coupled. The magnetically loose coupling does not mean that the magnetic flux interlinking with the primary winding N1 and the magnetic flux interlinking with the secondary winding N2 are common, but the primary winding N1 or the secondary winding N2. It means that there exists a magnetic flux that is linked only to either of the above. In this way, by using loose coupling, as described above, the primary winding N1 functions as the leakage inductor L1, and the secondary winding N2 functions as the leakage inductor L2. Here, the converter transformer PIT for loosely coupling the magnetic coupling between the primary winding N1 and the secondary winding N2 includes two core legs of the core CR1 and the core CR2 shown in FIG. This can be realized by making it shorter than the outer magnetic leg.

  Here, in the first embodiment, an EER type core is used as the ferrite core of the converter transformer PIT, and EER-40 (core model number) is used as the core size of the EER type core. The gap between the inner magnetic legs of the EER type core is set to a gap length of 1.4 mm. As a result, the value of the magnetic coupling coefficient k between the primary winding N1 and the secondary winding N2 is 0.7. To get a loosely coupled state. With this structure, the converter transformer PIT generates a predetermined leakage inductor L1 in the primary winding N1 and a predetermined leakage inductor L2 in the secondary winding N2.

  Since the primary side smoothing capacitor Co can be regarded as a short circuit at the switching frequency fs at which the switching element Q1 switches, the primary side parallel resonant capacitor C1 connected in parallel with the switching element Q1 is primary in terms of alternating current. It is connected in parallel with the winding N1. Therefore, a primary side parallel resonant circuit is formed by the leakage inductor L1 of the primary winding N1 and the capacitance of the primary side parallel resonant capacitor C1. Here, the number of turns of the primary winding N1 was 60T (turn), and the inductance value of the leakage inductor L1 was 540 μH (micro Henry). The capacitance of the primary side parallel resonant capacitor C1 was 7500 pF (pico farad). In this way, on the primary side of the switching power supply circuit of the first embodiment shown in FIG. 1, a voltage resonance circuit is formed by the leakage inductor L1 generated in the primary winding N1 and the primary side parallel resonance capacitor C1. The

  Next, the secondary side of the switching power supply circuit according to the first embodiment will be described. A secondary side parallel resonant capacitor C3 is connected in parallel to the secondary winding N2 of the converter transformer PIT, and a voltage resonant circuit is formed by the leakage inductor L2 and the secondary side parallel resonant capacitor C3. The number of turns of the secondary winding N2 is 45T, and the inductance value of the leakage inductor L2 is 330 μH. The capacitance value of the secondary side parallel resonant capacitor C3 is 0.022 μF.

  Further, a secondary side series resonance circuit that acts as a current resonance circuit is provided on the secondary side of the switching power supply circuit, and this secondary side resonance circuit is connected to the inductance value of the leakage inductor L2 and the secondary side series resonance circuit. The secondary series resonance frequency fso2 is controlled by the capacitance value of the resonance capacitor C4. As described above, the inductance value of the leakage inductor L2 is 330 μH, and the capacitance value of the secondary side series resonance capacitor C4 is 0.015 μF.

  A secondary side of the switching power supply circuit includes a high-speed diode Do1, an inductor Lo that functions as a low-pass filter, and a secondary-side smoothing capacitor Co. The inductance value of the inductor Lo was 400 μH, and the capacitance value of the secondary side smoothing capacitor Co was 1000 μF. The inductor Lo has the same structure as the converter transformer shown in FIG. 2, and only one winding is wound around the bobbin B as the winding. EE-25 (model number of the core material) was used as the ferrite core material, and the gap length of the inner magnetic leg was set to 0.8 mm.

  The primary side circuit and the secondary side circuit as described above are magnetically coupled via the converter transformer PIT. However, the winding direction of the primary winding N1 and the secondary winding N2 is an additional polarity. It is wound to be. In FIG. 1, the winding direction of the primary winding N1 and the secondary winding N2 indicates the winding direction of the winding with a black circle mark arranged in the vicinity of the winding end.

  When the voltage V2 (see FIG. 1) generated in the secondary winding N2 is positive, the current I5 (indicated by the arrow (→)) indicates the current direction from the secondary winding N2 to the secondary parallel resonant capacitor C3. As shown in FIG. 1, the current I4 (FIG. 1) indicates the current direction from the secondary winding N2 to the secondary side series resonance capacitor C4, the inductor Lo, and the secondary side smoothing capacitor Co. Flow).

  When the voltage V2 generated in the secondary winding N2 is negative, a current I5 flows in the opposite direction of the arrow (→) from the secondary winding N2 to the secondary parallel resonant capacitor C3, and the secondary The current I3 (see FIG. 1) in the direction opposite to the arrow (→) from the winding N2 to the high-speed diode Do1 and the secondary-side series resonant capacitor C4, and further from the inductor Lo to the secondary-side smoothing capacitor Co, A current I4 indicating a current direction by an arrow (→) flows through the diode Do1.

  Here, when the inductance value of the inductor Lo is selected to be larger than the inductance value of the leakage inductor L2, the waveform of the current I4 flowing through the inductor Lo can be formed into a sine wave shape in the current continuous mode. By setting it as such a current waveform, the magnitude | size of voltage (DELTA) Eo which is a ripple voltage which arises in the secondary side DC output voltage Eo of the both ends of the secondary side smoothing capacitor Co can be reduced. Further, by using such a current waveform, the magnitude of the spike voltage caused by the influence of the reverse recovery time (trr) of the high speed diode Do1 can be greatly reduced.

  3 and 4 show operation waveforms of each part of the power supply circuit shown in FIG. In these figures, FIG. 3 shows an operation waveform at a load power Po = 300 W, which is the maximum load power, and FIG. 4 shows an operation waveform at a load power Po = 0 W, which is no-load power. In these figures, experimental results in the case where the AC input voltage VAC = 100 V are shown.

  3 and 4, the voltage V1 (see FIG. 1) is a voltage across the switching element Q1, and shows a change depending on whether the switching element Q1 is turned on or off. That is, a current IQ1 (see FIG. 1) of the illustrated switching element Q1 flows during a period in which the voltage V1 is 0 level. The current I1 (see FIG. 1) is a resonance current that flows through the primary winding N1.

  When the resonance current flows through the primary winding N1, the illustrated voltage V2 (see FIG. 1) is excited in the secondary winding N2 of the converter transformer PIT. By obtaining such an alternating voltage V2, current I2 (see FIG. 1), current I3 (see FIG. 1), current I4 (see FIG. 1), current I5 (see FIG. 1) through the above-described path. ) Flows, and a voltage V3 (see FIG. 1) is generated, and a DC voltage of 175 V is obtained as the secondary side DC output voltage Eo. The voltage ΔEo indicates the ripple voltage of the secondary side DC output voltage Eo. The voltage V3 has a peak value of 630V, and the high speed diode Do1, the inductor Lo, and the secondary side smoothing capacitor Co function as components of the step-down converter, and the secondary side DC output voltage Eo is 175V. You can see that it is obtained. In the switching power supply circuit of the first embodiment, the range of the AC input voltage VAC output from the AC power supply AC is in the range of 85V to 144V.

  FIG. 5 shows the switching frequency fs and the power conversion efficiency ηAC in the range of the load power Po from 0 W to 300 W with the load power Po as the horizontal axis of the switching power supply circuit of the first embodiment shown in FIG. DC, a period TON in which the drain and source of the switching element Q1 are turned on (conducted), and a period TOFF in which the drain and source of the switching element Q1 are turned off (disconnected) are shown.

  The following can be understood from the facts obtained from the experimental results shown in FIGS. The ripple voltage values in the respective circuits shown in FIGS. 3 and 31 are compared. In the circuit shown in FIG. 31, when the filter circuit is simplified and the voltage Eo1 is used as the output DC voltage, the voltage is about 0.5 V as seen from the voltage Eo1 shown in FIG. Is as small as about 0.15 V as can be seen from the voltage ΔEo shown in FIG. The magnitude of the spike voltage is about 2.5 V as seen from the voltage Eo1 shown in FIG. 32, but is as small as about 0.25 V as seen from the voltage ΔEo shown in FIG. This indicates that good ripple characteristics and spike characteristics can be obtained without using a pi-type (π-type) filter as shown in FIG. 31, and the filter of the first embodiment has a smaller number of parts. It shows little and exhibits good characteristics.

  In the switching power supply circuit shown as the first embodiment, the primary side is a voltage resonance single-ended converter, the secondary side is a combination of a voltage resonance circuit and a current resonance circuit, and further configured as a step-down converter. The multiple resonance converter is constituted by the side and the secondary side. With this configuration, the magnitude of the voltage ΔEo, which is the ripple voltage of the switching period included in the secondary side DC output voltage Eo, is ½ compared to the ripple voltage included in the voltage Eo1 in the background art. The following can be done. Further, the magnitude of the spike voltage included in the secondary side DC output voltage Eo can be 1/10 or less as compared with the spike voltage included in the voltage Eo1 in the background art. Therefore, the secondary side smoothing capacitor Co2 in the background art becomes unnecessary.

  Further, in the Chuuk converter and Tesla converter, the secondary side DC output voltage is made constant by PWM control, so depending on the value of the DC voltage input to the converter and the magnitude of the load power supplied to the load, Although the magnitude of the ripple voltage included in the secondary side DC output voltage is large, since the switching power supply circuit shown as the first embodiment uses PFM control, the value of the DC voltage input to the converter is Even when it fluctuates, the magnitude of the ripple voltage included in the secondary side DC output voltage hardly changes, and as the load power decreases, the magnitude of the spike voltage included in the secondary side DC output voltage is Decrease.

  In addition, in the Tesla converter, for the purpose of functioning as an active clamp circuit and functioning as a synchronous rectification circuit, for example, three MOS-FETs and a set of high-speed diodes are required, and the circuit configuration becomes complicated. In the switching power supply circuit shown as the first embodiment, a switching power supply circuit having a small ripple voltage value can be configured with a simple circuit configuration.

"Modification of the first embodiment"
Various modifications of the switching power supply circuit according to the first embodiment will be described below. FIG. 6 shows a part of a circuit diagram of a circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 6, the polarity relationship between the primary winding N1 and the secondary winding N2 of the converter transformer PIT is different from that in FIG. That is, in FIG. 1, the polarity relationship between the primary winding N1 and the secondary winding N2 is additive, but in FIG. 6, the polarity between the primary winding N1 and the secondary winding N2 This relationship is depolarized. In the case of depolarization, the phase of each of the voltage V3 and the current I4 is advanced by 180 degrees in the switching cycle with respect to that in the circuit shown in FIG.

  FIG. 7 shows a part of a circuit diagram of another circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 7, the secondary parallel resonant capacitor C3 is connected in parallel to the high speed diode Do1. That is, the secondary side parallel resonant capacitor C3 is connected to the secondary winding in an AC manner via the secondary side series resonant capacitor C4 without directly connecting the secondary side parallel resonant capacitor C3 to the secondary winding in parallel. Connected in parallel. In this case, the capacitance value of the secondary side parallel resonance capacitor C3 is 0.039 μF, and the capacitance value of the secondary side series resonance capacitor C4 is 0.082 μF. In such a setting, the change range Δfs of the switching frequency fs required for the constant voltage control by changing the switching frequency fs is approximately 1/2 as compared with the circuit shown in FIG. We were able to.

  FIG. 8 shows a part of a circuit diagram of a circuit that replaces another part of the circuit shown in the circuit diagram of FIG. In FIG. 8, the secondary parallel resonant capacitor C3 is connected in parallel to the high speed diode Do1, and the polarity relationship between the primary winding N1 and the secondary winding N2 is depolarized.

  The switching power supply circuit shown as a modification of the first embodiment and the first embodiment includes a full-wave rectifier circuit composed of a primary side rectifying element Di and a primary side smoothing capacitor Ci, and supplies power from the AC power supply AC. Although the configuration for converting to DC power is adopted, DC power from DC power supply means for generating DC power, such as a storage battery or a solar battery, is converted into one of the converter transformer PIT without providing such a full-wave rectifier circuit. The effects described above can also be produced by supplying directly to the next winding N1.

“Second Embodiment”
The switching power supply circuit of the second embodiment shown in FIG. 9 is the same as the switching power supply circuit of the first embodiment with respect to the primary side connection mode, and the secondary side connection mode is different from that of the first embodiment. Is. Hereinafter, in the description of the second embodiment, the same parts as those in the first embodiment are denoted by the same reference numerals as those in the first embodiment, and the description thereof is omitted.

  As in the first embodiment, the primary side circuit includes a switching element Q1 whose switching frequency fs is controlled by the oscillation / drive circuit 2 and a leakage inductor that is a leakage inductance component generated in the primary winding N1 of the converter transformer PIT. The primary side resonance frequency fpo1 is formed by the capacitance of the primary side parallel resonance capacitor C1 connected in parallel to L1 and the switching element Q1, and the operation of the primary side circuit is a voltage resonance type 1 A secondary parallel resonant circuit is provided.

  The converter transformer PIT also includes a secondary winding N2 (first secondary winding N2) and a secondary winding N2 ′ (second secondary winding N2 ′) to which the respective windings are connected by a center tap. Are formed. Each of the secondary winding N2 and the secondary winding N2 ′ is loosely coupled to the primary winding N1, and as a result, the secondary side circuit generates a leakage inductor L2 generated in the secondary winding N2. And the secondary winding N2 ′ has a leakage inductor L2 ′ generated.

  In the secondary side circuit, the secondary side parallel resonance frequency fpo2 is dominated by the secondary side parallel resonance capacitor C3 connected to both ends of the series connection of the secondary winding N2 and the secondary winding N2 ′. The secondary side parallel resonant circuit formed in this way is provided. Here, the inductance value forming the secondary parallel resonant circuit is obtained by adding the mutual inductance generated by the secondary winding N2 and the secondary winding N2 ′ in addition to the leakage inductor L2 and the leakage inductor L2 ′. Thus, the connection point between the secondary winding N2 and the secondary winding N2 ′ is formed as a center tap, so that the secondary winding N2 and the secondary winding N2 ′ are in a positive polarity connection. That is, the mutual inductance value described above is a positive value.

  The secondary side circuit includes a secondary side series resonant capacitor C4 (first output) connected in series to the leakage inductor L2 (first leakage inductor) and the secondary winding N2 (first secondary winding). The secondary side series resonance circuit (first secondary side resonance capacitor) is formed such that the secondary side series resonance frequency fso21 (first secondary side series resonance frequency fso21) is dominated. Side series resonance circuit), and a secondary side series resonance capacitor C4 ′ connected in series to the leakage inductor L2 ′ (second leakage inductor) and the secondary winding N2 ′ (second secondary winding). The secondary side series resonant circuit (second secondary side series resonant frequency fso22) (second secondary side series resonant frequency fso22) is controlled by the (second secondary side series resonant capacitor). 2 secondary side series resonance Comprising the road).

  The secondary side circuit includes a high speed diode Do1 (second diode), an inductor Lo ′ (second inductor), and a secondary side smoothing capacitor Co for functioning as a step-down converter (second step-down converter). A high-speed diode Do2 (first diode), an inductor Lo (second inductor), and the above-described secondary-side smoothing capacitor Co for functioning as a step-down converter (first step-down converter). ing. Here, since the secondary side smoothing capacitor Co is shared by the two step-down converters (the first step-down converter and the second step-down converter), the operation equivalent to that the step-down converters are connected in parallel is achieved. To do.

  In this way, the switching power supply circuit according to the second embodiment has a primary side formed as a voltage resonance type single-ended converter, and a secondary side of two secondary windings connected in series by a center tap. A voltage resonance capacitor is connected to both ends, and a current resonance capacitor is connected in series with each secondary winding to constitute a multiple resonance type converter. In addition, this multiple resonance type converter is formed with two sets of step-down converters composed of a high speed diode, an inductor, and a secondary side smoothing capacitor on the secondary side.

  The primary circuit has the same connection mode as that of the first embodiment, and the operation thereof is also the same. Therefore, description of the operation is omitted, and specific constants will be described.

  In the second embodiment, an EER type core is used as the ferrite core of the converter transformer PIT, and EER-40 (core model number) is used as the core size of the EER type core. The gap of the inner magnetic leg of the EER type core is set to a gap length of 1.4 mm, thereby the value of the magnetic coupling coefficient k between the primary winding N1 and the secondary winding N2 and the primary winding N1. As a value of the magnetic coupling coefficient k between the second winding N2 ′ and the secondary winding N2 ′, a loosely coupled state of 0.7 is obtained. With such a structure, the converter transformer PIT generates a predetermined leakage inductor L1 in the primary winding N1, a predetermined leakage inductor L2 in the secondary winding N2, and a predetermined leakage in the secondary winding N2 ′. Inductor L2 'is generated.

  Here, the number of turns of the primary winding N1 was 45T, and the inductance value of the leakage inductor L1 was 330 μH. The capacitance of the primary side parallel resonant capacitor C1 was 8200 pF. In this way, on the primary side of the switching power supply circuit of the second embodiment shown in FIG. 9, a voltage resonance circuit is formed by the leakage inductor L1 generated in the primary winding N1 and the primary side parallel resonance capacitor C1. The

  The secondary side of the switching power supply circuit according to the second embodiment will be described. A secondary parallel resonant capacitor C3 is connected in parallel to the series connection circuit of the secondary winding N2 and the secondary winding N2 ′ of the converter transformer PIT, and the leakage inductor L2, the leakage inductor L2 ′, and the secondary winding N2 are connected. And the secondary winding N2 ′ form a voltage resonant circuit with the mutual inductor and the secondary side parallel resonant capacitor C3. The number of turns of the secondary winding N2 is 30T, and the number of turns of the secondary winding N2 'is 30T, and the inductance value of each of the leakage inductor L2 and the leakage inductor L2' is 270 μH. The capacitance value of the secondary side parallel resonant capacitor C3 is 0.015 μF.

  In addition, two sets of secondary side series resonant circuits (first secondary side series resonant circuit, second secondary side series resonant circuit) that function as current resonant circuits are provided on the secondary side of the switching power supply circuit. In this secondary side resonance circuit, the secondary side series resonance frequency fso21 is dominated by the inductance value of the leakage inductor L2 and the capacitance value of the secondary side series resonance capacitor C4, and the inductance of the leakage inductor L2 ′ is controlled. The secondary side series resonance frequency fso22 is governed by the value and the capacitance value of the secondary side series resonance capacitor C4 ′. As described above, the inductance value of each of the leakage inductor L2 and the leakage inductor L2 ′ is 270 μH, and the capacitance value of the secondary side series resonance capacitor C4 and each capacitance value of the secondary side series resonance capacitor C4 ′ are 0. 0.033 μF.

  The secondary side of the switching power supply circuit includes a high-speed diode Do1, an inductor Lo ′ that functions as a low-pass filter (second low-pass filter), and a secondary-side smoothing capacitor Co. The high-speed diode Do2 and the low-pass filter (first And the secondary smoothing capacitor Co described above. The inductance value of each of the inductor Lo and the inductor Lo ′ was 400 μH, and the capacitance value of the secondary side smoothing capacitor Co was 1000 μF. Each of the inductor Lo and the inductor Lo ′ has the same structure as the converter transformer shown in FIG. 2, and only one type of winding is wound around the bobbin B as the winding. EE-25 (model number of the core material) was used as the ferrite core material, and the gap length of the inner magnetic leg was set to 0.8 mm. Here, each of the first low-pass filter and the second low-pass filter also functions as a member constituting the first step-down converter and the second step-down converter described above. Suppress. Further, as described above, the output of the first low-pass filter and the output of the second low-pass filter are added by sharing the secondary side smoothing capacitor Co between the first step-down converter and the second step-down converter. To do.

  The primary side circuit and the secondary side circuit as described above are magnetically coupled via the converter transformer PIT, but the primary winding N1, the secondary winding N2, and the secondary winding N2 ' The winding direction is wound in such a way that it has a positive polarity. In FIG. 9, the winding direction of the primary winding N1, the secondary winding N2, and the secondary winding N2 ′ indicates the winding direction of the winding with a black circle mark arranged near the winding end. ing.

  When the voltage V2 (see FIG. 9) generated in the secondary winding N2 ′ is positive, a current flows from the secondary winding N2 to the high-speed diode Do2 and the secondary side series resonance capacitor C4, and the secondary A current flows from the winding N2 to the secondary winding N2 ′ and the secondary parallel resonant capacitor C3, and further from the secondary winding N2 ′ to the secondary series resonant capacitor C4 ′, the inductor Lo ′, and the secondary smoothing capacitor. A current I4 (see FIG. 9) flows in the direction indicated by an arrow (→) with respect to Co.

  Further, when the voltage V2 generated in the secondary winding N2 is negative, the current indicating the current direction with an arrow (→) from the secondary winding N2 ′ to the high-speed diode Do1 and the secondary side series resonant capacitor C4 ′. As I3 (see FIG. 9) flows, current flows from the secondary winding N2 ′ to the secondary winding N2 and the secondary side parallel resonant capacitor C3, and further from the secondary winding N2 to the secondary side series resonant capacitor. Current flows through C4, inductor Lo, and secondary smoothing capacitor Co.

  Here, if the inductance value of each of the inductor Lo and the inductor Lo ′ is selected to be larger than the inductance value of each of the leakage inductor L2 and the leakage inductor L2 ′, the waveform of the current flowing through the inductor Lo and the inductor Lo ′ is made current continuous. The mode can be a sine wave shape. By setting it as such a current waveform, the magnitude | size of voltage (DELTA) Eo which is a ripple voltage which arises in the secondary side DC output voltage Eo of the both ends of the secondary side smoothing capacitor Co can be reduced. Further, by using such a current waveform, the value of the spike voltage caused by the influence of the reverse recovery time (trr) of the high speed diode Do1 and the high speed diode Do2 can be greatly reduced.

  10 and 11 show operation waveforms of each part of the power supply circuit shown in FIG. In these drawings, FIG. 10 shows an operation waveform at a load power Po = 300 W, which is the maximum load power, and FIG. 11 shows an operation waveform at a load power Po = 0 W, which is no-load power. In these figures, experimental results in the case where the AC input voltage VAC = 100 V are shown.

  10 and 11, the voltage V1 (see FIG. 9) is the voltage across the switching element Q1, and during the period when the voltage V1 is at the 0 level, the current IQ1 (shown in FIG. 9) of the switching element Q1 shown in the figure. Flow). The current I1 (see FIG. 9) is a resonance current that flows through the primary winding N1.

  When the resonance current flows through the primary winding N1, the illustrated voltage V2 (see FIG. 9) is excited in the secondary winding N2 of the converter transformer PIT. By obtaining such an alternating voltage V2, current I2 (see FIG. 9), current I3 (see FIG. 9), and current I4 (see FIG. 9) flow through the above-described path, and voltage V3 (See FIG. 9) occurs, and a DC voltage of 175 V is obtained as the secondary side DC output voltage Eo. The voltage ΔEo indicates the ripple voltage of the secondary side DC output voltage Eo. The voltage V3 has a peak value of 680 V, and each of the high speed diode Do1, the inductor Lo ′, the secondary side smoothing capacitor Co, and the high speed diode Do2, the inductor Lo, and the secondary side smoothing capacitor Co is configured as a step-down converter. It can be seen that 175 V is obtained as the secondary side DC output voltage Eo functioning as a component. In the switching power supply circuit of the second embodiment, the range of the AC input voltage VAC output from the AC power supply AC is in the range of 85V to 144V.

  FIG. 12 shows the switching frequency fs and power conversion efficiency ηAC in the range of the load power Po from 0 W to 300 W with the load power Po as the horizontal axis of the switching power supply circuit of the second embodiment shown in FIG. DC, a period TON in which the drain and source of the switching element Q1 are turned on (conducted), and a period TOFF in which the drain and source of the switching element Q1 are turned off (disconnected) are shown.

  The following can be understood from the facts obtained from the experimental results shown in FIGS. The value of the ripple voltage in each circuit shown in FIG. 10 and FIG. 31 is compared. In the circuit shown in FIG. 31, when the filter circuit is simplified, the voltage is about 0.5 V as seen from the voltage Eo1 shown in FIG. 32. However, in the circuit shown in FIG. As can be seen from the voltage ΔEo, it is as small as about 0.05V. The magnitude of the spike voltage is about 2.5V as seen from the voltage Eo1 shown in FIG. 32, but is as small as about 0.15V as seen from the voltage ΔEo shown in FIG. This indicates that good ripple characteristics and spike characteristics can be obtained without using a pi-type (π-type) filter as shown in FIG. 31, and the filter of the second embodiment has a smaller number of parts. It shows little and exhibits good characteristics.

  In the switching power supply circuit shown as the second embodiment, the primary side is a voltage resonance single-ended converter, the secondary side is a combination of a voltage resonance circuit, a first current resonance circuit, and a second current resonance circuit. As a step-down converter, the primary side and the secondary side form a multiple resonance converter. With this configuration, the magnitude of the voltage ΔEo that is the ripple voltage of the switching period included in the secondary side DC output voltage Eo is 1/50 compared with the ripple voltage included in the voltage Eo1 in the background art. The following can be done. In addition, the magnitude of the spike voltage included in the secondary side DC output voltage Eo can be 1/15 or less compared to the spike voltage included in the voltage Eo1 in the background art. Therefore, the secondary side smoothing capacitor Co2 in the background art becomes unnecessary.

  Further, in the Chuuk converter and Tesla converter, the secondary side DC output voltage is made constant by PWM control, so depending on the value of the DC voltage input to the converter and the magnitude of the load power supplied to the load, Although the magnitude of the ripple voltage included in the secondary side DC output voltage is large, since the switching power supply circuit shown as the second embodiment uses PFM control, the value of the DC voltage input to the converter is Even when it fluctuates, the magnitude of the ripple voltage included in the secondary side DC output voltage hardly changes, and as the load power decreases, the magnitude of the spike voltage included in the secondary side DC output voltage is Decrease.

  In addition, in the Tesla converter, for the purpose of functioning as an active clamp circuit and functioning as a synchronous rectification circuit, for example, three MOS-FETs and a set of high-speed diodes are required, and the circuit configuration becomes complicated. In the switching power supply circuit shown as the second embodiment, a switching power supply circuit having a small ripple voltage value can be configured with a simple circuit configuration.

  Further, when the load power changes in the range of 300 W to 0 W, the value of the change range Δfs of the switching frequency fs required to maintain the constant voltage characteristic can be as small as 33 kHz (kilohertz). . This is advantageous when an oscillation / drive circuit is configured using a general-purpose IC having a limited variable range of the switching frequency fs.

“Modification of Second Embodiment”
Various modifications of the switching power supply circuit according to the second embodiment will be described below. FIG. 13 shows a part of a circuit diagram of a circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 13, the connection mode of the secondary side parallel resonant capacitor C3, the secondary side series resonant capacitor C4, and the secondary side series resonant capacitor C4 ′ is different from that in FIG. That is, in FIG. 13, one end of the secondary side series resonance capacitor C4 is connected to the secondary winding N2, and one end of the secondary side series resonance capacitor C4 'is connected to the secondary winding N2'. A secondary parallel resonant capacitor C3 is connected between the other end of the series resonant capacitor C4 and the other end of the secondary side series resonant capacitor C4 ′. That is, the secondary side parallel resonant capacitor C3 is connected in parallel to the secondary winding N2 and the secondary winding N2 ′ in an alternating manner via the secondary side series resonant capacitor C4 and the secondary side series resonant capacitor C4 ′. ing.

  FIG. 14 shows a part of a circuit diagram of another circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 14, the winding No and the winding No ′ are applied to the same core as the composite choke coil without configuring the inductor Lo and the inductor Lo ′ as separate parts, and the winding No and the winding No ′ are magnetized. Thus, it functions as an inductor Lo and an inductor Lo ′ as loose coupling.

  Note that the switching power supply circuit shown as a modification of the second embodiment and the second embodiment includes a full-wave rectifier circuit including a primary side rectifier element Di and a primary side smoothing capacitor Ci, and uses the power from the AC power supply AC. Although the configuration for converting to DC power is adopted, DC power from DC power supply means for generating DC power, such as a storage battery or a solar battery, is converted into one of the converter transformer PIT without providing such a full-wave rectifier circuit. The effects described above can also be produced by supplying directly to the next winding N1.

“Third Embodiment”
The switching power supply circuit of the third embodiment shown in FIG. 15 is the same as the switching power supply circuit of the first embodiment with respect to the primary side connection mode, and the secondary side connection mode is different from that of the first embodiment. Is. Hereinafter, in the description of the third embodiment, portions similar to those in the first embodiment are denoted by the same reference numerals as those in the first embodiment, and description thereof is omitted.

  As in the first embodiment, the primary side circuit includes a switching element Q1 whose switching frequency fs is controlled by the oscillation / drive circuit 2 and a leakage inductor that is a leakage inductance component generated in the primary winding N1 of the converter transformer PIT. The primary side resonance frequency fpo1 is formed by the capacitance of the primary side parallel resonance capacitor C1 connected in parallel to L1 and the switching element Q1, and the operation of the primary side circuit is a voltage resonance type 1 A secondary parallel resonant circuit is provided.

  In the above configuration, the winding direction relationship between the primary winding N1 and the secondary winding N2 is additive in the first embodiment, but in the third embodiment, the primary winding N1 and the secondary winding N2 are secondary. The difference is that the winding direction relationship with the winding N2 is depolarized.

  The secondary side circuit is formed so that the secondary side series resonance frequency fso2 is dominated by the leakage inductor L2 and the secondary side series resonance capacitor C4 connected in series to the secondary winding N2. A secondary series resonant circuit is provided. Further, the secondary side circuit includes a high speed diode Do1, an inductor Lo, and a secondary side smoothing capacitor Co for functioning as a step-down converter.

  In the first embodiment, the secondary parallel resonant capacitor C3 is provided. In the third embodiment, a snubber circuit is used instead of the secondary parallel resonant capacitor C3. The snubber circuit is configured by a series connection circuit of a capacitor Cs and a resistor Rs.

  Further, the secondary side circuit includes a high speed diode Do1, an inductor Lo, and a secondary side smoothing capacitor Co for functioning as a step-down converter.

  In this way, the switching power supply circuit according to the third embodiment is formed as a voltage resonance type single-ended converter on the primary side, and a current resonance capacitor is connected in series with the secondary winding to thereby provide a multiple resonance type converter. It is configured as. The multiple resonance type converter is formed with a step-down converter including a high speed diode, an inductor, and a secondary side smoothing capacitor on the secondary side.

  The primary circuit has the same connection mode as that of the first embodiment, and the operation thereof is also the same. Therefore, description of the operation is omitted, and specific constants will be described.

  In the third embodiment, an EER type core is used as the ferrite core of the converter transformer PIT, and EER-40 (core model number) is used as the core size of the EER type core. The gap between the inner magnetic legs of the EER type core is set to a gap length of 1.4 mm. As a result, the value of the magnetic coupling coefficient k between the primary winding N1 and the secondary winding N2 is 0.7. To get a loosely coupled state. With this structure, the converter transformer PIT generates a predetermined leakage inductor L1 in the primary winding N1 and a predetermined leakage inductor L2 in the secondary winding N2.

  Here, the number of turns of the primary winding N1 was 45T, and the inductance value of the leakage inductor L1 was 330 μH. The capacitance of the primary side parallel resonant capacitor C1 was 6800 pF. In this way, on the primary side of the switching power supply circuit of the third embodiment shown in FIG. 15, a voltage resonance circuit is formed by the leakage inductor L1 generated in the primary winding N1 and the primary side parallel resonance capacitor C1. The

  The secondary side of the switching power supply circuit according to the third embodiment will be described. A snubber circuit is connected in parallel to the secondary winding N2 of the converter transformer PIT so as to absorb a spike voltage generated in the secondary side circuit. The number of turns of the secondary winding N2 is 60T, and the inductance value of the leakage inductor L2 is 540 μH. The capacitance value of the capacitor Cs used in the snubber circuit is 220 pF, and the resistance value Rs used in the snubber circuit is 1 kΩ (kiloohm).

  Further, the secondary side series resonance circuit acting as a current resonance circuit is configured such that the secondary side series resonance frequency fso2 is dominated by the inductance value of the leakage inductor L2 and the capacitance value of the secondary side series resonance capacitor C4. Formed. As described above, the inductance value of the leakage inductor L2 is 540 μH, and the capacitance value of the secondary side series resonance capacitor C4 is 0.012 μF.

  The secondary side of the switching power supply circuit includes a high-speed diode Do1, an inductor Lo that functions as a low-pass filter, and a secondary-side smoothing capacitor Co. The inductance value of the inductor Lo was 585 μH, and the capacitance value of the secondary side smoothing capacitor Co was 1000 μF. The inductor Lo has the same structure as the converter transformer shown in FIG. 2, and only one type of winding is wound around the bobbin B as the winding. EE-25 (model number of the core material) was used as the ferrite core material, and the gap length of the inner magnetic leg was set to 0.8 mm.

  When the voltage V2 (see FIG. 15) generated in the secondary winding N2 is positive, the arrow (→) from the secondary winding N2 to the secondary side series resonant capacitor C4, the inductor Lo, and the secondary side smoothing capacitor Co. ) Flows a current I4 (see FIG. 15) indicating the current direction.

  When the voltage V2 generated in the secondary winding N2 is negative, a current I5 (indicated by an arrow (→)) indicates a current direction from the secondary winding N2 to the high-speed diode Do1 and the secondary series resonance capacitor C4. 15) flows, and current flows from the inductor Lo to the secondary smoothing capacitor Co and the high-speed diode Do1.

  Here, the current I3 (see FIG. 15), which is the current flowing through the snubber circuit, will be described. The magnitude of the current I3 is smaller than the magnitude of the current in each part described above, regardless of whether the voltage V2 is positive or negative. By adopting such a snubber circuit, it is possible to prevent a spike voltage from being generated due to the influence of the reverse recovery time (trr) of the high-speed diode Do1.

  Further, when the inductance value of the inductor Lo is selected to be larger than the inductance value of the leakage inductor L2, the waveform of the current flowing through the inductor Lo can be made into a sine wave shape of a continuous current mode. By setting it as such a current waveform, the magnitude | size of voltage (DELTA) Eo which is a ripple voltage which arises in the secondary side DC output voltage Eo of the both ends of the secondary side smoothing capacitor Co can be reduced.

  16 and 17 show operation waveforms of each part of the power supply circuit shown in FIG. In these figures, FIG. 16 shows an operation waveform at a load power Po = 250 W, which is the maximum load power, and FIG. 17 shows an operation waveform at a load power Po = 0 W, which is no-load power. In these figures, experimental results in the case where the AC input voltage VAC = 100 V are shown.

  16 and 17, the voltage V1 (see FIG. 15) is the voltage across the switching element Q1, and during the period when the voltage V1 is at the 0 level, the current IQ1 (shown in FIG. 15) of the switching element Q1 shown in the figure. Flow). The current I1 (see FIG. 15) is a resonance current that flows through the primary winding N1.

  When the resonance current flows through the primary winding N1, the illustrated voltage V2 (see FIG. 15) is excited in the secondary winding N2 of the converter transformer PIT. By obtaining such an alternating voltage V2, current I2 (see FIG. 15), current I3 (see FIG. 15), current I4 (see FIG. 15), current I5 (see FIG. 15) through the above-described path. Each of which flows, and a voltage V3 (see FIG. 15) is generated, and a DC voltage of 175 V is obtained as the secondary side DC output voltage Eo. The voltage ΔEo indicates the ripple voltage of the secondary side DC output voltage Eo. The voltage V3 has a peak value of 700V, and the high speed diode Do1, the inductor Lo, and the secondary side smoothing capacitor Co function as components of the step-down converter, and the secondary side DC output voltage Eo is 175V. You can see that it is obtained. In the switching power supply circuit of the third embodiment, the range of the AC input voltage VAC output from the AC power supply AC is in the range of 85V to 144V.

  FIG. 18 shows the switching frequency fs and power conversion efficiency ηAC in the range of the load power Po from 0 W to 250 W with the load power Po as the horizontal axis of the switching power supply circuit of the third embodiment shown in FIG. DC, a period TON in which the drain and source of the switching element Q1 are turned on (conducted), and a period TOFF in which the drain and source of the switching element Q1 are turned off (disconnected) are shown.

  The following can be understood from the facts obtained from the experimental results shown in FIGS. The value of the ripple voltage in each circuit shown in FIG. 15 and FIG. 31 is compared. In the circuit shown in FIG. 31, when the filter circuit is simplified, it is about 0.5 V as seen from the voltage Eo1 shown in FIG. 32, but in the circuit shown in FIG. 15, the circuit shown in FIG. As can be seen from the voltage ΔEo, the voltage is as small as about 0.1V. The magnitude of the spike voltage is 2.5 V as can be seen from the voltage Eo1 shown in FIG. 32. However, in the circuit shown in FIG. 15, the snubber circuit formed by the capacitor Cs and the resistor Rs causes the FIG. It disappears as can be seen from the indicated voltage ΔEo. This indicates that good ripple characteristics and spike characteristics can be obtained without using a pi-type (π-type) filter as shown in FIG. 31. The filter of the third embodiment has a smaller number of parts. It shows little and exhibits good characteristics.

  In the switching power supply circuit shown as the third embodiment, the primary side is a voltage resonance single-ended converter, the secondary side is a combination of a snubber circuit and a current resonance circuit, and is further configured as a step-down converter. And the secondary side constitute a multiple resonance converter. With this configuration, the magnitude of the voltage ΔEo, which is the ripple voltage of the switching period included in the secondary side DC output voltage Eo, is 1/5 compared with the ripple voltage included in the voltage Eo1 in the background art. The following can be done. Further, the spike voltage included in the secondary side DC output voltage Eo can be eliminated. Therefore, the secondary side smoothing capacitor Co2 in the background art becomes unnecessary.

  Further, in the Chuuk converter and Tesla converter, the secondary side DC output voltage is made constant by PWM control, so depending on the value of the DC voltage input to the converter and the magnitude of the load power supplied to the load, Although the magnitude of the ripple voltage included in the secondary side DC output voltage is large, since the switching power supply circuit shown as the third embodiment uses PFM control, the value of the DC voltage input to the converter is Even when it fluctuates, the magnitude of the ripple voltage included in the secondary side DC output voltage hardly changes, and as the load power decreases, the magnitude of the spike voltage included in the secondary side DC output voltage is Decrease.

  In addition, in the Tesla converter, for the purpose of functioning as an active clamp circuit and functioning as a synchronous rectification circuit, for example, three MOS-FETs and a set of high-speed diodes are required, and the circuit configuration becomes complicated. In the switching power supply circuit shown as a modified example of the third embodiment, a switching power supply circuit having a small ripple voltage value can be configured with a simple circuit configuration.

  Further, when the load power changes in the range of 250 W to 0 W, the value of the change range Δfs of the switching frequency fs required to maintain the constant voltage characteristic can be as small as 7 kHz (kilohertz). . This is advantageous when an oscillation / drive circuit is configured using a general-purpose IC having a limited variable range of the switching frequency fs.

“Modification of Third Embodiment”
Various modifications of the switching power supply circuit according to the third embodiment will be described below. FIG. 19 shows a part of a circuit diagram of a circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 15, the winding direction relationship between the primary winding N1 and the secondary winding N2 of the converter transformer PIT is depolarized, but in FIG. 19, the primary winding N1 and the secondary winding of the converter transformer PIT The relationship of the winding direction with the winding N2 is the additive polarity. In this case, the voltage V3 and the current I4 are advanced in phase by 180 degrees compared to those in the circuit shown in the circuit diagram of FIG.

  FIG. 20 shows a part of a circuit diagram of another circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 20, the relationship in the winding direction between the primary winding N1 and the secondary winding N2 of the converter transformer PIT is set as a polarity, and a snubber circuit formed by a capacitor Cs and a resistor Rs is in parallel with the high-speed diode Do1. It is connected to the. In other words, the snubber circuit formed by the capacitor Cs and the resistor Rs is not only directly connected in parallel to the secondary winding N2, but also in an alternating manner to the secondary winding N2 via the secondary side series resonance capacitor C4. Even in the case of connection, it has an effect of removing spikes.

  FIG. 21 shows a part of a circuit diagram of another circuit that replaces a part of the circuit shown in the circuit diagram of FIG. In FIG. 21, the relationship in the winding direction between the primary winding N1 and the secondary winding N2 of the converter transformer PIT is set as a positive polarity, and a secondary winding N2 'is further added. The winding direction of the secondary winding N2 and the secondary winding N2 'is determined so that the connection point between the secondary winding N2 and the secondary winding N2' is a center tap. One terminal of the secondary side series resonance capacitor C4 is connected to the secondary winding N2, and one terminal of the secondary side series resonance capacitor C4 'is connected to the secondary winding N2'. A high speed diode Do2 is connected between the other terminal of the secondary side series resonant capacitor C4 and the center tap, and a high speed diode is connected between the other terminal of the secondary side series resonant capacitor C4 ′ and the center tap. Do1 is connected. A snubber circuit formed of a capacitor Cs and a resistor Rs is connected in parallel to a series circuit of a high speed diode Do1 and a high speed diode Do2.

  Note that the switching power supply circuit shown as a modification of the third embodiment and the third embodiment includes a full-wave rectifier circuit including a primary side rectifier element Di and a primary side smoothing capacitor Ci, and uses the power from the AC power supply AC. Although the configuration for converting to DC power is adopted, DC power from DC power supply means for generating DC power, such as a storage battery or a solar battery, is converted into one of the converter transformer PIT without providing such a full-wave rectifier circuit. The effects described above can also be produced by supplying directly to the next winding N1.

  Further, the present invention is not limited to the configurations shown as the above embodiments. For example, regarding the main switching element (and auxiliary switching element), it is also conceivable to select an element other than the MOSFET, such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor.

It is a figure which shows the structure of the switching power supply circuit of 1st Embodiment. It is a figure which shows sectional drawing of the converter transformer used for the power supply circuit of embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 1st Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 1st Embodiment. It is a figure which shows the switching frequency with respect to the load power of the switching power supply circuit of 1st Embodiment, an ON period, an OFF period, and power conversion efficiency. It is a circuit diagram which shows the secondary side circuit as a modification of 1st Embodiment. It is a circuit diagram which shows the secondary side circuit as a modification of 1st Embodiment. It is a circuit diagram which shows the secondary side circuit as a modification of 1st Embodiment. It is a figure which shows the structure of the switching power supply circuit of 2nd Embodiment. It is a figure which shows the operation | movement waveform of each part of the power supply circuit of 2nd Embodiment. It is a figure which shows the operation | movement waveform of each part of the power supply circuit of 2nd Embodiment. It is a figure which shows the switching frequency with respect to the load power of the switching power supply circuit of 2nd Embodiment, an ON period, an OFF period, and power conversion efficiency. It is a circuit diagram which shows the secondary side circuit as a modification of 2nd Embodiment. It is a circuit diagram which shows the secondary side circuit as a modification of 2nd Embodiment. It is a figure which shows the structure of the switching power supply circuit of 3rd Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 3rd Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 3rd Embodiment. It is a figure which shows the switching frequency with respect to the load power of the switching power supply circuit of 3rd Embodiment, an ON period, an OFF period, and power conversion efficiency. It is a circuit diagram which shows the secondary side circuit as a modification of 3rd Embodiment. It is a circuit diagram which shows the secondary side circuit as a modification of 3rd Embodiment. It is a circuit diagram which shows the secondary side circuit as a modification of 3rd Embodiment. It is a figure which shows the Chuk converter of background art. It is a figure which shows the step-up converter of background art. It is a figure which shows the step-down converter of background art. It is a figure which shows the equivalent circuit of the Chuuk converter of background art. It is a figure which shows the equivalent circuit of the Chuuk converter of background art. It is a figure which shows the equivalent circuit of the Chuuk converter of background art. It is a figure which shows the equivalent circuit of the Chuuk converter of background art. It is a figure which shows the primary side electric current in the switching period of the switching power supply circuit of background art. It is a figure which shows the electric current of the secondary side of the switching power supply circuit of background art. It is a figure which shows the structure of another switching power supply circuit of background art. It is a figure which shows the operation | movement waveform of each part of another switching power supply circuit of background art.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Control circuit, 2 Oscillation drive circuit, AC alternating current power supply, B bobbin, C1 primary side parallel resonance capacitor, C2 primary side series resonance capacitor, C3 secondary side parallel resonance capacitor, C4 secondary side series resonance capacitor, Ci Primary side smoothing capacitor, CL filter capacitor, CMC common mode choke coil, Co, Co1, Co2 Secondary side smoothing capacitor, CR1, CR2 core, Cs capacitor, DD1 body diode, Di primary side rectifier, Do1, Do2 High speed Diode, Eo secondary side DC output voltage, L1, L2 leakage inductor, Lo, Lo ′ inductor, N1 primary winding, N2, N2 ′ secondary winding, PIT converter transformer, Q1 switching element, R load, Rs resistance VAC AC input voltage

Claims (4)

A primary circuit for generating AC power by inputting DC power;
A primary winding and a secondary winding for transmitting the AC power to the secondary circuit are wound around a core, and the primary winding and the secondary winding are magnetically loosely coupled. A converter transformer to be formed;
Constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied from the secondary side circuit to the load becomes a constant voltage, and a switching power supply circuit comprising:
The primary circuit is
A switching element whose switching frequency is controlled by the constant voltage control means;
1 formed such that the primary side parallel resonance frequency is governed by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side parallel resonance capacitor connected in parallel to the switching element. A secondary parallel resonant circuit,
The secondary circuit is
The secondary parallel resonant frequency is governed by the leakage inductance component generated in the secondary winding of the converter transformer and the capacitance of the secondary parallel resonant capacitor connected directly or in parallel to the secondary winding. A secondary side parallel resonant circuit formed as described above,
The secondary side series resonant frequency is governed by the leakage inductance component generated in the secondary winding of the converter transformer and the capacitance of the secondary side series resonant capacitor connected in series directly or AC to the secondary winding. A secondary side series resonant circuit formed as described above,
Secondary side DC output voltage generating means for rectifying AC power obtained in the secondary winding,
The secondary side DC output voltage generating means includes:
A diode connected in parallel to a series connection circuit of the secondary side series resonant capacitor and the secondary winding, an inductor having one end connected to the diode, and a secondary side connected to the other end of the inductor A switching power supply circuit comprising a smoothing capacitor and configured to step down a voltage generated in the secondary winding. A primary circuit for generating AC power by inputting DC power;
A primary winding and a secondary winding for transmitting the AC power to the secondary circuit are wound around a core, and the primary winding and the secondary winding are magnetically loosely coupled. A converter transformer to be formed;
Constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied from the secondary side circuit to the load becomes a constant voltage, and a switching power supply circuit comprising:
The primary circuit is
A switching element whose switching frequency is controlled by the constant voltage control means;
1 formed such that the primary side parallel resonance frequency is governed by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side parallel resonance capacitor connected in parallel to the switching element. A secondary parallel resonant circuit,
The secondary circuit is
The secondary side series resonant frequency is governed by the leakage inductance component generated in the secondary winding of the converter transformer and the capacitance of the secondary side series resonant capacitor connected in series directly or AC to the secondary winding. A secondary side series resonant circuit formed as described above,
A snubber circuit formed as a series connection circuit of a capacitor and a resistor connected in parallel directly or alternatingly to the secondary winding;
Secondary side DC output voltage generating means for rectifying AC power obtained in the secondary winding,
The secondary side DC output voltage generating means includes:
A diode connected in parallel to a series connection circuit of the secondary side series resonant capacitor and the secondary winding, an inductor having one end connected to the diode, and a secondary side connected to the other end of the inductor A switching power supply circuit comprising a smoothing capacitor and configured to step down a voltage generated in the secondary winding. The converter transformer is formed having a first secondary winding and a second secondary winding, each winding being connected by a center tap,
The secondary parallel resonant capacitor is connected directly or in parallel in parallel to both ends of the winding ends of the first secondary winding and the second secondary winding,
The secondary side series resonant circuit is:
A first secondary side series resonant circuit formed by a first secondary side series resonant capacitor connected in series with the first secondary winding; and in series with the second secondary winding. A second secondary side series resonant circuit formed by a second secondary side series resonant capacitor connected thereto,
The secondary side DC output voltage generating means includes:
A first diode connected in parallel to a series connection circuit of the first secondary side series resonant capacitor and the first secondary winding; and a first diode having one end connected to the first diode. An inductor, and the secondary smoothing capacitor connected to the other end of the first inductor;
A second diode connected in parallel to a series connection circuit of the second secondary side series resonant capacitor and the second secondary winding; and a second diode having one end connected to the second diode. It has an inductor and the secondary side smoothing capacitor connected to the other end of the second inductor, and is formed so as to step down a voltage generated in the secondary winding. Item 4. The switching power supply circuit according to Item 1.   2. The secondary side parallel resonant capacitor is connected in parallel with the secondary winding in an AC manner through the secondary side series resonant capacitor connected to the secondary winding. The switching power supply circuit according to claim 2.

Images