JP2007053823A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a power supply circuit having a power factor improving function, improving the power conversion efficiency and reducing the number of components for composing the circuit. <P>SOLUTION: The switching power supply has a rectifying/smoothing circuit for generating a rectified/smoothed voltage Ei from AC power, inputs the rectified/smoothed voltage Ei to a converter transformer PIT having a leakage inductor through a choke coil PCC, converts power by using an E-class converter circuit having a primary series resonance circuit comprising a primary series resonance capacitor C2 and a leakage inductor L1 and a secondary series resonance circuit comprising a secondary series resonance capacitor C3 and a leakage inductor L2, and includes a power factor improving circuit 10 for rectifying a primary series resonance current and returning it to a commercial power supply by using a power factor improving transformer VFT. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

  In recent years, most power supply circuits that rectify a commercial power supply to obtain a desired DC voltage are switching power supply circuits. The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

  By the way, in general, when a commercial power supply is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the utilization efficiency of the power supply is impaired. Further, there is a need for measures for suppressing harmonics generated by such a distorted current waveform. Therefore, a technique using a so-called active filter is known as a technique for improving the power factor in the past (see, for example, Patent Document 1).

  FIG. 14 shows a basic configuration of such an active filter. In FIG. 14, a bridge rectifier circuit Di is connected to a commercial AC power source AC. A step-up type converter is connected to the positive / negative line of the bridge rectifier circuit Di, and a smoothing capacitor Cout is connected in parallel to the output thereof, and a DC voltage Vout is obtained as a voltage across the both ends. This DC voltage Vout is supplied as an input voltage to a load 110 such as a DC-DC converter in the subsequent stage.

  As a configuration for improving the power factor, a step-up type including a inductor 111, a fast recovery type fast switching diode D, a step-up type converter including a switching element Q, and a multiplier 111 as main components. A control unit of the converter. The inductor L and the high-speed switching diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Cout. The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the smoothing capacitor Cout. The switching element Q is, for example, a MOS-FET, and is inserted between the connection point of the inductor L and the high-speed switching diode D and the primary side ground.

  To the multiplier 111, a current detection line LI and a waveform input line LW are connected, and a voltage detection line LV is further connected. Then, the multiplier 111 detects a signal corresponding to the rectified current Iin input from the current detection line LI and flowing in the negative output terminal of the bridge rectifier circuit Di, from both ends of the resistor Ri. Further, a signal corresponding to the rectified voltage Vin of the positive output terminal of the bridge rectifier circuit Di input from the waveform input line LW is detected. The rectified voltage Vin is an absolute value of the waveform of the AC input voltage VAC from the commercial AC power supply AC. Further, based on the DC voltage Vout of the smoothing capacitor Cout input from the voltage detection line LV, the fluctuation difference of the DC input voltage (a signal obtained by amplifying the difference between the predetermined reference voltage and the DC voltage Vout is referred to as a fluctuation difference). The same is used in the following). The multiplier 111 outputs a drive signal for driving the switching element Q.

  In the multiplier 111 (control unit of the step-up type converter) and the step-up type converter, a signal corresponding to the rectified current Iin detected from the current detection line LI and a DC input voltage detected from the voltage detection line LV The fluctuation difference is multiplied, and an error between the multiplication result and a signal corresponding to the rectified voltage Vin detected from the waveform input line LW is detected. After the error signal is amplified, PWM (Pulse Width Modulation) conversion is performed, and the switching element Q is controlled by a binary signal of a high level and a low level. In this way, a two-input feedback system is configured, the value of the DC voltage Vout is set to a predetermined value, and the rectified current Iin has a similar waveform with respect to the rectified voltage Vin. As a result, the AC voltage applied to the bridge rectifier circuit Di from the commercial AC power supply AC and the waveform of the AC current flowing into the bridge rectifier circuit Di are similar, and the power factor is improved so that the power factor approaches one. Will be planned.

  FIG. 15A shows the rectified voltage Vin and the rectified current Iin when the active filter circuit shown in FIG. 14 operates appropriately. FIG. 15B shows an energy (power) change Pchg input / output to / from the smoothing capacitor Cout. A line indicated by a broken line indicates an energy (power) average value Pin to be input / output. That is, the smoothing capacitor Cout stores energy when the rectified voltage Vin is high, and releases energy when the rectified voltage Vin is low, thereby maintaining the flow of output power. FIG. 15C shows a waveform of the charge / discharge current Ichg with respect to the smoothing capacitor Cout. FIG. 15D shows a DC voltage Vout that is a voltage across the smoothing capacitor Cout. In the DC voltage Vout, a ripple voltage mainly including the second harmonic component of the cycle of the rectified voltage Vin is superimposed on the DC voltage (for example, a DC voltage of 375 V).

  FIG. 16 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG. The power supply circuit shown in this figure employs a configuration that can cope with a load power Po in the range of 300 W to 0 W when the value of the AC input voltage VAC is in the range of 85 V to 264 V. Further, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

  The power supply circuit shown in FIG. 16 will be described in order from the AC input side. A common mode noise filter is provided by two line filter transformers LFT and three across capacitors CL, and a bridge rectifier circuit Di is connected to the subsequent stage. In addition, a normal mode noise filter 125 having a pi-type configuration including an inductor LN and a filter capacitor (film capacitor) CN is connected to the rectified output line of the bridge rectifier circuit Di.

  The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci via the inductor LN, the choke coil PCC (functioning as the inductor Lpc), and a series connection of a fast recovery type fast switching diode D20. The The smoothing capacitor Ci has the same function as the smoothing capacitor Cout in FIG. Further, the inductor Lpc and the high speed switching diode D20 of the choke coil PCC have the same functions as the inductor L and the high speed switching diode D shown in FIG. Further, an RC snubber circuit comprising a series connection of a capacitor Csn and a resistor Rsn is connected in parallel to the high-speed switching diode D20 in this figure.

  Switching element Q103 corresponds to switching element Q in FIG. In this case, the power factor / output voltage control IC 120 is an integrated circuit (IC) that controls the operation of the active filter for improving the power factor so that the power factor approaches 1. The multiplier, the divider, An error voltage amplifier, a PWM control circuit, and a drive circuit that outputs a drive signal for driving the switching element Q103 are configured. The voltage obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistor R5 and the voltage dividing resistor R6 is input to the terminal T1 of the power factor / output voltage control IC 120 so as to be rectified and smoothed. A first feedback control circuit having the voltage Ei as a predetermined value is formed.

  Further, a series connection of a voltage dividing resistor R101 and a voltage dividing resistor R102 is provided between the positive electrode output terminal of the bridge rectifier circuit Di and the primary side ground, and a connection point between the voltage dividing resistor R101 and the voltage dividing resistor R102 is a terminal. It connects to T5. Thereby, the rectified voltage of the bridge rectifier circuit Di is divided and inputted to the terminal T5. Further, the voltage of the resistor 103, that is, the voltage corresponding to the source current of the switching element Q103 is input to the terminal T2. Here, the source current of the switching element Q103 is a current contributing to storing magnetic energy in the current I1 flowing through the choke coil PCC. A signal corresponding to the rectified voltage input to the terminal T5 of the power factor / output voltage control IC 120 and a signal corresponding to the voltage envelope input to the terminal T2 (that is, the envelope of the current I1) are similar. A second feedback control circuit is formed.

  Further, the operating power of the power factor / output voltage control IC 120 is supplied to the terminal T4. At this terminal T4, the alternating voltage excited in the coil N5 in the choke coil PCC and transformer coupled to the inductor Lpc is converted into a low-voltage DC voltage by a half-wave rectifier circuit comprising a rectifier diode D11 and a capacitor C11 shown in the figure. Supplied. Further, the terminal T4 is connected to the positive output terminal of the bridge rectifier circuit Di via the starting resistor Rs. After the commercial AC power supply AC is turned on, the rectified output obtained at the positive output terminal of the bridge rectifier circuit Di is supplied to the terminal T4 via the starting resistor Rs during the rise time until the voltage is excited in the winding N5. Supplied. The power factor / output voltage control IC 120 starts operation using the rectified voltage supplied in this manner as a starting power source.

  A drive signal (gate voltage) for driving the switching element is output from the terminal T3 to the gate of the switching element Q103. That is, the envelope of the current I1 is made similar to the first feedback control circuit in which the voltage value divided by the voltage dividing resistor R5 and the voltage dividing resistor R6 is a predetermined value, and the rectified and smoothed voltage Ei. A drive signal for operating the two feedback control circuits with the second feedback control circuit is output to the gate of the switching element Q103. As a result, the waveform of the AC input current IAC flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC, and the power factor is controlled to be approximately 1. That is, power factor improvement is achieved.

  Here, regarding the power factor improvement operation of the active filter shown in FIG. 16, waveforms of respective parts are shown in FIG. 17 and FIG. First, FIG. 17 shows a switching operation (ON: conduction and OFF: disconnection operation) of the switching element Q103 according to the load variation, and a current I1 flowing through the inductor Lpc of the choke coil PCC. FIG. 17A shows the operation at a light load, FIG. 17B shows the operation at an intermediate load, and FIG. 17C shows the operation at a heavy load. As can be seen by comparing FIGS. 17 (a), 17 (b), and 17 (c), the switching element Q103 is turned on as the switching cycle becomes constant and the tendency to heavy load is increased. The period gets longer. In this way, by adjusting the current I1 flowing into the smoothing capacitor Ci through the inductor Lpc according to the load condition, the rectified smoothing voltage Ei is stabilized against the voltage fluctuation of the AC input voltage VAC and the load fluctuation. It is done. For example, the value of the rectified and smoothed voltage Ei is made constant at 380 V with respect to the range of the AC input voltage VAC from 85 V to 264 V. The rectified and smoothed voltage Ei is a voltage across the smoothing capacitor Ci, and is a DC input voltage for the subsequent current resonance type converter.

  FIG. 18 shows the waveforms of the AC input current IAC and the rectified and smoothed voltage Ei by comparison with the AC input voltage VAC. In this figure, experimental results when the value of the AC input voltage VAC is 100 V are shown. As shown in this figure, the waveform of the AC input voltage VAC and the waveform of the AC input current IAC are substantially similar to the passage of time. That is, the power factor is improved. Further, it is shown that the rectified and smoothed voltage Ei is stabilized at an average value of 380 V along with the improvement of the power factor. Further, as shown in the figure, it has a ripple fluctuation of 10 Vp-p with respect to 380 V.

  Returning to FIG. 16 again, the current resonance converter at the latter stage of the active filter will be described. The current resonance type converter performs the switching operation for power conversion by inputting the rectified and smoothed voltage Ei, and forms a current resonance type converter including a switching circuit connected in a half bridge by the switching elements Q101 and Q102. . The current resonance type converter in this case is a separately excited type, and MOS-FETs are used for the switching elements Q101 and Q102. A body diode DD101 and a body diode DD102 are connected in parallel to these MOS-FETs. The switching element Q101 and the switching element Q102 are switched and driven at a required switching frequency by the oscillation / drive circuit 102 at the timing when they are alternately turned on / off. The oscillation / drive circuit 2 is controlled by a signal from the control circuit 1, and the control circuit 1 operates so as to variably control the switching frequency according to the level of the secondary side DC output voltage Eo. The secondary side DC output voltage Eo is stabilized.

  Converter transformer PIT is provided to transmit the switching outputs of switching element Q101 and switching element Q102 from the primary side to the secondary side. One end of the primary winding N1 of the converter transformer PIT is connected to a connection point (switching output point) of the switching element Q101 and the switching element Q102 via the primary side series resonance capacitor C2, and the other end of the primary winding N1 is connected. The end is grounded. Here, a series resonance circuit is formed by the primary side series resonance capacitor C2 and the primary side leakage inductor L1. This series resonance circuit generates a resonance operation when a switching output is supplied by the switching element Q101 and the switching element Q102.

  A secondary winding N2 is wound around the secondary side of the converter transformer PIT. The secondary winding N2 in this case has a secondary winding portion N2A and a secondary winding portion N2B that are center-tapped as shown in the figure, and the center tap is connected to the secondary side ground. Thus, the secondary winding portion N2A and the secondary winding portion N2B are connected to the anodes of the rectifier diode Do1 and the rectifier diode Do2, respectively, and the cathodes of the rectifier diode Do1 and the rectifier diode Do2 are connected to the smoothing capacitor Co. By doing so, a double-wave rectifier circuit is formed. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is input to the control circuit 1 described above.

  FIG. 19 shows the characteristics of power conversion efficiency ηAC → DC (overall efficiency) from AC power to DC power with respect to load fluctuations, power factor PF, and rectified smoothing voltage Ei. In this figure, the characteristic with respect to the fluctuation | variation with the value of the load electric power Po when the value of AC input voltage VAC is 100V from 300W to 0W is shown. FIG. 16 shows characteristics of power conversion efficiency ηAC → DC (total efficiency), power factor PF, and rectified smoothing voltage Ei with respect to fluctuations in the AC input voltage VAC. This figure shows characteristics with respect to fluctuations in the value of the AC input voltage VAC from 85V to 264V under a constant load condition with a load power Po value of 300W.

  First, as shown in FIG. 19, the power conversion efficiency (total efficiency) decreases as the load power Po becomes a heavy load condition. Further, with respect to fluctuations in the AC input voltage VAC, under the same load condition, as shown in FIG. 20, the level of the AC input voltage VAC tends to increase as the level increases. ing. For example, when the load power Po is 300 W and the AC input voltage VAC is 100 V, the power conversion efficiency (total efficiency) is about 83.0%, and when the AC input voltage VAC is 230 V, the power conversion efficiency (total efficiency). Is about 89.0%, and when the AC input voltage VAC is 85V, the power conversion efficiency (total efficiency) is about 80.0%.

  Further, as shown in FIG. 19, the power factor PF has a characteristic that is substantially constant with respect to fluctuations in the load power Po. Further, as shown in FIG. 20, the fluctuation characteristic of the power factor PF with respect to the fluctuation of the AC input voltage VAC is a characteristic that may be regarded as almost constant, although it tends to decrease as the AC input voltage VAC increases. I understand that. For example, when the load power Po is 300 W and the AC input voltage VAC is 100 V, the power factor PF is about 0.96, and when the AC input voltage VAC is 230 V, the power factor PF is about 0.94. can get.

In addition, as shown in FIGS. 19 and 20, the rectified and smoothed voltage Ei has a result that is almost constant with respect to fluctuations in the load power Po and the AC input voltage VAC.
JP-A-6-327246

  As can be seen from the above description, the power supply circuit shown in FIG. 16 is configured by mounting the conventionally known active filter shown in FIG. 14, and the power factor is improved by adopting such a configuration. I am trying.

  However, the power supply circuit having the configuration shown in FIG. 16 has the following problems. First, as the power conversion efficiency in the power supply circuit shown in FIG. 16, the conversion efficiency from the AC power corresponding to the active filter in the previous stage to the DC power, and the conversion efficiency from the DC power to the DC power in the current resonance type converter in the subsequent stage Will be a synthesis of That is, the total power conversion efficiency (total efficiency) of the circuit shown in FIG. 16 is a value obtained by multiplying these power conversion efficiency values, and is a product of numbers that are each 1 or less. Overall efficiency will decrease.

  In addition, since the active filter circuit is a hard switching operation, noise generation is large, so that strict noise suppression measures are required. For this reason, in the circuit shown in FIG. 16, a noise filter including two line filter transformers and three across capacitors is formed for the line of the commercial AC power supply AC. In addition, a normal mode noise filter including one inductor LN and two filter capacitors CN is provided for the rectified output line. Furthermore, an RC snubber circuit is provided for the fast recovery type fast switching diode D20 for rectification. In this way, it is necessary to take measures against noise due to the large number of parts, resulting in an increase in cost and an increase in the mounting area of the power supply circuit board.

  Further, the switching frequency of the switching element Q103 operated by the power factor / output voltage control IC 120 as a general-purpose IC is fixed at 60 kHz, whereas the switching frequency of the subsequent current resonance type converter is in the range of 80 kHz to 200 kHz. Variable. Since the switching timings (clocks) of the two are thus independent of each other, the ground potential interferes and becomes unstable due to the switching operation of the two based on the respective clocks. For example, abnormal oscillation is likely to occur. . As a result, problems such as difficulty in circuit design and deterioration of reliability are also caused.

  The switching power supply circuit according to the present invention includes a rectifying element and a smoothing capacitor that input and rectifies and smoothes AC power from an AC power source, and generates a rectified and smoothed voltage as a voltage across the smoothing capacitor. A converter transformer having a circuit, a choke coil to which the rectified and smoothed voltage is applied to one terminal, a leakage inductor in which one terminal of the primary winding is connected to the other terminal of the choke coil, and the primary winding A primary-side series resonance circuit having a primary-side resonance frequency governed by a primary-side inductance component generated by the leakage inductor and a capacitance of the primary-side series resonance capacitor; , Connected to the other terminal of the choke coil to supply AC power to the primary side series resonance circuit An switching element, an oscillation / drive circuit for driving the switching element, and a secondary winding of the converter transformer is connected to a secondary side series resonance capacitor, and a secondary side inductance component generated by the leakage inductor and the secondary side A secondary side series resonant circuit having a secondary side resonance frequency governed by the capacitance of the side series resonant capacitor, and a secondary side DC output voltage output from the secondary side series resonant circuit by a secondary side rectifier circuit. A control circuit for supplying a control signal such as a predetermined value to the oscillation / drive circuit, and a current corresponding to a component in one direction of a current flowing through the primary side series resonance circuit from the AC power supply, A power factor correction circuit for flowing current from the rectifying element to the smoothing capacitor, the power factor correction circuit including a power factor correction diode and a leakage A power factor improving transformer having a inductor and a filter capacitor, wherein the primary winding of the power factor improving transformer is connected in series with the primary side series resonance circuit, and one of the secondary windings of the power factor improving transformer The terminal is connected to one terminal of the power factor improving diode, the other terminal of the secondary winding of the power factor improving transformer is connected to one terminal of the smoothing capacitor and the filter capacitor, and the power factor improving diode Is connected to the other terminal of the rectifying element and the filter capacitor.

  That is, a rectifying / smoothing circuit that includes a rectifying element that smoothes and rectifies and smoothes by inputting AC power from an AC power source and generates a rectifying and smoothing voltage as a voltage across the smoothing capacitor, and the rectifying and smoothing A converter transformer having a choke coil to which a voltage is applied to one terminal and a leakage inductor to which the other terminal of the primary winding is connected to the other terminal of the choke coil; The rectified and smoothed voltage is applied to the transformer, and the current flowing from the rectified and smoothed circuit to the choke coil can be a direct current. The other terminal of the primary winding is connected to a primary side series resonance capacitor, and has a primary side resonance frequency governed by a primary side inductance component generated by the leakage inductor and a capacitance of the primary side series resonance capacitor. A primary-side series resonance circuit; a switching element connected to the other terminal of the choke coil, for supplying AC power to the primary-side series resonance circuit; and an oscillation / drive circuit for driving the switching element. AC power can be supplied to the converter transformer. The secondary winding of the converter transformer is connected to a secondary side series resonant capacitor, and is controlled by the secondary side inductance component generated by the leakage inductor and the capacitance of the secondary side series resonant capacitor. A secondary side series resonance circuit having a resonance frequency, and a control signal for setting a value of a secondary side DC output voltage output from the secondary side series resonance circuit by a secondary rectifier circuit to a predetermined value are oscillated. Since the control circuit for supplying to the drive circuit is provided, the secondary side DC output voltage can be supplied to the load and the secondary side DC output voltage can be set to a predetermined value. The power factor correction circuit further includes a power factor correction transformer having a power factor correction diode and a leakage inductor, and a filter capacitor. A primary winding of the power factor correction transformer is the primary winding. The power factor improving transformer secondary terminal is connected in series with a side series resonant circuit, and the power factor improving transformer secondary terminal is connected to the other terminal of the power factor improving transformer secondary coil. The terminal is connected to one terminal of the smoothing capacitor and the filter capacitor, and the other terminal of the power factor improvement diode is connected to the other terminal of the rectifier element and the filter capacitor, and the power factor improvement is performed. it can.

  Another switching power supply circuit according to the present invention includes a rectifying element and a smoothing capacitor that receive and rectify and smoothen AC power from an AC power supply, and generates a rectified and smoothed voltage as a voltage across the smoothing capacitor. A rectifying and smoothing circuit; a choke coil to which the rectified and smoothed voltage is applied to one terminal; a converter transformer having a leakage inductor in which one terminal of a primary winding is connected to the other terminal of the choke coil; and the primary The other terminal of the winding is connected to the primary side series resonance capacitor, and the primary side series resonance having the primary side resonance frequency governed by the primary side inductance component generated by the leakage inductor and the capacitance of the primary side series resonance capacitor. Circuit and the other terminal of the choke coil to supply AC power to the primary side series resonance circuit Switching element, an oscillation / drive circuit for driving the switching element, and a secondary winding of the converter transformer is connected to a secondary side series resonant capacitor, and the secondary inductance component generated by the leakage inductor and the second A secondary side series resonance circuit having a secondary side resonance frequency governed by the capacitance of the secondary side series resonance capacitor, and a secondary side DC output voltage output from the secondary side series resonance circuit by a secondary side rectifier circuit A control circuit that supplies a control signal that sets the value of the current to a predetermined value to the oscillation / drive circuit, and a current corresponding to a unidirectional component of the current that flows in the primary side series resonance circuit is supplied from the AC power supply. A power factor correction circuit for flowing a current from the rectifying element to the smoothing capacitor, and the power factor correction circuit includes a power factor correction diode and A power factor improving variable inductor having a variable ductance value and a filter capacitor, and one terminal of the power factor improving variable inductor is two choke coils magnetically coupled to a primary winding of the choke coil. The other terminal of the power factor improving variable inductor is connected to one terminal of the power factor improving diode, and the other terminal of the secondary winding of the choke coil is connected to the smoothing terminal. A capacitor and one terminal of the filter capacitor, and the other terminal of the power factor improving diode is connected to the other terminal of the rectifying element and the filter capacitor. The inductance of the power factor improving variable inductor Varies depending on the magnitude of the current flowing through the rectifying element and / or the magnitude of the rectified smoothing voltage.

  That is, a rectifying / smoothing circuit that includes a rectifying element that smoothes and rectifies and smoothes by inputting AC power from an AC power source and generates a rectifying and smoothing voltage as a voltage across the smoothing capacitor, and the rectifying and smoothing A converter transformer having a choke coil to which a voltage is applied to one terminal and a leakage inductor to which the other terminal of the primary winding is connected to the other terminal of the choke coil; The rectified and smoothed voltage is applied to the transformer, and the current flowing from the rectified and smoothed circuit to the choke coil can be a direct current. The other terminal of the primary winding is connected to a primary side series resonance capacitor, and has a primary side resonance frequency governed by a primary side inductance component generated by the leakage inductor and a capacitance of the primary side series resonance capacitor. A primary-side series resonance circuit; a switching element connected to the other terminal of the choke coil, for supplying AC power to the primary-side series resonance circuit; and an oscillation / drive circuit for driving the switching element. AC power can be supplied to the converter transformer. The secondary winding of the converter transformer is connected to a secondary side series resonant capacitor, and is controlled by the secondary side inductance component generated by the leakage inductor and the capacitance of the secondary side series resonant capacitor. A secondary side series resonance circuit having a resonance frequency, and a control signal for setting a value of a secondary side DC output voltage output from the secondary side series resonance circuit by a secondary rectifier circuit to a predetermined value are oscillated. Since the control circuit for supplying to the drive circuit is provided, the secondary side DC output voltage can be supplied to the load and the secondary side DC output voltage can be set to a predetermined value. The power factor correction circuit further includes a power factor correction diode, a power factor correction variable inductor having a variable inductance value, and a filter capacitor, and one of the power factor correction variable inductors. Is connected to one terminal of the secondary winding of the choke coil that is magnetically coupled to the primary winding of the choke coil, and the other terminal of the power factor improving variable inductor is connected to the power factor improving diode. One terminal is connected, the other terminal of the secondary winding of the choke coil is connected to one terminal of the smoothing capacitor and the filter capacitor, the other terminal of the power factor correction diode is the rectifier element and An inductance value of the power factor improving variable inductor formed in the other terminal of the filter capacitor flows to the rectifying element. It is power factor correction changes according to the size of the size or / and the rectified smoothed voltage current.

  According to the switching power supply circuit of the present invention, it is possible to omit the active filter and to have a power factor improving function. By omitting the active filter, the power conversion efficiency characteristic of the switching power supply circuit is improved. Then, the heat sink and the like can be omitted and reduced. In addition, the number of parts is greatly reduced as compared with a configuration including an active filter, so that the circuit can be reduced in size and weight and the cost can be reduced. The active filter has a hard switching operation, whereas the switching converter of the present invention has a soft switching operation because it is based on a resonant converter. As a result, the switching noise is greatly reduced, which contributes to the reduction in size and weight and cost of the noise filter. Further, since there are no plural clocks having different frequencies, the problem of mutual interference due to plural clock frequencies does not occur, the reliability is improved, and the pattern design of the circuit board is facilitated.

  (First embodiment)

  The switching power supply circuit according to the first embodiment will be described with reference to FIG. The switching power supply circuit according to the first embodiment applies a so-called class E switching converter to the power supply circuit and improves the power factor based on the resonance current of the current resonance circuit.

  The switching power supply circuit of the first embodiment shown in FIG. 1 will be described below in order from the commercial AC power supply AC side. A two-phase input line of the commercial AC power supply AC is connected to a bridge rectifier circuit Di, which is a kind of rectifier element, through a common mode noise filter including a common mode choke coil CMC and two across capacitors CL. Here, the common mode noise filter has a function of removing common mode noise generated between the line of the commercial AC power supply AC and the secondary side of the switching power supply circuit.

  AC power is rectified by a bridge rectifier circuit Di formed by bridge-connecting four low-speed rectifier elements (diodes) to generate pulsating power, and the pulsating power is improved in power factor with a fast switching speed. The smoothing capacitor Ci is charged through the diode D1. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. That is, the bridge rectification circuit Di and the smoothing capacitor Ci constitute a rectification smoothing circuit. Here, the power factor correction diode D1 constitutes a part of a power factor correction circuit 10 to be described later, and the process in which the current from the AC power source is charged to the smoothing capacitor Ci via the power factor correction diode D1 will be described. The operation of the improvement circuit 10 will be described later. Here, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent class E switching converter.

  The class E switching converter is formed mainly of a choke coil PCC, a converter transformer PIT, a primary side series resonance capacitor C2, and a switching element Q1. That is, the class E switching converter is configured as follows. The rectified and smoothed voltage Ei is applied to one terminal of the choke coil PCC. Then, one terminal of the primary winding N1 of the converter transformer PIT having a leakage inductor is connected to the other terminal of the choke coil PCC. The other terminal of the primary winding N1 is connected to the primary side series resonant capacitor C2, and the primary side inductance component generated by the leakage inductor (inductance component corresponding to the equivalent inductor represented by the inductor L1 in FIG. 1) and the primary side A primary side series resonance circuit having a primary side resonance frequency that is dominated by the capacitance of the series resonance capacitor C2 is formed. And switching element Q1 which supplies alternating current power to a primary side series resonance circuit is connected to the other terminal of choke coil PCC. Here, the oscillation / drive circuit 2 drives the switching element Q1, the switching element Q1 is a MOS-FET, and the converter transformer PIT has a structure with a relatively large leakage inductance value by reducing the value of the coupling coefficient k. It is said that. The structure of the converter transformer PIT will be described later.

  The secondary winding N2 of the converter transformer is connected to the secondary side series resonant capacitor C3, and the secondary side inductance component generated by the leakage inductor (inductance component corresponding to the equivalent inductor represented by the inductor L2 in FIG. 1) and A secondary side series resonance circuit having a secondary side resonance frequency controlled by the capacitance of the secondary side series resonance capacitor is formed. Then, the secondary side DC output voltage Eo output from the secondary side series resonance circuit by the secondary side rectifier circuit (formed by the rectifier diode Do1, the rectifier diode Do2 and the smoothing capacitor Co) is set to a predetermined value. A control circuit 1 for supplying such a control signal to the oscillation / drive circuit 2 is provided.

  Further, a power factor improving circuit 10 is provided for flowing a current corresponding to a unidirectional component of the current flowing through the primary side series resonant circuit from the commercial AC power supply AC and flowing the current from the bridge rectifier circuit Di to the smoothing capacitor Co. ing. The power factor correction circuit 10 includes a power factor correction diode D1, a power factor correction transformer VFT, and a filter capacitor CN. The filter capacitor CN also functions as a filter for removing normal mode noise.

  That is, as described above, in the first embodiment, power transmission is performed by the class E switching converter, the primary side series resonant capacitor C2 constituting the class E switching converter is connected to the power factor correction circuit 10, A secondary side series resonance circuit is also configured on the secondary side, and a control circuit 1 is provided to oscillate a signal from the control circuit 1 in order to set the value of the secondary side DC output voltage to a predetermined value. A feature of the first embodiment is that the gate of the switching element Q1 is driven by a drive signal output from the oscillation / drive circuit 2 in addition to the drive circuit 2.

  Hereinafter, a more detailed configuration of the switching power supply circuit according to the first embodiment shown in FIG. 1 will be described.

  The converter transformer PIT has functions of insulating the primary side and the secondary side and performing voltage conversion, and also functions as an inductor L1 that constitutes a part of a resonance circuit for causing the class E switching converter to function. To do. Here, the inductor L1 is an inductance component formed by the leakage inductor of the converter transformer PIT. A specific structure of the converter transformer PIT in the first embodiment will be described along a cross-sectional view of the converter transformer PIT shown in FIG.

  The converter transformer PIT includes an EE type core (EE-shaped core) in which an E type core CR1 and an E type core CR2 made of a ferrite material are combined so that their magnetic legs face each other. The primary side and secondary side winding portions are divided so as to be independent from each other, and provided with a bobbin B formed of, for example, resin. The primary winding N1 is wound around one winding region of the bobbin B. Further, the secondary winding N2 is wound around the other winding region. By attaching the bobbin B on which the primary winding N1 and the secondary winding N2 are wound to the EE-shaped core in this way, the primary winding N1 and the secondary winding N2 have different EE-shaped shapes, respectively. It will be in the state wound by the central magnetic leg of a core. In this way, the overall structure of the converter transformer PIT is obtained.

  A gap G of 2.2 mm is formed with respect to the central magnetic leg of the EE-shaped core. As a result, 0.67 is obtained as the value of the coupling coefficient k between the primary side and the secondary side. In this way, the primary side and the secondary side are preferably loosely coupled with a coupling coefficient of 0.7 or less so as to obtain a leakage inductor L1 having a large inductance value. The gap G is formed by making the central magnetic legs of the E-type core CR1 and the E-type core CR2 shorter than the two outer magnetic legs. The number of turns of the primary winding N1 was 58T (turns), the number of turns of the primary winding N1 was 30T (turns), and the core material was EER-35 (core material name).

  As described above, a MOS-FET is selected as the switching element Q1, and a body diode DD is incorporated in parallel between the source and the drain. The inductance value of the inductor L10 (choke coil PCC) is 1 mH (Milli Henry). The value of the primary side series resonance capacitor C2 was 0.018 μF (micro farad), and the value of the capacitor C1 was 5600 pF (pico farad).

  On the secondary side of the converter transformer PIT, a voltage waveform similar to the alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. A secondary side series resonant capacitor C3 is connected in series to the secondary winding N2. Thus, a secondary side series resonance circuit is formed by the leakage inductor L2 and the secondary side series resonance capacitor C3 viewed from the secondary winding N2 side. The resonance frequency of the secondary side series resonance circuit is set to be substantially equal to the frequency of the primary side series resonance frequency determined by the primary side series resonance capacitor C2 and the leakage inductor L1.

  The secondary side rectifier circuit connects the rectifier diode Do1 and the rectifier diode Do2 and the smoothing capacitor Co, which operate at high speed, to the secondary winding N2 to which the secondary side series resonant capacitor C3 is connected in series. , Formed as a voltage doubler half-wave rectifier circuit. In the voltage doubler half-wave rectifier circuit, the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 are connected to one end of the secondary winding N2 via the secondary side series resonant capacitor C3. Further, the cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The other end of the secondary winding N2, the anode of the rectifier diode Do2, and the negative terminal of the smoothing capacitor Co are connected to the secondary side ground. Here, the value of the secondary side series resonance capacitor C3 was set to 0.082 μF.

  The control circuit 1 supplies a detection output corresponding to the difference between the input secondary side DC output voltage Eo and a predetermined reference voltage value to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching element Q <b> 1 is driven mainly by changing the switching frequency according to the input detection output of the control circuit 1. Further, the time ratio, which is the ratio of the time during which the switching element Q1 is turned on in one cycle, may be changed together with the switching frequency.

  In this way, the switching frequency of the switching element Q1 is variably controlled, whereby the primary and secondary resonance impedances in the power supply circuit change, and the converter transformer PIT changes from the primary winding N1 to the secondary winding N2 side. The amount of power to be transmitted and the amount of power to be supplied from the secondary side rectifier circuit to the load will change. As a result, an operation for matching the magnitude of the secondary side DC output voltage Eo with the reference voltage is obtained. That is, the secondary side DC output voltage Eo is stabilized.

  Next, the power factor correction circuit 10 will be described. This power factor correction circuit 10 is provided so as to be inserted into a rectified current path in a rectifying and smoothing circuit for obtaining a rectified and smoothed voltage Ei from a commercial AC power supply AC. The power factor correction circuit 10 of the first embodiment includes a power factor correction transformer VFT, a power factor correction diode D1, and a filter capacitor CN. In the power factor correction circuit 10, the primary winding N11 of the power factor improvement transformer VFT is connected in series with the primary side series resonance circuit, and one terminal of the secondary winding N12 of the power factor improvement transformer VFT is power factor improvement. The other terminal of the secondary winding N12 of the power factor improving transformer VFT is connected to one terminal of the smoothing capacitor Ci and the filter capacitor CN, and the other terminal of the power factor improving diode D1 is connected to one terminal of the diode D1. Is connected to the other terminal of the bridge rectifier circuit Di and the filter capacitor CN.

  In the first embodiment, the inductance value of the primary side leakage inductor (represented by an inductor L11 as an equivalent inductor) of the power factor improvement transformer VFT is 82 μH (micro Henry), and the secondary side leakage of the power factor improvement transformer VFT is The inductance value of the inductor (represented by the inductor L12 as an equivalent inductor) was 45 μH (micro Henry). At this time, EE-25 was used as the core material of the power factor improving transformer VFT. The electromagnetic coupling coefficient between the primary winding N11 and the secondary winding N12 of the power factor correction transformer VFT is set to 0.77. The value of the filter capacitor CN was 1 μF (micro farad).

  Next, the operation of each part of the switching power supply circuit of the first embodiment will be described in order. First, the operation of the class E switching converter, which is one main part of the first embodiment, will be described. For the sake of simplicity, the operation of the class E switching converter will be described assuming that there is no power factor correction circuit 10 at first. The class E switching converter is equivalent to a primary side converted load impedance obtained by converting a load impedance connected to the primary side primary resonance capacitor C2, the leakage inductor L1, and the secondary side into the primary side and the switching element Q1 ( In a series resonance circuit including a body diode DD, which is the same unless otherwise specified in the following description), a rectified smoothing voltage is connected to a connection point between the switching element Q1 and the leakage inductor L1 via an inductor L10. It can be considered that Ei is supplied.

  In the class E switching converter having such a configuration, the primary side primary resonance capacitor C2, the leakage inductor L1, the primary side equivalent load impedance (the secondary side load is the primary winding) by turning on and off the switching element Q1. And an equivalent impedance inserted as a load in series with the primary-side series resonance circuit by electromagnetic coupling between the secondary winding and the secondary winding in series, thereby forming a resonance circuit with a high Q value. At this time, the current flowing through the resonance circuit has a high Q value, so that it is almost a sinusoidal current.

  On the other hand, since the inductance value of the inductor L10 (choke coil PCC) has a very large impedance with respect to the resonance frequency of the resonance circuit, the current flowing through the inductor L10 is close to direct current. As a result, the voltage generated at both ends of the inductor L10 is also a sine wave, and in terms of alternating current, the voltage generated at both ends of the switching element Q1 and the capacitor C1 connected in parallel with the inductor L10 is also a sine wave. It becomes. Since the current flowing through the inductor L10 can be regarded as a direct current, both the waveform of the current flowing through the primary winding N1 of the converter transformer PIT and the waveform of the current flowing through the switching element Q1 and the capacitor C1 are close to a sine wave. The currents flowing through the inductor L10, which are phase-adjusted and added to each other, are close to direct current as described above. The voltage generated in primary winding N1 and secondary winding N2 of converter transformer PIT is also a sine wave. The inductor L10 and the capacitor C1 form an additional parallel resonant circuit to further improve the resonance characteristics.

  Next, the operation on the secondary side will be described. In a half cycle corresponding to one polarity of the alternating voltage obtained in the secondary winding N2, a forward voltage is applied to the rectifier diode Do2, so that the rectifier diode Do2 conducts and the rectified current is converted into the secondary voltage. An operation of charging the side series resonant capacitor C3 is obtained. As a result, a voltage across the secondary side series resonant capacitor C3 is generated at a level corresponding to an equal multiple of the alternating voltage level induced in the secondary winding N2. In a half cycle corresponding to the other polarity of the alternating voltage obtained in the secondary winding, a forward voltage is applied to the rectifier diode Do2 to conduct. At this time, the smoothing capacitor Co is charged with a potential in which the voltage generated in the secondary winding N2 and the voltage across the secondary side series resonance capacitor C3 are superimposed. As a result, the secondary side DC output voltage Eo having a level corresponding to twice the alternating voltage level induced in the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co. In this rectification operation, the smoothing capacitor Co is charged only in one half cycle of the alternating voltage excited by the secondary winding N2. That is, a rectification operation as a double voltage half wave is obtained. The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  Here, since the resonance frequency of the resonance circuit of the primary side and the resonance frequency of the resonance circuit of the secondary side are set to be approximately equal, the power transmission characteristic with respect to the frequency from the primary side to the secondary side is slight. It becomes very sensitive to frequency fluctuations. That is, in the first embodiment, a series resonance circuit having a resonance frequency similar to that of the primary side is also arranged on the secondary side of the class E switching converter, so that the amount of power transmitted due to a slight frequency difference is increased. It can be changed. This means that when the secondary side DC output voltage Eo is kept at a predetermined value, the control circuit 1 sends a signal for changing the frequency very slightly regardless of a wide range of load fluctuations. This means that a stable constant voltage characteristic can be obtained if the switching element Q1 is supplied to the switching element Q1. In this way, the switching frequency of the switching element Q1 is variably controlled, so that the primary and secondary resonance impedances in the power supply circuit change and are transmitted from the primary winding N1 to the secondary winding N2 of the converter transformer PIT. The amount of power to be changed will change. As a result, the value of the secondary side DC output voltage Eo is maintained at a predetermined value. In this embodiment, the resonance frequency of the resonance circuit of the primary side and the resonance frequency of the resonance circuit of the secondary side are set to be approximately equal, but the relationship between the frequencies can be determined as appropriate.

  Next, the operation of the power factor correction circuit 10, which is another main part of the first embodiment, will be described. In the above description of the operation of the class E switching converter, it has been described that the power factor correction circuit 10 does not exist, but actually, the current of the primary side series resonance circuit flows through the power factor correction circuit 10. Here, it is assumed that the high-frequency resonance current flows in the same direction in the same direction in both directions. As a low-frequency component, no current flowing in one direction is generated, and no AC input current IAC from the commercial AC power supply AC is generated. It does not give the effect of expanding the flow time. That is, there is no power factor improvement effect. Therefore, the power factor correction diode D1 and the power factor correction transformer VFT are provided so that only the current in one direction contributes to the increase of the AC input current IAC. Thereby, the resonance current flows through different current paths in the positive direction and the negative direction. That is, the inductor L1, the primary side series resonance capacitor C2, and the power factor improving transformer VFT are black circles (the power of FIG. 1) of the primary winding N11 of the power factor improving transformer VFT among the AC series resonance currents flowing through the primary winding N11. The current flowing in the direction from the factor improving transformer VFT is converted into the current I1 by the power factor improving transformer VFT and flows through the power factor improving diode D1, thereby improving the power factor.

  Further, since the power factor improving transformer VFT has the primary winding N11 and the secondary winding N12, adjusting the ratio of the primary winding N11 and the secondary winding N12 makes the power factor improving effect more appropriate. Can be achieved. Further, the primary winding N11 of the power factor correction transformer VFT is connected in series with the primary winding N1 of the converter transformer PIT to form a part of the primary side series resonant circuit, and therefore the ratio of the leakage inductor L11 and the leakage inductor L1 is set. By adjusting the power factor improvement characteristic by the leakage inductor L11, the power transmission characteristic between the primary side and the secondary side can be optimized by the leakage inductor L1.

  That is, the unidirectional flow of the primary side resonance current described above flows from the inductor L1 to the primary side series resonance capacitor C2, and the current from the primary winding N11 of the power factor correction transformer VFT and the secondary winding N12 ( The current I1 from the power factor correction diode D1) merges and further flows through the smoothing capacitor Ci. The current in the other direction flows from the smoothing capacitor Ci to the primary winding N11 of the power factor correction transformer VFT, the primary-side series resonance capacitor C2 on the primary side, and the inductor L1. Thus, without the power factor correction circuit 10, when the voltage of the smoothing capacitor Ci is larger than the magnitude of the pulsating voltage obtained from the bridge rectifier circuit Di, the current from the bridge rectifier circuit Di is interrupted and short. Although only the time flows, by providing the power factor correction circuit 10, a current based on the primary side DC resonance current flows from the bridge rectifier circuit Di to the smoothing capacitor Ci, and the conduction period of the pulsating current is increased. become longer. Then, the conduction period of the AC input current IAC is also substantially coincident with the conduction period of the rectified current, and the power factor is improved.

  (Operation waveform and measurement data of main part of the first embodiment)

  FIG. 3 shows an operation waveform of a main part of the switching power supply circuit according to the first embodiment shown in FIG. 1, and FIGS. 4 and 5 show measurement data.

  FIG. 3 shows the operation waveform of the main part of the power factor correction circuit 10 by the commercial AC power supply cycle. From the upper stage to the lower stage of FIG. 3, an AC input voltage VAC (see FIG. 1), an AC input current IAC (see FIG. 1), a voltage V2 (see FIG. 1), a current I1 (see FIG. 1), Each of voltage V3 (refer FIG. 1) and (DELTA) Eo which is the ripple component of the secondary side DC output voltage Eo are shown. In the waveform diagram of voltage V2 and current I1 in FIG. 3, the shaded portion indicates that switching is performed in the same cycle as the switching waveform of switching element Q1. Here, the current I1 flowing through the secondary winding N12 of the power factor improvement transformer VFT and the power factor improvement diode D1 is based on the resonance current as described above, and this current extends the distribution period of the AC input current IAC. FIG. 3 shows that this is done. Further, the secondary side DC output voltage Eo has an average value of 175 V (volt) as a predetermined value, and a ripple voltage of 80 mVp-p (peak-to-peak value) having a period of half of the AC input voltage VAC is superimposed thereon.

  FIG. 4 shows the rectified smoothing voltage Ei and force against load fluctuation when the value of the load power Po is in the range of 0 W (no load) to 300 W (maximum load power) under the input voltage condition where the value of the AC input voltage VAC is 100V. The power conversion efficiency ηAC → DC of the DC output power with respect to the rate PF and the AC input power is shown. FIG. 5 shows the rectified smoothing voltage Ei, power factor PF, and power in a range in which the value of the AC input voltage VAC is changed from 85 V to 144 V under a load condition where the load power is constant at 300 W (maximum load power). The conversion efficiency ηAC → DC is shown.

  As shown in FIG. 4, the rectified and smoothed voltage Ei has a tendency to gradually increase as the load becomes lighter. However, when the load power value is 100 W or more, the rectified and smoothed voltage Ei remains almost constant. On the other hand, it changes in proportion to the change of the AC input voltage VAC. Further, as shown in FIG. 4, the power factor PF is maintained at 0.75 or more in the range of 50 W to 300 W, and it can be said that a practically sufficient power factor value is obtained. As shown in FIG. 5, the characteristics of the AC input voltage VAC are almost constant as shown in FIG.

  As for the power conversion efficiency ηAC → DC, as shown in FIG. 4 and FIG. 5, the load power value peaks around 100 W, but with respect to load fluctuation, A power conversion efficiency ηAC → DC of 90% or more is obtained in the range of load power 50 W to load power 300 W. With respect to the fluctuation of the AC input voltage VAC, a substantially constant 90% or more is maintained in the range of 85V to 144V.

  In the switching power supply circuit of the first embodiment shown in FIG. 1, the power conversion efficiency ηAC → DC is improved as compared with the case of the switching power supply circuit shown as the background art in FIG. This is mainly because the configuration including the power factor correction circuit 10 eliminates the need for an active filter. That is, in this embodiment, the total efficiency is not reduced by the product of the two power conversion efficiency values of the former stage and the latter stage as in the case where the active filter is provided.

  In the switching power supply circuit according to the first embodiment, the number of circuit components can be reduced by eliminating the need for an active filter. In other words, as can be seen from the description with reference to FIG. 16, the active filter includes a switching element Q103 and a power factor / output voltage control IC 120 for driving the switching element Q103, and the like. On the other hand, in the switching power supply circuit of the first embodiment, a filter capacitor CN, a power factor improving diode D1, and a power factor improving transformer VFT may be provided as additional components necessary for improving the power factor. Compared with, the number of parts can be very small. As a result, the power supply circuit having the power factor correction function can be manufactured at a much lower cost than the circuit shown in FIG. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight.

  Further, in the switching power supply circuit of the first embodiment, the operation of the class E resonance converter and the power factor correction circuit 10 is a so-called soft switching operation. Therefore, switching noise is compared with the circuit using the active filter shown in FIG. The level of is greatly reduced. In particular, since the current input to the class E switching converter can be linked to a direct current, the level of switching noise can be made extremely small. Further, since the power factor improving transformer VFT has a primary winding N11 and a secondary winding N12, the power factor improving effect can be optimized by adjusting the ratio of the primary winding N11 and the secondary winding N12. You can plan. Further, the primary winding N11 of the power factor correction transformer VFT is connected in series with the primary winding N1 of the converter transformer PIT to form a part of the primary side series resonant circuit, and therefore the ratio of the leakage inductor L11 and the leakage inductor L1 is set. By adjusting, the optimization of the primary side series resonance circuit and the optimization of the power factor improvement characteristic can be independently achieved to some extent, so that the degree of freedom in the configuration of the switching power supply can be increased.

  In addition, since the switching circuit of the first embodiment includes the secondary side series resonance circuit together with the primary side series resonance circuit, the secondary side DC output voltage Eo is maintained at a predetermined voltage by a very slight frequency change. The noise filter can be designed easily. For this reason, if a one-stage noise filter comprising one common mode choke coil CMC and two across capacitors CL is provided, it is possible to sufficiently satisfy the power source disturbance standard. Moreover, sufficient measures can be taken with respect to normal mode noise of the rectified output line by using only one filter capacitor CN.

  Further, the switching element Q1, the rectifier diode Do1 and the rectifier diode Do2 on the secondary side, the power factor improving diode D1, the power factor improving transformer VFT, and the like operate in synchronization with the switching element Q1. Therefore, as in the power supply circuit of FIG. 16, the ground potential does not interfere with the active filter side and the subsequent switching converter, and can be stabilized regardless of changes in the switching frequency.

  Further, the power factor PF obtained by the switching power supply circuit of the first embodiment has extremely good characteristics as shown in FIGS. 4 and 5, and according to such power factor characteristics, the power harmonics The wave distortion regulation can be cleared, and it can be said that a practically sufficient power factor is obtained. In this way, the power supply circuit of the present embodiment shown in FIG. 1 obtains a power factor improving power supply that solves various problems of a power supply circuit including an active filter.

  (Second Embodiment)

  FIG. 6 shows a switching power supply circuit according to the second embodiment, which is a modification of the switching power supply circuit according to the first embodiment. In this figure, the same parts as those shown in FIG. In the switching power supply circuit of the second embodiment, a power factor correction circuit 11 is provided. In the power factor correction circuit 11, a low-speed bypass diode D1A is additionally provided in addition to the configuration of the power factor correction circuit 10 shown in FIG. The bypass diode D1A has an anode connected to the positive output terminal of the bridge rectifier circuit Di and a cathode connected to the positive terminal of the smoothing capacitor Ci. Therefore, in the power factor correction circuit 11, the bypass diode D1A is connected in parallel to the input side of the power factor correction circuit 10. By doing so, the rectified current from the bridge rectifier circuit Di also branches and flows to the bypass diode D1A, so the amount of rectified current flowing to the power factor correction diode D1 is reduced. Thereby, the switching loss in the power factor correction diode D1 is reduced and the power conversion efficiency is improved. In particular, since the amount of current flowing through the power factor correction diode D1 that tends to be heavy load increases, the efficiency improvement effect becomes remarkable.

  (Third embodiment)

  FIG. 7 shows a switching power supply circuit according to a third embodiment which is a modification of the switching power supply circuit according to the first embodiment. In this figure, the same parts as those shown in FIG. The difference between the switching power supply circuit of the first embodiment and the switching power supply circuit of the third embodiment is that a power factor correction circuit 12 is used instead of the power factor correction circuit 10. The power factor improvement circuit 10 is different from the power factor improvement circuit 10 in that a control transformer PRT that can change the inductance value is used instead of the power factor improvement transformer VFT that has a fixed inductance value. Is.

  In order to control the control transformer PRT, a circuit for detecting a signal SDi corresponding to the current flowing in the bridge rectifier circuit Di, a circuit for detecting a signal SEi corresponding to the magnitude of the rectified smoothing voltage Ei, and the magnitude of the signal SDi 1 and / or further differs from the switching power supply circuit of the first embodiment shown in FIG. 1 in that the power factor control circuit 3 that controls the inductance of the control transformer PRT according to the magnitude of the signal SEi is provided. As described above, the filter capacitor CN having the same reference numeral in the power factor correction circuit 10 has a capacitance of 1 μF, and the power factor correction diode D1 is the same as that in the power factor correction circuit 12.

  The signal SDi corresponding to the current flowing through the bridge rectifier circuit Di is detected by detecting the voltage across the resistor R51, and the signal SEi corresponding to the magnitude of the rectified and smoothed voltage Ei is detected using the rectified and smoothed voltage Ei. The voltage was divided by the resistor R52 and the resistor R53.

  FIG. 8 shows a structural example of the control transformer PRT. The control transformer PRT has four magnetic poles, a three-dimensional core (using a size of 16 mm × 16 mm × 2.6 mm) in which a core CR11 and a core CR12 having the same shape are combined, a primary winding N11 and a secondary winding It is formed having a winding N12 and a winding Nc. Then, the primary winding N11 (12T (number of turns)), the secondary winding N12 (15T) and the winding Nc (1000T) are orthogonally wound around the magnetic poles as shown in FIG. Has been. The primary winding N11 and the secondary winding N12 of the control transformer PRT having such a structure function as a power factor correction transformer VFT, and the primary side leakage of the power factor improvement transformer VFT depends on the magnitude of the current flowing through the winding Nc. The inductance values of the inductor L11 and the secondary side leakage inductor L12 can be controlled. In the control transformer PRT used in the present embodiment, when the value of the current flowing through the winding Nc is 5 mA (milliampere), the primary side leakage inductor of the power factor improving transformer VFT formed by the winding No. of the control transformer PRT. The value of L11 is 80 μH (micro henry), and when the value of the current flowing through the winding Nc is 50 mA, the value of the primary leakage inductor L11 of the power factor improving transformer VFT is 20 μH. .

  The power factor control circuit 3 is a normal voltage-current converter, and controls the magnitude of the current flowing through the winding Nc of the control transformer PRT according to the magnitude of the signal SDi and / or the magnitude of the signal SEi. As a result, the value of the leakage inductance of the power factor correction transformer VFT is controlled according to the magnitude of the signal SDi and / or the magnitude of the signal SEi, and according to the change of the current flowing through the bridge rectifier circuit Di, or rectified and smoothed. The power factor improving transformer VFT is adjusted so that the power factor becomes good according to the change in the magnitude of the voltage Ei, and further according to the change in the current flowing in the bridge rectifier circuit Di and the change in the magnitude of the rectified smoothing voltage Ei. The value of the leakage inductance can be adjusted.

  FIG. 9 and FIG. 10 show the characteristics of the switching power supply circuit according to the third embodiment having the power factor correction circuit 12 having such a control transformer PRT as a component. In addition, since the operation waveform of each part is the same as in FIG. 3, description is abbreviate | omitted.

  FIG. 9 shows the rectified smoothing voltage Ei and force against load fluctuation when the value of the load power Po is in the range of 0 W (no load) to 300 W (maximum load power) under the input voltage condition where the value of the AC input voltage VAC is 100V. The power conversion efficiency ηAC → DC of the DC output power with respect to the rate PF and the AC input power is shown. FIG. 10 shows a rectified smoothing voltage Ei, power factor PF, and power in a range where the value of the AC input voltage VAC is changed from 85 V to 144 V under a load condition in which the load power is constant at 300 W (maximum load power). The conversion efficiency ηAC → DC is shown. Here, in FIG. 9 and FIG. 10, the solid line is a characteristic when the control transformer PRT in which the value of the leakage inductance changes is used as the power factor improving transformer VFT, and the broken line is fixed instead of the power factor improving transformer VFT. This is a characteristic when a fixed inductor having an inductance value is used.

  As shown in FIG. 9, when the control transformer PRT is used, the rectified and smoothed voltage Ei has a higher rectified voltage in the low load region than the case where the fixed inductor is used for load fluctuation. . This is because the period during which current flows from the bridge rectifier circuit Di to the smoothing capacitor Ci rises and the value of the rectified smoothing voltage Ei also rises in response to the power factor being greatly improved at low load. It is done. On the other hand, the rectified and smoothed voltage Ei changes substantially in proportion to the change in the AC input voltage VAC. Further, as shown in FIG. 9, the power factor PF is greatly improved in the region where the load power amount is small, and reaches 0.85 regardless of the load fluctuation in the range of the load size from 50 W to 300 W. It can be said that a practically sufficient power factor value is obtained. As shown in FIG. 10, the AC input voltage VAC is substantially constant and has a high power factor characteristic of 0.9 or more.

  As for the power conversion efficiency ηAC → DC, as shown in FIG. 9 and FIG. 10, the load power value peaks around 100 W with respect to the load fluctuation, but it can be regarded as almost constant, A power conversion efficiency ηAC → DC of 90% or more is obtained in the range of load power 50 W to load power 300 W. With respect to the fluctuation of the AC input voltage VAC, a substantially constant 90% or more is maintained in the range of 85V to 144V.

  (Fourth embodiment)

  FIG. 11 shows a switching power supply circuit according to a fourth embodiment which is a switching power supply circuit according to another embodiment. In this figure, the same parts as those in the first embodiment shown in FIG. The switching power supply circuit of the first embodiment and the switching power supply circuit of the fourth embodiment are common in that a class E converter is used. However, it is different in that the voltage generated in the choke coil PCC is used instead of using the current of the primary side series resonance circuit for power factor improvement. Accordingly, the power factor improvement circuit 13 adopts a configuration different from that of the power factor improvement circuit 10.

  That is, the power factor correction circuit 13 includes a power factor correction diode D1, a power factor correction variable inductor Lo (consisting of a primary winding No and a core) whose inductance value is variable, and a filter capacitor CN. One terminal of the power factor improving variable inductor Lo is connected to one terminal of the secondary winding N4 of the choke coil magnetically coupled to the primary winding N3 of the choke coil PCC, and the power factor improving variable inductor Lo is connected. Is connected to one terminal of the power factor improving diode D1, and the other terminal of the secondary winding N4 of the choke coil PCC is connected to one terminal of the smoothing capacitor Co and the filter capacitor CN. The other terminal of the improvement diode D1 is connected to the other terminal of the bridge rectifier circuit Di and the filter capacitor CN. Here, the value of the inductance of the power factor improving variable inductor Lo changes according to the magnitude of the current flowing through the bridge rectifier circuit Di and / or the magnitude of the rectified smoothing voltage Ei.

  Here, the power factor improving variable inductor Lo is provided with a winding No in place of the primary winding N11 and the secondary winding N12 in the control transformer PRT having the same structure as shown in FIG. The value of the power factor improving variable inductor Lo having such a structure can be controlled by the magnitude of the current flowing through the winding Nc.

  Here, the relationship between the operation of the class E converter in the fourth embodiment and the effect of power factor improvement will be described. Unlike the first embodiment, in the fourth embodiment, a current corresponding to the resonance current from the primary side series resonance circuit does not flow directly into the power factor correction circuit 13. As described above, the primary winding N1 (inductor L1) of the converter transformer PIT, the primary side series resonance capacitor C2, and the series winding of the series circuit of the choke coil PCC and the series winding of the load converted to the primary side. A class E converter is configured by N3 (inductor L3). Here, as described above, the voltage waveform generated in the primary winding N3 of the choke coil PCC is close to the sine wave of the frequency of the primary side series resonance frequency, so that it is generated in the secondary winding N4 of the choke coil PCC. The voltage waveform to be used is also a sine wave having a frequency of the primary side series resonance frequency.

  The polarity of the sine wave generated in the secondary winding N4 of the choke coil PCC is alternately switched between positive and negative. However, only when the polarity is one, the current I1 flows through the power factor improving diode D1 and the power factor improving variable inductor Lo, The power factor is improved by the current I1. Further, in order to control the value of the power factor improving variable inductor Lo, the signal SDi corresponding to the current flowing through the bridge rectifier circuit Di is detected, the signal SEi corresponding to the magnitude of the rectified smoothing voltage Ei is detected, and the signal SDi The inductance of the control transformer PRT is controlled according to the magnitude of the signal SEi and / or the magnitude of the signal SEi. As a result, a good power factor improvement characteristic can be obtained regardless of the power of the load.

  The signal SDi corresponding to the current flowing through the bridge rectifier circuit Di is detected by detecting the voltage across the resistor R51, and the signal SEi corresponding to the magnitude of the rectified and smoothed voltage Ei is detected using the rectified and smoothed voltage Ei. This is performed by dividing the voltage between the resistor R52 and the resistor R53. The power factor control circuit 3 is a normal voltage-current converter, and controls the magnitude of the current flowing through the winding Nc of the control transformer PRT according to the magnitude of the signal SDi and / or the magnitude of the signal SEi. . Thus, the value of the power factor improving variable inductor Lo formed by the control transformer PRT is controlled according to the magnitude of the signal SDi and / or the magnitude of the signal SEi. That is, according to a change in the current flowing through the bridge rectifier circuit Di or according to a change in the magnitude of the rectified smoothing voltage Ei, and further, a change in the current flowing through the bridge rectifier circuit Di and the magnitude of the rectified smoothing voltage Ei. The leakage inductance value of the power factor improving variable inductor Lo can be adjusted so that the power factor becomes good according to the change in the power factor.

  Although characteristic data of the switching power supply circuit of the fourth embodiment is omitted, the power of the load has a good power factor characteristic from a low region to a high region as in the third embodiment. The power conversion efficiency ηAC → DC is improved as compared with the case of the switching power supply circuit shown as the technology. This is mainly because the configuration including the power factor correction circuit 13 eliminates the need for an active filter. That is, in this embodiment, the total efficiency is not reduced by the product of the two power conversion efficiency values of the former stage and the latter stage as in the case where the active filter is provided.

  In the switching power supply circuit according to the fourth embodiment, the number of circuit components can be reduced by eliminating the need for an active filter. In other words, as can be seen from the description with reference to FIG. 16, the active filter includes a switching element Q103 and a power factor / output voltage control IC 120 for driving the switching element Q103, and the like. On the other hand, in the switching power supply circuit of the fourth embodiment, a filter capacitor CN, a power factor improving diode D1, and a power factor improving variable inductor Lo may be provided as additional components necessary for improving the power factor. Compared with a filter, the number of parts can be very small. As a result, the power supply circuit having the power factor correction function can be manufactured at a much lower cost than the circuit shown in FIG. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight.

  In the switching power supply circuit of the fourth embodiment, the operation of the class E resonant converter and the power factor correction circuit 13 is a so-called soft switching operation. Therefore, the level of switching noise is lower than that of the active filter shown in FIG. It is greatly reduced. In particular, since the current input to the class E switching converter can be linked to a direct current, the level of switching noise can be made extremely small. In addition, the switching power supply circuit according to the fourth embodiment includes a secondary side series resonance circuit together with a primary side series resonance circuit, so that the secondary side DC output voltage Eo is changed to a predetermined voltage by a very slight frequency change. The noise filter can be easily designed. For this reason, if a one-stage noise filter comprising one common mode choke coil CMC and two across capacitors CL is provided, it is possible to sufficiently satisfy the power source disturbance standard. Moreover, sufficient measures can be taken with respect to normal mode noise of the rectified output line by using only one filter capacitor CN.

Further, the switching element Q1, the rectifier diode Do1 and the rectifier diode Do2 on the secondary side, the power factor correction diode D1, the control transformer PRT, and the like operate in synchronization with the switching element Q1. Therefore, as in the power supply circuit of FIG. 16, the ground potential does not interfere with the active filter side and the subsequent switching converter, and can be stabilized regardless of changes in the switching frequency.
In this way, the switching power supply circuit according to the fourth embodiment obtains a power factor improving power supply that solves various problems of a power supply circuit including an active filter.

  (Modification of secondary circuit)

  12 and 13 show modifications of the secondary side circuit that can be replaced in any of the other first to fourth embodiments.

  The secondary side rectifier circuit shown in FIG. 12 constitutes a voltage doubler full wave rectifier circuit. That is, a center tap is applied to the secondary winding N2, and the secondary winding N2A and the secondary winding N2B are divided into two with the center tap as a boundary. The same predetermined number of turns (number of turns) is set in the secondary winding part N2A and the secondary winding part N2B. The center tap of the secondary winding N2 is connected to the secondary side ground. Further, a secondary side series resonance capacitor C3A is connected in series to the end of the secondary winding N2 on the secondary winding portion N2A side, and the end of the secondary winding N2 on the secondary winding portion N2B side is connected. The secondary side series resonant capacitor C3B is connected in series to the unit. As a result, the first secondary side series resonance circuit composed of the leakage inductance component of the secondary winding part N2A and the capacitance of the secondary side series resonance capacitor C3A, the leakage inductance component of the secondary winding part N2B, and the secondary side A second secondary side series resonance circuit composed of the capacitance of the series resonance capacitor C3B is formed.

  The end of the secondary winding N2 side of the secondary winding N2 is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the series connection of the secondary side series resonant capacitor C3A. Connect. Further, the end of the secondary winding N2 on the secondary winding N2B side is connected to the connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4 via the series connection of the secondary side series resonant capacitor C3B. Connect to each other. The cathodes of the rectifier diode Do1 and the rectifier diode Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground. Further, the connection point of each anode of the rectifier diode Do2 and the rectifier diode Do4 is connected to the secondary side ground.

  In this way, the first multiplier including the first secondary series resonance circuit including the secondary winding N2A, the secondary series resonance capacitor C3A, the rectifier diode Do1, the rectifier diode Do2, and the smoothing capacitor Co. A second half-voltage rectifier circuit and a second secondary-side series resonant circuit comprising a secondary winding N2B, a secondary-side series resonant capacitor C3B, a rectifier diode Do1, a rectifier diode Do2, and a smoothing capacitor Co. Thus, a double voltage half-wave rectifier circuit is formed. Thus, for the smoothing capacitor Co, in the half cycle of one polarity of the alternating voltage of the secondary winding N2, the induced voltage of the secondary winding portion N2B and the voltage across the secondary side series resonant capacitor C3B The rectified current is charged by the superimposed potential of the rectified current. In the half cycle of the other polarity, the rectified current is charged by the superimposed potential of the induced voltage of the secondary winding portion N2A and the voltage across the secondary side series resonant capacitor C3A. Will be. Accordingly, a level corresponding to twice the induced voltage level of the secondary winding portion N2A and the secondary winding portion N2B is obtained as the secondary side DC output voltage Eo that is the voltage across the smoothing capacitor Co. Become. That is, a voltage doubler full wave rectifier circuit is obtained.

  The secondary side rectifier circuit shown in FIG. 13 constitutes a bridge full-wave rectifier circuit. That is, four rectifier diodes Do1, rectifier diodes Do2, rectifiers are used as secondary rectifier circuits connected to a series connection circuit (secondary series resonant circuit) of secondary winding N2 and secondary side series resonant capacitor C3. A bridge rectifier circuit including a diode Do3 and a rectifier diode Do4 and a bridge full-wave rectifier circuit including one smoothing capacitor Co are provided. The winding end of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the secondary side series resonant capacitor C3. Further, the winding start end portion of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4. The cathode of the rectifier diode Do1 and the cathode of the rectifier diode Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the connection point between the anode of the rectifier diode Do2 and the anode of the rectifier diode Do4 at the secondary side ground potential.

  Depending on the full-wave rectifier circuit thus formed, in one half cycle of the alternating voltage induced in the secondary winding N2, the pair of the rectifier diode Do1 and the rectifier diode Do4 of the bridge rectifier circuit is conductive, An operation for charging the smoothing capacitor Co with a rectified current is obtained. Further, in the other half cycle of the alternating voltage induced in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining a set of the rectifying diode Do2 and the rectifying diode Do3 is obtained. As a result, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the alternating voltage level induced in the secondary winding N2 is generated as the voltage across the smoothing capacitor Co.

  In addition, the specific design example of the power supply circuit of the embodiment described so far is based on the assumption that the AC input voltage VAC is input with a commercial AC power supply of 100V. The value of VAC is not particularly limited. For example, the same effect can be obtained by adopting the configuration based on the present invention even in the case of a design corresponding to a 200V commercial AC power input. Further, for example, a detailed circuit configuration of the primary side voltage resonance type converter, a configuration of a secondary side rectifier circuit formed including a secondary side series resonance circuit, and the like are also conceivable. As the switching element, it may be considered to select an element other than the MOS-FET such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor. Further, in each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

It is a circuit diagram which shows the structural example of the switching power supply circuit of 1st Embodiment. It is sectional drawing which shows the structural example of the converter transformer with which the switching power supply circuit of embodiment is provided. It is a wave form diagram which shows operation | movement of the principal part in the power supply circuit of 1st Embodiment by a commercial alternating current power supply period. It is a figure which shows the characteristic of the rectification | straightening voltage with respect to load fluctuation | variation, a power factor, and AC-> DC power conversion efficiency about the power supply circuit of 1st Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, power factor, and AC-> DC power conversion efficiency with respect to alternating current input voltage fluctuation | variation about the power supply circuit of 1st Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of 2nd Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of 3rd Embodiment. It is a structural diagram of a power factor improving transformer. It is a figure which shows the characteristic of the rectification | straightening smoothing voltage with respect to load fluctuation | variation, a power factor, and AC-> DC power conversion efficiency about the power supply circuit of 3rd Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, power factor, and AC-> DC power conversion efficiency with respect to alternating current input voltage fluctuation | variation about the power supply circuit of 3rd Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of 4th Embodiment. It is a modification of the secondary side circuit of embodiment. It is a modification of the secondary side circuit of embodiment. It is a block diagram of the active filter shown in background art. It is a wave form diagram explaining operation | movement of the active filter shown to background art. It is a circuit diagram which shows the structural example of the switching power supply circuit shown in background art. It is a wave form diagram explaining operation | movement of the active filter shown to background art. It is a wave form diagram which shows operation | movement of the principal part in the power supply circuit which mounted the active filter shown in background art by a commercial alternating current power supply period. It is the characteristic view shown about each characteristic of the power conversion efficiency with respect to the load fluctuation | variation of the power supply circuit which mounted the active filter shown in background art, a power factor, and a rectification smoothing voltage. It is the characteristic view shown about each characteristic of the power conversion efficiency with respect to the alternating current input voltage fluctuation | variation of a power supply circuit which mounted the active filter shown in background art, a power factor, and a rectification smoothing voltage.

Explanation of symbols

1 control circuit, 2 oscillation / drive circuit, 3 power factor control circuit,
10, 11, 12, 13 Power factor correction circuit AC Commercial AC power supply, C2 Primary side series resonant capacitor, C3, C3A, C3B Secondary side series resonant capacitor, CL across capacitor, CMC common mode choke coil CN Filter capacitor, Ci, Co smoothing capacitor, C1 capacitor CR1, CR2, CR11, CR12 core, D1 power factor correction diode D11, Do1, Do2, Do3, Do4 rectifier diode D1A bypass diode, DD body diode Di bridge rectifier circuit, Ei rectified smooth voltage Eo secondary Side DC output voltage, G gap, I1, I2 current, IAC AC input current L1, L2, L3, L10, Inductor, Lo Power factor improving variable inductor LFT Line filter transformer, N1, N3, N11 Primary windings N2, N4, N 2 Secondary winding, N2A Secondary winding, N2B Secondary winding Nc Winding, PCC choke coil, PF power factor, PIT converter transformer PRT Control transformer, Q101, Q102, Q103 Switching elements R51, R52, R53 Resistance, SDi, SEi Signals T1, T2, T3, T4 terminals, V1, V2, V3 voltage, VAC AC input voltage IAC AC input current, VFT power factor improving transformer, ηAC → DC Power conversion efficiency

Claims (5)

A rectifying / smoothing circuit that is formed by including a rectifying element and a smoothing capacitor that input and rectifies and smoothes AC power from an AC power source, and generates a rectified and smoothed voltage as a voltage across the smoothing capacitor;
A choke coil to which the rectified and smoothed voltage is applied to one terminal;
A converter transformer having a leakage inductor in which one terminal of a primary winding is connected to the other terminal of the choke coil;
The other side of the primary winding is connected to a primary side series resonance capacitor, and the primary side has a primary side resonance frequency governed by the primary side inductance component generated by the leakage inductor and the capacitance of the primary side series resonance capacitor. A series resonant circuit;
A switching element connected to the other terminal of the choke coil and supplying AC power to the primary side series resonance circuit;
An oscillation / drive circuit for driving the switching element;
The secondary winding of the converter transformer is connected to a secondary side series resonance capacitor, and the secondary side resonance frequency governed by the secondary side inductance component generated by the leakage inductor and the capacitance of the secondary side series resonance capacitor A secondary side series resonant circuit having
A control circuit for supplying a control signal to the oscillation / drive circuit such that the value of the secondary side DC output voltage output from the secondary side series resonance circuit by the secondary side rectifier circuit is a predetermined value;
A power factor improving circuit for flowing a current corresponding to a unidirectional component of a current flowing through the primary side series resonant circuit from the AC power supply and flowing a current from the rectifying element to the smoothing capacitor;
The power factor correction circuit includes a power factor correction transformer having a power factor correction diode and a leakage inductor, and a filter capacitor, and a primary winding of the power factor correction transformer is connected in series with the primary side series resonance circuit, One terminal of the secondary winding of the power factor correction transformer is connected to one terminal of the power factor correction diode, and the other terminal of the secondary winding of the power factor correction transformer is the smoothing capacitor and the filter capacitor. A switching power supply circuit, wherein the other terminal of the power factor correction diode is connected to the other terminal of the rectifying element and the filter capacitor.   The switching power supply circuit according to claim 1, wherein the primary side resonance frequency and the secondary side resonance frequency are set to substantially equal frequencies.   The switching power supply circuit according to claim 1, further comprising: a bypass diode that bypasses a rectified current from the rectifying element to the smoothing capacitor, between the rectifying element and the smoothing capacitor.   2. The switching power supply according to claim 1, wherein a value of the leakage inductance of the power factor improving transformer changes in accordance with a magnitude of a current flowing through the rectifying element and / or a magnitude of the rectified smoothing voltage. circuit. A rectifying / smoothing circuit that is formed by including a rectifying element and a smoothing capacitor that input and rectifies and smoothes AC power from an AC power source, and generates a rectified and smoothed voltage as a voltage across the smoothing capacitor;
A choke coil to which the rectified and smoothed voltage is applied to one terminal;
A converter transformer having a leakage inductor in which one terminal of a primary winding is connected to the other terminal of the choke coil;
The other side of the primary winding is connected to a primary side series resonance capacitor, and the primary side has a primary side resonance frequency governed by the primary side inductance component generated by the leakage inductor and the capacitance of the primary side series resonance capacitor. A series resonant circuit;
A switching element connected to the other terminal of the choke coil and supplying AC power to the primary side series resonance circuit;
An oscillation / drive circuit for driving the switching element;
The secondary winding of the converter transformer is connected to a secondary side series resonance capacitor, and the secondary side resonance frequency governed by the secondary side inductance component generated by the leakage inductor and the capacitance of the secondary side series resonance capacitor A secondary side series resonant circuit having
A control circuit for supplying a control signal to the oscillation / drive circuit such that the value of the secondary side DC output voltage output from the secondary side series resonance circuit by the secondary side rectifier circuit is a predetermined value;
A power factor improving circuit for flowing a current corresponding to a unidirectional component of a current flowing through the primary side series resonant circuit from the AC power supply and flowing a current from the rectifying element to the smoothing capacitor;
The power factor correction circuit includes a power factor correction diode, a power factor correction variable inductor having a variable inductance value, and a filter capacitor. One terminal of the power factor correction variable inductor is a primary of the choke coil. And connected to one terminal of a secondary winding of a choke coil magnetically coupled to a winding, and the other terminal of the power factor improving variable inductor is connected to one terminal of the power factor improving diode, The other terminal of the secondary winding of the coil is connected to one terminal of the smoothing capacitor and the filter capacitor, and the other terminal of the power factor correction diode is connected to the other terminal of the rectifier element and the filter capacitor. Being formed,
A switching power supply circuit, wherein an inductance value of the power factor improving variable inductor varies depending on a magnitude of a current flowing through the rectifying element and / or a magnitude of the rectified smoothing voltage.

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