JP2007053841A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a power supply circuit having a power factor improving function, improving the power conversion efficiency, and reducing the number of components for composing the circuit. <P>SOLUTION: The switching power supply has a rectifying/smoothing circuit for generating a rectified/smoothed voltage Ei from AC power, inputs the rectified/smoothed voltage Ei to a converter transformer PIT having a leakage inductor through a choke coil PCC, converts power by using an E-class switching converter having a primary series resonance circuit comprising a primary series resonance capacitor C2 and a leakage inductor L1 and a secondary series resonance circuit comprising a secondary series resonance capacitor C3 and a leakage inductor L2, and returns a voltage from the primary series resonance capacitor C2 to a power factor improving circuit 10. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

  In recent years, most power supply circuits that rectify a commercial power supply to obtain a desired DC voltage are switching power supply circuits. The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

  By the way, the commercial power supply is a sinusoidal AC voltage. However, when the commercial power supply is rectified and smoothed in a smoothing / rectifying circuit using a rectifying element and a smoothing capacitor, the peak holding action of the smoothing / rectifying circuit is required. The current flows from the commercial power supply to the switching power supply circuit for a short time near the peak voltage of the AC voltage, resulting in a distorted waveform greatly different from the sine wave. And the problem that the power factor which shows the utilization efficiency of a power supply is impaired arises. In addition, it is necessary to take measures to suppress the harmonics of the commercial power supply cycle that are generated by such a distorted current waveform. In order to solve these problems, a technique using a so-called active filter is conventionally known as a technique for improving the power factor (see, for example, Patent Document 1).

  FIG. 13 shows a basic configuration of such an active filter. In FIG. 13, a bridge rectifier circuit Di is connected to a commercial AC power supply AC. A step-up type converter is connected to the positive / negative line of the bridge rectifier circuit Di, and a smoothing capacitor Cout is connected in parallel to the output thereof, and a DC voltage Vout is obtained as a voltage across the both ends. This DC voltage Vout is supplied as an input voltage to a load 110 such as a DC-DC converter in the subsequent stage.

  As a configuration for improving the power factor, a step-up type including a inductor 111, a fast recovery type fast switching diode D, a step-up type converter including a switching element Q, and a multiplier 111 as main components. A control unit of the converter. The inductor L and the high-speed switching diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Cout. The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the smoothing capacitor Cout. The switching element Q is, for example, a MOS-FET, and is inserted between the connection point of the inductor L and the high-speed switching diode D and the primary side ground.

  To the multiplier 111, a current detection line LI and a waveform input line LW are connected, and a voltage detection line LV is further connected. Then, the multiplier 111 detects a signal corresponding to the rectified current Iin input from the current detection line LI and flowing in the negative output terminal of the bridge rectifier circuit Di, from both ends of the resistor Ri. Further, a signal corresponding to the rectified voltage Vin of the positive output terminal of the bridge rectifier circuit Di input from the waveform input line LW is detected. The rectified voltage Vin is an absolute value of the waveform of the AC input voltage VAC from the commercial AC power supply AC. Further, based on the DC voltage Vout of the smoothing capacitor Cout input from the voltage detection line LV, the fluctuation difference of the DC input voltage (a signal obtained by amplifying the difference between the predetermined reference voltage and the DC voltage Vout is referred to as a fluctuation difference). The same is used in the following). The multiplier 111 outputs a drive signal for driving the switching element Q.

  In the multiplier 111 (control unit of the step-up type converter) and the step-up type converter, a signal corresponding to the rectified current Iin detected from the current detection line LI and a DC input voltage detected from the voltage detection line LV The fluctuation difference is multiplied, and an error between the multiplication result and a signal corresponding to the rectified voltage Vin detected from the waveform input line LW is detected. After the error signal is amplified, PWM (Pulse Width Modulation) conversion is performed, and the switching element Q is controlled by a binary signal of a high level and a low level. In this way, a two-input feedback system is configured, the value of the DC voltage Vout is set to a predetermined value, and the rectified current Iin has a similar waveform with respect to the rectified voltage Vin. As a result, the AC voltage applied to the bridge rectifier circuit Di from the commercial AC power supply AC and the waveform of the AC current flowing into the bridge rectifier circuit Di are similar, and the power factor is improved so that the power factor approaches one. Will be planned.

  FIG. 14A shows the rectified voltage Vin and the rectified current Iin when the active filter circuit shown in FIG. 13 operates appropriately. FIG. 14B shows an energy (power) change Pchg input / output to / from the smoothing capacitor Cout. A line indicated by a broken line indicates an energy (power) average value Pin to be input / output. That is, the smoothing capacitor Cout stores energy when the rectified voltage Vin is high, and releases energy when the rectified voltage Vin is low, thereby maintaining the flow of output power. FIG. 14C shows the waveform of the charge / discharge current Ichg with respect to the smoothing capacitor Cout. FIG. 14D shows a DC voltage Vout that is a voltage across the smoothing capacitor Cout. In the DC voltage Vout, a ripple voltage mainly including the second harmonic component of the cycle of the rectified voltage Vin is superimposed on the DC voltage (for example, a DC voltage of 375 V).

  FIG. 15 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG. The power supply circuit shown in this figure employs a configuration that can cope with a load power Po in the range of 300 W to 0 W when the value of the AC input voltage VAC is in the range of 85 V to 264 V. Further, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

  The power supply circuit shown in FIG. 15 will be described in order from the AC input side. A common mode noise filter is provided by two line filter transformers LFT and three across capacitors CL, and a bridge rectifier circuit Di is connected to the subsequent stage. In addition, a normal mode noise filter 125 having a pi-type configuration including an inductor LN and a filter capacitor (film capacitor) CN is connected to the rectified output line of the bridge rectifier circuit Di.

  The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci via the inductor LN, the choke coil PCC (functioning as the inductor Lpc), and a series connection of a fast recovery type fast switching diode D20. The The smoothing capacitor Ci has the same function as the smoothing capacitor Cout in FIG. Further, the inductor Lpc and the high speed switching diode D20 of the choke coil PCC have the same functions as the inductor L and the high speed switching diode D shown in FIG. Further, an RC snubber circuit comprising a series connection of a capacitor Csn and a resistor Rsn is connected in parallel to the high-speed switching diode D20 in this figure.

  Switching element Q103 corresponds to switching element Q in FIG. In this case, the power factor / output voltage control IC 120 is an integrated circuit (IC) that controls the operation of the active filter for improving the power factor so that the power factor approaches 1. The multiplier, the divider, An error voltage amplifier, a PWM control circuit, and a drive circuit that outputs a drive signal for driving the switching element Q103 are configured. The voltage obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistor R5 and the voltage dividing resistor R6 is input to the terminal T1 of the power factor / output voltage control IC 120 so as to be rectified and smoothed. A first feedback control circuit having the voltage Ei as a predetermined value is formed.

  Further, a series connection of a voltage dividing resistor R101 and a voltage dividing resistor R102 is provided between the positive electrode output terminal of the bridge rectifier circuit Di and the primary side ground, and a connection point between the voltage dividing resistor R101 and the voltage dividing resistor R102 is a terminal. It connects to T5. Thereby, the rectified voltage of the bridge rectifier circuit Di is divided and inputted to the terminal T5. Further, the voltage of the resistor 103, that is, the voltage corresponding to the source current of the switching element Q103 is input to the terminal T2. Here, the source current of the switching element Q103 is a current contributing to storing magnetic energy in the current I1 flowing through the choke coil PCC. A signal corresponding to the rectified voltage input to the terminal T5 of the power factor / output voltage control IC 120 and a signal corresponding to the voltage envelope input to the terminal T2 (that is, the envelope of the current I1) are similar. A second feedback control circuit is formed.

  Further, the operating power of the power factor / output voltage control IC 120 is supplied to the terminal T4. At this terminal T4, the alternating voltage excited in the coil N5 in the choke coil PCC and transformer coupled to the inductor Lpc is converted into a low-voltage DC voltage by a half-wave rectifier circuit comprising a rectifier diode D11 and a capacitor C11 shown in the figure. Supplied. Further, the terminal T4 is connected to the positive output terminal of the bridge rectifier circuit Di via the starting resistor Rs. After the commercial AC power supply AC is turned on, the rectified output obtained at the positive output terminal of the bridge rectifier circuit Di is supplied to the terminal T4 via the starting resistor Rs during the rise time until the voltage is excited in the winding N5. Supplied. The power factor / output voltage control IC 120 starts operation using the rectified voltage supplied in this manner as a starting power source.

  A drive signal (gate voltage) for driving the switching element is output from the terminal T3 to the gate of the switching element Q103. That is, the envelope of the current I1 is made similar to the first feedback control circuit in which the voltage value divided by the voltage dividing resistor R5 and the voltage dividing resistor R6 is a predetermined value, and the rectified and smoothed voltage Ei. A drive signal for operating the two feedback control circuits with the second feedback control circuit is output to the gate of the switching element Q103. As a result, the waveform of the AC input current IAC flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC, and the power factor is controlled to be approximately 1. That is, power factor improvement is achieved.

  Here, regarding the power factor improvement operation of the active filter shown in FIG. 15, waveforms of respective parts are shown in FIG. 16 and FIG. 17. First, FIG. 16 shows the switching operation (ON: conduction and OFF: disconnection operation) of the switching element Q103 according to the load variation, and the current I1 flowing through the inductor Lpc of the choke coil PCC. 16A shows the operation at a light load, FIG. 16B shows the operation at an intermediate load, and FIG. 16C shows the operation at a heavy load. As can be seen by comparing FIGS. 16 (a), 16 (b), and 16 (c), the switching element Q103 is turned on as the switching period becomes constant and the load becomes heavy. The period gets longer. In this way, by adjusting the current I1 flowing into the smoothing capacitor Ci through the inductor Lpc according to the load condition, the rectified smoothing voltage Ei is stabilized against the voltage fluctuation of the AC input voltage VAC and the load fluctuation. It is done. For example, the value of the rectified and smoothed voltage Ei is made constant at 380 V with respect to the range of the AC input voltage VAC from 85 V to 264 V. The rectified and smoothed voltage Ei is a voltage across the smoothing capacitor Ci, and is a DC input voltage for the subsequent current resonance type converter.

  FIG. 17 shows the waveforms of the AC input current IAC and the rectified and smoothed voltage Ei by comparison with the AC input voltage VAC. In this figure, experimental results when the value of the AC input voltage VAC is 100 V are shown. As shown in this figure, the waveform of the AC input voltage VAC and the waveform of the AC input current IAC are substantially similar to the passage of time. That is, the power factor is improved. Further, it is shown that the rectified and smoothed voltage Ei is stabilized at an average value of 380 V along with the improvement of the power factor. Further, as shown in the figure, it has a ripple fluctuation of 10 Vp-p with respect to 380 V.

  Returning to FIG. 15 again, the current resonance converter at the latter stage of the active filter will be described. The current resonance type converter performs a switching operation for power conversion by inputting the rectified and smoothed voltage Ei, and forms a current resonance type converter including a switching circuit connected in a half bridge by the switching elements Q101 and Q102. The current resonance type converter in this case is a separately excited type, and MOS-FETs are used for the switching elements Q101 and Q102. A body diode DD101 and a body diode DD102 are connected in parallel to these MOS-FETs. The switching element Q101 and the switching element Q102 are switched and driven at a required switching frequency by the oscillation / drive circuit 102 at the timing when they are alternately turned on / off. The oscillation / drive circuit 2 is controlled by a signal from the control circuit 1, and the control circuit 1 operates so as to variably control the switching frequency according to the level of the secondary side DC output voltage Eo. The secondary side DC output voltage Eo is stabilized.

  Converter transformer PIT is provided to transmit the switching outputs of switching element Q101 and switching element Q102 from the primary side to the secondary side. One end of the primary winding N1 of the converter transformer PIT is connected to a connection point (switching output point) of the switching element Q101 and the switching element Q102 via the primary side series resonance capacitor C2, and the other end of the primary winding N1 is connected. The end is grounded. Here, a series resonance circuit is formed by the primary side series resonance capacitor C2 and the primary side leakage inductor L1. This series resonance circuit generates a resonance operation when a switching output is supplied by the switching element Q101 and the switching element Q102.

  A secondary winding N2 is wound around the secondary side of the converter transformer PIT. The secondary winding N2 in this case has a secondary winding portion N2A and a secondary winding portion N2B that are center-tapped as shown in the figure, and the center tap is connected to the secondary side ground. Thus, the secondary winding portion N2A and the secondary winding portion N2B are connected to the anodes of the rectifier diode Do1 and the rectifier diode Do2, respectively, and the cathodes of the rectifier diode Do1 and the rectifier diode Do2 are connected to the smoothing capacitor Co. By doing so, a double-wave rectifier circuit is formed. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is input to the control circuit 1 described above.

  FIG. 18 shows characteristics of AC power to DC power conversion efficiency ηAC → DC (total efficiency), power factor PF, and rectified smoothing voltage Ei with respect to load fluctuation. In this figure, the characteristic with respect to the fluctuation | variation with the value of the load electric power Po when the value of AC input voltage VAC is 100V from 300W to 0W is shown. FIG. 15 shows characteristics of power conversion efficiency ηAC → DC (total efficiency), power factor PF, and rectified smoothing voltage Ei with respect to fluctuations in the AC input voltage VAC. This figure shows characteristics with respect to fluctuations in the value of the AC input voltage VAC from 85V to 264V under a constant load condition with a load power Po value of 300W.

  First, as shown in FIG. 18, the power conversion efficiency (total efficiency) decreases as the load power Po becomes a heavy load condition. Further, with respect to fluctuations in the AC input voltage VAC, under the same load condition, as shown in FIG. 19, the level of the AC input voltage VAC tends to increase as the level increases. ing. For example, when the load power Po is 300 W and the AC input voltage VAC is 100 V, the power conversion efficiency (total efficiency) is about 83.0%, and when the AC input voltage VAC is 230 V, the power conversion efficiency (total efficiency). Is about 89.0%, and when the AC input voltage VAC is 85V, the power conversion efficiency (total efficiency) is about 80.0%.

  Further, as shown in FIG. 18, the power factor PF has a characteristic that is substantially constant with respect to the fluctuation of the load power Po. Further, as shown in FIG. 19, the fluctuation characteristic of the power factor PF with respect to the fluctuation of the AC input voltage VAC is a characteristic that can be regarded as almost constant although it tends to decrease as the AC input voltage VAC increases. I understand that. For example, when the load power Po is 300 W and the AC input voltage VAC is 100 V, the power factor PF is about 0.96, and when the AC input voltage VAC is 230 V, the power factor PF is about 0.94. can get.

Further, as shown in FIGS. 18 and 19, the rectified and smoothed voltage Ei has a result that is almost constant with respect to fluctuations in the load power Po and the AC input voltage VAC.
JP-A-6-327246

  As can be seen from the above description, the power supply circuit shown in FIG. 15 is configured by mounting the conventionally known active filter shown in FIG. 13, and by adopting such a configuration, the power factor is improved. I am trying.

  However, the power supply circuit having the configuration shown in FIG. 15 has the following problems. First, as the power conversion efficiency in the power supply circuit shown in FIG. 15, the conversion efficiency from the AC power corresponding to the active filter in the previous stage to the DC power, and the conversion efficiency from the DC power to the DC power in the current resonance type converter in the subsequent stage, Will be a synthesis of That is, the total power conversion efficiency (total efficiency) of the circuit shown in FIG. 15 is a value obtained by multiplying the values of these power conversion efficiencies, and is a product of numbers that are each 1 or less. Overall efficiency will decrease.

  In addition, since the active filter circuit is a hard switching operation, noise generation is large, so that strict noise suppression measures are required. For this reason, in the circuit shown in FIG. 15, a noise filter including two line filter transformers and three across capacitors is formed for the line of the commercial AC power supply AC. In addition, a normal mode noise filter including one inductor LN and two filter capacitors CN is provided for the rectified output line. Furthermore, an RC snubber circuit is provided for the fast recovery type fast switching diode D20 for rectification. In this way, it is necessary to take measures against noise due to the large number of parts, resulting in an increase in cost and an increase in the mounting area of the power supply circuit board.

  Further, the switching frequency of the switching element Q103 operated by the power factor / output voltage control IC 120 as a general-purpose IC is fixed at 60 kHz, whereas the switching frequency of the subsequent current resonance type converter is in the range of 80 kHz to 200 kHz. Variable. Since the switching timings (clocks) of the two are thus independent of each other, the ground potential interferes and becomes unstable due to the switching operation of the two based on the respective clocks. For example, abnormal oscillation is likely to occur. . As a result, problems such as difficulty in circuit design and deterioration of reliability are also caused.

  The switching power supply circuit according to the present invention includes a rectifying element and a smoothing capacitor that input and rectifies and smoothes AC power from an AC power source, and generates a rectified and smoothed voltage as a voltage across the smoothing capacitor. A converter transformer having a circuit, a choke coil to which the rectified and smoothed voltage is applied to one terminal, a leakage inductor in which one terminal of a primary winding is connected to the other terminal of the choke coil, The one terminal of the primary winding is connected to a primary side series resonance capacitor, and a primary side having a primary side series resonance frequency governed by a primary side inductance component generated by the leakage inductor and a capacitance of the primary side series resonance capacitor. A side series resonant circuit, the primary winding and the primary side parallel resonant capacitor are connected in parallel; A primary side parallel resonant circuit having a primary side parallel resonant frequency governed by a primary side inductance component generated by the inductor and the capacitance of the primary side parallel resonant capacitor, the primary side series resonant circuit, and the primary side parallel resonant circuit A switching element connected to the other terminal of the primary winding to supply AC power to the oscillation circuit, an oscillation / drive circuit for driving the switching element, and a secondary winding of the converter transformer in a secondary side series resonance. A secondary side series resonant circuit connected to a capacitor and having a secondary side resonance frequency controlled by a secondary side inductance component generated by the leakage inductor and a capacitance of the secondary side series resonant capacitor; and the secondary side The secondary DC output voltage output from the series resonant circuit by the secondary rectifier circuit is controlled to a predetermined value. A power circuit in which one terminal is connected to the other terminal of the primary side series resonant capacitor and the choke coil, and the other terminal is connected to the rectifying element; a control circuit that supplies a signal to the oscillation / drive circuit; A power factor improvement circuit comprising: an improvement diode; and a filter capacitor connected to each of the other terminal of the power factor improvement diode and the rectifying and smoothing capacitor.

  That is, in this switching power supply circuit, the rectifying and smoothing circuit receives the AC power from the AC power supply and generates a rectified and smoothed voltage. Then, the rectified and smoothed voltage is input to the converter transformer via the choke coil. A primary side series resonant circuit with a converter transformer leakage inductor and a primary side series resonant capacitor, a primary side parallel resonant circuit with a converter transformer leakage inductor and a primary side parallel resonant capacitor, a converter transformer leakage inductor and a secondary side series resonant capacitor To form a primary side series resonant circuit. The switching element controlled by the oscillation / drive circuit drives the primary side series resonance circuit and the primary side parallel resonance circuit, and the control circuit outputs a control signal for setting the value of the secondary side DC output voltage to a predetermined value. Supply to the oscillation / drive circuit. Then, the voltage from the connection point between the primary side series resonance capacitor and the choke coil is voltage-feedbacked to the power factor correction circuit to improve the power factor.

  According to the switching power supply circuit of the present invention, it is possible to omit the active filter and to have a power factor improving function. By omitting the active filter, the power conversion efficiency characteristic of the switching power supply circuit is improved. Then, the heat sink and the like can be omitted and reduced. In addition, the number of parts is greatly reduced as compared with a configuration including an active filter, so that the circuit can be reduced in size and weight and the cost can be reduced. The active filter has a hard switching operation, whereas the switching converter of the present invention has a soft switching operation because it is based on a resonant converter. As a result, the switching noise is greatly reduced, which contributes to the reduction in size and weight and cost of the noise filter. Further, since there are no plural clocks having different frequencies, the problem of mutual interference due to plural clock frequencies does not occur, the reliability is improved, and the pattern design of the circuit board is facilitated.

  (First embodiment)

  The switching power supply circuit according to the first embodiment will be described with reference to FIG. The switching power supply circuit according to the first embodiment applies a so-called class E switching converter to the power supply circuit, and improves the power factor by voltage feedback of a resonance voltage resulting from superposition of current resonance and voltage resonance to a smoothing capacitor. It is.

  The switching power supply circuit of the first embodiment shown in FIG. 1 will be described below in order from the commercial AC power supply AC side. A two-phase input line of the commercial AC power supply AC is connected to a bridge rectifier circuit Di, which is a kind of rectifier element, through a common mode noise filter including a common mode choke coil CMC and two across capacitors CL. Here, the common mode noise filter has a function of removing common mode noise generated between the line of the commercial AC power supply AC and the secondary side of the switching power supply circuit.

  AC power is rectified by a bridge rectifier circuit Di formed by bridge-connecting four low-speed rectifier elements (diodes) to generate pulsating power, and the pulsating power is improved in power factor with a fast switching speed. The smoothing capacitor Ci is charged through the diode D1 and the choke coil PCC. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. That is, the bridge rectification circuit Di and the smoothing capacitor Ci constitute a rectification smoothing circuit. Here, the power factor correction diode D1 constitutes a part of a power factor correction circuit 10 to be described later, and the process in which the current from the AC power source is charged to the smoothing capacitor Ci via the power factor correction diode D1 will be described. The operation of the improvement circuit 10 will be described later. Here, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent class E switching converter.

  The class E switching converter is formed mainly of a choke coil PCC, a converter transformer PIT, a primary side series resonance capacitor C2, a primary side parallel resonance capacitor C1, and a switching element Q1. That is, the class E switching converter is configured as follows. The rectified and smoothed voltage Ei is applied to one terminal of the choke coil PCC. Then, one terminal of the primary winding N1 of the converter transformer PIT having a leakage inductor is connected to the other terminal of the choke coil PCC. Further, this one terminal of the primary winding N1 is connected to the primary side series resonant capacitor C2, and the primary side inductance component generated by the leakage inductor (inductance component corresponding to the equivalent inductor represented by the inductor L1 in FIG. 1) and the primary side A primary side series resonance circuit having a primary side series resonance frequency that is dominated by the capacitance of the side series resonance capacitor C2 is formed. Further, the primary winding and the primary side parallel resonant capacitor C1 are connected in parallel. And switching element Q1 which supplies alternating current power to a primary side series resonance circuit and a primary side parallel resonance circuit is connected to the other terminal of a primary winding. Here, the oscillation / drive circuit 2 drives the switching element Q1, the switching element Q1 is a MOS-FET, and the converter transformer PIT has a structure with a relatively large leakage inductance value by reducing the value of the coupling coefficient k. It is said that. The structure of the converter transformer PIT will be described later.

  The secondary winding N2 of the converter transformer is connected to the secondary side series resonant capacitor C3, and the secondary side inductance component generated by the leakage inductor (inductance component corresponding to the equivalent inductor represented by the inductor L2 in FIG. 1) and A secondary side series resonance circuit having a secondary side resonance frequency controlled by the capacitance of the secondary side series resonance capacitor is formed. Then, the secondary side DC output voltage Eo output from the secondary side series resonance circuit by the secondary side rectifier circuit (formed by the rectifier diode Do1, the rectifier diode Do2 and the smoothing capacitor Co) is set to a predetermined value. A control circuit 1 for supplying such a control signal to the oscillation / drive circuit 2 is provided.

  In addition, a power factor improving circuit 10 is provided to improve the power factor by flowing the current from the bridge rectifier circuit Di to the smoothing capacitor Ci and feeding back the voltage generated at both ends of the primary side series resonance capacitor C2 to the smoothing capacitor Ci. Yes. The power factor correction circuit 10 is formed of a power factor correction diode D1 and a filter capacitor CN. The filter capacitor CN also functions as a filter for removing normal mode noise.

  That is, as described above, in the first embodiment, power transmission is performed by a class E switching converter using a voltage / current resonant converter, and the voltage of the primary side series resonant capacitor C2 constituting the voltage / current resonant converter is smoothed. The primary side series resonance capacitor C2 is connected to the power factor correction circuit 10 so that the voltage is fed back to the capacitor Ci, and the secondary side series resonance circuit is configured on the secondary side, and the secondary side In order to set the value of the side DC output voltage to a predetermined value, a control circuit 1 is provided. A feature of the first embodiment is that the gate of Q1 is driven.

  Hereinafter, a more detailed configuration of the switching power supply circuit according to the first embodiment shown in FIG. 1 will be described.

  The converter transformer PIT has functions of insulating the primary side and the secondary side and performing voltage conversion, and also functions as an inductor L1 that constitutes a part of a resonance circuit for causing the class E switching converter to function. To do. Here, the inductor L1 is an inductance component formed by the leakage inductor of the converter transformer PIT. A specific structure of the converter transformer PIT in the first embodiment will be described along a cross-sectional view of the converter transformer PIT shown in FIG.

  The converter transformer PIT includes an EE type core (EE-shaped core) in which an E type core CR1 and an E type core CR2 made of a ferrite material are combined so that their magnetic legs face each other. The primary side and secondary side winding portions are divided so as to be independent from each other, and provided with a bobbin B formed of, for example, resin. The primary winding N1 is wound around one winding region of the bobbin B. Further, the secondary winding N2 is wound around the other winding region. By attaching the bobbin B on which the primary winding N1 and the secondary winding N2 are wound to the EE-shaped core in this way, the primary winding N1 and the secondary winding N2 have different EE-shaped shapes, respectively. It will be in the state wound by the central magnetic leg of a core. In this way, the overall structure of the converter transformer PIT is obtained.

  A gap G of 2.2 mm is formed with respect to the central magnetic leg of the EE-shaped core. As a result, 0.67 is obtained as the value of the coupling coefficient k between the primary side and the secondary side. In this way, the primary side and the secondary side are preferably loosely coupled with a coupling coefficient of 0.7 or less so as to obtain a leakage inductor L1 having a large inductance value. The gap G is formed by making the central magnetic legs of the E-type core CR1 and the E-type core CR2 shorter than the two outer magnetic legs. The number of turns of the primary winding N1 was 36T (turn), the number of turns of the primary winding N2 was 30T (turn), and the core material was EER-35 (core material name).

  As described above, a MOS-FET is selected as the switching element Q1, and a body diode DD is incorporated in parallel between the source and the drain. The inductance value of the inductor Lo (choke coil PCC) is 1 mH (Milli Henry). The value of the primary side series resonant capacitor C2 was 0.033 μF (micro farad), and the value of the primary side parallel resonant capacitor C1 was 7500 pF (pico farad).

  On the secondary side of the converter transformer PIT, a voltage waveform similar to the alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. A secondary side series resonant capacitor C3 is connected in series to the secondary winding N2. Thus, a secondary side series resonance circuit is formed by the leakage inductor L2 and the secondary side series resonance capacitor C3 viewed from the secondary winding N2 side. The resonance frequency of the secondary side series resonance circuit is set to be substantially equal to the frequency of the primary side series resonance frequency determined by the primary side series resonance capacitor C2 and the leakage inductor L1.

  The secondary side rectifier circuit connects the rectifier diode Do1 and the rectifier diode Do2 and the smoothing capacitor Co, which operate at high speed, to the secondary winding N2 to which the secondary side series resonant capacitor C3 is connected in series. , Formed as a voltage doubler half-wave rectifier circuit. In the voltage doubler half-wave rectifier circuit, the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 are connected to one end of the secondary winding N2 via the secondary side series resonant capacitor C3. Further, the cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The other end of the secondary winding N2, the anode of the rectifier diode Do2, and the negative terminal of the smoothing capacitor Co are connected to the secondary side ground. Here, the value of the secondary side series resonance capacitor C3 was set to 0.082 μF.

  The control circuit 1 supplies a detection output corresponding to the difference between the input secondary side DC output voltage Eo and a predetermined reference voltage value to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching element Q <b> 1 is driven mainly by changing the switching frequency according to the input detection output of the control circuit 1. Further, the time ratio, which is the ratio of the time during which the switching element Q1 is turned on in one cycle, may be changed together with the switching frequency.

  In this way, the switching frequency of the switching element Q1 is variably controlled, whereby the primary and secondary resonance impedances in the power supply circuit change, and the converter transformer PIT changes from the primary winding N1 to the secondary winding N2 side. The amount of power to be transmitted and the amount of power to be supplied from the secondary side rectifier circuit to the load will change. As a result, an operation for matching the magnitude of the secondary side DC output voltage Eo with the reference voltage is obtained. That is, the secondary side DC output voltage Eo is stabilized.

  Next, the power factor correction circuit 10 will be described. As described above, the power factor correction circuit 10 is provided so as to be inserted into the rectification current path in the rectification / smoothing circuit for obtaining the rectification / smoothing voltage Ei from the commercial AC power supply AC. The power factor correction circuit 10 of the first embodiment includes a power factor correction diode D1 and a filter capacitor CN. The primary side series resonant capacitor C2 and the other terminal of the choke coil are connected to one terminal of the power factor correction diode D1, and the bridge rectifier circuit Di is connected to the other terminal of the power factor correction diode D1. The terminals of the filter capacitor CN are connected to the bridge rectifier circuit Di and the smoothing capacitor Ci, respectively. Here, the value of the filter capacitor CN was set to 1 μF (micro farad).

  Next, the operation of each part of the switching power supply circuit of the first embodiment will be described in order. First, the operation of the class E switching converter, which is one main part of the first embodiment, will be described. For the sake of simplicity, the operation of the class E switching converter will be described assuming that there is no power factor correction circuit 10 at first. The class E switching converter is equivalent to a primary side conversion in which the primary side series resonance capacitor C2, the primary side parallel resonance capacitor C1, the leakage inductor L1, and the load impedance connected to the secondary side are converted to the primary side. A rectified and smoothed voltage Ei is supplied via an inductor Lo to a voltage / current resonance converter including a load impedance and a switching element Q1 (including the body diode DD, unless otherwise specified). Can be considered a thing.

  In the class E switching converter having such a configuration, the primary side primary resonance capacitor C2, the leakage inductor L1, the primary side equivalent load impedance (the secondary side load is the primary winding) by turning on and off the switching element Q1. Current resonance occurs due to electromagnetic coupling between the primary winding and the secondary winding and equivalent impedance inserted as a load in series with the primary side series resonance circuit, and the primary side parallel column resonant capacitor C1, the leakage inductor L1, and the primary side conversion A voltage resonance occurs due to the load impedance, and as a result, a current due to the current resonance and a current due to the voltage resonance flow through the primary side series resonance capacitor C2, but the Q value of the resonance circuit is high. The current flowing through the series resonant capacitor C2 is substantially a sine wave current.

  On the other hand, since the inductance value of the inductor Lo (choke coil PCC) has a very large impedance with respect to the resonance frequency of the resonance circuit, the current flowing through the inductor Lo is close to DC. As a result, the voltage generated across the inductor Lo is also a sine wave. Here, in terms of alternating current, the primary side series resonant capacitor C2 and the inductor Lo are connected in parallel. Further, the voltage generated in the primary winding N1 and the secondary winding N2 of the converter transformer PIT is also substantially a sine wave.

  Next, the operation on the secondary side will be described. In a half cycle corresponding to one polarity of the alternating voltage obtained in the secondary winding N2, a forward voltage is applied to the rectifier diode Do2, so that the rectifier diode Do2 conducts and the rectified current is converted into the secondary voltage. An operation of charging the side series resonant capacitor C3 is obtained. As a result, a voltage across the secondary side series resonant capacitor C3 is generated at a level corresponding to an equal multiple of the alternating voltage level induced in the secondary winding N2. In a half cycle corresponding to the other polarity of the alternating voltage obtained in the secondary winding, a forward voltage is applied to the rectifier diode Do2 to conduct. At this time, the smoothing capacitor Co is charged with a potential in which the voltage generated in the secondary winding N2 and the voltage across the secondary side series resonance capacitor C3 are superimposed. As a result, the secondary side DC output voltage Eo having a level corresponding to twice the alternating voltage level induced in the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co. In this rectification operation, the smoothing capacitor Co is charged only in one half cycle of the alternating voltage excited by the secondary winding N2. That is, a rectification operation as a double voltage half wave is obtained. The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  Here, since the resonance frequency of the resonance circuit of the primary side and the resonance frequency of the resonance circuit of the secondary side are set to be approximately equal, the power transmission characteristic with respect to the frequency from the primary side to the secondary side is slight. It becomes very sensitive to frequency fluctuations. That is, in the first embodiment, the Q value of the primary side resonance circuit is increased by using the class E switching converter, and a series resonance circuit having a resonance frequency substantially the same as that of the primary side is arranged on the secondary side. By doing so, the amount of electric power to be transmitted can be greatly changed by a slight difference in frequency. This means that when the secondary side DC output voltage Eo is kept at a predetermined value, the control circuit 1 sends a signal for changing the frequency very slightly regardless of a wide range of load fluctuations. This means that a stable constant voltage characteristic can be obtained if the switching element Q1 is supplied to the switching element Q1. In this way, the switching frequency of the switching element Q1 is variably controlled, so that the primary and secondary resonance impedances in the power supply circuit change and are transmitted from the primary winding N1 to the secondary winding N2 of the converter transformer PIT. The amount of power to be changed will change. As a result, the value of the secondary side DC output voltage Eo is maintained at a predetermined value. In this embodiment, the resonance frequency of the resonance circuit of the primary side and the resonance frequency of the resonance circuit of the secondary side are set to be approximately equal, but the relationship between the frequencies can be determined as appropriate.

  Next, the operation of the power factor correction circuit 10, which is another main part of the first embodiment, will be described. In the above description of the operation of the class E switching converter, it has been described that the power factor correction circuit 10 does not exist. However, in practice, the function of feeding back the voltage to the power factor correction circuit 10 is also the E of the first embodiment. Class switching converter has. That is, the primary side series resonance capacitor C2 and the other terminal of the inductor Lo are connected, and in terms of AC, the potential at both ends of the primary side series resonance capacitor C2 and the voltage at both ends of the winding of the inductor Lo are the same potential. Become. This resonance voltage is fed back between the cathode of the power factor correction diode D1 and the smoothing capacitor Ci. When the voltage V2 becomes lower than the voltage V3, a current I1 flows, and the low frequency component of the current I1 is AC input current IAC. As a result, even when the value of the rectified smoothing voltage Ei, which is the voltage across the smoothing capacitor Ci, is larger than the voltage V3, the current I2 acts so as to improve the power factor.

  That is, without the power factor correction circuit 10, when the voltage of the smoothing capacitor Ci is larger than the magnitude of the pulsating voltage obtained from the bridge rectifier circuit Di, the current from the bridge rectifier circuit Di is interrupted and flows only for a short time. However, by providing the power factor correction circuit 10, a current based on the voltage across the winding of the inductor Lo flows from the bridge rectifier circuit Di to the smoothing capacitor Ci, and the conduction period of the pulsating current is increased. become longer. Then, the conduction period of the AC input current IAC is also substantially coincident with the conduction period of the rectified current, and the power factor is improved.

  (Operation waveform and measurement data of main part of the first embodiment)

  FIG. 3 shows an operation waveform of a main part of the switching power supply circuit according to the first embodiment shown in FIG. 1, and FIGS. 4 and 5 show measurement data.

  FIG. 3 shows the operation waveform of the main part of the power factor correction circuit 10 by the commercial AC power supply cycle. From the upper stage to the lower stage of FIG. 3, an AC input voltage VAC (see FIG. 1), an AC input current IAC (see FIG. 1), a voltage V2 (see FIG. 1), a current I1 (see FIG. 1), Each of current I2 (refer to FIG. 1), voltage V3 (refer to FIG. 1), and ΔEo that is a ripple component of secondary side DC output voltage Eo is shown. In the waveform diagram of voltage V2 and current I1 in FIG. 3, the shaded portion indicates that switching is performed in the same cycle as the switching waveform of switching element Q1. Here, the current I1 flowing through the power factor correction diode D1 is based on the resonance voltage as described above, and FIG. 3 shows that the distribution period of the AC input current IAC is expanded by this voltage. The secondary side DC output voltage Eo is superimposed with a ripple voltage of 100 mVp-p (peak-to-peak value) having a half period of the AC input voltage VAC, with a predetermined value of 175 V (volt) as an average value.

  FIG. 4 shows the rectified smoothing voltage Ei and force against load fluctuation when the value of the load power Po is in the range of 0 W (no load) to 300 W (maximum load power) under the input voltage condition where the value of the AC input voltage VAC is 100V. The power conversion efficiency ηAC → DC of the DC output power with respect to the rate PF and the AC input power is shown. FIG. 5 shows the rectified smoothing voltage Ei, power factor PF, and power in a range in which the value of the AC input voltage VAC is changed from 85 V to 144 V under a load condition where the load power is constant at 300 W (maximum load power). The conversion efficiency ηAC → DC is shown.

  As shown in FIG. 4, the rectified and smoothed voltage Ei has a tendency to gradually increase as the load becomes lighter, but the change is as small as 10V. On the other hand, it changes in proportion to the change of the AC input voltage VAC. Further, as shown in FIG. 4, the power factor PF is maintained at 0.75 or more in the range of 50 W to 300 W, and it can be said that a practically sufficient power factor value is obtained. As shown in FIG. 5, the characteristics of the AC input voltage VAC are almost constant as shown in FIG.

  Further, as shown in FIGS. 4 and 5, the power conversion efficiency ηAC → DC is a characteristic that can be considered to be almost constant with respect to the load fluctuation, and is 90% or more in the range from the load power of 25 W to the load power of 300 W. The power conversion efficiency ηAC → DC is obtained. With respect to the fluctuation of the AC input voltage VAC, a substantially constant 90% or more is maintained in the range of 85V to 144V.

  In the switching power supply circuit of the first embodiment shown in FIG. 1, the power conversion efficiency ηAC → DC is improved as compared with the case of the switching power supply circuit shown as the background art in FIG. This is mainly because the configuration including the power factor correction circuit 10 eliminates the need for an active filter. That is, in this embodiment, the total efficiency is not reduced by the product of the two power conversion efficiency values of the former stage and the latter stage as in the case where the active filter is provided.

  In the switching power supply circuit according to the first embodiment, the number of circuit components can be reduced by eliminating the need for an active filter. In other words, as can be seen from the description with reference to FIG. 15, the active filter includes a switching element Q103 and a power factor / output voltage control IC 120 for driving the switching element Q103, and the like. On the other hand, in the switching power supply circuit of the first embodiment, a filter capacitor CN and a power factor correction diode D1 may be provided as additional components necessary for power factor improvement, which is very much compared with an active filter. The number of parts can be reduced. As a result, the power supply circuit having the power factor correction function can be manufactured at a much lower cost than the circuit shown in FIG. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight.

  Further, in the switching power supply circuit of the first embodiment, the operation of the class E switching converter and the power factor correction circuit 10 is a so-called soft switching operation, so that switching noise is reduced as compared with the circuit using the active filter shown in FIG. The level is greatly reduced. In particular, since the current input to the class E switching converter can be linked to a direct current, the level of switching noise can be made extremely small.

  In addition, the switching circuit of the first embodiment includes a secondary side series resonant circuit together with a primary side series resonant circuit and a primary side parallel resonant circuit. The voltage Eo can be maintained at a predetermined voltage, and the noise filter can be easily designed. For this reason, if a one-stage noise filter comprising one common mode choke coil CMC and two across capacitors CL is provided, it is possible to sufficiently satisfy the power source disturbance standard. Moreover, sufficient measures can be taken with respect to normal mode noise of the rectified output line by using only one filter capacitor CN.

  Further, the switching element Q1, the secondary side rectifier diode Do1, the rectifier diode Do2, and the power factor correction diode D1 and the like operate in synchronization with the switching element Q1. Therefore, the ground potential does not interfere with the active filter side and the subsequent switching converter as in the power supply circuit of FIG. 15, and can be stabilized regardless of changes in the switching frequency.

  Further, the power factor PF obtained by the switching power supply circuit of the first embodiment has extremely good characteristics as shown in FIGS. 4 and 5, and according to such power factor characteristics, the power harmonics The wave distortion regulation can be cleared, and it can be said that a practically sufficient power factor is obtained. Thus, the power supply circuit of this embodiment shown in FIG. 1 provides a power factor correction power supply that solves various problems of a power supply circuit including an active filter.

  (Second Embodiment)

  FIG. 6 shows a switching power supply circuit according to the second embodiment, which is a modification of the switching power supply circuit according to the first embodiment. In this figure, the same parts as those shown in FIG. In the switching power supply circuit of the second embodiment, a power factor correction circuit 11 is provided. In the power factor correction circuit 11, a low-speed bypass diode D1A is additionally provided in addition to the configuration of the power factor correction circuit 10 shown in FIG. The bypass diode D1A has an anode connected to the positive output terminal of the bridge rectifier circuit Di and a cathode connected to the positive terminal of the smoothing capacitor Ci. Therefore, in the power factor correction circuit 11, the bypass diode D1A is connected in parallel to the input side of the power factor correction circuit 10. By doing so, the rectified current from the bridge rectifier circuit Di also branches and flows to the bypass diode D1A, so the amount of rectified current flowing to the power factor correction diode D1 is reduced. Thereby, the switching loss in the power factor correction diode D1 is reduced and the power conversion efficiency is improved. In particular, since the amount of current flowing through the power factor correction diode D1 that tends to be heavy load increases, the efficiency improvement effect becomes remarkable.

  (Third embodiment)

  FIG. 7 shows a switching power supply circuit according to a third embodiment which is a modification of the switching power supply circuit according to the first embodiment. In this figure, the same parts as those shown in FIG. The difference between the switching power supply circuit of the first embodiment and the switching power supply circuit of the third embodiment is that an auxiliary resonance capacitor C4 connected in parallel to the primary side series resonance capacitor C2 and the auxiliary resonance capacitor C4 are connected to the primary side series resonance capacitor C2. And a capacitance adjusting switching element Q2 that controls whether or not to be connected in parallel with each other by conduction and disconnection, and further, a control winding Ng arranged in the converter transformer PIT for controlling the capacitance adjusting switching element Q2, a capacitor Cg, resistor Rg1, resistor Rg2, power factor control circuit 3, and further, a signal RDi that detects the signal SEi by dividing the rectified and smoothed voltage Ei and a signal SDi corresponding to the current flowing through the bridge rectifier circuit Di are detected. The resistor R51 is provided.

  Here, as the capacitance adjusting switching element Q2, a MOS-FET is selected, and a body diode DD is incorporated in parallel between the source and the drain. The voltage from the control winding Ng is applied to the gate of the capacity adjustment switching element Q2 by being divided by the resistor Rg2 via the capacitor Cg and the resistor Rg1. Here, the polarity of the voltage of the control winding Ng is such that the capacity adjustment switching element Q2 is turned on when the switching element Q1 is disconnected, and the capacity adjustment switching element Q2 is disconnected when the switching element Q1 is turned on, that is, The polarity is selected such that the switching element Q1 and the capacity adjustment switching element Q2 repeat conduction and disconnection in a complementary manner. Further, the time during which the capacitance adjusting switching element Q2 is turned on is controlled by the power factor control circuit 3 so that the time when the rectified and smoothed voltage Ei increases and becomes shorter as the power supplied to the load decreases. The For this purpose, the signal SEi is input to the power factor control circuit 3 as a signal corresponding to the rectified and smoothed voltage Ei, and the signal SDi is supplied as a signal corresponding to the power supplied to the load.

  Here, when the conduction time of the capacity adjustment switching element Q2 controlled by the power factor control circuit 3 is controlled by the signal SEi, the power factor is optimized with respect to the fluctuation of the input AC voltage VAC, and the signal When controlled by SDi, the power factor can be optimized with respect to fluctuations in load power. Further, when the conduction time of the capacity adjustment switching element Q2 is controlled by the signal SEi and the signal SDi, the power factor can be optimized with respect to the fluctuation of the input AC voltage VAC and the fluctuation of the load power. That is, the value of the combined capacitance of the primary side series resonance capacitor C2 and the equivalent auxiliary resonance capacitor C4 is changed by changing the equivalent capacitance value of the auxiliary resonance capacitor C4 according to the conduction time of the capacitance adjusting switching element Q2. In order to improve the power factor, the amount of voltage feedback to the smoothing capacitor Ci is changed in accordance with the change in the value of the resonance voltage generated by changing.

  Here, the constant setting of each part in 3rd Embodiment was performed as follows. The converter transformer PIT uses EER-35 as the core material, the gap G is 2.2 mm, the primary winding is 55 T, the secondary winding is 30 T, the coupling coefficient is 0.67, and the inductance of the control winding Ng The value is 360 μH, the value of the primary side parallel resonant capacitor C1 is 3900 pF, the value of the primary side series resonant capacitor C2 is 0.033 μF, the value of the secondary side series resonant capacitor C3 is 0.068 μF, and the value of the auxiliary resonant capacitor C4 is It was set to 0.033 μF. The inductor Lo (choke coil PCC) uses EER-28 as a core material and a gap of 1.2 mm. The value of the filter capacitor CN was 1 μH.

  The characteristics of the switching power supply circuit of the third embodiment having such a power factor correction circuit 12, power factor control circuit 3, control winding Ng, auxiliary resonant capacitor C4, and capacitance adjusting switching element Q2 are shown in FIGS. . In addition, since the operation waveform of each part is the same as in FIG. 3, description is abbreviate | omitted.

  FIG. 8 shows the rectified smoothing voltage Ei and force against load fluctuation when the value of the load power Po is in the range of 0 W (no load) to 300 W (maximum load power) under the input voltage condition of the AC input voltage VAC of 100V. The power conversion efficiency ηAC → DC of the DC output power with respect to the rate PF and the AC input power is shown. FIG. 9 shows the rectified smoothing voltage Ei, power factor PF, and power in a range in which the value of the AC input voltage VAC is changed from 85 V to 144 V under a load condition in which the load power is constant at 300 W (maximum load power). The conversion efficiency ηAC → DC is shown. Here, in FIG. 8, the broken line indicates the characteristic when the auxiliary resonance capacitor C4 is not used.

  As shown in FIG. 9, when the auxiliary resonant capacitor C4 is used, the rectified and smoothed voltage Ei has a rectified voltage in a low load region with respect to load fluctuations compared to the case where the auxiliary resonant capacitor C4 is not used. It is rising. This is because the period during which current flows from the bridge rectifier circuit Di to the smoothing capacitor Ci rises and the value of the rectified smoothing voltage Ei also rises in response to the power factor being greatly improved at low load. It is done. On the other hand, the rectified and smoothed voltage Ei changes substantially in proportion to the change in the AC input voltage VAC. Further, as shown in FIG. 9, the power factor PF is greatly improved in a region where the amount of power of the load is small, and reaches 0.91 in the range of 50 W to 300 W regardless of the load variation. It can be said that a practically sufficient power factor value is obtained. As shown in FIG. 10, the AC input voltage VAC is substantially constant and has a high power factor characteristic of 0.9 or more.

  As for the power conversion efficiency ηAC → DC, as shown in FIG. 9 and FIG. 10, the load power value peaks around 100 W with respect to the load fluctuation, but it can be regarded as almost constant, A power conversion efficiency ηAC → DC of 90% or more is obtained in the range of load power 25 W to load power 300 W. With respect to the fluctuation of the AC input voltage VAC, a substantially constant 90% or more is maintained in the range of 85V to 144V.

  (Fourth embodiment)

  FIG. 10 shows a switching power supply circuit according to a fourth embodiment which is a modification of the switching power supply circuit according to the third embodiment. In FIG. 10, the same portions as those in the third embodiment shown in FIG. In the switching power supply circuit according to the fourth embodiment, an inductor Lo (choke coil PCC) is constituted by a winding N3 and a winding N4, and a resonance voltage is output from the middle point between the winding N3 and the winding N4, thereby voltage feedback. The amount is adjusted. This makes it possible to simultaneously optimize the voltage feedback amount while optimizing the operation as a class E switching converter. In the switching power supply circuit of the fourth embodiment, by providing the winding N3 and the winding N4, the value of power conversion efficiency ηAC → DC can be improved by about 0.5% compared to that in the third embodiment. did it.

  (Modification of secondary circuit)

  11 and 12 show modifications of the secondary side circuit that can be replaced in any of the other first to fourth embodiments.

  The secondary side rectifier circuit shown in FIG. 11 constitutes a voltage doubler full wave rectifier circuit. That is, a center tap is applied to the secondary winding N2, and the secondary winding N2A and the secondary winding N2B are divided into two with the center tap as a boundary. The same predetermined number of turns (number of turns) is set in the secondary winding part N2A and the secondary winding part N2B. The center tap of the secondary winding N2 is connected to the secondary side ground. Further, a secondary side series resonance capacitor C3A is connected in series to the end of the secondary winding N2 on the secondary winding portion N2A side, and the end of the secondary winding N2 on the secondary winding portion N2B side is connected. The secondary side series resonant capacitor C3B is connected in series to the unit. As a result, the first secondary side series resonance circuit composed of the leakage inductance component of the secondary winding part N2A and the capacitance of the secondary side series resonance capacitor C3A, the leakage inductance component of the secondary winding part N2B, and the secondary side A second secondary side series resonance circuit composed of the capacitance of the series resonance capacitor C3B is formed.

  The end of the secondary winding N2 side of the secondary winding N2 is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the series connection of the secondary side series resonant capacitor C3A. Connect. Further, the end of the secondary winding N2 on the secondary winding N2B side is connected to the connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4 via the series connection of the secondary side series resonant capacitor C3B. Connect to each other. The cathodes of the rectifier diode Do1 and the rectifier diode Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground. Further, the connection point of each anode of the rectifier diode Do2 and the rectifier diode Do4 is connected to the secondary side ground.

  In this way, the first multiplier including the first secondary series resonance circuit including the secondary winding N2A, the secondary series resonance capacitor C3A, the rectifier diode Do1, the rectifier diode Do2, and the smoothing capacitor Co. A second half-voltage rectifier circuit and a second secondary-side series resonant circuit comprising a secondary winding N2B, a secondary-side series resonant capacitor C3B, a rectifier diode Do1, a rectifier diode Do2, and a smoothing capacitor Co. Thus, a double voltage half-wave rectifier circuit is formed. Thus, for the smoothing capacitor Co, in the half cycle of one polarity of the alternating voltage of the secondary winding N2, the induced voltage of the secondary winding portion N2B and the voltage across the secondary side series resonant capacitor C3B The rectified current is charged by the superimposed potential of the rectified current. In the half cycle of the other polarity, the rectified current is charged by the superimposed potential of the induced voltage of the secondary winding portion N2A and the voltage across the secondary side series resonant capacitor C3A. Will be. Accordingly, a level corresponding to twice the induced voltage level of the secondary winding portion N2A and the secondary winding portion N2B is obtained as the secondary side DC output voltage Eo that is the voltage across the smoothing capacitor Co. Become. That is, a voltage doubler full wave rectifier circuit is obtained.

  The secondary side rectifier circuit shown in FIG. 12 constitutes a bridge full-wave rectifier circuit. That is, four rectifier diodes Do1, rectifier diodes Do2, rectifiers are used as secondary rectifier circuits connected to a series connection circuit (secondary series resonant circuit) of secondary winding N2 and secondary side series resonant capacitor C3. A bridge rectifier circuit including a diode Do3 and a rectifier diode Do4 and a bridge full-wave rectifier circuit including one smoothing capacitor Co are provided. The winding end of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the secondary side series resonant capacitor C3. Further, the winding start end portion of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4. The cathode of the rectifier diode Do1 and the cathode of the rectifier diode Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the connection point between the anode of the rectifier diode Do2 and the anode of the rectifier diode Do4 at the secondary side ground potential.

  Depending on the full-wave rectifier circuit thus formed, in one half cycle of the alternating voltage induced in the secondary winding N2, the pair of the rectifier diode Do1 and the rectifier diode Do4 of the bridge rectifier circuit is conductive, An operation for charging the smoothing capacitor Co with a rectified current is obtained. Further, in the other half cycle of the alternating voltage induced in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining a set of the rectifying diode Do2 and the rectifying diode Do3 is obtained. As a result, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the alternating voltage level induced in the secondary winding N2 is generated as the voltage across the smoothing capacitor Co.

  In addition, the specific design example of the power supply circuit of the embodiment described so far is based on the assumption that the AC input voltage VAC is input with a commercial AC power supply of 100V. The value of VAC is not particularly limited. For example, the same effect can be obtained by adopting the configuration based on the present invention even in the case of a design corresponding to a 200V commercial AC power input. Further, for example, a detailed circuit configuration of the primary side voltage resonance type converter, a configuration of a secondary side rectifier circuit formed including a secondary side series resonance circuit, and the like are also conceivable. As the switching element, it may be considered to select an element other than the MOS-FET such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor. Further, in each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

It is a circuit diagram which shows the structural example of the switching power supply circuit of 1st Embodiment. It is sectional drawing which shows the structural example of the converter transformer with which the switching power supply circuit of embodiment is provided. It is a wave form diagram which shows operation | movement of the principal part in the power supply circuit of 1st Embodiment by a commercial alternating current power supply period. It is a figure which shows the characteristic of the rectification | straightening voltage with respect to load fluctuation | variation, a power factor, and power conversion efficiency about the power supply circuit of 1st Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, power factor, and power conversion efficiency with respect to alternating current input voltage fluctuation | variation about the power supply circuit of 1st Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of 2nd Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of 3rd Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage with respect to load fluctuation | variation, a power factor, and power conversion efficiency about the power supply circuit of 3rd Embodiment. It is a figure which shows the characteristic of the rectification | straightening voltage, power factor, and power conversion efficiency with respect to alternating current input voltage fluctuation | variation about the power supply circuit of 3rd Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of 4th Embodiment. It is a modification of the secondary side circuit of embodiment. It is a modification of the secondary side circuit of embodiment. It is a block diagram of the active filter shown in background art. It is a wave form diagram explaining operation | movement of the active filter shown to background art. It is a circuit diagram which shows the structural example of the switching power supply circuit shown in background art. It is a wave form diagram explaining operation | movement of the active filter shown to background art. It is a wave form diagram which shows the alternating current input voltage in the power supply circuit which mounted the active filter shown in background art, an alternating current, and a smooth voltage with a commercial alternating current power supply period. It is the characteristic view shown about each characteristic of the power conversion efficiency with respect to the load fluctuation | variation of the power supply circuit which mounted the active filter shown in background art, a power factor, and a rectification smoothing voltage. It is the characteristic view shown about each characteristic of the power conversion efficiency with respect to the alternating current input voltage fluctuation | variation of a power supply circuit which mounted the active filter shown in background art, a power factor, and a rectification smoothing voltage.

Explanation of symbols

1 control circuit, 2 oscillation / drive circuit, 3 power factor control circuit,
10, 11, 12 Power factor correction circuit AC Commercial AC power supply, C2 Primary side series resonant capacitor, C3, C3A, C3B Secondary side series resonant capacitor, CL across capacitor, CMC common mode choke coil CN Filter capacitor, Ci, Co smoothing Capacitor, C1 Capacitor CR1, CR2, CR11, CR12 Core, D1 Power factor improving diode D11, Do1, Do2, Do3, Do4 Rectifier diode D1A Bypass diode, DD Body diode Di Bridge rectifier circuit, Ei Rectified smoothed voltage Eo Secondary side DC Output voltage, G gap, I1, I2 current, IAC AC input current L1, L2 inductor, Lo inductor (PCC choke coil)
LFT line filter transformer, N1 primary winding, N3, N4 winding N2, secondary winding, N2A secondary winding, N2B secondary winding PF power factor, PIT converter transformer Q1, Q2 switching elements R51, R52, R53, Rg1, Rg2 resistance, SDi, SEi signal V1, V2, V3 voltage, VAC AC input voltage IAC AC input current, ηAC → DC Power conversion efficiency

Claims (4)

A rectifying / smoothing circuit that is formed by including a rectifying element and a smoothing capacitor that input and rectifies and smoothes AC power from an AC power source, and generates a rectified and smoothed voltage as a voltage across the smoothing capacitor;
A choke coil to which the rectified and smoothed voltage is applied to one terminal;
A converter transformer having a leakage inductor in which one terminal of the primary winding is connected to the other terminal of the choke coil;
The one terminal of the primary winding of the converter transformer is connected to a primary side series resonance capacitor, and the primary side series resonance is governed by the primary side inductance component generated by the leakage inductor and the capacitance of the primary side series resonance capacitor. A primary side series resonant circuit having a frequency;
A primary side having a primary side parallel resonant frequency that is connected in parallel with the primary winding and a primary side parallel resonant capacitor, and is governed by a primary side inductance component generated by the leakage inductor and a capacitance of the primary side parallel resonant capacitor. A parallel resonant circuit;
A switching element connected to the other terminal of the primary winding to supply AC power to the primary side series resonant circuit and the primary side parallel resonant circuit;
An oscillation / drive circuit for driving the switching element;
The secondary winding of the converter transformer is connected to a secondary side series resonance capacitor, and the secondary side resonance frequency governed by the secondary side inductance component generated by the leakage inductor and the capacitance of the secondary side series resonance capacitor A secondary side series resonant circuit having
A control circuit for supplying a control signal to the oscillation / drive circuit such that the value of the secondary side DC output voltage output from the secondary side series resonance circuit by the secondary side rectifier circuit is a predetermined value;
A power factor correction diode having one terminal connected to the other terminal of the primary side series resonance capacitor and the choke coil, and the other terminal connected to the rectifier element, and the other terminal of the power factor correction diode And a filter capacitor having respective terminals connected to the rectifying and smoothing capacitor, and a power factor improving circuit,
A switching power supply circuit comprising:   The switching power supply circuit according to claim 1, wherein the primary side series resonance frequency and the secondary side resonance frequency are set to substantially equal frequencies.   The switching power supply circuit according to claim 1, further comprising: a bypass diode that bypasses a rectified current from the rectifier element to the smoothing capacitor, between the rectifier element and the smoothing capacitor. Further, a series connection circuit of an auxiliary resonance capacitor and a capacity adjustment switching element connected in parallel with the primary side series resonance capacitor is provided, and the switching element and the capacity adjustment switching element repeat conduction and disconnection in a complementary manner. The switching power supply circuit according to claim 1, wherein a conduction time of the capacitance adjusting switching element is controlled according to a magnitude of a current flowing through the rectifying element and / or a magnitude of the rectified smoothing voltage.

Images