JP2007288823A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply circuit that copes with widening of the range, as well as, that is high in efficiency. <P>SOLUTION: The switching power supply circuit comprises a primary series resonance circuit by the leakage inductance L1 of a converter transformer PIT and a primary-side series resonance capacitor C1; and a secondary parallel resonance circuit by an intermediate tap formed in the secondary winding; and a secondary parallel resonance capacitor C3 forming a full-wave rectifier circuit, having diodes Do1 and Do2 for rectifying AC power obtained between the intermediate tap and each end of the secondary winding and connected to the intermediate tap and the secondary winding N2 via a smoothing capacitor Co. The secondary parallel resonance frequency is set equal to about three times the primary series resonance frequency. With such an arrangement, ZVS region is extended, efficiency is improved, and wide range DC input voltage can be coped. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

  Previously, the present applicant has proposed various power supply circuits including a resonant converter on the primary side (see Patent Document 1). FIG. 34 is a circuit diagram showing an example of a switching power supply circuit including a resonance type converter configured based on the invention previously filed by the present applicant. The switching converter of the power supply circuit shown in FIG. 34 has a configuration in which a partial voltage resonance circuit that performs a voltage resonance operation only at the time of turn-off during switching is combined with a separately excited current resonance type converter using a half-bridge coupling method. take. The switching power supply circuit shown in this figure is used as a power supply for a printer device, for example. For example, the printer apparatus is subjected to a load fluctuation condition in a relatively wide range from about 100 W or more to no load.

  First, in the switching power supply circuit shown in FIG. 34, a common mode noise filter including two sets of filter capacitors CL and one set of common mode choke coil CMC is connected to commercial AC power supply AC. As a rectifying / smoothing circuit for generating a DC input voltage from the commercial AC power supply AC, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci is provided for the subsequent stage of the common mode noise filter. The rectified output of the bridge rectifier circuit Di is charged to the smoothing capacitor Ci, whereby a rectified and smoothed voltage Ei (DC input voltage) corresponding to an equal level of the AC input voltage VAC is applied to both ends of the smoothing capacitor Ci. Will be obtained.

  As shown in the figure, the current resonance type converter that switches by inputting the rectified and smoothed voltage Ei includes a switching circuit system in which two switching elements Q1 and switching elements Q2 by MOS-FETs are connected by half bridge coupling. Between each drain and source of the switching element Q1 and the switching element Q2, a damper diode DD1 and a damper diode DD2 by body diodes are connected in parallel according to the direction shown in the drawing.

  A primary side partial voltage resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the primary side partial voltage resonance capacitor Cp and the leakage inductance (leakage inductance) L1 generated in the primary winding N1. By this partial voltage resonance circuit, a partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1 and Q2 are turned off can be obtained. The leakage inductance L1 is an inductance generated by a magnetic flux (leakage magnetic flux) generated by the primary winding N1 and not interlinked with the secondary winding N2. Similarly, the secondary winding N2 generates a leakage inductance (leakage inductance) L2, which is an inductance generated by the secondary winding N2 and generated by a magnetic flux (leakage magnetic flux) not linked to the primary winding N1. .

  In this power supply circuit, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided in order to perform switching driving of the switching elements Q1 and Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and applies a drive signal (gate voltage) having a required frequency to the gates of the switching element Q1 and the switching element Q2. Thereby, the switching element Q1 and the switching element Q2 perform a switching operation so as to be alternately turned on / off at a required switching frequency.

  A converter transformer PIT (Powerwr Isolation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. In this case, one end of the primary winding N1 of the converter transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, thereby obtaining a switching output. The The other end of the primary winding N1 is connected to the primary side ground via a primary side series resonance capacitor C1 as shown in the figure.

  In this case, the primary side series resonant capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the primary side series resonant capacitor C1 and the leakage inductance generated in the primary winding N1 of the converter transformer PIT. L1 forms a primary side series resonance circuit for making the operation of the switching converter into a current resonance type.

  According to the description so far, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1 and C1) and the partial voltage resonance circuit (Cp and L1) described above. Thus, a partial voltage resonance operation is obtained. That is, the power supply circuit shown in this figure adopts a form in which another resonance circuit is combined with a resonance circuit for making the primary side switching converter a resonance type. Hereinafter, such a switching converter is referred to as a “composite resonance type converter”.

  The structure of the converter transformer PIT will be described with reference to FIG. The converter transformer PIT includes an EE type core obtained by combining an E type core CR1 and an E type core CR2 made of a ferrite material. Then, the bobbin B is used to divide the winding site on the primary side and the secondary side, and then the primary winding N1 and the secondary winding N2 are wound around the inner magnetic legs of the EE core. Disguise. A gap G of about 0.8 mm is formed with respect to the inner magnetic leg of the EE type core of the converter transformer PIT. Thus, a coupling coefficient of about 0.8 to 0.85 is obtained by the primary winding N1 and the secondary winding N2.

  The secondary winding N2 of the converter transformer PIT is provided with a double-wave rectifier circuit formed by a rectifier diode Do1, a rectifier diode Do2, and a smoothing capacitor Co. As a result, the secondary side DC output voltage Eo, which is a DC voltage corresponding to the same level of the alternating voltage induced in the secondary winding N2, is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied as a DC power source to a load (not shown) and is also given to the control circuit 1 as a detection voltage for constant voltage control.

  The control circuit 1 outputs a control signal as a voltage or current whose level is variable in accordance with the level of the secondary side DC output voltage Eo to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the oscillation signal frequency generated by the oscillation circuit in the oscillation / drive circuit 2 is varied based on the control signal input from the control circuit 1. The frequency of the switching drive signal applied to the gate is changed. As a result, the switching frequency is varied. As described above, the switching frequency of the switching element Q1 and the switching element Q2 is variably controlled in accordance with the level of the secondary side DC output voltage Eo, whereby the resonance impedance of the primary side series resonance circuit is changed and the primary side is changed. The energy transmitted from the primary winding N1 forming the series resonance circuit to the secondary side is also variable, and the level of the secondary side DC output voltage Eo is also variably controlled. Thereby, constant voltage control of the secondary side DC output voltage Eo is achieved. In the following, the constant voltage control method that achieves stabilization by variably controlling the switching frequency will be referred to as a “switching frequency control method”.

  The waveform diagram of FIG. 36 shows a waveform during operation of the main part of the power supply circuit shown in FIG. At this time, the AC input voltage VAC = 100 V is constant, and the load power Po is 200 W (watts). From the top, the waveform shows voltage V1 (see FIG. 34), current IQ2 (see FIG. 34), current I1 (see FIG. 34), and current I2 (see FIG. 34). The waveform shown in FIG. 37 is described on the same page for easy understanding of the correspondence with the waveform of FIG. 36, which will be described later.

  The rectangular wave voltage V1 is a voltage across the switching element Q2, and changes according to the on / off timing of the switching element Q2. The period during which the voltage V1 is 0 level is an ON period in which the switching element Q2 is conductive. In this ON period, the switching circuit IQ comprising the switching element Q2 and the damper diode DD2 includes a switching current IQ2 having a waveform shown in the figure. Flows. Further, the period during which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei is a period during which the switching element Q2 is turned off, and the current IQ2 is at 0 level as shown in the figure.

  Although not shown, the voltage across the other switching element Q1 and the current flowing through the switching circuit (Q1, DD1) are obtained as a waveform in which the voltage V1 and the current IQ2 are 180 ° out of phase. That is, as described above, the switching element Q1 and the switching element Q2 perform the switching operation at the timing of turning on / off alternately.

  Further, as the primary side series resonance current flowing through the primary side series resonance circuit (C1-L1), the currents flowing through the switching circuit (Q1, DD1) and the switching circuit (Q2, DD2) are combined. , As shown in FIG.

  FIG. 38 shows DC power conversion efficiency ηAC → DC (hereinafter abbreviated as power conversion efficiency ηAC → DC) and switching frequency with respect to AC AC with respect to load fluctuation under an input voltage condition of AC input voltage VAC = 100V. The characteristics are shown. The curves shown by the alternate long and short dash line in FIG. 38 indicate each of the power supply circuit shown in FIG. The switching frequency fs is lower as the constant voltage control operation is performed, and the load frequency Po is not a change characteristic that is linear with respect to the load fluctuation. In the range from about 50 W to 0 W, the switching frequency fs tends to increase sharply. Specifically, the value of the switching frequency fs is in the range of 76 kHz to 173 kHz, and the value of the change width Δfs is 97 kHz.

  Similarly, the power conversion efficiency ηAC → DC shown by the one-dot chain line in FIG. 38 tends to increase as the load power Po increases.

  According to the characteristics shown in FIG. 38, since the value of the switching frequency fs changes greatly, for example, the operation of the AC input voltage VAC in the range of 85 V to 264 V, so-called wide range compatibility, It turns out to be difficult.

  FIG. 39 shows another switching power supply circuit that supports a wide range from such a viewpoint. In the following, the same parts as those of the switching power supply circuit shown in FIG. The difference between the switching power supply circuit shown in FIG. 39 and the switching power supply circuit shown in FIG. 34 is that the switching power supply circuit shown in FIG. 39 has a secondary side series resonant capacitor C2 on the secondary side. Thus, a converter having a resonance circuit on the secondary side is referred to as a “multiple resonance type converter”. Here, since the primary side of the switching power supply circuit shown in FIG. 39 is also a “composite resonance type converter”, such a converter is referred to as a “multiple complex resonance type converter”.

  In the switching power supply circuit shown in FIG. 39, each part is set as follows. With respect to the gap with respect to the inner magnetic leg of the EE type core of the converter transformer PIT, the gap is further expanded to about 2 mm than the gap of about 0.8 mm in the switching power supply circuit shown in FIG. As a result, a coupling coefficient of 0.75 or less is obtained.

  Further, the resonance frequency of the primary side series resonance circuit determined by the leakage inductance L1 generated in the primary winding N1 and the primary side series resonance capacitor C1 is set to the leakage inductance L2 generated in the resonance frequency fso1 and the secondary winding N2 and the secondary. When the resonance frequency fso2 is set as the resonance frequency of the secondary side series resonance circuit determined by the side series resonance capacitor C2, a wide range can be achieved by setting the relationship such that the resonance frequency fso1 <the resonance frequency fso2.

  Here, the waveform diagram of FIG. 37 shows waveforms during operation of the main part of the power supply circuit shown in FIG. 39 described above. At this time, the AC input voltage VAC = 100 V is constant, and the load power Po is 200 W (watts). From the top, the waveforms show voltage V1 (see FIG. 39), current IQ2 (see FIG. 39), current I1 (see FIG. 39), and current I2 (see FIG. 39). It should be noted that the values of current IQ2 and current I1 shown in FIG. 37 are more suppressed than the values of current IQ2 and current I1 shown in FIG.

  The solid line and the broken line in FIG. 38 indicate the characteristics at the AC input voltage VAC = 100 V, and the broken line indicates the characteristic at the AC input voltage VAC = 230 V, for the power supply circuit shown in FIG. . That is, the characteristics of the power conversion efficiency ηAC → DC and the switching frequency with respect to the load variation under the condition of each AC input voltage are shown. The switching frequency fs changes according to the constant voltage control operation. However, when the load power Po is in the range from about 200 W to 0 W, the value of the change width Δfs of the switching frequency fs is as small as 9 kHz. is there.

  As described above, since the value of the change width Δfs of the switching frequency fs can be suppressed to be small by using the multiple composite resonance type converter, a wide range is possible as long as the characteristics of the change width Δfs of the switching frequency fs are observed. it is conceivable that.

JP 2003-235259 A

  However, it can be seen from FIG. 38 that the power conversion efficiency ηAC → DC shown by the solid and broken lines in FIG. 38 has the following problems. That is, according to the characteristics shown in FIG. 38, the power conversion efficiency ηAC → DC at the AC input voltage VAC = 230 V indicated by the broken line is that in the switching power supply circuit shown in FIG. 34 represented by the alternate long and short dash line at the load power Po = 200 W. However, the load power Po = 150 W or less is lower than that in the switching power supply circuit shown in FIG.

  The present invention provides a switching power supply circuit that solves the above-described problems and copes with a wide range and is highly efficient.

  A switching power supply circuit according to the present invention includes a primary side circuit that generates a DC power by inputting a DC input voltage, and a primary winding and a secondary winding for transmitting the AC power to a secondary side circuit. A converter transformer formed by being wound around a constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied from the secondary side circuit to the load becomes a constant voltage; A switching power supply circuit comprising: a switching means formed with a switching element whose switching frequency is controlled by the constant voltage control means; and a leakage generated in a primary winding of the converter transformer. It is formed such that the primary side series resonance frequency is dominated by the inductance component and the capacitance of the primary side series resonance capacitor connected in series to the primary winding. A primary-side series resonant circuit whose operation is the current resonance type, and the secondary-side circuit includes an intermediate tap formed in the secondary winding and both ends of the secondary winding. A secondary-side DC output voltage generating means for forming a full-wave rectifier circuit having a rectifier diode for rectifying the AC power obtained between each of the two, and an intermediate tap formed on the secondary winding; A secondary side parallel resonant circuit formed such that the secondary side parallel resonant frequency is dominated by a secondary side parallel resonant capacitor connected directly or AC to one end of the secondary winding. The secondary side parallel resonance frequency is set to a frequency approximately three times the primary side series resonance frequency.

  In this switching power supply circuit, the primary side series resonant frequency is controlled by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side series resonant capacitor connected in series to the primary winding. And a primary-side series resonance circuit that is formed as described above and has the operation of the switching means as a current resonance type. In addition, the secondary side circuit is a full-wave rectifier circuit including an intermediate tap formed in the secondary winding and a rectifier diode that rectifies AC power obtained between both ends of the secondary winding. And the secondary parallel resonant frequency is dominated by the secondary parallel resonant capacitor that is directly or AC connected to the intermediate tap and one end of the secondary winding. A secondary parallel resonant circuit. The secondary side parallel resonance frequency is set to a frequency that is approximately three times the primary side series resonance frequency. As a result, the ZVS region can be expanded, efficiency can be improved, and a wide range of DC input voltages can be handled.

  Another switching power supply circuit according to the present invention includes a primary side circuit that receives a DC input voltage to generate AC power, and a primary winding and a secondary winding for transmitting the AC power to the secondary side circuit. A converter transformer formed by being wound around a core, and constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied from the secondary side circuit to the load becomes a constant voltage; The primary side circuit includes a switching means formed with a switching element whose switching frequency is controlled by the constant voltage control means, and a primary winding of the converter transformer. Formed such that the primary side series resonance frequency is dominated by the leakage inductance component generated and the capacitance of the primary side series resonance capacitor connected in series to the primary winding. And a primary side series resonance circuit that makes the operation of the switching means a current resonance type, and the secondary side circuit obtains a voltage that is twice the AC voltage obtained in the secondary winding. Formed so that the secondary side parallel resonant frequency is dominated by the secondary side DC output voltage generating means having a voltage rectifier circuit and the secondary side parallel resonant capacitor connected directly or AC to the secondary winding. A secondary side parallel resonant circuit, and the secondary side parallel resonant frequency is set to a frequency that is approximately twice the primary side series resonant frequency.

  In this switching power supply circuit, the primary side series resonance frequency is dominated by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding. And a primary side series resonance circuit having a current resonance type operation of the switching means. Further, the secondary side circuit includes a secondary side DC output voltage generating means having a voltage doubler rectifier circuit for obtaining a voltage twice as high as the AC voltage obtained in the secondary winding, and directly or AC in the secondary winding. A secondary side parallel resonant circuit formed such that the secondary side parallel resonant frequency is dominated by the connected secondary side parallel resonant capacitor. The secondary side parallel resonance frequency is set to a frequency that is approximately twice the primary side series resonance frequency. As a result, the ZVS region can be expanded, efficiency can be improved, and a wide range of DC input voltages can be handled.

  According to the present invention, it is possible to provide a switching power supply circuit that can cope with a wide range and has high efficiency.

“First Embodiment”
The switching power supply circuit according to the first embodiment shown in FIG. 1 includes a primary side circuit that generates a high-frequency AC power by inputting a DC input voltage obtained as a rectified and smoothed voltage Ei, and this AC power to a secondary side circuit. A converter transformer PIT formed by winding a primary winding N1 and a secondary winding N2 for transmission on a core, and a secondary side DC output supplied from a secondary side circuit to a load (not shown) The switching power supply circuit includes a control circuit 1 and an oscillation / drive circuit 2 that function as constant voltage control means for controlling the primary circuit so that the voltage Eo becomes a constant voltage. Here, the primary side circuit and the secondary side circuit are separated by the primary winding N1 and the secondary winding N2 of the converter transformer, and the left side of the primary winding N1 in FIG. A circuit on the right side of the secondary winding N2 in FIG. 1 is a secondary circuit.

  The primary side circuit includes a switching element Q1 and a switching element Q2 whose switching frequency is controlled by the oscillation / drive circuit 2 constituting the constant voltage control means, and a primary of the converter transformer PIT. Formed such that the primary side series resonance frequency is governed by the leakage inductance L1 which is a leakage inductance component generated in the winding N1 and the capacitance of the primary side series resonance capacitor C1 connected in series to the primary winding N1, A primary-side series resonance circuit in which the operation of the switching means is a current resonance type. Here, the fact that the primary-side series resonance frequency is dominant means that the leakage inductance L1 and the primary-side series resonance capacitor C1 are the main factors that determine the resonance frequency fso1.

  In addition, the secondary side circuit rectifies the AC power obtained in the secondary winding N2 and the secondary winding N2 ′ to generate a secondary DC output voltage Eo, and forms a double-wave rectifier circuit Do1 and Secondary side DC output voltage generating means having a rectifier diode Do2 and a smoothing capacitor Co, an intermediate tap formed by connection of the secondary winding N2 and the secondary winding N2 ′, and the secondary winding N2 or secondary winding. A secondary side parallel resonant circuit formed such that the secondary side parallel resonant frequency fpo2 is dominated by the secondary side parallel resonant capacitor C3 that is directly or AC connected to one end of the line N2 ′. And. Here, the AC connection is not only directly connected to the secondary winding N2 ′ but also connected to the secondary winding N2 ′ via the smoothing capacitor Co1 as shown in FIG. This includes cases where That is, when the value of the capacitance of the smoothing capacitor Co1 is an order of magnitude larger than the capacitance of the secondary side parallel resonant capacitor C3, the smoothing capacitor Co1 can be regarded as a short circuit in terms of AC. .

  When the connection mode of the secondary side parallel resonant capacitor C3 is viewed from another point of view, the capacitance of the secondary side parallel resonant capacitor C3 is connected in parallel with the rectifier diode Do2 forming the double-wave rectifier circuit. Can also be seen. Here, the AC connection in parallel includes not only the case of being directly connected in parallel to the rectifier diode Do2, but also the case of being connected in parallel via the smoothing capacitor Co. In FIG. 1, since the value of the capacitance of the smoothing capacitor Co is larger than the value of the capacitance of the secondary side parallel resonant capacitor C3, the smoothing capacitor Co can be regarded as a short circuit in terms of AC, and the secondary side parallel resonant capacitor C3. Corresponds to a state of being connected in parallel to the rectifier diode Do2 in an alternating manner.

  Further, the fact that the secondary side parallel resonance frequency is dominated means that the leakage inductance L2 'and the secondary side parallel resonance capacitor C3 are the main factors that determine the resonance frequency fpo2. In FIG. 1, the value of the smoothing capacitor Co also affects the secondary side parallel resonance frequency, but as described above, the value of the capacitance of the smoothing capacitor Co becomes the value of the capacitance of the secondary side parallel resonance capacitor C3. Since it is relatively large, it does not become a main factor for determining the resonance frequency fpo2. Here, in the first embodiment, the secondary side parallel resonance frequency is set to be higher than the primary side series resonance frequency, and the secondary side parallel resonance frequency is substantially equal to the primary side series resonance frequency. The frequency is set to 3 times. Note that “approximately 3 times” includes an error range of 20%. Hereinafter, the first embodiment will be described in more detail.

  FIG. 1 is a diagram illustrating a configuration of the switching power supply circuit according to the first embodiment. The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a separately excited current resonance converter using a half-bridge coupling method as a primary side basic configuration. In addition, the power supply circuit of the first embodiment is either a 100V system (a voltage system in which the value of the AC input voltage VAC is around 100V) or a 200V system (a voltage system in which the value of the AC input voltage VAC is around 200V). A so-called wide-range configuration that operates in response to commercial AC power input is adopted. Moreover, as corresponding load electric power, for example, it corresponds to the fluctuation range from about load electric power Po = 200W to Po = 0W (no load).

  In the switching power supply circuit shown in FIG. 1, the power path from the input side to the output side will be described in order below. First, a common mode noise filter including a filter capacitor CL and a common mode choke coil CMC is formed for the commercial AC power supply AC.

  A full-wave rectifier circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci is connected to the commercial AC power supply AC that is a subsequent stage of the noise filter. When this full-wave rectifier circuit inputs a commercial AC power supply AC and performs a full-wave rectification operation, a rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC.

  As a current resonance type converter for switching (intermittently) by inputting the rectified and smoothed voltage Ei, as shown in the figure, a switching circuit in which two switching elements Q1, MOS-FETs are connected by a half-bridge coupling. Is provided. Damper diodes DD1 and DD2 are connected in parallel between the drain and source of switching element Q1 and switching element Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes included in the switching element Q1 and the switching element Q2, respectively, and are naturally formed in the MOS-FET manufacturing process.

  A primary side partial voltage resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the primary side partial voltage resonance capacitor Cp and the leakage inductance (leakage inductance) L1 of the primary winding N1. In addition, a partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1 and Q2 are turned off is obtained.

  In addition, an oscillation / drive circuit 2 is provided for switching the switching elements Q1 and Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. In this case, for example, a general-purpose IC can be used. The oscillation circuit of the oscillation / drive circuit 2 generates an oscillation signal having a required frequency, and the drive circuit generates a switching drive signal that is a gate voltage for switching the MOS-FET using the oscillation signal, The voltage is applied to the gates of the switching element Q1 and the switching element Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation continuously so that both switching elements are alternately turned on / off according to the switching frequency according to the cycle of the switching drive signal.

  The converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side. This converter transformer PIT has a configuration similar to that shown in FIG. 35 as background art. One end of the primary winding N1 is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, so that the switching output is transmitted. Yes. The other end of the primary winding N1 is connected to the primary side ground via the primary side series resonance capacitor C1.

  Here, as the gap of the inner magnetic leg of the EE type core of the converter transformer PIT, for example, a gap length of about 1.6 mm is set, so that the value of the coupling coefficient k between the primary side and the secondary side is 0. A loosely coupled state of about 72 is obtained. The gap can be formed by making the inner magnetic legs of the E-type core CR1 and the E-type core CR2 shown in FIG. 35 shorter than the two outer magnetic legs. The core material of the E-type core CR1 and E-type core CR2 was EER-35 (core material model number). The converter transformer PIT generates a predetermined leakage inductance L1 in the primary winding N1 by such a structure. In the first embodiment, as the value of the coupling coefficient k, the same performance can be obtained in the range of about 0.7 to 0.8, and the gap at this time is in the range of about 1 mm to 2 mm. there were.

  The primary winding N1 and the primary side series resonance capacitor C1 are connected in series. Therefore, a series resonance circuit (primary side series resonance circuit) is formed by the leakage inductance L1 of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1. Here, the number of turns of the primary winding N1 was 30 T (turns), and the capacity of the primary side series resonance capacitor C1 was 0.082 μF (micro farads). The resonance frequency fso1 of the primary side series resonance circuit at this time was 54.3 kHz (kilohertz).

  And this primary side series resonance circuit is connected with respect to the switching output point of switching element Q1 and switching element Q2. Accordingly, the switching outputs of the switching elements Q1 and Q2 are transmitted to the primary side series resonance circuit. In the primary side series resonant circuit, the operation of the primary side switching converter is changed to the current resonance type by performing the resonance operation by the transmitted switching output.

  The primary side of the switching converter shown in FIG. 1 is a primary side partial voltage resonance which is a parallel resonance circuit formed by a leakage inductance L1 generated in the primary winding N1 and a primary side partial voltage resonance capacitor Cp. It has a circuit. Here, the value of the primary side partial voltage resonance capacitor Cp is 680 pF (pico farad).

  In this way, on the primary side of the switching power supply circuit of the first embodiment shown in FIG. 1, a current resonance circuit is formed by the leakage inductance L1 generated in the primary winding N1 and the primary side series resonance capacitor C1. And a “composite resonance converter” having a primary side partial voltage resonance circuit which is a parallel resonance circuit formed by the leakage inductance L1 of the primary winding N1 and the primary side partial voltage resonance capacitor Cp. Yes. The resonance frequency fso1 of the primary side series resonance circuit of the primary side series resonance circuit is substantially determined by the leakage inductance L1 and the primary side series resonance capacitor C1, and the resonance frequency fpo1 of the primary side partial voltage resonance circuit is the leakage. It is substantially determined by the inductance L1 and the primary side partial voltage resonance capacitor Cp.

  Next, the secondary side of the switching power supply circuit according to the first embodiment will be described. A rectifier diode Do2 is connected to the secondary winding N2 of the converter transformer PIT, and a rectifier diode Do1 is connected to the secondary winding N2 'so as to obtain a secondary side DC output voltage Eo. Here, the secondary side of converter transformer PIT has secondary winding N2 ′ having the same number of turns as secondary winding N2 and secondary winding N2, and secondary winding N2 and secondary winding. The connection point of N2 ′ is an intermediate tap. That is, the voltage at one end of the secondary winding N2 with respect to the intermediate tap (a point not on the intermediate tap side) and the end of the secondary winding N2 ′ with the intermediate tap as a reference (a point not on the intermediate tap side) The phase is 180 ° different from the voltage. Further, the secondary parallel resonant capacitor C3 is connected between one end of the secondary winding N2 of the converter transformer PIT and the grounding point on the secondary side, and is connected in an alternating manner, that is, through the smoothing capacitor Co. The side parallel resonant capacitor C3 is connected in parallel to the rectifier diode Do2.

  By such a connection mode on the secondary side, the current I2 flowing through the intermediate tap between the secondary winding N2 and the secondary winding N2 ′ is equal to the current I3 flowing through the secondary parallel resonant capacitor C3 and the two-wave rectification. When the voltage at the one end of the secondary winding is the voltage V2, which is the sum of the current I4 flowing to the load side, which is the side to which the load (not shown) is connected via the rectifier diode Do2 constituting the circuit The voltage V2 (see FIG. 1) flows as follows according to the sign of the voltage V2 (see FIG. 1).

  When the voltage V2 is positive, the current I4 flows from one end of the secondary winding N2 in the order of the rectifier diode Do2, the smoothing capacitor Co, the secondary winding N2, and the intermediate tap of the secondary winding N2 ′. When the diode Do2 is turned off, a current I3 that is a parallel resonance current flows from one end of the secondary winding N2 in the order of the secondary side parallel resonance capacitor C3, the secondary winding N2, and the intermediate tap of the secondary winding N2 ′. Here, the current I2 is the sum of the current I4 and the current I3 as described above.

  On the other hand, when the voltage V2 is negative, the current I4 flows from one end of the secondary winding N2 ′ in the order of the rectifier diode Do1, the smoothing capacitor Co, the secondary winding N2, and the intermediate tap of the secondary winding N2 ′. When the rectifier diode Do1 is turned off, the current I3, which is a parallel resonance current, flows from the intermediate tap of the secondary winding N2 and the secondary winding N2 ′ to one end of the secondary parallel resonance capacitor C3 and the secondary winding N2. It flows in order. Here, the current I2 is the sum of the current I4 and the current I3 as described above.

  Here, as described above, the capacitance value of the secondary side parallel resonant capacitor C3 and the capacitance value of the smoothing capacitor Co are Co >> C3 (the capacitance value of the smoothing capacitor Co is secondary). That is much larger than the capacitance value of the side parallel resonant capacitor C3). Therefore, the secondary side voltage resonance circuit has a resonance frequency fpo2 of the secondary side voltage resonance circuit by the leakage inductance L2 generated in the secondary winding N2 and the electrostatic capacitance of the secondary side parallel resonance capacitor C3. Is formed so as to be substantially determined. Here, the capacitance of the secondary parallel resonant capacitor C3 is set to 5600 pF, and the value of the resonant frequency fpo2 is set to 174.2 kHz.

  As described above, the resonance frequency fpo2 is set to be three times or more the resonance frequency fso1, the value of the resonance frequency fso1 is set to 54.3 kHz, and the value of the resonance frequency fpo2 is set to 174.2 kHz in the present embodiment. Yes. In such a setting, a stable ZVS operation can be obtained with respect to a wide range of fluctuations between the AC input voltage VAC and the load power Po. The current I3 flowing through the secondary winding N2 or the secondary winding N2 ′ and the secondary side parallel resonant capacitor C3 is a current I4 flowing toward the smoothing capacitor Co when the rectifier diode Do1 or the rectifier diode Do2 is conducted. The current I4 decreases as the load power Po decreases, and at the same time the current I3 decreases. Thus, when the values of the peak currents of the currents I4 and I3 are reduced, the value of the power conversion efficiency ηAC → DC becomes closer to 1 when the value of the load power Po is small, and the power conversion efficiency is improved. To do. That is, the loss caused by the current I3 is the loss of the secondary parallel resonant capacitor C3, the copper loss generated in the secondary winding N2, and the loss of the rectifier diode Do2, and the loss caused by the current I4 is the loss of the rectifier diode Do2. The loss of the smoothing capacitor Co and the copper loss generated in the secondary winding N2. Therefore, the magnitude of the loss proportional to the square of the current is reduced overall when viewed as a whole device by distributing the current to many elements without concentrating the current on some elements. . Further, at the maximum load, the period during which the current I3 flows through the secondary winding N2 or the secondary winding N2 ′ and the secondary parallel resonant capacitor C3 is shortened, so that compared with the switching power supply circuit shown in the background art. Thus, the value of the power conversion efficiency ηAC → DC becomes better.

  As described above, in addition to the primary side series resonance circuit and the secondary side parallel resonance circuit in which each resonance frequency is set so that the resonance frequency fpo2 is three times or more of the resonance frequency fso1, the resonance frequency fpo1. Is provided, and the loss in the switching element is reduced by setting the resonance frequency fso1 <resonance frequency fpo1, and further excellent characteristics are obtained.

  The control circuit 1 will be described. The control circuit 1 is provided to stabilize the secondary side DC output voltage Eo by the switching frequency control method. In this case, the control circuit 1 supplies the oscillation / drive circuit 2 with a detection output corresponding to the level change of the secondary side DC output voltage Eo as a detection input. The oscillation / drive circuit 2 drives the switching element Q1 and the switching element Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. For this purpose, the frequency of the oscillation signal generated by the internal oscillation circuit is varied. By changing the switching frequency of the switching element Q1 and the switching element Q2, the resonance impedance of the primary side series resonance circuit changes and is transmitted from the primary winding N1 of the converter transformer PIT to the secondary winding N2 side. Although the electric energy changes, the operation is performed so as to stabilize the level of the secondary side DC output voltage Eo.

  2 and 3 show operation waveforms of each part of the power supply circuit shown in FIG. In these drawings, FIG. 2 shows an operation waveform when the load power Po = 200 W (maximum load power), and FIG. 3 shows an operation waveform when the load power Po = 0 W (no load power). In these figures, experimental results are shown when the AC input voltage VAC = 100 V is constant.

  2 and 3, the voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2. That is, the current IQ2 of the illustrated switching element Q2 flows during the period in which the voltage V1 is 0 level, and the switching element Q2 is turned on during this period. Further, it can be seen that the current IQ2 becomes 0 level during the period in which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei as shown, and the switching element Q2 is turned off during this period. Although not shown, the voltage across one switching element Q1 is obtained as a waveform obtained by shifting the phase of the voltage V1 by 180 degrees. Similarly, a waveform obtained by shifting the phase of the current IQ2 by 180 degrees is obtained as the current of the switching element Q1. That is, the switching element Q1 and the switching element Q2 are alternately turned on / off.

  2 and 3, the current I1 is a resonance current that flows through the primary side series resonance circuit, and has a waveform as shown in the figure, in which the waveforms of the current of the switching element Q1 and the current IQ2 are combined. Is.

  When the waveforms of voltage V1 and current IQ2 shown in FIGS. 2 and 3 are compared with each other, the period of these waveforms shown in FIG. 3 is shorter than the period of these waveforms shown in FIG. ing. This indicates that the switching frequency is controlled so as to increase as the load changes from heavy to light. That is, as a stabilization control, when the level of the secondary side DC output voltage Eo decreases due to heavy load, the switching frequency is lowered and the level of the secondary side DC output voltage Eo decreases due to light load. This indicates that a “switching frequency control method” in which the control is performed so as to increase the switching frequency when it increases is employed. When the switching frequency is controlled to be low under heavy load conditions, the peak level of the current IQ2 is 4.8 A (ampere) as shown in FIG. On the other hand, when the switching is performed so that the switching frequency is increased under light load conditions, the peak level of the current IQ2 is 1.5A. The two phases are different from each other by 90 °, and it can be seen that no power loss occurs.

  When the primary side series resonance current flows, the illustrated voltage V2 is excited in the secondary winding N2 of the converter transformer PIT. The level of the secondary side DC output voltage Eo is obtained as being equal to the positive and negative peak levels of the voltage V2. By obtaining such an AC voltage V2, in the double-side rectifier circuit on the secondary side, current flows in the secondary winding N2 by the action of the rectifier diode Do1, the rectifier diode Do2, and the secondary parallel resonant capacitor C3. Run I2. In FIGS. 2 and 3, the current I2, the current I3, and the current I4 flow in accordance with the sign of the voltage V2. When the load power Po = 0W, the value of the current I4 is zero.

  FIG. 4 shows the switching frequency fs and power conversion efficiency ηAC in the range of the load power Po from 0 W to 200 W with the load power Po as the horizontal axis of the switching power supply circuit of the first embodiment shown in FIG. The value of DC is shown. Here, the alternate long and short dash line indicates each characteristic at the AC input voltage VAC = 100 V of the switching power supply circuit shown as the background art in FIG. 34, and the solid line indicates each characteristic at the AC input voltage VAC = 100 V. Indicates respective characteristics at an AC input voltage VAC = 230V. As shown in the figure, it has the following characteristics. The value of power conversion efficiency ηAC → DC at the AC input voltage VAC = 100 V and the load power Po = 200 W is 91.3%. Further, the value of the switching frequency fs in the range of the AC input voltage VAC = 100 V and the load power Po = 200 W to 0 W is 71.3 kHz to 92.4 kHz. Further, the value of power conversion efficiency ηAC → DC at the AC input voltage VAC = 230 V and the load power Po = 200 W is 92%. Further, the value of the switching frequency fs in the range of the AC input voltage VAC = 230 V and the load power Po = 200 W to 0 W is 140.3 kHz to 160.8 kHz.

  That is, the switching frequency fs has a narrower variable range than that in the switching power supply circuit shown as the background art in FIG. 34, and the value of power conversion efficiency ηAC → DC is the switching power supply shown as the background art in FIG. It is better in the entire range from 0 W to 150 W of the load power Po than in the circuit.

  FIG. 5 shows values of the switching frequency fs and the power conversion efficiency ηAC → DC in the range of the AC input voltage VAC = 80V to the AC input voltage VAC = 260V. Thus, for a wide range of AC input voltage VAC, the range of switching frequency fs is sufficiently narrow, and the value of power conversion efficiency ηAC → DC becomes good, thereby achieving a wide range.

  FIG. 6 and FIGS. 8 to 12 show other examples of the secondary side circuit suitable in combination with the first embodiment. In FIG. 6, a low-value resistor R3 is connected in series with the secondary side parallel resonant capacitor C3. FIG. 7 shows the current I3 in the connection of FIG. 6. By doing so, it can be seen that the high-frequency ringing current as shown in the current I3 in FIG. 2 is suppressed. This high-frequency ringing current is generated by the depletion layer capacitance of the rectifier diode Do2 and the equivalent inductance (ESL) of the smoothing capacitor Co. By connecting the resistor R3 in series with the secondary parallel resonant capacitor C3, So-called Q dump is performed, and this effect is produced. The waveform shown in FIG. 7 is an example in which 3.3Ω (ohms) is adopted as the value of the resistor R3. In this case, 1 W of power loss occurs. However, it is possible to prevent the generation of noise given to the external device by the high frequency generated by the ringing current.

  FIG. 8 shows a case where the secondary side parallel resonant capacitor C3 is AC-parallel with the rectifier diode Do1. From another viewpoint, the secondary parallel resonant capacitor C3 is directly connected in parallel to the secondary winding N2 '. FIG. 9 shows a case where the secondary side parallel resonant capacitor C3 is directly in parallel with the rectifier diode Do1, and from another viewpoint, the secondary winding N2 ′ is connected in an alternating manner, that is, through the smoothing capacitor Co. A secondary side parallel resonant capacitor C3 is connected in parallel. FIG. 10 shows a case where the secondary side parallel resonant capacitor C3 is directly connected to the rectifier diode Do2 in parallel. From another viewpoint, the secondary side parallel resonant capacitor C3 is connected in parallel to the secondary winding N2. Is connected to.

  FIG. 11 shows a case where the secondary side parallel resonant capacitor C3 is directly connected in parallel to the rectifier diode Do2, and the secondary side parallel resonant capacitor C3 ′ is directly connected in parallel to the rectifier diode Do1. From the perspective, the secondary side parallel resonant capacitor C3 ′ is connected in parallel to the secondary winding N2 ′, and the secondary side parallel resonant capacitor C3 is connected in parallel to the secondary winding N2. It is what is done. FIG. 12 shows a case where the secondary parallel resonant capacitor C3 is directly connected to the rectifier diode Do2 in parallel, and the secondary parallel resonant capacitor C3 ′ is connected to the rectifier diode Do1 in parallel in an alternating manner. From the viewpoint, the secondary side parallel resonant capacitor C3 ′ is directly connected in parallel to the secondary winding N2 ′, and the secondary side parallel resonant capacitor C3 is connected in parallel to the secondary winding N2. It is what is done.

  Here, the operation when the connection mode of the secondary side parallel resonant capacitor is parallel to the rectifier diode will be described. The anode (or cathode) of the rectifier diode is connected to one of the secondary windings, and the cathode (or anode) of the rectifier diode is connected to a smoothing capacitor. Further, the value of the capacitance of the smoothing capacitor is much larger than the value of the capacitance of the secondary side parallel resonant capacitor. Therefore, according to such a connection mode, the secondary side parallel resonant capacitor is connected in parallel with the secondary winding having a leakage inductance at least in terms of an alternating current to form a secondary side parallel resonant circuit. Therefore, each of the secondary side circuits shown in FIGS. 8 to 12 described above has the same effect as the case where the secondary side parallel resonant capacitor C3 is parallel to the rectifier diode Do2, as shown in FIG. Similar effects can be obtained.

  That is, in the switching circuit of the first embodiment, a current resonance circuit is formed by the leakage inductance L1 generated in the primary winding N1 and the primary side series resonance capacitor C1, and the primary winding The primary side is configured as a “composite resonance type converter” having a primary side partial voltage resonance circuit which is a parallel resonance circuit formed by the leakage inductance L1 of N1 and the primary side partial voltage resonance capacitor Cp, and the leakage inductance L2 And a secondary-side parallel resonance capacitor C3, the secondary side is a multiple composite resonance type converter configured as a parallel resonance circuit, and the resonance frequency fpo2 of the secondary-side parallel resonance circuit is a primary-side series resonance circuit. Is set to approximately three times the resonance frequency fso1. When such a switching power supply circuit having the configuration of the first embodiment is compared with the switching power supply circuit shown in FIG. 34 as the background art, the following effects are produced.

  Compared with the background art, the first embodiment expands the ZVS region by setting the resonance frequency fpo2 of the secondary side parallel resonance circuit to approximately three times the resonance frequency fso1 of the primary side series resonance circuit. Also, as a background art, the current I1 flowing in the primary side series resonance circuit and the current I2 flowing in the secondary side parallel resonance circuit are reduced by the action of the secondary side parallel resonance capacitor C3 which does not exist in the switching power supply circuit shown in FIG. Thus, in the load range from no load to the maximum load, the value of power conversion efficiency ηAC → DC is improved. In the first embodiment, AC power that is a loss at no load is reduced from 7.5 W to 6.5 W.

  Further, as a background art, the specifications of the converter transformer PIT in the switching power supply circuit shown in FIG. 34 are as follows: EER-40 as a core material, gap is 0.8 mm, primary winding N1 is 20T, and secondary winding N2 is 25T. In the first embodiment, the core material is EER-35, which has a smaller amount of core material than EER-40, the gap is 1.6 mm, the primary winding N1 is 30T, and the secondary material in the first embodiment. Each of the winding N2 and the secondary winding N2 ′ is 45T and can be reduced in size and weight.

  When the voltage value of the AC input voltage VAC is reduced from 100 V to 85 V, in the switching power supply circuit shown in FIG. 34 as the background art, the value of the power conversion efficiency ηAC → DC is reduced by about 2%, but in the first embodiment, By increasing the current I3 flowing through the secondary parallel resonant circuit, the value of power conversion efficiency ηAC → DC can be suppressed to a decrease of about 0.5%.

“Second Embodiment, Third Embodiment”
The switching power supply circuit according to the second and third embodiments will be described with reference to FIGS. These switching power supply circuits include a primary side circuit that inputs a DC input voltage as a rectified and smoothed voltage Ei to generate high-frequency AC power, a primary winding N1 for transmitting AC power to the secondary side circuit, and A converter transformer PIT formed by winding the secondary winding N2 around the core, and a constant voltage for controlling the primary side circuit so that the secondary side DC output voltage Eo obtained from the secondary side circuit becomes a constant voltage. A switching power supply circuit including a control circuit 1 and an oscillation / drive circuit 2 as control means.

  The primary side circuit includes a switching means formed by including a switching element Q1 and a switching element Q2 whose switching frequency is controlled by the constant voltage control means, and a leakage inductance component generated in the primary winding N1 of the converter transformer PIT. The primary side series resonance frequency is controlled by the leakage inductance L1 and the capacitance of the primary side series resonance capacitor C1 connected in series to the primary winding N1, and the operation of the switching means is controlled by current. A primary-side series resonance circuit having a resonance type. Here, the fact that the primary side series resonance frequency is dominated means that the leakage inductance L1 and the primary side series resonance capacitor C1 are the main factors that determine the resonance frequency fso1.

  The secondary side circuit includes secondary side DC output voltage generation means configured as a voltage doubler rectifier circuit. In the second embodiment, the secondary-side DC output voltage generating means is configured as a voltage doubler full-wave rectifier circuit having an intermediate tap that is a connection point between the secondary winding N2 and the secondary winding N2 ′. In the third embodiment, only a secondary winding N2 is provided and configured as a voltage doubler half-wave rectifier circuit. And a secondary parallel resonant circuit formed such that the secondary parallel resonant frequency is dominated by a secondary parallel resonant capacitor connected directly or alternatingly to the secondary winding. It is. Further, when the connection mode of the secondary side parallel resonant capacitor is viewed from another viewpoint, the secondary side parallel resonant capacitor C3 is AC connected to either the voltage doubler full wave rectifier circuit or the voltage doubler half wave rectifier circuit. The leakage inductance L2 that is the leakage inductance component of the secondary winding N2 of the converter transformer PIT or the leakage inductance L2 ′ that is the leakage inductance component of the secondary winding N2 ′ and the secondary side It can be seen that a secondary side parallel resonance circuit formed so that the secondary side parallel resonance frequency is dominated by the capacitance of the parallel resonance capacitor C3 is provided.

  Here, the fact that the secondary side parallel resonance frequency is dominant means that the leakage inductance L2 and / or the leakage inductance L2 ′ and the secondary side parallel resonance capacitor C3 are the main factors that determine the resonance frequency fpo2. is there.

  The secondary parallel resonance frequency (fpo2) is set to be approximately twice the primary side series resonance frequency (fso1). Here, “substantially double” includes an error range of 20%. Hereinafter, the second embodiment and the third embodiment will be described in more detail.

“Second Embodiment”
FIG. 13 is a diagram illustrating a configuration of a switching power supply circuit according to the second embodiment. As in the switching power supply circuit of the first embodiment shown in FIG. 1, the power supply circuit shown in FIG. A configuration having a combination of circuits and a parallel resonance circuit on the secondary side is adopted. In addition, a series resonance circuit is further provided on the secondary side. In addition, the power supply circuit of the second embodiment includes either a 100V system (a voltage system in which the value of the AC input voltage VAC is around 100V) or a 200V system (a voltage system in which the value of the AC input voltage VAC is around 200V). A so-called wide-range configuration that operates in response to commercial AC power input is adopted. Moreover, as corresponding load electric power, for example, it corresponds to the fluctuation range from about load electric power Po = 200W to Po = 0W (no load).

  In the following description of the second embodiment, the same parts as those in the switching power supply circuit of the first embodiment shown in FIG.

  First, regarding the primary side portion, basically the same configuration as that of the first embodiment is adopted, and the operation and effect thereof are also the same. The specific constant value of each part on the primary side in the second embodiment was determined as follows. The core material of the converter transformer PIT was EER-35. In addition, a value of the coupling coefficient k in the range of about 0.7 to 0.8 can be obtained in the range of the gap of 1 mm to 2 mm, but the value of the coupling coefficient k is 0.7 and the gap is 1.6 mm. It was. The number of turns of the primary winding N1 was 25T, the capacitance of the primary side series resonant capacitor C1 was 0.1 μF, and the value of the primary side partial voltage resonant capacitor Cp was 680 pF. The resonance frequency fso1 of the primary side series resonance circuit at this time was 51.7 kHz (kilohertz).

  Next, the secondary side of the switching power supply circuit according to the second embodiment will be described. The secondary side of the converter transformer PIT has a secondary winding N2 'having the same number of turns as the secondary winding N2 and the secondary winding N2, and the secondary winding N2 and the secondary winding N2' The connection point is an intermediate tap. That is, the voltage at one end of the secondary winding N2 with respect to the intermediate tap (a point not on the intermediate tap side) and the end of the secondary winding N2 ′ with the intermediate tap as a reference (a point not on the intermediate tap side) The phase differs from the voltage by 180 °. A secondary side parallel resonant capacitor C3 is connected to both ends of the rectifier diode Do2. Further, the smoothing capacitor Co1 is connected to the intermediate tap, and the other end of the smoothing capacitor Co1 is connected to the secondary ground point. The secondary winding N2 is connected to the cathode of the rectifier diode Do4, and the secondary winding N2 'is connected to the cathode of the rectifier diode Do2. Both the anode of the rectifier diode Do4 and the anode of the rectifier diode Do2 are grounded. The anode of the rectifier diode Do3 is connected to the secondary winding N2, and the anode of the rectifier diode Do1 is connected to the secondary winding N2 '. Both the cathode of the rectifier diode Do3 and the cathode of the rectifier diode Do1 are connected to the smoothing capacitor Co2 so as to obtain the secondary side DC output voltage Eo. The secondary parallel resonant capacitor C3 is connected to the secondary winding N2 ′ via the smoothing capacitor Co1 connected to the intermediate tap, and leakage is a leakage inductance component of the secondary winding N2 ′ of the converter transformer PIT. A secondary side parallel resonance circuit is formed so that the secondary side parallel resonance frequency is dominated by the inductance L2 ′ and the capacitance of the secondary side parallel resonance capacitor C3.

  Further, when the connection mode of the secondary side parallel resonant capacitor C3 is viewed from another viewpoint, the secondary side parallel resonant capacitor C3 is connected in parallel to one rectifier diode Do2 constituting the voltage doubler full wave rectifier circuit. 2 formed so that the secondary parallel resonant frequency is dominated by the leakage inductance L2 ′, which is the leakage inductance component of the secondary winding N2 ′ of the converter transformer PIT, and the capacitance of the secondary parallel resonant capacitor C3. It can also be regarded as having a secondary parallel resonant circuit.

  According to such a connection mode on the secondary side, the current I2 flowing through the secondary winding N2 is changed to the bridge rectifier circuit Do via the current I3 flowing through the secondary parallel resonant capacitor C3 and the secondary series resonant capacitor C2. When the voltage at one end of the secondary winding is set to voltage V2 (see FIG. 13), depending on whether the voltage V2 is positive or negative, as follows: Flowing.

  When the voltage V2 is positive, the currents I2 and I4 are supplied from one end of the secondary winding N2 ′ to the rectifier diode Do1, the smoothing capacitor Co2, the rectifier diode Do4, one end of the secondary winding N2, and the secondary winding. At the same time, the current I3 flows from one end of the secondary winding N2 ′ to the other end of the secondary parallel resonance capacitor C3, the smoothing capacitor Co1, and the secondary winding N2 ′.

  On the other hand, when the voltage V2 is negative, the current I2 and the current I4 are supplied from one end of the secondary winding N2 to one end of the rectifier diode Do3, the smoothing capacitor Co2, the rectifier diode Do2, and the secondary winding N2 ′ At the same time, the current I3 flows from one end of the secondary winding N2 ′ to the other end of the secondary parallel resonant capacitor C3, the smoothing capacitor Co1, and the secondary winding N2 ′.

  Here, the capacitance value of the secondary side parallel resonant capacitor C3 and the capacitance value of the smoothing capacitor Co are Co1, Co2 >> C3 (the capacitance value of the smoothing capacitor Co1 and the static capacitance of the smoothing capacitor Co2). The capacitance value is determined to be much larger than the capacitance value of the secondary parallel resonant capacitor C3). Therefore, the resonance of the secondary side voltage resonance circuit is caused by the leakage inductance L2 generated in the secondary winding N2 or the leakage inductance L2 ′ generated in the secondary winding N2 ′ and the electrostatic capacitance of the secondary parallel resonance capacitor C3. The frequency fpo2 is formed so as to be substantially determined. Here, the value of the resonance frequency fpo2 is set to be approximately twice the resonance frequency fso1, and the value of the resonance frequency fpo2 is set to 135.2 kHz. Here, the number of turns of the secondary winding N2 and the number of turns of the secondary winding N2 'are each 18T, and the value of the secondary side parallel resonant capacitor C3 is 0.033 μF.

  As described above, when the value of the resonance frequency fpo2 is set to be approximately twice or more the resonance frequency fso1, the ZVS operation is performed in a wide range of load power. Further, the current I3 that is the secondary side resonance current does not flow during the period in which the rectifier diode Do1 or the rectifier diode Do2 conducts and the current flows toward the smoothing capacitor Co2, and the load power Po decreases. The value of the current I4 decreases, and at the same time, the value of the current I3 also decreases, and the efficiency at light load is improved. In addition, since the time during which the current I3 flows through the secondary parallel resonant capacitor C3 is shortened at the maximum load, the power conversion efficiency ηAC → DC has a better value than the switching power supply circuit shown in the background art. It becomes.

  In addition, as described above, in addition to the primary side series resonance circuit and the secondary side parallel resonance circuit in which each resonance frequency is set so that the value of the resonance frequency fpo2 is approximately twice or more than the resonance frequency fso1, By providing a primary side partial voltage resonance circuit having a resonance frequency fpo1 and setting the resonance frequency fso1 <resonance frequency fpo1, loss in the switching element is reduced, and further excellent characteristics are obtained.

  Since the configuration and operation of the control circuit 1 and the oscillation / drive circuit 2 are the same as those in the first embodiment, description thereof will be omitted.

  14 and 15 show operation waveforms of each part of the power supply circuit shown in FIG. In these drawings, FIG. 14 shows an operation waveform when the load power Po = 200 W (maximum load power), and FIG. 15 shows an operation waveform when the load power Po = 0 W (no load power). In these figures, experimental results are shown when the AC input voltage VAC = 100 V is constant.

  14 and 15, the voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2. That is, the current IQ2 of the illustrated switching element Q2 flows during the period in which the voltage V1 is 0 level, and the switching element Q2 is turned on during this period. Further, it can be seen that the current IQ2 becomes 0 level during the period in which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei as shown, and the switching element Q2 is turned off during this period. Although not shown, the voltage across one switching element Q1 is obtained as a waveform obtained by shifting the phase of the voltage V1 by 180 degrees. Similarly, a waveform obtained by shifting the phase of the current IQ2 by 180 degrees is obtained as the current of the switching element Q1. That is, the switching element Q1 and the switching element Q2 are alternately turned on / off.

  14 and 15, the current I1 is a resonance current that flows through the primary side series resonance circuit, and is substantially the waveform as shown in the figure, in which the waveforms of the current of the switching element Q1 and the current IQ2 are combined. Is.

  When the waveforms of the voltage V1 and the current IQ2 shown in FIGS. 14 and 15 are compared with each other, the period of these waveforms shown in FIG. 15 is shorter than the period of these waveforms shown in FIG. ing. This indicates that the switching frequency is controlled to be higher as the trend is from heavy load to light load. That is, as stabilization control, when the level of the secondary side DC output voltage Eo decreases due to a heavy load, the switching frequency is lowered, and the level of the secondary side DC output voltage Eo decreases due to a light load. When it rises, it indicates that the control is performed so as to increase the switching frequency. When the switching frequency is controlled to be low under heavy load conditions, the peak level of the current IQ2 is 4.6 A (ampere) as shown in FIG. On the other hand, when the switching is performed so that the switching frequency becomes high under light load conditions, the peak level of the current IQ2 is 1A. When the waveform of the current IQ2 and the waveform of the voltage V1 at this time are compared, It can be seen that the phases are different by 90 ° and no power loss occurs.

  When the primary series resonance current flows, the illustrated voltage V2 is excited in the secondary winding N2 of the converter transformer PIT. The level of the secondary side DC output voltage Eo is obtained as being equal to twice the positive / negative peak level of the voltage V2. As shown in FIGS. 14 and 15, the current I2, the current I3, and the current I4 flow positively and negatively according to whether the voltage V2 is positive or negative. When the load power Po = 0W, the value of the current I4 is zero.

  FIG. 16 shows the switching frequency fs and the power conversion efficiency ηAC in the range of the load power Po from 0 W to 200 W with the load power Po as the horizontal axis of the switching power supply circuit of the second embodiment shown in FIG. The value of DC is shown. Here, the alternate long and short dash line indicates each characteristic at the AC input voltage VAC = 100 V of the switching power supply circuit shown as the background art in FIG. 34, and the solid line indicates each characteristic at the AC input voltage VAC = 100 V. Indicates respective characteristics at an AC input voltage VAC = 230V. As shown in the figure, it has the following characteristics. The value of power conversion efficiency ηAC → DC at the AC input voltage VAC = 100 V and the load power Po = 200 W is 91.9%. In addition, the value of the switching frequency fs in the range of the AC input voltage VAC = 100 V and the load power Po = 200 W to 0 W is 64.9 kHz to 83.2 kHz. Further, the value of power conversion efficiency ηAC → DC at the AC input voltage VAC = 230 V and the load power Po = 200 W is 92.5%. Further, the value of the switching frequency fs in the range from the AC input voltage VAC = 230 V and the load power Po = 200 W to 0 W is 134.5 kHz to 153.8 kHz.

  That is, the switching frequency fs has a narrower variable range than that in the switching power supply circuit shown as the background art in FIG. 34, and the value of power conversion efficiency ηAC → DC is the switching power supply shown as the background art in FIG. It is better in the entire range from 0 W to 200 W of load power Po than in the circuit.

  FIG. 17 shows values of the switching frequency fs and the power conversion efficiency ηAC → DC in the range of the AC input voltage VAC = 80V to the AC input voltage VAC = 260V. Thus, for a wide range of AC input voltage VAC, the range of switching frequency fs is sufficiently narrow, and the value of power conversion efficiency ηAC → DC becomes good, thereby achieving a wide range.

  FIG. 18 and FIGS. 20 to 25 show other examples of the secondary circuit suitable in combination with the second embodiment. FIG. 18 shows a case where a low-value resistor R3 is connected in series to the secondary side parallel resonant capacitor C3. FIG. 19 shows the current I3 in the connection of FIG. 18. By doing so, it can be seen that the high-frequency ringing current shown in the current I3 in FIG. 14 is suppressed. This high-frequency ringing current is generated by the depletion layer capacitance of the rectifier diode and the equivalent inductance (ESL) of the smoothing capacitor. By connecting the resistor R3 in series with the secondary side parallel resonant capacitor C3, so-called A Q dump is made and this effect is produced. The waveform shown in FIG. 19 is an example in which 0.47Ω (ohms) is adopted as the value of the resistor R3. In this case, 1 W of power loss occurs. However, it is possible to prevent the generation of noise given to the external device by the high frequency generated by the ringing current.

  FIG. 20 shows a case where the secondary side parallel resonant capacitor C3 is arranged in parallel with the rectifier diode Do1, and from another viewpoint, the secondary winding N2 ′ is connected in an alternating manner, that is, the smoothing capacitor Co1 and the smoothing capacitor Co2 are connected. The secondary side parallel resonant capacitor C3 is connected in parallel. FIG. 21 shows a case where the secondary side parallel resonant capacitor C3 is placed in parallel with the rectifier diode Do4. From another viewpoint, the secondary winding N2 is connected to the secondary side through the smoothing capacitor Co1 in an alternating manner. A parallel resonant capacitor C3 is connected in parallel.

  FIG. 22 shows a case where the secondary side parallel resonant capacitor C3 is in parallel with the rectifier diode Do3. From another viewpoint, the secondary winding N2 is connected to the secondary winding N2 in an alternating manner, that is, through the smoothing capacitor Co1 and the smoothing capacitor Co2. The secondary parallel resonant capacitor C3 is connected in parallel. FIG. 23 shows a case where the secondary side parallel resonant capacitor C3 ′ is placed in parallel with the rectifier diode Do2 and the secondary side parallel resonant capacitor C3 is placed in parallel with the rectifier diode Do3. From another viewpoint, the secondary winding A secondary parallel resonant capacitor C3 ′ is connected to N2 ′ in an alternating manner, that is, through a smoothing capacitor Co1, and is further connected to the secondary winding N2 in an alternating manner, that is, the smoothing capacitor Co1 and the smoothing capacitor Co2. The secondary side parallel resonant capacitor C3 is connected in parallel via

  FIG. 24 shows a case where the secondary parallel resonant capacitor C3 ′ is in parallel with the rectifier diode Do1 and the secondary parallel resonant capacitor C3 is in parallel with the rectifier diode Do4. From another viewpoint, the secondary winding A secondary side parallel resonant capacitor C3 ′ is connected in parallel to N2 ′ via a smoothing capacitor Co1 and a smoothing capacitor Co2, and further connected to the secondary winding N2 in an AC manner, ie, a smoothing capacitor Co1. The secondary side parallel resonant capacitor C3 is connected in parallel via

  Here, the operation when the connection mode of the secondary side parallel resonant capacitor is parallel to the rectifier diode will be described. The anode (or cathode) of the rectifier diode is connected to one of the secondary windings, and the cathode (or anode) of the rectifier diode is connected to a smoothing capacitor. Further, the value of the capacitance of the smoothing capacitor is much larger than the value of the capacitance of the secondary side parallel resonant capacitor. Therefore, according to such a connection mode, the secondary side parallel resonant capacitor is connected in parallel with the secondary winding having the leakage inductance at least in terms of an alternating current to form a secondary side parallel resonant circuit. Therefore, in each of the above-described FIGS. 20 to 24, as shown in FIG. 13, the same effect is obtained as in the case where the secondary parallel resonant capacitor C3 is paralleled with the rectifier diode Do2, and the same effect is obtained. It is something that can be done.

  That is, in the switching circuit of the second embodiment, a current resonance circuit is formed by the leakage inductance L1 generated in the primary winding N1 and the primary side series resonance capacitor C1, and the primary winding. The primary side is configured as a “composite resonance type converter” having a primary side partial voltage resonance circuit which is a parallel resonance circuit formed by the leakage inductance L1 of N1 and the primary side partial voltage resonance capacitor Cp, and the leakage inductance L2 And a secondary-side parallel resonant capacitor C3, the secondary side is a multiple composite resonance type converter having a secondary resonance circuit, and the value of the resonance frequency fpo2 of the secondary side parallel resonance circuit is the primary The resonance frequency fso1 of the side series resonance circuit is set to be twice or more. When such a switching power supply circuit having the configuration of the second embodiment is compared with the switching power supply circuit shown in FIG. 34 as the background art, the following effects are produced.

  As the background art, the current I1 flowing in the primary side series resonance circuit and the current I2 flowing in the secondary side parallel resonance circuit are reduced by the action of the secondary side parallel resonance capacitor C3 which does not exist in the switching power supply circuit shown in FIG. In the load range from no load to the maximum load, the value of power conversion efficiency ηAC → DC is improved. In the second embodiment, the AC power, which is a loss at no load, is reduced from 7.5 W to 4.2 W.

  Further, as a background art, the specifications of the converter transformer PIT in the switching power supply circuit shown in FIG. 34 are as follows: EER-40 as a core material, gap is 0.8 mm, primary winding N1 is 20T, and secondary winding N2 is 25T. In the first embodiment, the core material is EER-35, the gap is 1.6 mm, the primary winding N1 is 25T, the secondary winding N2 and the secondary winding N2 ′ are 22T. Therefore, the size and weight can be reduced.

  When the voltage value of the AC input voltage VAC is reduced from 100 V to 85 V, in the switching power supply circuit shown in FIG. 34 as the background art, the value of the power conversion efficiency ηAC → DC is reduced by about 2%, but in the second embodiment, By increasing the current I3 flowing through the secondary parallel resonant circuit, the value of power conversion efficiency ηAC → DC can be suppressed to a decrease of about 0.5%.

“Third Embodiment”
FIG. 25 is a diagram illustrating a configuration of a switching power supply circuit according to the third embodiment. As in the switching power supply circuit of the second embodiment shown in FIG. 13, the power supply circuit shown in this figure has partial voltage resonance as compared with a separately excited type current resonance type converter using a half-bridge coupling system as a basic configuration on the primary side. A combination of circuits is provided, and a secondary voltage rectifier circuit and a secondary side parallel resonant circuit are provided on the secondary side. In addition, the power supply circuit of the third embodiment includes either a 100V system (a voltage system in which the value of the AC input voltage VAC is around 100V) or a 200V system (a voltage system in which the value of the AC input voltage VAC is around 200V). It adopts a configuration for so-called wide-range operation that operates in response to commercial AC power supply input, and the corresponding load power corresponds to, for example, a fluctuation range from about load power Po = 200 W to Po = 0 W (no load). This is the same as in the second embodiment. However, the difference between the third embodiment and the second embodiment is that in the second embodiment, a voltage doubler full-wave rectifier circuit is used, whereas in the third embodiment, the secondary side rectifier circuit is replaced with a voltage doubler half. This is different in that it is formed as a wave rectifier circuit.

  A series connection circuit of a smoothing capacitor Co2 and a secondary side parallel resonant capacitor C3 is connected to each terminal of the secondary winding N2. Further, when looking at the connection mode of the secondary side parallel resonant capacitor C3 from another viewpoint, the secondary side parallel resonant capacitor C3 is connected in parallel to one rectifier diode Do2 constituting the voltage doubler half-wave rectifier circuit. The secondary side parallel resonant circuit formed in parallel with the leakage inductance L2 ′ that is the leakage inductance component of the secondary winding N2 ′ of the converter transformer PIT via the smoothing capacitor Co2 may be considered. It can be done.

  In the following description of the third embodiment, the same parts as those in the switching power supply circuit according to the first embodiment shown in FIG. 1 or the switching power supply circuit according to the second embodiment shown in FIG. May be omitted.

  First, regarding the primary side portion, basically the same configuration as in the first embodiment and the second embodiment is adopted, and the operation and effect thereof are also the same. The specific constant value of each part on the primary side in the third embodiment was determined as follows. The core material of the converter transformer PIT was EER-35. A coupling coefficient k value in the range of about 0.7 to 0.8 can be obtained in the gap range of 1 mm to 2 mm, but the coupling coefficient k value is 0.7 and the gap is 1.6 mm. . The number of turns of the primary winding N1 was 25T, the capacity of the primary side series resonant capacitor C1 was 0.082 μF, and the value of the primary side partial voltage resonant capacitor Cp was 680 pF. At this time, the resonance frequency fso1 of the primary side series resonance circuit was 57.0 kHz (kilohertz).

  Next, the secondary side of the switching power supply circuit according to the third embodiment will be described. The anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 are connected to one end of the secondary winding N2 of the converter transformer PIT. A secondary parallel resonant capacitor C3 is connected in parallel with the rectifier diode Do2. The anode of the rectifier diode Do2 is grounded to the secondary ground point, and the cathode of the rectifier diode Do1 is connected to the positive electrode side of the smoothing capacitor Co1. The other end of the secondary winding N2 is connected to the negative electrode side of the smoothing capacitor Co1 and the positive electrode side of the smoothing capacitor Co2, and the negative electrode side of the smoothing capacitor Co2 is connected to the secondary ground point. By such a connection mode, a voltage twice as large as the peak value of the voltage generated in the secondary winding N2 is obtained as the secondary side DC output voltage Eo.

  When the voltage V2 (see FIG. 25) is positive according to such a connection state on the secondary side, the current I2 flowing through the secondary winding N2 is represented by the sum of the current I4 and the current I3. The current I4 flows from one end of the secondary winding N2 to the rectifier diode Do1, the smoothing capacitor Co1, and the other end of the secondary winding N2. At the same time, the current I3 flows from one end of the secondary winding N2 to the secondary side. It flows to the other end of the parallel resonant capacitor C3, the smoothing capacitor Co2, and the secondary winding N2.

  Even when the voltage V2 is negative, the current I2 flowing through the secondary winding N2 is represented by the sum of the current I4 and the current I3, and the current I4 is supplied from the other end of the secondary winding N2 to the smoothing capacitor Co2, The rectifier diode Do2 flows to one end of the secondary winding N2, and at the same time, the current I3 flows from the other end of the secondary winding N2 to the smoothing capacitor Co2, the secondary parallel resonant capacitor C3, and one end of the secondary winding N2. And flow.

  Here, the capacitance value of the secondary side parallel resonance capacitor C3, the capacitance value of the smoothing capacitor Co1, and the capacitance value of the smoothing capacitor Co2 are Co1, Co2 >> C3 (smoothing capacitor Co1 and smoothing capacitor Co1). The capacitance value of the capacitor Co2 is determined to be much larger than the capacitance value of the secondary side parallel resonant capacitor C3). Therefore, in the secondary side parallel resonance circuit, the resonance frequency fpo2 of the secondary side voltage resonance circuit is substantially determined by the leakage inductance L2 generated in the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C3. Formed. Here, the value of the resonance frequency fpo2 was set to 140.5 kHz. Here, the number of turns of the secondary winding N2 is 18T, and the value of the secondary side parallel resonant capacitor C3 is 0.027 μF.

  As described above, the value of the resonance frequency fpo2 is 140.5 kHz, and the resonance frequency fso1 is at least twice the value of 57.0 kHz, so that the ZVS region is subjected to load fluctuations and a wide range of fluctuations in the AC input voltage VAC. It was wide. Further, the current I3 that is the secondary parallel resonance current flowing through the secondary winding N2 and the secondary parallel resonance capacitor C3 does not flow during the period when the rectifier diode Do1 or the rectifier diode Do2 is conducted and the current I4 flows. As the load power Po decreases, the value of the current I4 decreases, and at the same time, the value of the current I3 also decreases. And the value of power conversion efficiency (eta) AC-> DC becomes favorable at the time of light load. At the maximum load, since the period during which the current I3 flows through the secondary winding N2 and the secondary parallel resonant capacitor C3 is shortened, the power conversion efficiency ηAC → DC is higher than that in the switching power supply circuit shown in the background art. The value of becomes better.

  As described above, a primary-side partial voltage resonance circuit having the resonance frequency fpo1 is provided, and the loss in the switching element is reduced by setting the resonance frequency fso1 <resonance frequency fpo1. Yes.

  Since the configuration and operation of the control circuit 1 and the oscillation / drive circuit 2 are the same as those in the first and second embodiments, description thereof will be omitted.

  26 and 27 show operation waveforms of the respective parts of the power supply circuit shown in FIG. In these drawings, FIG. 25 shows an operation waveform when the load power Po = 200 W (maximum load power), and FIG. 26 shows an operation waveform when the load power Po = 0 W (no load power). In these figures, experimental results are shown when the AC input voltage VAC = 100 V is constant.

  In FIG. 26 and FIG. 27, the voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2. That is, the current IQ2 of the illustrated switching element Q2 flows during the period in which the voltage V1 is 0 level, and the switching element Q2 is turned on during this period. Further, it can be seen that the current IQ2 becomes 0 level during the period in which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei as shown, and the switching element Q2 is turned off during this period. Although not shown, the voltage across one switching element Q1 is obtained as a waveform obtained by shifting the phase of the voltage V1 by 180 degrees. Similarly, a waveform obtained by shifting the phase of the current IQ2 by 180 degrees is obtained as the current of the switching element Q1. That is, the switching element Q1 and the switching element Q2 are alternately turned on / off.

  In FIG. 26 and FIG. 27, the current I1 is a resonance current flowing through the primary side series resonance circuit, and has a waveform as shown in the figure, in which the waveforms of the current of the switching element Q1 and the current IQ2 are combined. Is.

  When the waveforms of voltage V1 and current IQ2 shown in FIG. 26 and FIG. 27 are compared with each other, the period of these waveforms shown in FIG. 27 is shorter than the period of these waveforms shown in FIG. ing. This indicates that the switching frequency is controlled to be higher as the trend is from heavy load to light load. That is, as stabilization control, when the level of the secondary side DC output voltage Eo decreases due to a heavy load, the switching frequency is lowered, and the level of the secondary side DC output voltage Eo decreases due to a light load. When it rises, it indicates that the control is performed so as to increase the switching frequency. When the switching frequency is controlled to be low under heavy load conditions, the peak level of the current IQ2 is 4.6 A (ampere) as shown in FIG. Further, in the case where the switching is performed so that the switching frequency becomes high under light load conditions, when the waveform of the current IQ2 and the waveform of the voltage V1 are compared, the phases of the two are different by 90 °, resulting in power loss. I understand that there is no.

  When the primary series resonance current flows, the illustrated voltage V2 is excited in the secondary winding N2 of the converter transformer PIT. Assuming that the voltage V2 is equal to the positive / negative peak level, a double voltage is obtained as the level of the secondary side DC output voltage Eo. Such an alternating voltage V2 is obtained. As shown in FIGS. 26 and 27, the current I2, the current I3, and the current I4 flow in the positive and negative directions according to the positive and negative of the voltage V2. When the load power Po = 0W, the value of the current I4 is zero.

  FIG. 28 shows the switching frequency fs and the power conversion efficiency ηAC in the range of the load power Po from 0 W to 200 W with the load power Po as the horizontal axis of the switching power supply circuit of the third embodiment shown in FIG. The value of DC is shown. Here, the alternate long and short dash line indicates each characteristic of the switching power supply circuit shown in FIG. 34 at the AC input voltage VAC = 100V, and the solid line indicates each waveform at the AC input voltage VAC = 100V. Indicates respective waveforms at an AC input voltage VAC = 230V. As shown in the figure, it has the following characteristics. The value of power conversion efficiency ηAC → DC at AC input voltage VAC = 100 V and load power Po = 200 W is 91.2%. Further, the value of the switching frequency fs in the range of the AC input voltage VAC = 100 V and the load power Po = 200 W to 0 W is 71.3 kHz to 92.4 kHz. The value of power conversion efficiency ηAC → DC at AC input voltage VAC = 230 V and load power Po = 200 W is 91.7%. Further, the value of the switching frequency fs in the range from the AC input voltage VAC = 230 V and the load power Po = 200 W to 0 W is 144.5 kHz to 166.7 kHz.

  That is, the switching frequency fs has a narrower variable range than that in the switching power supply circuit shown as the background art in FIG. 34, and the value of power conversion efficiency ηAC → DC is the switching power supply shown as the background art in FIG. It is better in the entire range from 0 W to 200 W of load power Po than in the circuit.

  FIG. 29 shows values of the switching frequency fs and the power conversion efficiency ηAC → DC in the range of the AC input voltage VAC = 80V to the AC input voltage VAC = 260V. Thus, for a wide range of AC input voltage VAC, the range of switching frequency fs is sufficiently narrow, and the value of power conversion efficiency ηAC → DC becomes good, thereby achieving a wide range.

  FIGS. 30, 32 and 33 show other examples of the secondary circuit suitable in combination with the third embodiment. In FIG. 30, a low-value resistor R3 is connected in series to the secondary side parallel resonant capacitor C3. FIG. 31 shows the current I3 in the connection of FIG. 30, and it can be seen that a high-frequency ringing current as shown in the current I3 in FIG. 26 is suppressed by doing so. This high-frequency ringing current is generated by the depletion layer capacitance of the rectifier diode and the equivalent inductance (ESL) of the smoothing capacitor. By connecting the resistor R3 in series with the secondary side parallel resonant capacitor C3, so-called A Q dump is made and this effect is produced. The waveform shown in FIG. 31 is an example in which 0.47Ω (ohms) is adopted as the value of the resistor R3. In this case, 1 W of power loss occurs. However, it is possible to prevent the generation of noise given to the external device by the high frequency generated by the ringing current.

  FIG. 32 shows a case where the secondary parallel resonant capacitor C3 is connected in parallel to the rectifier diode Do2 and the secondary parallel resonant capacitor C3 ′ is parallel to the rectifier diode Do1. A secondary side parallel resonant capacitor C3 ′ is connected in parallel to the secondary winding N2 via a smoothing capacitor Co1, and further connected to the secondary winding N2 in an alternating manner, ie via a smoothing capacitor Co2. The secondary parallel resonant capacitor C3 is connected in parallel. FIG. 33 shows a case where the secondary side parallel resonant capacitor C3 is placed in parallel with the rectifier diode Do1, and from another viewpoint, the secondary winding N2 is connected to the secondary side through the smoothing capacitor Co1 in an alternating manner. A parallel resonant capacitor C3 is connected in parallel.

  Here, the operation when the connection mode of the secondary side parallel resonant capacitor is parallel to the rectifier diode will be described. The anode (or cathode) of the rectifier diode is connected to one of the secondary windings, and the cathode (or anode) of the rectifier diode is connected to a smoothing capacitor. Further, the value of the capacitance of the smoothing capacitor is much larger than the value of the capacitance of the secondary side parallel resonant capacitor. Therefore, according to such a connection mode, the secondary side parallel resonant capacitor is connected in parallel with the secondary winding having the leakage inductance at least in terms of an alternating current to form a secondary side parallel resonant circuit. Therefore, in each of the above-described FIG. 30, FIG. 32 and FIG. 33, as shown in FIG. An effect can be obtained.

  That is, in the switching circuit of the third embodiment, a current resonance circuit is formed by the leakage inductance L1 generated in the primary winding N1 and the primary side series resonance capacitor C1, and the primary winding. The primary side is configured as a “composite resonance type converter” having a primary side partial voltage resonance circuit which is a parallel resonance circuit formed by the leakage inductance L1 of N1 and the primary side partial voltage resonance capacitor Cp, and the leakage inductance L2 And a secondary composite resonance type converter having a secondary resonance circuit having a parallel resonance circuit formed by a secondary parallel resonance capacitor C3, and the value of the resonance frequency fpo2 of the secondary parallel resonance circuit is the primary It is configured by setting as twice the resonance frequency fso1 of the side series resonance circuit. When the switching power supply circuit having the configuration of the third embodiment as described above is compared with the switching power supply circuit shown in FIG. 34 as the background art, the following effects are produced.

  As the background art, the current I1 flowing in the primary side series resonance circuit and the current I2 flowing in the secondary side parallel resonance circuit are reduced by the action of the secondary side parallel resonance capacitor C3 which does not exist in the switching power supply circuit shown in FIG. In the load range from no load to the maximum load, the value of power conversion efficiency ηAC → DC is improved. In 3rd Embodiment, the alternating current power which is a loss at the time of no load is reducing from 7.5W to 4.8W.

  Further, as a background art, the specifications of the converter transformer PIT in the switching power supply circuit shown in FIG. 34 are: EER-40 as a core material, gap is 0.8 mm, primary winding N1 is 20T, and secondary winding N2 is 25T. In the third embodiment, the core material is EER-35, the gap is 1.6 mm, the primary winding N1 is 20T, and the secondary winding N2 is 25T, so that the size and weight can be reduced. .

  When the voltage value of the AC input voltage VAC is reduced from 100 V to 85 V, in the switching power supply circuit shown in FIG. 34 as the background art, the value of the power conversion efficiency ηAC → DC is reduced by about 2%, but in the first embodiment, By increasing the current I3 flowing through the secondary parallel resonant circuit, the value of power conversion efficiency ηAC → DC can be suppressed to a decrease of about 0.5%.

It is a circuit diagram which shows the structural example of the power supply circuit of 1st Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 1st Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 1st Embodiment. It is a figure which shows the switching frequency with respect to load electric power, and power conversion efficiency of the switching power supply circuit of 1st Embodiment. It is a figure which shows the switching frequency with respect to the alternating current input voltage of the switching power supply circuit of 1st Embodiment, and power conversion efficiency. It is a circuit diagram which shows the other secondary side circuit of 1st Embodiment. It is a figure which shows operation | movement of the secondary side circuit shown in FIG. It is a circuit diagram which shows the other secondary side circuit of 1st Embodiment. It is a circuit diagram which shows the other secondary side circuit of 1st Embodiment. It is a circuit diagram which shows the other secondary side circuit of 1st Embodiment. It is a circuit diagram which shows the other secondary side circuit of 1st Embodiment. It is a circuit diagram which shows the other secondary side circuit of 1st Embodiment. It is a figure which shows the structure of the switching power supply circuit of 2nd Embodiment. It is a figure which shows the operation | movement waveform of each part of the power supply circuit of 2nd Embodiment. It is a figure which shows the operation | movement waveform of each part of the power supply circuit of 2nd Embodiment. It is a figure which shows the switching frequency with respect to load electric power, and power conversion efficiency of the switching power supply circuit of 2nd Embodiment. It is a figure which shows the switching frequency with respect to the alternating current input voltage of the switching power supply circuit of 2nd Embodiment, and power conversion efficiency. It is a circuit diagram which shows the other secondary side circuit of 2nd Embodiment. It is a figure which shows operation | movement of the secondary side circuit shown in FIG. It is a circuit diagram which shows the other secondary side circuit of 2nd Embodiment. It is a circuit diagram which shows the other secondary side circuit of 2nd Embodiment. It is a circuit diagram which shows the other secondary side circuit of 2nd Embodiment. It is a circuit diagram which shows the other secondary side circuit of 2nd Embodiment. It is a circuit diagram which shows the other secondary side circuit of 2nd Embodiment. It is a figure which shows the structure of the switching power supply circuit of 3rd Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 3rd Embodiment. It is a figure which shows the operation waveform of each part of the power supply circuit of 3rd Embodiment. It is a figure which shows the switching frequency with respect to the load electric power of the switching power supply circuit of 3rd Embodiment, and power conversion efficiency. It is a figure which shows the switching frequency with respect to the alternating current input voltage of the switching power supply circuit of 3rd Embodiment, and power conversion efficiency. It is a circuit diagram which shows the other secondary side circuit of 3rd Embodiment. FIG. 31 is a diagram showing an operation of the secondary side circuit shown in FIG. 30. It is a circuit diagram which shows the other secondary side circuit of 3rd Embodiment. It is a circuit diagram which shows the other secondary side circuit of 3rd Embodiment. It is a figure which shows the structure of the switching power supply circuit of background art. It is a figure which shows the structure of the converter transformer of background art. It is a figure which shows the operation | movement waveform of each part of the switching power supply circuit of background art. It is a figure which shows the operation | movement waveform of each part of the switching power supply circuit of background art. It is a figure which shows the switching frequency with respect to the load electric power of the switching power supply circuit of background art, and power conversion efficiency. It is a figure which shows the structure of the other switching power supply circuit of background art.

Explanation of symbols

1 control circuit, 2 oscillation / drive circuit, AC commercial AC power supply, C1 primary side series resonant capacitor, C3, C3 ′ secondary side parallel resonant capacitor, Ci smoothing capacitor, CL filter capacitor, CMC common mode choke coil, Co, Co1, Co2 smoothing capacitor, Cp primary side partial voltage resonance capacitor, CR1, CR2 E type core, DD1, DD2 damper diode, Di, Do bridge rectifier circuit, Do1, Do2, Do3, Do4 rectifier diode, Ei rectified smoothing voltage, Eo secondary side DC output voltage, L1, L2 leakage inductance, N1 primary winding, N2, N2 ′ secondary winding, PIT converter transformer, Q1, Q2 switching element, R3 resistance, VAC AC input voltage

Claims (5)

A primary circuit for generating AC power by inputting a DC input voltage;
A converter transformer formed by winding a primary winding and a secondary winding for transmitting the AC power to a secondary side circuit around a core;
A constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied from the secondary side circuit to the load becomes a constant voltage;
The primary circuit is
Switching means formed with a switching element whose switching frequency is controlled by the constant voltage control means;
It is formed such that the primary side series resonance frequency is dominated by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding. A primary-side series resonant circuit in which the operation of the switching means is a current resonant type,
The secondary circuit is
A secondary side that forms a full-wave rectifier circuit that includes a rectifier diode that rectifies AC power obtained between an intermediate tap formed in the secondary winding and both ends of the secondary winding. DC output voltage generating means;
The secondary side parallel resonant frequency is dominated by the intermediate side tap formed in the secondary winding and the secondary side parallel resonant capacitor connected directly or AC to one end of the secondary winding. A secondary side parallel resonant circuit formed as described above,
2. The switching power supply circuit according to claim 1, wherein the secondary side parallel resonance frequency is set to a frequency approximately three times the primary side series resonance frequency. A primary circuit for generating AC power by inputting a DC input voltage;
A converter transformer formed by winding a primary winding and a secondary winding for transmitting the AC power to a secondary side circuit around a core;
A constant voltage control means for controlling the primary side circuit so that the secondary side DC output voltage supplied from the secondary side circuit to the load becomes a constant voltage;
The primary circuit is
Switching means formed with a switching element whose switching frequency is controlled by the constant voltage control means;
It is formed such that the primary side series resonance frequency is dominated by the leakage inductance component generated in the primary winding of the converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding. A primary-side series resonant circuit in which the operation of the switching means is a current resonant type,
The secondary circuit is
Secondary-side DC output voltage generating means having a voltage doubler rectifier circuit for obtaining a voltage twice that of the AC voltage obtained in the secondary winding;
A secondary side parallel resonant circuit formed such that a secondary side parallel resonant frequency is dominated by a secondary side parallel resonant capacitor connected directly or alternatingly to the secondary winding,
2. The switching power supply circuit according to claim 1, wherein the secondary side parallel resonance frequency is set to a frequency that is approximately twice the primary side series resonance frequency.   The switching power supply circuit according to claim 1, wherein a resistance is added in series to the secondary side parallel resonant capacitor. The voltage doubler rectifier circuit is formed as a voltage doubler full wave rectifier circuit in which an intermediate tap is provided in the secondary winding,
The switching power supply circuit according to claim 2, wherein the secondary parallel resonant capacitor is connected to the secondary winding through a smoothing capacitor connected to the intermediate tap. The voltage doubler rectifier circuit is formed as a voltage doubler half wave rectifier circuit,
The switching power supply circuit according to claim 2, wherein a series connection circuit of a smoothing capacitor and the secondary side parallel resonant capacitor is connected to the secondary winding in parallel.

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