JP2006074897A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce a control range necessary for switching frequency control, to obtain configuration compatible with a wide range, and improve power conversion efficiency with respect to the condition of a load that is heavier than a certain level, in a power supply circuit that performs constant voltage control by the switching frequency control, and comprises a power factor improving function. <P>SOLUTION: A coupling type resonance circuit constituted by the electromagnetic coupling of an insulated converter transformer PIT is formed by comprising a primary-side series resonance circuit that forms a current resonance type converter, and a secondary-side series resonance circuit that is formed of at least a secondary winding N2 and a secondary-side series resonance capacitor C2. Then, in order to obtain a single peak characteristic of the coupling type resonance circuit, a general coupling coefficient kt of the insulated converter transformer PIT is set as kt≤0.65. For this arrangement, a circuit is configured such that an inductor is inserted in series into at least either of a primary winding N1 and the secondary winding N2, while reducing a gap G of an internal magnetic leg into a range of about 1.6mm with respect to the insulated converter transformer PIT itself. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

JP 2003-235259 A

The present applicant has previously proposed various power supply circuits including a resonance type converter on the primary side.
FIG. 20 is a circuit diagram showing an example of a switching power supply circuit including a resonant converter configured based on the invention previously filed by the present applicant.
The switching converter of the power supply circuit shown in FIG. 20 has a configuration in which a partial voltage resonance circuit that performs a voltage resonance operation only at the time of turn-off during switching is combined with a separately excited current resonance converter using a half-bridge coupling method. take.

First, in the power supply circuit shown in FIG. 20, a common mode noise filter including two sets of filter capacitors CL and CL and one set of common mode choke coil CMC is connected to the commercial AC power supply AC.
As a rectifying / smoothing circuit for generating a DC input voltage from the commercial AC power supply AC, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci is provided for the subsequent stage of the common mode noise filter.
The rectified output of the bridge rectifier circuit Di is charged to the smoothing capacitor Ci, whereby a rectified smoothing voltage Ei (DC input voltage) corresponding to an equal level of the AC input voltage VAC is applied to both ends of the smoothing capacitor Ci. Will be obtained.

  As shown in the figure, the current resonance type converter for switching by inputting the DC input voltage includes a switching circuit system in which two switching elements Q1 and Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 formed of body diodes are connected in parallel with each other between the drains and sources of the switching elements Q1 and Q2, respectively, in the direction shown in the drawing.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. With this partial voltage resonance circuit, a partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off can be obtained.

  In this power supply circuit, in order to switch the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and applies a drive signal (gate voltage) having a required frequency to the gates of the switching elements Q1 and Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side.
In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the source of the switching element Q1 and the drain of the switching element Q2 via the primary side series resonance capacitor C1. A switching output is obtained.
The other end of the primary winding N1 is connected to the primary side ground as shown in the figure.

  In this case, the series resonant capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonant capacitor C1 and the leakage inductance (leakage) of the primary winding N1 (series resonant winding) of the insulating converter transformer PIT. The primary side series resonance circuit for making the operation of the switching converter into a current resonance type is formed by the inductance L1.

According to the description so far, the primary side switching converter shown in this figure includes the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the partial voltage resonance circuit (Cp // L1) described above. Thus, partial voltage resonance operation due to is obtained.
That is, the power supply circuit shown in this figure adopts a form in which another resonance circuit is combined with a resonance circuit for making the primary side switching converter into a resonance type. Here, such a switching converter is referred to as a composite resonance type converter.

Although the description by illustration here is abbreviate | omitted, as a structure of the above-mentioned insulation converter transformer PIT, the EE type | mold core which combined the E type | mold core by a ferrite material is provided, for example. Then, after the winding part is divided between the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the inner magnetic leg of the EE core.
Further, a gap of 1.5 mm or less is formed with respect to the inner magnetic leg of the EE type core of the insulating converter transformer PIT. As a result, a coupling coefficient of 0.75 or more is obtained between the primary winding N1 and the secondary winding N2.

The secondary winding N2 of the insulating converter transformer PIT is provided with a full-wave rectifier circuit formed by a bridge rectifier circuit including rectifier diodes Do1 to Do4 and a smoothing capacitor Co.
As a result, the secondary side DC output voltage Eo, which is a DC voltage corresponding to the same level of the alternating voltage induced in the secondary winding N2, is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied as a main DC power source to a main load (not shown) and is also branched and input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 outputs a control signal as a voltage or current whose level is variable in accordance with the level of the secondary side DC output voltage Eo to the oscillation / drive circuit 2.
In the oscillation / drive circuit 2, the oscillation signal frequency generated by the oscillation circuit in the oscillation / drive circuit 2 is varied based on the control signal input from the control circuit 1. The frequency of the switching drive signal to be applied is changed. As a result, the switching frequency is varied. In this way, the switching frequency of the switching elements Q1, Q2 is variably controlled according to the level of the secondary side DC output voltage Eo, so that the resonance impedance of the primary side series resonance circuit changes and the primary side series resonance circuit is changed. The energy transmitted from the primary winding N1 to be formed to the secondary side is also varied, and the level of the secondary side DC output voltage Eo is also variably controlled. Thereby, constant voltage control of the secondary side DC output voltage Eo is achieved.
In the following, the constant voltage control method that achieves stabilization by variably controlling the switching frequency will be referred to as a “switching frequency control method”.

  The waveform diagram of FIG. 21 shows the operation of the main part of the power supply circuit shown in FIG. In this figure, the operation when the load power Po = 150 W is shown on the left side, and the operation when the load power Po = 25 W is shown for the same part on the right side. The input voltage condition is constant at AC input voltage VAC = 100V.

First, in FIG. 11, a rectangular waveform voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2.
The period during which the voltage V1 is 0 level is the ON period in which the switching element Q2 is conductive. In this ON period, the switching current IQ2 having the waveform shown in FIG. Flowing. The period during which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei is a period during which the switching element Q2 is turned off, and the switching current IQ2 is at 0 level as shown in the figure.
Although not shown, the voltage across the other switching element Q1 and the switching current flowing through the switching circuit (Q1, DD1) are obtained as a waveform obtained by shifting the voltage V1 and the switching current IQ2 by 180 °. . That is, as described above, the switching element Q1 and the switching element Q2 perform the switching operation at the timing of turning on / off alternately.
Further, as the primary side series resonance current Io flowing through the primary side series resonance circuit (C1-N1 (L1)), the switching currents flowing through these switching circuits (Q1, DD1) (Q2, DD2) are synthesized. , And flows according to the waveform shown.

  Further, for example, as can be seen by comparing the waveform of the voltage V1 shown in this figure between when the load power Po = 150 W and when the load power Po = 25 W, as the switching frequency, the secondary side DC output voltage Eo is It can be seen that the primary side switching frequency is controlled to be lower in the heavy load condition (Po = 150 W) than in the light load condition (Po = 25 W). That is, as the level of the secondary side DC output voltage Eo decreases due to a heavy load, the switching frequency is lowered and the level of the secondary side DC output voltage Eo increases due to a light load. In response to this, the switching frequency is increased. This indicates that a constant voltage control operation by upper side control is performed as a switching frequency control method.

  Further, by obtaining the above-described primary side operation, an alternating voltage V2 having a waveform shown in the figure is induced in the secondary winding N2 of the insulating converter transformer PIT. In the half-cycle period in which the alternating voltage V2 is positive, the secondary side rectifier diodes [Do1, Do4] are turned on, and the rectified current ID1 flows with the waveform and timing shown in the figure. Further, in the other half cycle period in which the alternating voltage V2 is negative, the secondary side rectifier diodes [Do2, Do3] are turned on, and the rectified current ID2 flows with the waveform and timing shown in the figure. The secondary winding current I2 flowing through the secondary winding N2 is a composite of the rectified currents ID1 and ID2 as shown in the figure.

FIG. 22 shows the AC → DC power conversion efficiency and switching frequency characteristics with respect to load fluctuations under the input voltage condition of the AC input voltage VAC = 100 V for the power supply circuit shown in FIG.
First, the switching frequency fs has a characteristic that decreases as the tendency of a heavy load is increased in accordance with the constant voltage control operation. However, it is not a change characteristic that is linear with respect to the load fluctuation, and the switching frequency fs tends to increase sharply, for example, in the range of the load power Po = 25 W to Po = 0 W or less.
Further, the AC → DC power conversion efficiency (ηAC → DC) tends to increase as the load power Po increases, and ηAC → DC = 91.0% when the load power Po = 150 W. The degree is obtained.

By the way, in the case of adopting a configuration as a resonance type converter that stabilizes the secondary side DC output voltage by the switching frequency control method as in the power supply circuit shown in FIG. 20, the switching frequency for stabilization is variable. The control range tends to be relatively wide.
This will be described with reference to FIG. FIG. 23 shows the constant voltage control characteristics of the power supply circuit shown in FIG. 20 by the relationship between the switching frequency fs and the level of the secondary side DC output voltage Eo.
In the description of this figure, it is assumed that the power supply circuit of FIG. 20 employs so-called upper side control as a switching frequency control method. Here, the upper side control means that the switching frequency is variably controlled in a frequency range higher than the resonance frequency fo of the primary side series resonance circuit, and the change of the resonance impedance generated thereby makes it possible to control the secondary side DC output voltage Eo. Control that controls the level.

As a general matter, the series resonant circuit has the smallest resonance impedance at the resonance frequency fo. As a result, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs in the upper side control, the level of the secondary side DC output voltage Eo increases as the switching frequency fs approaches the resonance frequency fo1. As it moves away from fo1, it will decrease.
Therefore, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs under the condition that the load power Po is constant, the switching frequency fs is the same as the resonance frequency fo1 of the primary side series resonance circuit as shown in FIG. It sometimes becomes a peak, and shows a quadratic curve change that decreases as the frequency moves away from the resonance frequency fo1.

  Further, a characteristic is obtained that the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs is shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  Under such characteristics, when the secondary side DC output voltage Eo is to be stabilized by Eo = tg by the upper side control, it is necessary in the power supply circuit shown in FIG. The variable range (necessary control range) of the switching frequency is a range indicated as Δfs.

In practice, the power supply circuit shown in FIG. 20 has an input fluctuation range of AC input voltage VAC = 85V to 120V as an AC 100V system, maximum load power Pomax = 150 W of secondary side DC output voltage Eo which is a main DC power source, minimum In response to the load condition of load power Pomin = 0 W (no load), constant voltage control is performed by the switching frequency control method so as to stabilize at the secondary side DC output voltage Eo = 135V.
In this case, the variable range of the switching frequency fs that the power supply circuit shown in FIG. 20 varies for constant voltage control is fs = 80 kHz to 200 kHz or more, and Δfs is also a correspondingly wide range of 120 kHz or more.

As a power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V is possible so as to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V system region such as Europe. In addition, what is known as a wide range is known.
Therefore, it is considered that the power supply circuit shown in FIG.
In the wide range correspondence, as described above, for example, it corresponds to the AC input voltage range of AC85V to 288V. Therefore, for example, the level fluctuation range of the secondary side DC output voltage Eo is larger than that corresponding to a single range of only the AC 100 V system or only the AC 200 V system. In order to perform constant voltage control on the secondary side DC output voltage Eo whose level fluctuation range is expanded corresponding to such an AC input voltage range, a wider switching frequency control range is required. For example, in the case of the circuit shown in FIG. 20, the control range of the switching frequency fs needs to be expanded to about 80 kHz to 500 kHz.

  However, for an IC (oscillation / drive circuit 2) for driving a current switching element, the upper limit of the drive frequency that can be handled is about 200 kHz. Even if the switching drive IC capable of driving at a high frequency as described above is configured and mounted, the power conversion efficiency is significantly reduced when the switching element is driven at such a high frequency. Therefore, it is not practical as an actual power supply circuit. Incidentally, for example, the upper limit of the AC input voltage VAC level that can be stabilized by the power supply circuit shown in FIG. 20 is about 100V.

  For this reason, it is known that the switching power supply circuit that is stabilized by the switching frequency control system, for example, adopts the following configuration in order to actually support a wide range.

For one, a rectifier circuit system for generating a DC input voltage (Ei) by inputting a commercial AC power source, a voltage doubler rectifier circuit, a full-wave rectifier circuit, A function is given so that switching is performed with.
In this case, a commercial AC power supply level is detected, and a voltage doubler rectifier circuit or a full-wave rectifier circuit is formed according to the detected level, and a rectifier circuit is switched by a switch using an electromagnetic relay. The circuit is configured to switch the circuit connection in the system.

  However, such a rectifying circuit system switching configuration requires a required number of electromagnetic relays as described above. Further, it is necessary to provide at least two sets of smoothing capacitors in order to form a voltage doubler rectifier circuit. For this reason, the number of parts is increased and the cost is increased, and the mounting area of the power supply circuit board is increased and the size is increased. In particular, since these smoothing capacitors and electromagnetic relays are large among the components forming the power supply circuit, the substrate size becomes considerably large.

  Further, when the full-wave rectification operation and the voltage doubler rectification operation are switched, an instantaneous power failure occurs or the AC input voltage drops below the rating when an AC 200V commercial AC power supply is input. For example, when the level is lower than that corresponding to the AC200 system, it is assumed that a malfunction occurs in which the AC100V system is detected and switched to the voltage doubler rectifier circuit. When such a malfunction occurs, voltage doubler rectification is performed on the AC input voltage of AC200V system level, so that there is a possibility that the switching elements Q1, Q2, etc., for example, will be broken and destroyed.

  Therefore, as an actual circuit, in order to prevent the above-described malfunction, not only the DC input voltage of the main switching converter but also the DC input voltage of the converter circuit on the standby power supply side is detected. It is made to take. As a result, the addition of components for detecting the converter circuit on the standby power supply side further promotes the above-described cost increase and increase in circuit board size.

  In addition, the detection of the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunctions means that a wide range compatible power supply circuit having a circuit for switching the rectifying operation includes a standby power supply in addition to the main power supply. This means that it cannot actually be used unless it is an electronic device. That is, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and the use range is narrowed accordingly.

Also, as a configuration for wide range compatibility, the primary side current resonance type converter may be switched between half-bridge coupling and full-bridge coupling in accordance with AC 100 V / AC 200 V commercial AC power input. Are known.
With this configuration, even if the AC 200V system AC input voltage drops to the AC 100V system level due to, for example, the momentary power failure described above, the switching operation only changes from the half-bridge operation to the full-bridge operation. The switching element or the like does not exceed the breakdown voltage. For this reason, it is not necessary to detect the DC input voltage on the standby power supply side, and therefore, it can be adopted for an electronic device that does not have a standby power supply. In addition, since it is not switching in the commercial power supply line, switching of the circuit form by a semiconductor switch is possible, so that a large switch part such as an electromagnetic relay is not necessary.

However, in this configuration, it is necessary to provide at least four switching elements in order to form a full bridge coupling corresponding to the AC100V system. That is, it is necessary to add two switching elements as compared with the converter configuration using only the half-bridge coupling method that can be formed by two switching elements.
In this configuration, four stones perform switching operation in the full bridge operation, and three stone switching elements perform switching operation in the half bridge operation. Although the resonance type converter has low switching noise, as the number of switching elements that perform switching in this way increases, the switching noise becomes disadvantageous.

  In this way, in any of the configurations described above that are compatible with a wide range, when compared with a configuration compatible with a single range, an increase in circuit scale and cost increase due to an increase in the number of parts cannot be avoided. . In addition, there are inherent problems that the former configuration does not have, such as limitations on the range of use for equipment, and the latter configuration increases switching noise.

Further, as in the power supply circuit shown in FIG. 20, when the control range of the switching frequency is correspondingly wide, there is a problem that the high-speed response characteristic of stabilization for the secondary side DC output voltage Eo is deteriorated. Arise.
Some electronic devices involve, for example, an operation that fluctuates so that the load condition is instantaneously switched between a state of maximum load and a state of almost no load. Such a load variation is also called a switching load. As a power supply circuit mounted on such a device, it is necessary to appropriately stabilize the secondary side DC output voltage in response to the load fluctuation that is the switching load.
However, as described above with reference to FIG. 23, when the control range of the switching frequency is wide, the secondary side DC output voltage is set to a required level corresponding to the load fluctuation such as the switching load. It takes a relatively long time to vary the switching frequency to That is, an unsatisfactory result is obtained as a response characteristic of constant voltage control.
In particular, as shown in FIG. 22, the power supply circuit shown in FIG. 20 has a large switching frequency in the load range from about load power Po = 25 W or less to 0 W as a switching frequency characteristic according to constant voltage control. It can be seen that the constant voltage control response to the switching load as described above is disadvantageous.

Therefore, the present invention is configured as follows as a switching power supply circuit in consideration of the above-described problems.
That is, a switching unit formed by including a switching element that performs switching by inputting a DC input voltage, and a switching drive unit that switches the switching element.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the primary winding is wound, and the primary side is formed. An insulating converter transformer is provided in which a gap length formed at a predetermined position of the core is set so that a predetermined coupling coefficient is obtained on the secondary side.
Further, the primary side series is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type. A resonance circuit is provided.
Further, a first inductance circuit is formed by at least the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary side series resonance capacitor connected in series to the secondary winding, and is set in the primary side series resonance circuit. A secondary side series resonance circuit in which a predetermined second resonance frequency lower than the resonance frequency of 1 is set.
Also, a secondary side DC output voltage generating means for generating a secondary side DC output voltage by inputting a resonance output obtained in the secondary side series resonance circuit and performing a rectification operation, and a level of the secondary side DC output voltage. Accordingly, the switching drive means is controlled to vary the switching frequency of the switching means, thereby providing constant voltage control means for performing constant voltage control on the secondary side DC output voltage.
In addition, with respect to the electromagnetic coupling type resonance circuit formed by including the primary side series resonance circuit and the secondary side series resonance circuit, the output characteristic with respect to the input of the frequency signal having the switching frequency is a single peak characteristic. , Inserted so as to be connected in series with at least one of the primary winding and the secondary winding so that a total coupling coefficient of a predetermined primary side and secondary side regarded as loose coupling is set And an inductor to be provided.

The switching power supply circuit having the above configuration adopts a basic configuration as a current resonance type converter including a primary side series resonance circuit, and also on the secondary side, series resonance is caused by the secondary winding and the secondary side series resonance capacitor. A circuit is to be formed. In this case, the resonance frequency (second resonance frequency) of the secondary side series resonance circuit is set to a predetermined value lower than the resonance frequency (first resonance frequency) of the primary side series resonance circuit.
By adopting such a configuration, a coupled resonance circuit by electromagnetic coupling of an insulating converter transformer is formed as the switching power supply circuit of the present invention. The insulating converter transformer itself is set in series with at least one of the primary winding and the secondary winding of the insulating converter transformer after setting a predetermined coupling coefficient by forming a gap with a predetermined gap length. By inserting the connected inductor and combining the inductance of this inductor with the leakage inductance of the isolated converter transformer itself, the overall coupling coefficient (total coupling coefficient) of the isolated converter transformer in the power circuit is loosely coupled. A predetermined value that is considered to be set is set.
In this way, by setting a value that is loosely coupled to the overall coupling coefficient of the insulating converter transformer, the output characteristics for the frequency signal (switching output) of the switching frequency that is the input to the coupled resonance circuit are steep. It is possible to obtain a simple unimodal characteristic. As a result, the variable range (necessary control range) of the switching frequency required for stabilization can be reduced as compared with the case where the series resonance circuit is formed only on the primary side.
In addition, in order to obtain a loosely coupled state sufficient to obtain the above single peak characteristic, the inductance inductance of the insulating converter transformer itself is combined with the leakage inductance of the insulating converter transformer itself, so that the leakage inductance of the insulating converter transformer itself is obtained. Can be kept below a certain level. That is, the gap length, which is a determining factor of the leakage inductance of the insulating converter transformer, can be set to a certain value or less, and it is not particularly necessary to increase the gap length.

As described above, according to the present invention, the variable control range (necessary control range) of the switching frequency necessary for constant voltage control is reduced.
As a result, first, it is easy to obtain a resonant converter capable of handling a wide range only by switching frequency control. And it becomes possible to support a wide range only by switching frequency control, for example, switching the rectifier circuit system according to the rated level of commercial AC power supply, or switching between half-bridge coupling and full-bridge coupling There is no need to adopt a configuration for switching the circuit system. As a result, it is possible to reduce the number of circuit components and the area of the board, and to obtain an effect that the application range of the power supply circuit to the electronic device is expanded and that it is advantageous for switching noise.

  Further, as a basic configuration for obtaining the effect of the present invention as described above, a secondary side series resonance capacitor is added to the configuration of the current resonance type converter including the primary side series resonance circuit, and the primary winding And / or a configuration for providing an inductor connected in series with the secondary winding may be adopted, and the addition or change of the number of parts may be very small. .

  Further, by reducing the required control range of the switching frequency as described above, the response of the constant voltage control is also improved. This is because, for example, the load power fluctuates so as to switch between the maximum / no-load conditions, that is, a so-called switching load, a constant voltage with a higher response than before. Control can be performed, and the reliability of the device is improved accordingly.

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit as a first embodiment in the best mode for carrying out the present invention (hereinafter also referred to as an embodiment). The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a separately excited current resonance converter using a half-bridge coupling method as a basic configuration on the primary side.
The power supply circuit shown in FIG. 1 adopts a so-called wide-range configuration that operates in accordance with any AC 100 V system or AC 200 V system commercial AC power input. Moreover, as corresponding load electric power, for example, it corresponds to the fluctuation range from about load electric power Po = 200W to Po = 0W (no load).

First, in the power supply circuit shown in FIG. 1, a common mode noise filter including filter capacitors CL and CL and a common mode choke coil CMC is formed for the commercial AC power supply AC.
A full-wave rectifying / smoothing circuit including a bridge rectifier circuit Di and one smoothing capacitor Ci is connected to the commercial AC power supply AC that is a subsequent stage of the noise filter.
When this full-wave rectifying / smoothing circuit inputs a commercial AC power supply AC and performs a full-wave rectifying operation, a rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to the AC input voltage VAC.

  As shown in the figure, the current resonance type converter for switching (intermittently) by inputting the DC input voltage includes a switching circuit in which two switching elements Q1, Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A primary side partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the primary side partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1 (and the inductance of the high frequency inductor L11). A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In addition, an oscillation / drive circuit 2 is provided for switching the switching elements Q1, Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. In this case, for example, a general-purpose IC can be used. The oscillation circuit of the oscillation / drive circuit 2 generates an oscillation signal having a required frequency, and the drive circuit generates a switching drive signal that is a gate voltage for switching the MOS-FET by using the oscillation signal. The voltage is applied to the gates of the switching elements Q1, Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be continuously turned on / off at alternate timings according to the switching frequency according to the cycle of the switching drive signal.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 to the secondary side.
In this case, one end of the primary winding N1 of the insulating converter transformer PIT is switched with the source of the switching element Q1 via a series connection of a high frequency inductor (high frequency choke coil) L11 and a primary side series resonance capacitor C1. Switching output is transmitted by being connected to a connection point (switching output point) with the drain of the element Q2. The other end of the primary winding N1 is connected to the primary side ground.

Here, the insulating converter transformer PIT has a structure as shown in the cross-sectional view of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding (N1) and the secondary side winding (N2) are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side Due to the different winding regions, the windings are wound around the inner magnetic legs of the EE type core. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G is formed as shown in the figure for the inner magnetic leg of the EE type core. In this case, as the gap G, for example, a gap length of about 1.6 mm is set, and as a result, a coupling coefficient k between the primary side and the secondary side is set to, for example, about k = 0.75. Yes. Accordingly, the degree of coupling of the insulating converter transformer PIT is the same as that of the power supply circuit as the prior art shown in FIG. Note that k = 0.73 was set as the actual coupling coefficient k in the power supply circuit of the present embodiment. The gap G can be formed by making the inner magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

Returning to FIG.
The insulating converter transformer PIT generates a predetermined leakage inductance L1 in the primary winding N1 by the structure described with reference to FIG. Further, depending on the connection mode described above, the primary winding N1 and the primary side series resonance capacitor C1 are connected in series via the high frequency inductor L11. Accordingly, a series resonance circuit (primary side series resonance circuit) is formed by the leakage inductance L1 of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1.
In the present embodiment, since the high frequency inductor L11 is inserted in series between the primary winding N1 and the primary side series resonance capacitor C1, the inductance component that forms the primary side series resonance circuit described above is used. Is a combination of the inductance of the high-frequency inductor L11 and the leakage inductance L1 of the primary winding N1 (L1 + L11).
The primary side series resonance circuit is connected to the switching output points of the switching elements Q1 and Q2. Therefore, the switching outputs of the switching elements Q1 and Q2 are transmitted to the primary side series resonance circuit. Become. In the primary side series resonance circuit, the resonance operation is performed by the transmitted switching output, so that the operation of the primary side switching converter is a current resonance type.

In the present embodiment, as described above, the high frequency inductor L11 is connected in series to the primary winding N1. In this case, since the high-frequency inductor L11 is in a series relationship with the primary winding N1, equivalently, the inductance of the high-frequency inductor L11 can be regarded as a leakage inductance component of the primary winding N1. Therefore, the leakage inductance on the primary side in the insulating converter transformer PIT is L11 + L1.
For this reason, the coupling coefficient k as the insulating converter transformer PIT itself is k = 0.73 as described above. However, as described above, the leakage inductance on the primary side is equal to that of the high-frequency inductor L11. As a result of an apparent increase due to the combined inductance, a value lower than 0.73 is obtained as the overall coupling coefficient (total coupling coefficient) kt of the insulating converter transformer PIT in the power supply circuit. That is, the degree of coupling of the insulating converter transformer PIT in the power supply circuit is set lower than the coupling coefficient k due to the structure of the insulating converter transformer PIT itself. In the present embodiment, by setting a predetermined inductance value for the high-frequency inductor L11, the total coupling coefficient kt is set to about 0.65, and in practice, kt = 0.64 is set. It is said.

By the way, according to the description so far, as the primary side switching converter shown in this figure, the operation as the current resonance type by the primary side series resonance circuit (L1-L11-C1) and the above-described primary side partial voltage resonance circuit ( Cp // L1) and partial voltage resonance operation can be obtained.
That is, on the primary side of the power supply circuit shown in this figure, a configuration in which the resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit is adopted. Here, a switching converter in which two resonance circuits are combined in this way is referred to as a “composite resonance type converter”.

An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited (induced) in the secondary winding N2 of the insulating converter transformer PIT.
In this case, for the secondary winding N2, first, a secondary side series resonance capacitor C2 is connected in series with one end side. As a result, a secondary side series resonance circuit is formed by the capacitance of the secondary side series resonance capacitor C2 and the leakage inductance L2 of the secondary winding N2. That is, in this embodiment, a series resonant circuit is formed on each of the primary side and the secondary side of the insulating converter transformer PIT.

For the secondary side series resonant circuit (L2-C2), full-wave rectification formed by a bridge rectifier circuit formed by connecting four rectifier diodes Do1 to Do4 as shown in the figure and a smoothing capacitor Co. The circuit is connected.
Depending on the full-wave rectifier circuit, in one half cycle of the alternating voltage excited in the secondary winding N2, the pair of rectifier diodes [Do1, Do4] of the bridge rectifier circuit is turned on, and is connected to the smoothing capacitor Co. The operation of charging the rectified current is obtained. Further, in the other half cycle of the alternating voltage excited by the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining the set of the rectifier diodes [Do2, Do3] conductive.
As a result, the secondary side DC output voltage Eo having a level corresponding to the same level as the level of the alternating voltage excited in the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co.
The secondary side DC output voltage Eo obtained in this way is supplied to a load (not shown) and is also branched and input as a detection voltage for the control circuit 1 described later.
Further, since the full-wave rectifier circuit performs a rectifying and smoothing operation on the resonance output of the secondary side series resonant circuit, the secondary side rectifying operation by the full-wave rectifier circuit is also of a current resonance type. That is, the rectified current waveform includes a sine waveform based on the resonance frequency of the secondary side series resonance circuit.

Here, in the present embodiment, the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit are as follows.
fo1> fo2
The relationship expressed by In practice, the resonance frequency fo2 of the secondary side series resonance circuit is set to be about 2/3 of the resonance frequency fo1 of the primary side series resonance circuit. As the actual power supply circuit shown in FIG. 1, since the resonance frequency fo1 of the primary side series resonance circuit is set to about 60 kHz, the resonance frequency fo2 of the secondary side series resonance circuit is set to about 40 kHz. .

Further, on the secondary side of the power supply circuit shown in FIG. 1, a secondary side partial resonance capacitor Cp2 is inserted in parallel with the secondary side series resonance circuit (L2-C2).
A parallel resonance circuit as a secondary side partial voltage resonance circuit is formed by the capacitance of the secondary side partial resonance capacitor Cp2 and the leakage inductance L2 of the secondary winding N2. Depending on the secondary side partial voltage resonance circuit, at the timing when the pair of rectifier diodes [Do1, Do4] and the pair of rectifier diodes [Do2, Do3] forming the full-wave rectifier circuit on the secondary side are turned off. Only the secondary side partial voltage resonance operation in which voltage resonance occurs is obtained.
By this partial voltage resonance operation, a path through which a reverse current generated at the time of turn-off of each pair of rectifier diodes [Do1, Do4] [Do2, Do3] is formed, and the reactive power at this time is reduced. As a result, power loss in the secondary rectifier circuit can be reduced.

According to the description so far, the switching power supply circuit according to the present embodiment has a primary side series resonance circuit (L1-L11-C1) and a primary side partial voltage resonance circuit (L1 (-L11) // Cp) on the primary side. The secondary side is provided with a secondary side series resonance circuit (L2-C2) and a secondary side partial voltage resonance circuit (L2 // Cp2).
As described above, the switching converter in which the two resonance circuits including the series resonance circuit and the partial voltage resonance circuit on the primary side are combined is referred to as a composite resonance type converter. A switching converter in which three or more resonance circuits are combined as described above is referred to as a multiple resonance type converter.

The control circuit 1 is provided to stabilize the secondary side DC output voltage Eo by the switching frequency control method.
In this case, the control circuit 1 supplies the oscillation / drive circuit 2 with a detection output corresponding to the level change of the secondary side DC output voltage Eo as a detection input. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. For this purpose, the frequency of the oscillation signal generated by the internal oscillation circuit is varied.
By changing the switching frequency of the switching elements Q1 and Q2, the resonance impedance of the primary side series resonance circuit changes, and the amount of electric power transmitted from the primary winding N1 to the secondary winding N2 side of the insulating converter transformer PIT is reduced. Although it changes, this operates so as to stabilize the level of the secondary side DC output voltage Eo.

Although details will be described later, the switching frequency control method in the power supply circuit of the present embodiment is based on the resonance frequency fo1 of the primary side series resonance circuit and the intermediate resonance frequency fo determined by the resonance frequency fo2 of the secondary side series resonance circuit. A higher frequency range is set as a variable range of the switching frequency. That is, a so-called upper side control method is adopted.
As a general matter, a series resonance circuit has the lowest resonance impedance at the resonance frequency. Therefore, when the upper side control method based on the resonance frequency of the series resonance circuit is adopted as in the present embodiment, the resonance impedance is increased as the switching frequency fs increases. become.
Therefore, for example, when the secondary side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. This lowers the resonance impedance and increases the amount of power transmitted from the primary side to the secondary side, so that the secondary side DC output voltage Eo rises.
On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the switching frequency is controlled to be increased. As a result, the resonance impedance is increased and the power transmission amount is reduced, so that the secondary side DC output voltage Eo is lowered. Thus, the secondary side DC output voltage Eo is stabilized by changing the switching frequency.

  In the power supply circuit of FIG. 1 having the above-described configuration, on the primary side and the secondary side, series resonance circuits (primary side series resonance circuit (L1-L11-C1) and secondary side series resonance circuit (L2-C2)) are respectively provided. It is made to prepare. In this embodiment, by adopting such a configuration, as a power circuit based on a current resonance type converter, it operates in response to a commercial AC power input of AC100V system and AC200V system, and is practically used for a so-called wide range. It becomes possible. This point will be described with reference to FIGS.

The circuit diagram of FIG. 3 shows an equivalent circuit when the power supply circuit of the present embodiment shown in FIG. 1 is viewed from the relationship between the primary side series resonant circuit and the secondary side series resonant circuit. In this equivalent circuit diagram, the same parts as those in FIG.
In this figure, an insulating converter transformer PIT is shown in which a primary winding N1 and a secondary winding N2 of a predetermined number of turns having a winding ratio of 1: n are wound. In this figure, the coupling coefficient indicating the degree of coupling between the primary side and the secondary side in the insulating converter transformer PIT is regarded as the above-described total coupling coefficient kt.
On the primary side of the insulating converter transformer PIT, L1l and L1e indicate a leakage (leakage) inductance of the primary winding N1 and an excitation inductance of the primary winding N1, respectively. Further, L2l and L2e on the secondary side of the insulating converter transformer PIT respectively indicate a leakage (leakage) inductance of the secondary winding N2 and an excitation inductance of the secondary winding N2. For confirmation, the leakage inductance L1l of the primary winding N1 shown here is a combination of the leakage inductance L1 of the primary winding N1 itself and the inductance of the high-frequency inductor L11.

In the equivalent circuit diagram shown in FIG. 3, an alternating current (frequency signal) with a switching frequency fs is input on the primary side of the insulating converter transformer PIT. That is, the switching output of the primary side switching converter (switching elements Q1, Q2) is an input.
Then, on the primary side of the insulating converter transformer PIT, an AC input with the switching frequency fs is supplied to the primary side series resonance circuit. In the primary side series resonance circuit, as shown in the figure, the primary side series resonance capacitor C1−leakage inductance L1l is connected in series to the primary winding N1, and the excitation inductance L1e is parallel to the primary winding N1. Can be viewed as connected to
Similarly, as the secondary side series resonance circuit of the insulating converter transformer PIT, the secondary side series resonance capacitor C2−leakage inductance L2l is connected in series to the secondary winding N2, and the excitation inductance L2e is set to two. It can be viewed as being connected in parallel to the next winding N2. In this figure, the output of the secondary side series resonance circuit formed as described above is output to the load RL. The load RL here is a circuit and a load after the secondary-side full-wave rectifier circuit.

In the equivalent circuit of FIG. 3 as the above connection mode, assuming that the total coupling coefficient kt of the insulating converter transformer PIT and the self-inductance of the primary winding N1 are L1, the leakage inductance L1l of the primary winding N1 is L1l = (1− kt ² ) L1 (Formula 1)
Can be represented by
Also, regarding the exciting inductance L1e of the primary winding N1,
L1e = kt ² × L1 (Formula 2)
Can be represented by
Similarly, regarding the leakage inductance L2l and excitation inductance L2e of the secondary winding N2, if the self-inductance of the primary winding N2 is L2, respectively,
L2l = (1−kt ² ) L2 (Formula 3)
L2e = kt ² × L2 (Formula 4)
It is represented by

  Here, the equivalent circuit shown in FIG. 3 includes a primary side series resonant circuit on the primary side and a secondary side series resonant circuit on the secondary side via electromagnetic induction of the insulating converter transformer PIT. It is shown. Therefore, the circuit shown in this figure can be regarded as forming a coupled resonance circuit by electromagnetic coupling. For this reason, the constant voltage control characteristic for the secondary side DC output voltage Eo in the power supply circuit shown in FIG. 1 differs depending on the degree of coupling (coupling coefficient k) of the insulating converter transformer PIT. This point will be described with reference to FIG.

FIG. 4 shows the output characteristics with respect to the input (switching frequency signal) for the equivalent circuit of FIG. That is, the control characteristic for the secondary side DC output voltage Eo is shown by the relationship with the switching frequency fs. In this figure, the horizontal axis represents the switching frequency, and the vertical axis represents the level of the secondary side DC output voltage Eo.
As described with reference to FIG. 1, in the present embodiment, the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit are higher. In FIG. 4, the horizontal axis indicating the switching frequency fs is shown to correspond to the resonance frequencies fo1 and fo2, but also in this FIG. 4, the resonance frequency fo1 is higher than the resonance frequency fo2.

Here, it is assumed that the coupling degree of the insulating converter transformer PIT is set to a tight coupling state where the total coupling coefficient kt = 1. Then, the leakage inductance L1l of the primary winding N1 and the leakage inductance L2l of the secondary winding N2 in this case are obtained by substituting kt = 1 into the above (formula 1) and (formula 3), respectively.
L1l = L2l = 0 (Formula 5)
Will be represented as That is, it is shown that the leakage inductance of the primary winding N1 and the secondary winding N2 does not exist because the insulating converter transformer PIT is tightly coupled.

で表され、
周波数ｆ２は、

で表される。
また、上記（数１）（数２）における項の１つであるｆｏは、一次側直列共振回路の共振周波数ｆｏ１と、二次側直列共振回路の共振周波数ｆｏ２との中間に存在する中間共振周波数であり、１次側のインピーダンスと２次側のインピーダンスと、一次側と二次側とで共通となるインピーダンス（相互結合インダクタンスＭ）により決定される周波数である。
なお、相互結合インダクタンスＭについては、

により表される。 As described above, as a constant voltage control characteristic in a state where the primary side and the secondary side of the insulating converter transformer PIT are tightly coupled, as shown as a characteristic curve 1 in FIG. The resonance frequency fo1 and the resonance frequency fo2 of the secondary side series resonance circuit have so-called bimodal characteristics in which the secondary side DC output voltage Eo peaks at frequencies f1 and f2.
Here, the frequency f1 is

Represented by
The frequency f2 is

It is represented by
Further, fo which is one of the terms in the above (Equation 1) and (Equation 2) is an intermediate resonance that exists between the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side series resonance circuit. The frequency is determined by the impedance on the primary side, the impedance on the secondary side, and the impedance (mutual coupling inductance M) that is common to the primary side and the secondary side.
For the mutual coupling inductance M,

It is represented by

  Further, assuming that the total coupling coefficient kt is gradually reduced from the state of kt = 1, that is, when the degree of loose coupling is gradually increased from the tightly coupled state, FIG. The characteristic curve 1 shows a change in which the bimodal tendency gradually diminishes and becomes flat near the intermediate resonance frequency fo. Then, when the total coupling coefficient kt is reduced to a certain level, a so-called critical coupling state is obtained. In this critical coupling state, as shown by the characteristic curve 2, there is no tendency as a bimodal characteristic, and the curve shape becomes flat with the intermediate resonance frequency fo as the center.

Further, if the overall coupling coefficient kt is decreased from the critical coupling state and the loose coupling state is strengthened, the peak is obtained only at the intermediate frequency fo as shown by the characteristic curve 3 in FIG. A unimodal characteristic is obtained. Further, when comparing the characteristic curve 3 with the characteristic curves 1 and 2, the characteristic curve 3 has a peak level itself lower than that of the characteristic curves 1 and 2, but as a quadratic curve shape thereof, It can be seen that it has a steeper slope.
Insulating converter transformer PIT of the present embodiment is set in a loosely coupled state in which total coupling coefficient kt ≦ 0.75. In the setting of the coupling coefficient k, the operation is based on the single peak characteristic shown as the characteristic curve 3.

  Here, when the unimodal characteristic shown in FIG. 4 is compared with the constant voltage control characteristic of the composite resonant converter of the prior art power supply circuit (FIG. 20) shown in FIG. On the other hand, the characteristic shown in FIG. 23 with respect to 4 has a considerably gentle slope in terms of a quadratic function.

  In the power supply circuit shown in FIG. 20, since the characteristic shown in FIG. 23 is gradual as described above, the necessary control range of the switching frequency for performing the constant voltage control on the secondary side DC output voltage Eo. For example, even under conditions corresponding to a single range, Δfs = 120 kHz or more when fs = 80 kHz to 200 kHz or more. For this reason, as described above, it is very difficult to adapt to a wide range only by constant voltage control by switching frequency control.

On the other hand, the constant voltage control characteristic of the present embodiment is a single-peak characteristic indicated by the characteristic curve 3 in FIG. 4, and the constant voltage control operation is as shown in FIG.
5, the characteristic curves A and B for the maximum load power Pomax and the minimum load power Pomin when the AC input voltage VAC = 100 V (AC 100 V system) for the power supply circuit of the present embodiment shown in FIG. Four characteristic curves are shown, which are characteristic curves C and D at the maximum load power Pomax and at the minimum load power Pomin when the AC input voltage VAC = 230 V (AC 200 V system).

As can be seen from FIG. 5, first, when the AC input voltage VAC = 100V corresponding to the AC 100V system input, the switching required for making the secondary side DC output voltage Eo constant at the required rated level tg. The frequency variable control range (necessary control range) is indicated by Δfs1. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve A to the switching frequency fs at level tg in the characteristic curve B.
In addition, when the AC input voltage VAC = 230 V corresponding to the AC 200 V system input, the variable control range (necessary control) of the switching frequency necessary for making the secondary side DC output voltage Eo constant at the required rated level tg. (Range) is indicated by Δfs2. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve C to the switching frequency fs at level tg in the characteristic curve D.

As described above, the unimodal characteristic that is the control characteristic of the secondary side DC output voltage Eo in the present embodiment is considerably steep in a quadratic function curve as compared with the control characteristic shown in FIG. It is.
For this reason, Δfs1 and Δfs2 that are the respective required control ranges when the AC input voltage VAC = 100V and VAC = 230V are considerably reduced as compared with Δfs shown in FIG. . For example, Δfs1 and Δfs2 actually measured are within 4 kHz, and are reduced to about 1/30 of the actual Δfs shown in FIG.
In addition, the frequency variable range (ΔfsA) from the lowest switching frequency at Δfs1 (switching frequency fs at level tg in characteristic curve A) to the highest switching frequency at Δfs2 (switching frequency fs at level tg in characteristic curve A) is also used. , It is correspondingly narrow.

  Here, the actual frequency variable range ΔfsA in the power supply circuit of the present embodiment shown in FIG. 1 is sufficiently within the variable range of the switching frequency corresponding to the current switching drive IC (oscillation / drive circuit 2). It has become a thing. That is, in the power supply circuit shown in FIG. 1, the switching frequency can be actually variably controlled within the frequency variable range ΔfsA. This means that the power supply circuit shown in FIG. 1 can stabilize the secondary side DC output voltage Eo in accordance with any commercial AC power supply input of AC100V system and AC200V system. In other words, the power supply circuit shown in FIG. 1 can support a wide range only by switching frequency control.

  Incidentally, a coupled resonance circuit using electromagnetic coupling is already known as a technique for expanding the amplification bandwidth of an amplifier circuit using a transistor, for example, as an intermediate frequency transformer amplifier. However, in such a field, the bimodal characteristic in the tight coupling or the flat characteristic in the critical coupling is used, and the single peak characteristic in the loose coupling is not used. In the present embodiment, in such a coupled resonant circuit technology using electromagnetic coupling, a single-peak characteristic with loose coupling that has not been employed in the field of communication technology is actively used in the field of resonant switching converters. It can be said that. As a result, as described above, the variable range (necessary control range) of the switching frequency necessary for stabilizing the secondary side DC output voltage Eo is reduced, and the wide range only by the constant voltage control in the switching frequency control. This is possible.

By the way, when the loose coupling state equivalent to the total coupling coefficient kt = 0.65 or less in the present embodiment is obtained only by the structure of the insulating converter transformer PIT without the high frequency inductor L11, for example, The gap G of the inner magnetic leg of the EE type core of the insulating converter transformer PIT can be expanded to about 2.8 mm, and the insulating converter transformer PIT itself can be configured as a loosely coupled transformer having a coupling coefficient k = 0.65 or less. Conceivable.
By adopting such a configuration, the unimodal characteristics described with reference to FIG. 4 can be obtained. Therefore, as described with reference to FIG. 5, the necessary control range of the switching frequency is reduced, and the AC100V system and AC200V are reduced. The secondary side DC voltage can be stabilized corresponding to the commercial AC power input of the system.

However, in the case of such an insulating converter transformer PIT structure, the eddy current loss near the gap G of the core of the insulating converter transformer PIT increases, and the AC → DC power conversion efficiency (ηAC → DC) decreases correspondingly. Will occur. The tendency for the AC → DC power conversion efficiency to decrease due to this eddy current loss becomes more prominent as the level of the AC input voltage VAC increases. Therefore, the power supply circuit corresponding to the wide range causes a problem that the AC → DC power conversion efficiency is lowered when used in the AC 200V system than when used in the AC 100V system.
However, since the increase in the eddy current loss is within an allowable range under load conditions up to, for example, the maximum load power Pomax = 150 W or less, as described above, the high-frequency inductor L11 is omitted, and only the isolated converter transformer PIT is used. Even if a loosely coupled state with a coupling coefficient k = 0.65 or less is set by this, a practical power supply circuit compatible with a wide range can be obtained. However, when the maximum load power Pomax = 200 W is to be dealt with as in the present embodiment, the above increase in eddy current loss becomes significant to the extent that it cannot be ignored. For this reason, it becomes difficult to put the insulation converter transformer PIT itself into a power supply circuit compatible with a wide range by setting the coupling coefficient k = 0.65 or less.

Therefore, in the present embodiment, as described above, by connecting the high frequency inductor L11 to the primary winding N1, the leakage inductance of the primary winding N1 is equivalently increased by the inductance of the high frequency inductor L11. Thus, kt = 0.65 or less is set for the total coupling coefficient kt of the insulating converter transformer PIT in the power supply circuit.
In this case, since the coupling coefficient k of the insulating converter transformer PIT itself can be set to about k = 0.75, which is equivalent to the power circuit of the prior art, the gap length of the gap G is also as described above. It can be about 1.6mm. That is, it is possible to suppress the gap length to a certain level or less so as not to cause an increase in eddy current.
As a result, the power supply circuit according to the embodiment does not increase the eddy current loss described above, so that the AC → DC power conversion efficiency is not reduced due to this increase. Therefore, even when used in an AC 200V system, AC to DC power conversion efficiency characteristics that are practically satisfactory as a power supply circuit compatible with a wide range can be obtained.

Further, in the power supply circuit of the present embodiment, a series resonance circuit (secondary side series resonance circuit) is also formed on the secondary side, which is also a factor for improving the power conversion efficiency. ing.
In other words, by providing the secondary side series resonance circuit, it is possible to supply power as the secondary side DC output voltage Eo including the increase in energy obtained by the resonance operation, and the coupling is loosely coupled. The decrease in power conversion efficiency due to is compensated. Furthermore, as described above, by forming a secondary side partial voltage resonance circuit on the secondary side, switching loss in the rectifier diode on the secondary side is reduced, which also improves power conversion efficiency. Has contributed.

  FIG. 6 shows the operation of the main part of the power supply circuit of FIG. 1 configured for wide range as described above. In this figure, as AC input voltage (VAC) conditions / load conditions, VAC = 100 V (AC 100 V system) / Pomax (maximum load power) = 200 W, VAC = 100 V (AC 100 V system) / Pomin (minimum load power: no load) ) = 0 W, VAC = 230 V (AC 200 V system) / Pomax (maximum load power) = 200 W, VAC = 230 V (AC 200 V system) / Pomin (minimum load power: no load) = 0 W are shown. .

The rectangular wave voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2.
The voltage V1 has a waveform clamped at the level of the rectified and smoothed voltage Ei in the on period in which the switching element Q2 is turned on when the switching element Q2 is turned on and in the off period in which the switching element Q2 is turned off.
During the ON period of the switching element Q2, a switching current IQ2 having a waveform shown in the figure flows through the switching circuit system including the switching element Q2 and the clamp diode DD2. Further, the switching current IQ2 becomes 0 level during the OFF period of the switching element Q2.
Although not shown, the voltage across the other switching element Q1 and the switching current flowing through the switching circuit (Q1, DD1) are obtained as a waveform obtained by shifting the voltage V1 and the switching current IQ2 by 180 °. . That is, the switching element Q1 and the switching element Q2 perform the switching operation at the same cycle timing so as to be alternately turned on / off.
Further, as the primary side series resonance current Io flowing through the primary side series resonance circuit (L1-L11-C1), the switching currents flowing through these switching circuits (Q1, DD1) (Q2, DD2) are combined, It flows according to the waveform shown in the figure.

Here, as described above, the voltage V1 indicates the switching timing. When the waveform of the voltage V1 is compared between the AC input voltage VAC = 100 V and VAC 230 V under the same load condition, One cycle is longer when the AC input voltage VAC = 100V than when the AC input voltage VAC = 230V. This is because the switching frequency varies in such a way that the secondary side DC voltage Eo tends to decrease as the input level of the commercial AC power supply (VAC) decreases. As the input level of the commercial AC power supply (VAC) increases, the switching frequency on the primary side increases as the secondary side DC voltage Eo fluctuates as it increases. It is shown that.
Further, when the voltage V1 is compared between the maximum load power Pomax = 200 W and the minimum load power Pomin = 0 W under the same AC input voltage VAC level condition, the maximum load power Pomax = 200 W However, one cycle is longer than when the minimum load power Pomin = 0 W. That is, when the level of the secondary side DC output voltage Eo decreases due to heavy load, the switching frequency decreases, and when the level of the secondary side DC output voltage Eo increases due to light load. It shows a change that the switching frequency becomes higher.
This indicates that the constant voltage control operation by the switching frequency control method (upper side control) is performed as the constant voltage control operation of the secondary side DC voltage with respect to the load fluctuation and the fluctuation of the commercial AC power supply input level. Yes.

Further, according to the above-described primary side operation (V1, IQ2, Io), an alternating voltage V2 having a waveform shown in the figure is induced in the secondary winding N2 of the insulating converter transformer PIT. The period length of one period of the alternating voltage V2 corresponds to the switching period on the primary side.
Then, in one half cycle period of the alternating voltage V2, the rectifier diodes [Do1, Do4] on the secondary side are turned on and a rectified current flows. The rectifier diodes [Do2, Do3] on the next side are turned on and a rectified current flows. The secondary winding current I2 flowing through the secondary winding N2 is obtained by synthesizing the rectified currents flowing every half cycle of the alternating voltage V2, and the waveform shown in the figure is obtained.

FIG. 7 shows the characteristics of the power supply circuit shown in FIG. ). In this figure, the characteristics when the AC input voltage VAC = 100 V corresponding to the AC 100 V system is indicated by a solid line, and the characteristics when the AC input voltage VAC = 230 V corresponding to the AC 200 V system is indicated by a broken line.
Further, the characteristics shown in FIG. 7 are obtained by conducting experiments by selecting the main parts of the power supply circuit shown in FIG. 1 as follows.
First, with respect to the insulating converter transformer PIT, the gap length of the gap G of the EE type core was 1.6 mm, and the primary winding N1 = 31T and the secondary winding N2 = 28T were wound. With this structure, k = 0.73 is obtained as the coupling coefficient k of the insulating converter transformer PIT itself. In addition, the total coupling coefficient kt = 0.64 is set by selecting the high-frequency inductor L11 = 37 μH.
Further, each resonance capacitor for forming the primary side series resonance circuit and the secondary side series resonance circuit was selected as follows.
・ Primary side series resonant capacitor C1 = 0.033μF
・ Secondary side series resonant capacitor C2 = 0.15μF
Further, as the secondary side DC output voltage Eo, the rated level was 135V.
Note that the waveform diagram shown in FIG. 6 also shows the experimental results under the configuration in which the parts are selected as described above.

First, the switching frequency fs is overlapped from load power Po = 0 W (minimum load power: no load) to Po = 200 W (maximum load power) in both conditions of AC input voltage VAC = 100 V and VAC = 230 V. It tends to change so as to decrease in accordance with the load condition.
In addition, the variable range of the switching frequency fs with respect to the load power Po = 0W to 200W changes in the range where the AC input voltage VAC = 230V is higher than the AC input voltage VAC = 100V. That is, this figure also shows that constant voltage control is performed with respect to AC input voltage fluctuation and load fluctuation by the switching frequency control method (upper side control).
In this embodiment, the switching frequency fs changes linearly in the range of load power Po = 0W to 200W, and abrupt changes occur in the load power range below a certain level as in the prior art power supply circuit. Not shown.

The specific value of the switching frequency fs is a change in the range of 69.4 kHz to 66.7 kHz (required control range) when the AC input voltage VAC is 100 V with respect to the fluctuation of the load power Po = 0 W to 200 W. ) And Δfs1 shown in FIG. 5 is Δfs1 = 2.7 kHz (69.4 kHz−66.7 kHz). Further, when the AC input voltage VAC is 230 V, it is measured that the required control range is 100 kHz to 91.7 kHz, and Δfs2 shown in FIG. 5 is Δfs2 = 8.3 kHz (100 kHz-91.7 kHz). ing. In this way, the necessary control range of the switching frequency fs for each range in the AC100V system and AC200V system is less than 9 kHz, which is greatly reduced as compared with the power circuit of the prior art (FIG. 20). Yes. In FIG. 7, the frequency difference between the maximum value of the switching frequency fs of 100 kHz (VAC = 230 V / Pomin = 0 W) and the minimum value of 66.7 kHz (VAC = 100 V / Pomax = 200 W). 33.3 kHz (100 kHz to 66.7 kHz), the necessary control range of the switching frequency fs is greatly reduced as compared with the power supply circuit of the prior art (FIG. 20) even when the wide range is considered.
Further, as described above, the upper limit of the switching drive frequency of the switching element driving IC (oscillation / drive circuit 2) at present is about 200 kHz. Therefore, the above-mentioned range of the switching frequency fs of about 66.7 kHz to 100 kHz is sufficiently obtained by the current switching drive IC. In other words, as described above, the power supply circuit according to the present embodiment has the configuration of the current switching drive circuit system because the necessary control range of the switching frequency fs is greatly reduced in a relatively low frequency region. In this way, it is possible to cope with a wide range only by switching frequency control.

  Further, the AC → DC power conversion efficiency (ηAC → DC) tends to increase as the load increases, but when the maximum load power Pomax = 200 W, the AC input voltage VAC = 100 V Is 89.0%, and 91.0% when the AC input voltage VAC = 230V. That is, in this embodiment, it can be seen that the power conversion efficiency when the commercial AC power input level is high is particularly improved.

As can be understood from the above description, the power supply circuit according to the present embodiment shown in FIG. 1 can support a wide range only by switching frequency control.
As a result, for example, when the wide range is supported, the rectification operation is switched for the rectifier circuit system for generating the DC input voltage (Ei) according to the rated level of the commercial AC power supply, or the half bridge coupling method and the full bridge There is no need to adopt a configuration for switching the type of the switching converter with the coupling method.
If such a circuit switching configuration is not necessary, for example, only one smoothing capacitor Ci can be used, and at least two switching elements required for half-bridge coupling can be used. Accordingly, it is possible to reduce the number of circuit components, the circuit scale, and the switching noise.
Further, if the circuit switching configuration is not required, it is not necessary to provide a special configuration for preventing malfunction due to switching, and in this respect also, the increase in the number of components and the suppression of the cost increase can be achieved. Furthermore, since a standby power supply is not essential to prevent malfunctions, the range of devices in which the power supply circuit can be used can be expanded.

  Further, in order to obtain the effect as such an embodiment, the minimum necessary component to be added to the configuration of the current resonance type converter having the series resonance circuit only on the primary side is the secondary side. There are only two points: a series resonant capacitor and a high-frequency inductor L11. That is, it is possible to realize a wide range with much fewer parts than in the case of adopting the configuration based on the conventional circuit switching method. Incidentally, since the high-frequency inductor L11 only needs to be about several tens of μH, the size can be kept small, and the power loss due to the resistance is almost negligible.

In addition, since the required control range Δfs of the switching frequency is greatly reduced as described above, the responsiveness of the constant voltage control is greatly increased regardless of whether it is compatible with a wide range or a single range. Will be improved.
That is, some electronic devices perform an operation of changing the load power Po so as to be switched (switched) at a relatively high speed between a maximum and no load, which is called a so-called switching load. An example of a device that performs such an operation as a switching load is a printer that is a peripheral device of a personal computer.
When a power supply circuit having a relatively wide necessary control range Δfs as shown in FIG. 20 is mounted on a device that operates as such a switching load, for example, as described above, the device has a steep Following the change in the load power, the switching frequency fs is variably controlled by a correspondingly large amount of change. For this reason, it has been difficult to obtain high-speed constant voltage control response.
On the other hand, in the present embodiment, since the necessary control range Δfs is greatly reduced particularly in the region for each single range, with respect to the steep fluctuation between the maximum load power Po and no load, It is possible to stabilize the secondary side DC voltage Eo in response to high speed. That is, the response performance of the constant voltage control with respect to the switching load is greatly improved.

The circuit diagram of FIG. 8 shows a configuration example of a power supply circuit according to the second embodiment of the present invention. In this figure, the same parts as those in FIG.
In the power supply circuit shown in this figure, the configuration of the primary side current resonance type converter is a full bridge coupling system including four switching elements Q1 to Q4.
As shown in the figure, the full bridge coupling method is configured such that the half bridge connection of the switching elements Q3 and Q4 is connected in parallel to the half bridge connection of the switching elements Q1 and Q2.
As for the switching elements Q3 and Q4, similarly to the switching elements Q1 and Q2, a damper diode DD3 and a damper diode DD4, which are body diodes, are connected in parallel to the drain-source.

In addition, in this case, a primary side series resonance circuit comprising a series connection of the primary winding N1 of the insulating converter transformer PIT, the high frequency inductor L11, and the primary side series resonance capacitor C1 is connected as follows.
First, one end (winding start end) of the primary winding N1, which is one end of the primary side series resonance circuit, is connected to a connection point between the source of the switching element Q1 and the drain of the switching element Q2. The connection point between the source of the switching element Q1 and the drain of the switching element Q2 is one switching output point in a full-bridge coupled switching circuit system.
For the other end of the primary side series resonance circuit, the other end (winding end) of the primary winding N1 is connected to the other end via a series connection of the high frequency inductor L11 and the primary side series resonance capacitor C1. The switching output point is connected to a connection point between the source of the switching element Q3 and the drain of the switching element Q4.

  In this case, the primary-side partial resonance capacitor Cp1 is connected in parallel with the source and drain of the switching element Q4. As this primary side partial resonance capacitor Cp1, a parallel resonance circuit (partial voltage resonance circuit) is formed by its own capacitance and the leakage inductance L1 of the primary winding N1 (and the inductance of the high frequency inductor L11), and switching elements Q3, Q4 A partial voltage resonance operation in which voltage resonance occurs only at the turn-off time is obtained.

  The oscillation / drive circuit 2 in this case is configured to drive four switching elements Q1 to Q4. Depending on the oscillation / drive circuit 2, switching driving is performed such that the group of switching elements [Q1, Q4] and the group of switching elements [Q3, Q4] are alternately turned on / off.

The reason why the full-bridge coupling method is used for the configuration of the primary side current resonance type converter as in the second embodiment is to cope with the heavy load condition. As the load becomes heavier, the current flowing through the switching converter increases, the burden on the circuit components increases, and the power loss also decreases.
Therefore, if full-bridge coupling is used, the necessary load current is provided by four switching elements. Therefore, the burden on each component is lighter than in the case of a half-bridge coupling system composed of two switching elements, for example. In addition, power loss is reduced, which is advantageous for heavy load conditions. For example, by adopting the configuration shown in FIG. 8, it is possible to obtain a wide-range power supply circuit that can handle a maximum load power (Pomax) of about 300 W or more.

  The circuit diagram of FIG. 9 shows a configuration example of a power supply circuit as the third embodiment. In this figure, the same parts as those in FIG. 1 and FIG. Further, the insulating converter transformer PIT itself has a structure similar to that described with reference to FIG. 2, for example, so that a coupling coefficient k = about 0.75 is set.

In this figure, the high-frequency inductor L11 connected in series with the primary winding N1 in the first and second embodiments is omitted, and instead, the high-frequency inductor in series with the secondary winding N2 is omitted. A form in which L12 is connected is shown.
In this case, the high frequency inductor L12 is inserted in series between the secondary winding N2 and the secondary side series resonance capacitor C2.

When such a connection form is adopted, first, the primary side series resonance circuit is formed by the leakage inductance L1 of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1.
On the other hand, the secondary side series resonant circuit is formed by the leakage inductance L2 of the secondary winding N2, the inductance of the high frequency inductor L12, and the capacitance of the secondary side series resonant capacitor C2.

When such a circuit configuration is adopted, the apparent leakage inductance on the secondary winding side in the insulating converter transformer PIT increases due to the inductance of the high-frequency inductor L12. Even if the leakage inductance on the secondary side of the insulating converter transformer PIT is increased in this way, the total coupling coefficient kt of the insulating converter transformer PIT is lower than the coupling coefficient k of the insulating converter transformer PIT itself.
That is, in place of the primary winding N1, a circuit configuration in which a high-frequency inductor is connected in series to the secondary winding N2, the coupling degree of the insulating converter transformer PIT in the power supply circuit is shown in FIGS. It is possible to reduce the required control range of the switching frequency by setting a loosely coupled state such that the unimodal characteristics as described can be obtained. Therefore, in the third embodiment as well, The same effect as the embodiment is obtained.

The circuit diagram of FIG. 10 shows a configuration example of a power supply circuit as the fourth embodiment.
In this figure, the same parts as those in FIG. 1, FIG. 8, and FIG.
The insulation converter transformer PIT itself in the fourth embodiment also has a structure similar to that described with reference to FIG.
Further, the power supply circuit of the fourth embodiment also adopts a basic configuration as a multiple composite resonance type converter in the same manner as each of the previous embodiments. Therefore, the secondary winding N2 and the second winding A secondary series resonance circuit including a secondary resonance capacitor is provided. In addition, in order to set the total coupling coefficient kt of the insulating converter transformer PIT to kt = 0.65 or less, the primary circuit is the same as in the power supply circuits of the first and second embodiments shown in FIGS. A high frequency inductor L11 is connected in series with the winding N1.
In addition, the fourth embodiment includes a voltage doubler full wave rectifier circuit as a rectifier circuit connected to the secondary winding N2.

  In this voltage doubler full-wave rectifier circuit, first, a center tap is applied to the secondary winding N2 to divide it into secondary winding portions N2A and N2B. In this case, the center tap of the secondary winding N2 is grounded to the secondary side ground.

In this case, the end of the secondary winding N2A, which is the winding end of the secondary winding N2, is connected to the anode of the rectifier diode Do1 via the series connection of the secondary side series resonant capacitor C2A. The rectifier diode Do2 is connected to the connection point with the cathode.
The end of the secondary winding N2B, which is the winding start end of the secondary winding N2, is connected to the anode of the rectifier diode Do3 and the rectifier diode Do4 via a series connection of the secondary side series resonant capacitor C2B. Connected to the connection point of the cathode.
The node of the anodes of the rectifier diodes Do2 and Do4 is connected to the secondary side ground. The connection point of the cathodes of the rectifier diode Do1 and the rectifier diode Do2 is connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground.

In this case, two secondary side partial resonance capacitors Cp2A and Cp2B are provided, and the secondary side partial resonance capacitor Cp2A is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2. Insert between secondary grounds.
The secondary side partial resonance capacitor Cp2B is inserted between the connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4 and the secondary side ground.

The rectification operation of the voltage doubler full-wave rectifier circuit formed by the above connection mode is as follows.
First, the voltage doubler full-wave rectifier circuit includes a first voltage doubler formed by [secondary winding portion N2A, secondary side series resonant capacitor C2A, rectifier diodes Do1, Do2, and secondary side partial resonant capacitor Cp2A]. A second half-voltage rectifier circuit formed by a half-wave rectifier circuit and [secondary winding portion N2B, secondary-side series resonant capacitor C2B, rectifier diodes Do3, Do4, secondary-side partial resonant capacitor Cp2B] Can be divided into
In addition, in the first voltage doubler half-wave rectifier circuit, a leakage current component of the secondary winding portion N2A is formed by forming a series connection circuit of the secondary winding portion N2A and the secondary side series resonance capacitor C2A. A first secondary side series resonance circuit is formed by (L2A) and the capacitance of the secondary side series resonance capacitor C2A.
Similarly, in the second voltage doubler half-wave rectifier circuit, a leakage current component of the secondary winding portion N2B is formed by forming a series connection circuit of the secondary winding portion N2B-secondary side series resonance capacitor C2B. (L2B) and the capacitance of the secondary side series resonance capacitor C2B form a second secondary side series resonance circuit.

  The secondary side partial resonance capacitor Cp2A inserted as described above is connected in parallel to the series connection of the secondary winding part N2A and the secondary side series resonance capacitor C2A. Therefore, the secondary side partial resonance capacitor Cp2A forms a secondary side partial voltage resonance circuit corresponding to the first voltage doubler half-wave rectifier circuit by its own capacitance and the leakage inductance L2A of the secondary winding N2A. . Similarly, the secondary side partial resonance capacitor Cp2B is connected in parallel to the series connection of the secondary winding portion N2B-secondary side series resonance capacitor C2B, and its own capacitance and the secondary winding N2A. The secondary side partial voltage resonance circuit corresponding to the second voltage doubler half-wave rectifier circuit is formed by the leakage inductance L2A.

The rectifying operation of the first voltage doubler half-wave rectifier circuit is as follows.
First, in one half cycle period of the alternating voltage induced in the secondary winding N2, a rectified current flows through the path of the secondary winding portion N2A-rectifier diode Do2-secondary series resonance capacitor C2A. The secondary side series resonant capacitor C2A is charged with a rectified current. By this rectifying operation, a voltage across the level corresponding to the same voltage as the alternating voltage induced in the secondary winding portion N2A is generated in the secondary side series resonance capacitor C2A.
Further, in the subsequent half-cycle period of the alternating voltage of the secondary winding N2, a rectified current flows through the path of the secondary winding portion N2A-secondary side series resonant capacitor C2A-rectifier diode Do1-smoothing capacitor Co. . At this time, the voltage across the secondary side series resonant capacitor C2A obtained by the rectification operation during the half cycle of the alternating voltage of the previous secondary winding N2 with respect to the induced voltage of the secondary winding N2A is In the superposed state, the smoothing capacitor Co is charged. As a result, a voltage across the level that is twice the alternating voltage of the secondary winding N2A is generated in the smoothing capacitor Co.
In other words, the first voltage half-wave rectifier circuit has a secondary side at a level corresponding to an equal multiple of the alternating voltage of the secondary winding N2A in one half cycle of the alternating voltage of the secondary winding N2A. A voltage across the series resonant capacitor C2A is generated, and the alternating voltage of the secondary winding N2A and the voltage across the secondary side series resonant capacitor C2A are generated during the other half cycle of the alternating voltage of the secondary winding N2A. Double voltage half-wave rectification operation in which the smoothing capacitor Co is charged at the superposition level to obtain a voltage at both ends corresponding to twice the alternating voltage of the secondary winding N2A as the voltage across the smoothing capacitor Co. I do.
Further, in the above-described voltage doubler half-wave rectification operation, a current flows through the secondary side series resonant capacitor C2A every half cycle in both positive and negative pole directions. The secondary side series resonance circuit performs a resonance operation.

  Further, the partial voltage generated by the secondary partial voltage resonance circuit including the secondary partial resonance capacitor Cp2A according to the timing when the rectification diodes Do1 and Do2 are turned off in accordance with the rectification operation of the first voltage doubler half-wave rectification circuit. Resonant operation is obtained.

  The second voltage doubler half-wave rectifier circuit is similar to the first voltage doubler half-wave rectifier circuit by the [secondary winding section N2B, secondary side series resonant capacitor C2B, rectifier diodes Do3, Do4]. The voltage doubler half-wave rectification operation is executed at a cycle timing just shifted by a half cycle with respect to the rectification operation of the first voltage doubler half-wave rectifier circuit. Further, by this rectification operation, the second secondary side series resonance circuit obtains a resonance operation. Further, with this rectification operation, a partial voltage resonance operation by the secondary side partial voltage resonance circuit including the secondary side partial resonance capacitor Cp2B is obtained according to the timing when the rectification diodes Do3 and Do4 are turned off.

  By executing such a rectification operation, the smoothing capacitor Co is charged by the first voltage doubler half-wave rectifier circuit and the second voltage doubler half-wave rectifier circuit by the secondary winding. This is repeatedly executed every half cycle of the alternating voltage of the line N2. That is, in the entire rectifier circuit connected to the secondary winding N2, the alternating voltage of the secondary winding N2 is applied by the charging potential corresponding to twice the alternating voltage induced in the secondary winding portions N2A and N2B. Is performing a voltage doubler full wave rectification operation in which the smoothing capacitor Co is charged in each positive / negative half-wave period. By this rectification operation, the smoothing capacitor Co is provided with the secondary side DC output voltage Eo, which is a rectification smoothing voltage corresponding to twice the alternating voltage induced in the secondary winding portions N2A and N2B.

  Note that the power supply circuit of the fourth embodiment shown in FIG. 10 has the same AC input voltage conditions, load conditions, and secondary side DC output voltage Eo as those of the power supply circuit of the first embodiment shown in FIG. In the case of a specification with a set level, the number of turns of each of the secondary winding portions N2A and N2B is set to 14T, which is ½ of the number of turns of the secondary winding N2 of the power supply circuit of FIG. it can. The selection of other parts may be the same as in the case of obtaining the experimental results of FIGS. 6 and 7 for the power supply circuit of the first embodiment.

Further, as a result of experiments on the power supply circuit of the fourth embodiment shown in FIG. 10, the same operation waveforms as those shown in FIG. 6 were obtained.
Further, the change characteristic of the switching frequency fs with respect to the load change is a necessary control range of 70.5 kHz to 68.2 kHz when the AC input voltage VAC is 100 V with respect to the change of the load power Po = 0 W to 200 W. Δfs1 shown in FIG. 5 is Δfs1 = 2.3 kHz.
Further, when the AC input voltage VAC = 230 V, the required control range is 101.3 kHz to 93.2 kHz, and Δfs2 shown in FIG. 5 is Δfs2 = 8.1 kHz. Also in this embodiment, the necessary control range of the switching frequency fs for each range in the AC100V system and the AC200V system is less than 9 kHz, and the current switching element driving IC (oscillation / drive circuit 2) The switching drive frequency is sufficiently lower than the upper limit of 200 kHz.
In addition, regarding AC → DC power conversion efficiency (ηAC → DC), ηAC → DC = 88.8% and AC input voltage VAC = 230V under the load condition of maximum load power Pomax = 200 W when AC input voltage VAC = 100V. In some cases, a measurement result of ηAC → DC = 91.5% was obtained. From this result, even in the present embodiment, the effect of improving the AC → DC power conversion efficiency under the condition that the input level of the commercial AC power supply is high is sufficiently obtained.

FIG. 11 shows a configuration example of a power supply circuit according to the fifth embodiment. In this figure, the same parts as those in FIG. 1 and FIGS.
The insulating converter transformer PIT itself in the fifth embodiment also has a structure similar to that described with reference to FIG. 2, so that a coupling coefficient k = 0.75 is set.
The power supply circuit of the fifth embodiment also adopts a basic configuration as a multiple composite resonance type converter in the same manner as the previous embodiments, and includes a secondary winding N2, a secondary side resonance capacitor, A secondary side series resonant circuit.

In this case, when setting the total coupling coefficient kt = 0.65 or less, a high-frequency inductor is provided on the secondary side in the same manner as in the third embodiment shown in FIG.
In this case, since the secondary side rectifier circuit is a voltage doubler full wave rectifier circuit as in the fourth embodiment of FIG. 10, the secondary side high frequency inductor is the first voltage doubler. Two high-frequency inductors L12A and L12B are provided corresponding to each of the half-wave rectifier circuit and the second voltage doubler half-wave rectifier circuit.
The high frequency inductor L12A is inserted in series between the secondary winding portion N2A and the secondary side series resonant capacitor C2A, so that it is connected in series with the secondary winding portion N2A in the first voltage doubler half-wave rectifier circuit. To have a relationship. Similarly, the high frequency inductor L12B is inserted in series between the secondary winding portion N2B and the secondary side series resonant capacitor C2B, so that the secondary winding portion in the second voltage doubler half-wave rectifier circuit is provided. N2B is connected in series.

  By providing the high-frequency inductors L12A and L12B as described above, the apparent leakage inductance of the secondary winding portions N2A and N2B increases, and the total coupling coefficient kt of the insulating converter transformer PIT is kt = A loosely coupled state of about 0.65 is obtained.

In the secondary side voltage doubler full-wave rectifier circuit shown in this figure, the secondary side partial resonance capacitor is only one of the secondary side partial resonance capacitors Cp2, and this secondary side partial resonance capacitor Cp2 is used. Are inserted between the connection point of the anode of the rectification diode Do1 and the cathode of the rectification diode Do2, and the connection point of the anode of the rectification diode Do3 and the cathode of the rectification diode Do4.
By adopting such a configuration, the secondary side partial voltage resonance capacitor Cp2 has the first voltage doubler half-wave rectifier circuit and the second capacitance as capacitance for forming the secondary side partial voltage resonance circuit (parallel resonance circuit). It is shared by the double voltage half-wave rectifier circuit of 2.
In other words, the secondary side partial resonance capacitor Cp2 is the first in the first voltage doubler half-wave rectifier circuit. A partial voltage resonance circuit is formed by the secondary side leakage inductance component. Similarly, in the second voltage doubler half-wave rectifier circuit, due to the secondary side leakage inductance component of the insulating converter transformer PIT obtained by combining the leakage inductance L2B of the secondary winding portion N2B and the inductance of the high frequency inductor L12B, A partial voltage resonance circuit is formed.
Thus, a partial voltage resonance operation can be appropriately obtained at the timing when the rectifier diodes Do1 and Do4 are turned off and the timing when the rectifier diodes Do2 and Do3 are turned off.

The circuit diagram of FIG. 12 shows a configuration example of a power supply circuit as a sixth embodiment. In this figure, the same parts as those in FIG. 1 and FIGS.
The insulation converter transformer PIT itself in the sixth embodiment also has the same structure as that described with reference to FIG. 2, so that a coupling coefficient k = 0.75 is set.
The power supply circuit of the sixth embodiment also adopts a basic configuration as a multiple composite resonance type converter, as in the previous embodiments, and includes a secondary winding N2, a secondary side resonance capacitor, A secondary side series resonant circuit. Further, in order to set the total coupling coefficient kt of the insulating converter transformer PIT to kt = 0.65 or less, the primary winding is similar to the power supply circuits of the first and second embodiments shown in FIGS. A high frequency inductor L11 is connected in series with N1.

In addition, the sixth embodiment includes a voltage doubler half-wave rectifier circuit as a rectifier circuit connected to the secondary winding N2.
As the voltage doubler half-wave rectifier circuit in this case, first, the anode of the rectifier diode Do1 is connected to one end of the secondary winding N2 through the series connection of the secondary side series resonant capacitor C2. . The cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground.
The other end of the secondary winding N2 is grounded to the secondary side ground, and is also connected to the anode of the rectifier diode Do2. The cathode of the rectifier diode Do2 is connected to the connection point between the anode of the rectifier diode Do1 and the secondary side series resonant capacitor C2A.
Also in this case, the secondary side partial resonance capacitor Cp2 is connected in parallel to the series connection circuit of the secondary winding N2 and the secondary side series resonance capacitor C2 forming the secondary side series resonance circuit. The secondary side partial voltage resonance circuit is formed together with the leakage inductance L2 of the secondary winding N2.

  The rectifying operation of the above-described voltage doubler half-wave rectifier circuit is the same as that of the first or second voltage doubler half-wave rectifier circuit described with reference to FIG. 10 or FIG. 11, for example, and detailed description thereof is omitted here. . In addition, the secondary side partial voltage resonance circuit including the secondary side partial resonance capacitor Cp2 performs a partial voltage resonance operation according to the timing when the rectifier diodes Do1 and Do2 are turned off.

  For the power supply circuit of the sixth embodiment, the same AC input voltage condition, load condition, and level of the secondary side DC output voltage Eo as the power supply circuit of the first embodiment shown in FIG. 1 are set. In the case of the specification to be performed, the number of turns of the secondary winding portion N2 can be set to 14T that is ½ of that of the first embodiment. For example, the secondary side rectifier circuit is a normal full-wave rectifier. The number of turns as the secondary winding N2 is reduced as compared with the case of the circuit or the voltage doubler full wave rectifier circuit. In this case, for example, 0.39 μF is selected for the secondary side series resonance capacitor C2. The selection of other parts may be the same as in the case of obtaining the experimental results of FIGS. 6 and 7 for the power supply circuit of the first embodiment.

Also, as a result of experiments on the power supply circuit of the sixth embodiment shown in FIG. 12, the same operation waveforms as those in FIG. 6 were obtained. However, since the secondary side is a voltage doubler half-wave rectification operation, the current I2 (rectified current) flowing through the secondary winding N2 has a peak level almost twice as high.
Further, as a change characteristic of the switching frequency fs with respect to the load change, the required control range is 68.1 kHz to 65.2 kHz when the AC input voltage VAC is 100 V with respect to the change of the load power Po = 0 W to 200 W. Δfs1 shown in FIG. 5 is Δfs1 = 2.9 kHz.
Further, when the AC input voltage VAC = 230 V, the required control range is 98.7 kHz to 90.2 kHz, and Δfs2 shown in FIG. 5 is Δfs2 = 8.5 kHz. Also in this embodiment, the necessary control range of the switching frequency fs for each range in the AC100V system and the AC200V system is less than 9 kHz, and the current switching element driving IC (oscillation / drive circuit 2) The switching drive frequency is sufficiently lower than the upper limit of 200 kHz.
As for AC → DC power conversion efficiency (ηAC → DC), ηAC → DC = 89.0% and AC input voltage VAC = 230V when the AC input voltage VAC = 100V under the load condition of maximum load power Pomax = 200W. In some cases, a measurement result of ηAC → DC = 91.7% was obtained, and the effect of improving the AC → DC power conversion efficiency under the condition that the input level of the commercial AC power supply is high, as in the previous embodiments. Is sufficiently obtained.

The circuit diagram of FIG. 13 shows a configuration example of a power supply circuit as the seventh embodiment.
In this figure, the same parts as those in FIG. 1 and FIGS.
The insulating converter transformer PIT itself in the fifth embodiment also has a structure similar to that described with reference to FIG. 2, so that a coupling coefficient k = 0.75 is set.
The power supply circuit of the fifth embodiment also adopts a basic configuration as a multiple composite resonance type converter in the same manner as the previous embodiments, and includes a secondary winding N2, a secondary side resonance capacitor, A secondary side series resonant circuit.
Further, the secondary side rectifier circuit includes a voltage doubler half-wave rectifier circuit as in the sixth embodiment shown in FIG. In addition, instead of connecting a high-frequency inductor in series with the primary winding N1, a high-frequency inductor L12 is connected in series with the secondary winding N2.
In this case, the high frequency inductor L12 is inserted in series between one end of the secondary winding N2 and the secondary side series resonance capacitor C2 as shown in the figure. By inserting the high-frequency inductor L12 in this way, the apparent leakage inductance on the secondary winding N2 side is increased, and the total coupling coefficient kt of the insulating converter transformer PIT is kt = 0.65 or less. It is possible to obtain a loosely coupled state with a predetermined value.

  In the fourth to seventh embodiments of FIGS. 10 to 13, the secondary side rectifier circuit is configured to include a voltage doubler full wave rectifier circuit or a voltage doubler half wave rectifier circuit. In these embodiments as well, as in the second embodiment shown in FIG. 8, it is possible to adopt a configuration in which the primary-side current resonance type converter is a full-bridge coupling method. Thereby, similarly to the case of FIG. 8, a power supply circuit corresponding to, for example, the maximum load power Pomax = about 300 W or more can be obtained.

Subsequently, an eighth embodiment of the present invention will be described with reference to FIGS.
First, as the circuit configuration of the power supply circuit as the eighth embodiment, for example, the high frequency inductors (L11, L12, L12A) from the configurations of the embodiments shown in FIG. 1 and FIGS. , L12B) can be omitted. That is, in terms of a circuit diagram, a basic configuration as a multiple resonance type converter is sufficient.
In addition, in the eighth embodiment, as the insulating converter transformer PIT, for example, the structure shown in the sectional view of FIG. 14 is adopted.
Also in the insulating converter transformer PIT shown in FIG. 14, as in FIG. 2, for example, E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other to form an EE-type core (EE-shaped core). To be done.
Also in this case, the primary winding N1 and the secondary winding N2 are wound around different winding portions in the bobbin B, respectively. The bobbin B in this case is also formed of a resin or the like, for example, in a shape in which the primary side and secondary side winding portions are divided so as to be independent from each other. Then, the bobbin B around which the primary winding N1 and the secondary winding N2 are wound is attached to the EE type core (CR1, CR2). As a result, the primary winding N1 and the secondary winding N2 are wound around the inner magnetic legs of the EE core by different winding regions.

  Further, also in this insulating converter transformer PIT, a gap G is formed as shown in the figure with respect to the inner magnetic leg of the EE type core. As the gap length Ln1 of the gap G in this case, for example, a gap length of about 1.6 mm is set, and in a form in which the ferrite sea cores FSC1 and FSC2 described below are not inserted, the coupling coefficient k of the insulating converter transformer PIT itself As a structure, k = about 0.75 is obtained.

As the insulating converter transformer PIT shown in FIG. 14, the center portion of the two outer magnetic legs of the EE core, that is, the winding position of the primary winding N1 and the winding position of the secondary winding N2 are provided. The ferrite sheet cores FSC1 and FSC2 are provided so as to be sandwiched between the portions corresponding to the boundaries. These ferrite sheet cores FSC1 and FSC2 each have a predetermined thickness Ln2.
Moreover, the ferrite sheet cores FSC1 and FSC2 in this case are made of the same ferrite as the E-shaped cores CR1 and CR2, as can be seen from their names, and are provided in a state of being sandwiched between outer magnetic legs. In addition, the ferrite sheet cores FSC1 and FSC2 further have a protruding portion (magnetic path generating portion) with a predetermined length Ln3 from the inner end surface portion of the outer magnetic leg to the inner magnetic leg side of the EE core. Provided. As a result, the end portions of the ferrite sheet cores FSC1 and FSC2 are inserted into the portion between the winding portions of the primary winding N1 and the secondary winding N2 in the bobbin B by a predetermined length. Become.

Here, for example, in the case of a simple EE type core structure in which the ferrite sheet cores FSC1 and FSC2 are not inserted as in the insulating converter transformer PIT shown in FIG. 2, the magnetic flux φ1 in FIG. , Φ 2, a magnetic path is formed. In FIG. 15, the bobbin B is not shown for easy understanding.
As can be seen from this figure, the magnetic paths of the magnetic fluxes φ1 and φ2 pass through the outer magnetic legs so as to straddle the primary winding N1 side and the secondary winding N2 side. For this reason, the degree of coupling between the primary winding N1 and the secondary winding N2 is reasonably high, and the coupling coefficient k of the insulating converter transformer PIT itself described above is about 0.75. This was obtained by forming a gap G of about 1.6 mm on the inner magnetic leg.

On the other hand, when the ferrite sheet cores FSC1 and FSC2 are provided as shown in FIG. 14, the E-shaped cores CR1 and CR2 are ferrites of the same material as the ferrite sheet cores FSC1 and FSC2. In essence, as shown in FIG. 15 (b), the shape of the EE core is assumed to have a shape in which the center of the outer magnetic leg protrudes toward the center of the inner magnetic leg. Can do. The shape of such an EE core is different from that of the EE core in FIG. 15A by the length Ln3 of the protruding portion (magnetic path generation site) and the central portion of the outer magnetic leg. The spatial distance from the center part of the magnetic leg is shortened.
As these portions approach, the insulating converter transformer PIT shown in FIG. 15B generates magnetic fluxes indicated by φ11 and φ12 indicated by broken lines in FIG. The components of the magnetic fluxes φ11 and φ12 increase as the length Ln3 of the protruding portion of the ferrite sheet cores FSC1 and FSC2 becomes longer and the spatial distance between the central portion of the outer magnetic leg and the central portion of the inner magnetic leg becomes shorter. However, the component amount of one of the magnetic fluxes φ11 and φ12 decreases.

  Here, the magnetic paths of the magnetic fluxes φ11 and φ12 are respectively formed corresponding to the primary winding N1 side and the secondary winding N2 side. Accordingly, when the component amounts of the magnetic fluxes φ11 and φ12 are decreased, the magnetic fluxes φ11 and φ12 tend to increase, which reduces the degree of coupling between the primary winding N1 and the secondary winding N2. As a result, the total coupling coefficient kt increases. As the total coupling coefficient kt increases, equivalently, the leakage inductance value on the primary winding N1 side and the secondary winding N2 side of the insulating converter transformer PIT is a value determined by the gap G of the inner magnetic leg. Will be more than.

FIGS. 16 to 18 show equivalent circuit diagrams of the power supply circuit including the insulating converter transformer PIT shown in FIG.
First, FIG. 16 shows an equivalent circuit in the case of a power supply circuit including a full-wave rectifier circuit as a secondary side rectifier circuit, as shown in FIGS. 1, 8, 9 and the like. In this figure, the same parts as those in FIGS. 1, 8, and 9 are denoted by the same reference numerals.
The insulating converter transformer PIT shown in this figure is treated as having the leakage inductance L1 of the primary winding N1 and the leakage inductance L2 of the secondary winding N2 determined by the gap G as a factor. This also applies to FIGS. 17 and 18 described later.
In addition, with respect to the increase in equivalent leakage inductance due to the provision of the ferrite sheet cores FSC1 and FSC2 in the insulating converter transformer PIT, the inductor L1l connected in series with the primary winding N1 and the secondary winding N2 are connected in series. Can be represented as an inductor L2l. That is, it can be seen as a circuit configuration including the primary-side high-frequency inductor L11 and the secondary-side high-frequency inductor L12 shown in the previous embodiment.

FIG. 17 shows an equivalent circuit in the case of a power supply circuit including a voltage doubler full-wave rectifier circuit for the secondary side rectifier circuit shown in FIGS. 10 and 11, for example. In this figure, the same parts as those in FIGS. 10 and 11 are denoted by the same reference numerals. Further, in this figure, the same voltage doubler full wave rectifier circuit as shown in FIG. 11 is shown, but a similar equivalent circuit is formed even in the voltage doubler full wave rectifier circuit shown in FIG. It is what is done.
Also in this case, when the insulating converter transformer PIT having the structure shown in FIG. 14 is provided, the equivalent leakage inductance increase due to the provision of the ferrite sheet cores FSC1 and FSC2 is connected in series with the next winding N1. And inductors L2la and L2lb connected in series to the secondary windings N2A and N2B, respectively. Therefore, in this case, it can be considered that the circuit configuration includes the high-frequency inductors L12A and L12B corresponding to the primary-side high-frequency inductor L11 and the secondary-side voltage doubler full-wave rectifier circuit.

FIG. 18 shows an equivalent circuit in the case of a power supply circuit including a voltage doubler half wave rectifier circuit for the secondary side rectifier circuit shown in FIGS. In this figure, the same parts as those in FIGS. 12 and 13 are denoted by the same reference numerals.
In this case, when the insulating converter transformer PIT having the structure shown in FIG. 14 is provided, the increase in equivalent leakage inductance due to the provision of the ferrite sheet cores FSC1 and FSC2 is connected in series with the next winding N1. The inductor L1l and the secondary winding N2 can be expressed as an inductor L2l connected in series. In this case as well, the high-frequency inductor L12 corresponding to the primary-side high-frequency inductor L11 and the secondary-side voltage doubler half-wave rectifier circuit It can be considered that the circuit configuration is provided with.

Based on the formation of the equivalent circuits shown in FIGS. 16 to 18, as the eighth embodiment, for example, about kt = 0.65 or less is set for the total coupling coefficient kt of the insulating converter transformer PIT. As described above, the inductances of the primary side inductor L1l and the secondary side inductor L2l (L2la, L2lb) are set. These inductances can be set mainly by the length Ln3 (and thickness Ln2) of the protruding portion (magnetic path generating portion) of the ferrite sheet cores FSC1 and FSC2.
In this way, by setting the total coupling coefficient kt of the insulating converter transformer PIT to about kt = 0.65 or less, the primary-side high-frequency inductor L11 or the secondary-side high-frequency inductor that has been described so far Similar to the power supply circuit of the embodiment having L12 (L12A, L12B), the single-side characteristics described with reference to FIGS. 4 and 5 can be obtained as the secondary-side DC voltage control characteristics, and the wide range This is possible.

  For confirmation, in the structure of the isolated converter transformer PIT according to the eighth embodiment, as described with reference to FIG. The gap length (Ln1) is about 1.6 mm in the same manner as the first embodiment. In addition, depending on the ferrite sheet cores FSC1 and FSC2 provided on the outer magnetic legs, in actuality, only a change in the magnetic flux state occurs, and even though there is an increase in the equivalent leakage inductance, the actual increase in the leakage inductance. There is no. Therefore, even in the insulating converter transformer PIT, there is no problem of an increase in eddy current loss in the vicinity of the inner magnetic leg.

Further, as understood from the above description, the total coupling coefficient kt may be such that the part having the length Ln3 as the magnetic path generation part is formed with respect to the basic shape of the EE type core. Instead of using a ferrite sheet core, for example, an E-shaped core may be molded and combined so that an actual core shape as shown in FIG. 15B can be obtained.
However, in the present embodiment, the ferrite sheet cores (FSC1 and FSC2 are connected to the outer magnetic legs in consideration of the fact that the manufacturing process becomes simpler than molding the E-shaped core as described above. It is made into the form inserted | pinched between.

FIG. 19 shows the configuration of a power supply circuit as the ninth embodiment. In this figure, only the rectifying / smoothing circuit system for generating the rectified and smoothed voltage Ei (DC input voltage) by inputting the commercial AC power supply AC is shown, but the subsequent stages are described above. Any configuration of the embodiment may be adopted.
The rectifier circuit system shown in FIG. 19 is a voltage doubler rectifier circuit. This voltage doubler rectifier circuit comprises two rectifier diodes DA and DB and two smoothing capacitors Ci1 and Ci2.
The anode of the rectifier diode DA and the cathode of the rectifier diode DB are connected to the positive line of the commercial AC power supply AC that is the subsequent stage of the common mode noise filter (CMC, CL, CL). The cathode of the rectifier diode DA is connected to the positive terminal of the smoothing capacitor Ci1, and the anode of the rectifier diode DB is connected to the primary side ground.
The smoothing capacitors Ci1, Ci2 are connected in series so that the negative terminal of the smoothing capacitor Ci1 and the positive terminal of the smoothing capacitor Ci2 are connected, and the positive terminal of the smoothing capacitor Ci1 is a rectifier diode as described above. Connected to the cathode of DA. The negative terminal of the smoothing capacitor Ci2 is connected to the primary side ground. Further, the connection point of the smoothing capacitors Ci1-Ci2 is connected to the negative line of the commercial AC power supply AC, which is the subsequent stage of the common mode noise filter (CMC, CL, CL).

In the voltage doubler rectifier circuit thus formed, during the half-wave period in which the commercial AC power supply AC (AC input voltage VAC) is positive, the rectifier diode DA is turned on to charge the smoothing capacitor Ci1 with the rectified current. Operation is obtained. As a result, as the voltage across the smoothing capacitor Ci1, a DC voltage (rectified smoothing voltage) at a level corresponding to the same magnification as the commercial AC power supply AC (AC input voltage VAC) is obtained.
Further, during the half-wave period in which the AC input voltage VAC is negative, the operation of charging the rectified current to the smoothing capacitor Ci2 by turning on the rectifier diode DB is obtained, so that the voltage across the smoothing capacitor Ci2 can be obtained. As a result, a DC voltage (rectified smoothing voltage) at a level corresponding to the same size as the commercial AC power supply AC (AC input voltage VAC) can be obtained.
As a result, a rectified and smoothed voltage Ei corresponding to twice the commercial AC power supply AC is obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, the voltage doubler rectification operation by the voltage doubler rectifier circuit is performed. Then, the switching converter in the subsequent stage is configured to perform a switching operation by inputting the rectified and smoothed voltage Ei generated as described above as a DC input voltage.

In this way, the configuration in which the rectifying and smoothing circuit system that generates the DC input voltage (rectified and smoothing voltage Ei) is a voltage doubler rectifying circuit is a configuration corresponding to a single range of only the A100V system. For example, by applying the configuration of the ninth embodiment to a power supply circuit that is assumed to be used only in a single range of an AC 100V system, for example, the maximum load power Pomax = 300 W or more can be achieved under heavy load conditions. This is advantageous when responding.
In other words, the switching power supply circuit tends to increase the current flowing through the switching converter and increase power loss in response to a heavy load tendency. However, the voltage doubler rectifier circuit increases the rectified smoothed voltage Ei (DC input voltage). ) By making the level high, the amount of current flowing through the switching converter can be reduced under the same load condition. Thereby, the increase in the power loss accompanying a heavy load tendency is suppressed.
In this way, the ninth embodiment is compatible with a single range, but the constant voltage control characteristics are the same as those shown in FIGS. 4 and 5 as in the previous embodiments. Peak characteristics can be obtained, and therefore the response of constant voltage control is also improved.

The present invention should not be limited to the embodiments described so far.
For example, based on the equivalent circuit diagrams shown in FIGS. 16 to 18, for example, with respect to the insulating converter transformer PIT itself, the coupling coefficient k = 0.65 is set and the high-frequency inductors L11 and L12 having predetermined inductance values are set. The actual parts may be connected in series to the primary winding N1 and the secondary winding N2, respectively.
As for the insulating converter transformer PIT, as long as a necessary magnetic path is formed, the structure thereof including the core type may be appropriately changed.
In addition, the switching converter exemplified in the embodiment is based on a separately excited type current resonant converter, but may be configured to include, for example, a self excited type current resonant converter. In addition, as a switching element selected in the switching converter, for example, an element other than a MOS-FET such as a bipolar transistor or an IGBT (Insulated Gate Bipolar Transistor) may be employed.
Further, the constants of the component elements described above may be appropriately changed according to actual conditions and the like.

It is a circuit diagram showing an example of composition of a power circuit of a 1st embodiment of the present invention. It is sectional drawing which shows the structural example of the insulation converter transformer with which the switching power supply circuit of embodiment is provided. It is the equivalent circuit diagram which looked at the power supply circuit of embodiment as an electromagnetic coupling type resonance circuit. It is a figure which shows the constant voltage control characteristic about the power supply circuit of this Embodiment. It is a figure which shows the switching frequency control range (required control range) according to alternating current input voltage conditions and load fluctuation | variation as constant voltage control operation | movement of the power supply circuit of embodiment. FIG. 2 is a waveform diagram showing an operation of a main part in the power supply circuit shown in FIG. It is a figure which shows the characteristic of the switching frequency with respect to load fluctuation | variation, and AC-> DC power conversion efficiency about the power supply circuit shown in FIG. It is a circuit diagram which shows the structural example of the power supply circuit of 2nd Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit of 3rd Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit of 4th Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit of 5th Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit of 6th Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit of 7th Embodiment. It is sectional drawing which shows the structural example of the insulating converter transformer corresponding to 8th Embodiment. It is a figure explaining the magnetic path of the insulation converter transformer corresponding to 8th Embodiment. It is a circuit diagram which shows the equivalent circuit corresponding to the power supply circuit as 8th Embodiment. It is a circuit diagram which shows the equivalent circuit corresponding to the power supply circuit as 8th Embodiment. It is a circuit diagram which shows the equivalent circuit corresponding to the power supply circuit as 8th Embodiment. It is a circuit diagram which shows the structure as 9th Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art. FIG. 21 is a waveform diagram showing an operation of a main part in the power supply circuit shown in FIG. 20. It is a figure which shows the characteristic of the switching frequency with respect to load fluctuation | variation, and AC-> DC power conversion efficiency about the power supply circuit shown in FIG. It is a figure which shows the constant voltage control characteristic about the power supply circuit shown in FIG.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, Di bridge rectifier circuit, Ci, Ci1, Ci2 smoothing capacitor, Q1, Q2, Q3, Q4 switching element, PIT isolation converter transformer, C1 primary side series resonance capacitor, C2, C2A, C2B Secondary side series resonance capacitor, Cp primary side partial resonance capacitor, Cp2, Cp2A, Cp2B Secondary side partial resonance capacitor, N1 primary winding, N2 secondary winding, secondary winding part N2A, N2B, Do1 to Do4 ( Secondary side) Rectifier diode, Co (Secondary side) Smoothing capacitor, L11, L12, L12A, L12B High frequency inductor, FSC1, FSC2 Ferrite sheet core

Claims (5)

Switching means formed with a switching element for switching by inputting a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the primary winding are wound to form a primary side and a secondary side. An insulating converter transformer in which a gap length formed at a predetermined position of the core is set so that a predetermined coupling coefficient is obtained on the next side;
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
At least the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary side series resonance capacitor connected in series to the secondary winding are set in the primary side series resonance circuit. A secondary side series resonance circuit in which a predetermined second resonance frequency lower than the first resonance frequency is set;
A secondary side DC output voltage generating means for inputting a resonance output obtained by the secondary side series resonance circuit and performing a rectifying operation to generate a secondary side DC output voltage;
Constant voltage control means for performing constant voltage control on the secondary side DC output voltage by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. When,
An electromagnetic coupling type resonance circuit formed by including the primary side series resonance circuit and the secondary side series resonance circuit so that an output characteristic with respect to an input of a frequency signal having the switching frequency is a single peak characteristic. A total coupling coefficient between a predetermined primary side and a secondary side regarded as loose coupling is set so that at least one of the primary winding and the secondary winding is connected in series. And an inductor to be inserted
A switching power supply circuit.   The switching power supply circuit according to claim 1, wherein an inductance element as the inductor is connected in series to the primary winding.   The switching power supply circuit according to claim 1, wherein an inductance element as the inductor is connected in series to the secondary winding.   2. The switching power supply circuit according to claim 1, wherein a magnetic path generating portion for equivalently forming the inductor is provided at a predetermined position of the insulating converter transformer.   5. The magnetic path generating portion is a protruding portion formed so as to protrude toward the inner magnetic leg side at a predetermined position of the outer magnetic leg of the EE core constituting the insulating converter transformer. The switching power supply circuit described.

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