JP2006149016A - Switching regulator circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To realize an arrangement for wide range by reducing the required control range of switching frequency control in a power supply circuit performing constant voltage control by switching frequency control. <P>SOLUTION: In a current resonance converter, the secondary rectifier circuit is constituted of a circuit other than a both wave rectification circuit, the gap G being formed in the core of a power insulation transformer PIT is set at 2.0 mm or above, and the coupling coefficient k is set at about 0.70. Consequently, switching frequency variable control range can be reduced and an arrangement for wide range can be realized. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

JP 2003-235259 A

The present applicant has previously proposed various power supply circuits including a resonance type converter on the primary side.
FIG. 11 is a circuit diagram showing an example of a switching power supply circuit including a resonant converter configured based on the invention previously filed by the present applicant.
The switching converter of the power supply circuit shown in FIG. 11 has a configuration in which a partial voltage resonance circuit that performs a voltage resonance operation only at the time of turn-off during switching is combined with a separately-excited current resonance converter using a half-bridge coupling method. take.

First, in the power supply circuit shown in FIG. 11, a common mode noise filter including two sets of filter capacitors CL and CL and one set of common mode choke coil CMC is connected to the commercial AC power supply AC.
As a rectifying / smoothing circuit for generating a DC input voltage from the commercial AC power supply AC, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci is provided for the subsequent stage of the common mode noise filter.
The rectified output of the bridge rectifier circuit Di is charged to the smoothing capacitor Ci, whereby a rectified smoothing voltage Ei (DC input voltage) corresponding to an equal level of the AC input voltage VAC is applied to both ends of the smoothing capacitor Ci. Will be obtained.

  As shown in the figure, the current resonance type converter for switching by inputting the DC input voltage includes a switching circuit system in which two switching elements Q1 and Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 formed of body diodes are connected in parallel with each other between the drains and sources of the switching elements Q1 and Q2, respectively, in the direction shown in the drawing.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. With this partial voltage resonance circuit, a partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off can be obtained.

  In this power supply circuit, in order to switch the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and applies a drive signal (gate voltage) having a required frequency to the gates of the switching elements Q1 and Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side.
In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the source of the switching element Q1 and the drain of the switching element Q2 via the primary side series resonance capacitor C1. A switching output is obtained.
The other end of the primary winding N1 is connected to the primary side ground as shown in the figure.

  In this case, the series resonant capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonant capacitor C1 and the leakage inductance (leakage) of the primary winding N1 (series resonant winding) of the insulating converter transformer PIT. The primary side series resonance circuit for making the operation of the switching converter into a current resonance type is formed by the inductance L1.

According to the description so far, the primary side switching converter shown in this figure includes the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the partial voltage resonance circuit (Cp // L1) described above. Thus, partial voltage resonance operation due to is obtained.
That is, the power supply circuit shown in this figure adopts a form in which another resonance circuit is combined with a resonance circuit for making the primary side switching converter into a resonance type. Here, such a switching converter is referred to as a composite resonance type converter.

Although the description by illustration here is abbreviate | omitted, as a structure of the above-mentioned insulation converter transformer PIT, the EE type | mold core which combined the E type | mold core by a ferrite material is provided, for example. Then, after the winding part is divided between the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the inner magnetic leg of the EE core.
Further, a gap of 1.0 mm or less is formed with respect to the inner magnetic leg of the EE type core of the insulating converter transformer PIT, and the primary winding N1 and the secondary winding N2 are about 0.80 to 0.90. The coupling coefficient is obtained.
Actually, the gap G is about 0.8 mm, and the number of turns (number of turns) of the primary winding N1 and the secondary winding N2 is the primary winding N1 = 20T (turns), and the secondary winding N2 = 50T (25T + 25T). Thus, a coupling coefficient k = 0.85 is obtained.

The secondary winding N2 of the insulating converter transformer PIT is provided with a center tap connected to the secondary side ground as shown in the figure, so that the secondary winding N2A and the secondary winding N2B are connected. It is divided. In addition, a rectifier diode Do1 and a rectifier diode Do2 are connected in series to the secondary winding portion N2A and the secondary winding portion N2B, respectively, and the rectified output by the rectifier diodes Do1 and Do2 is smoothed. A double-wave rectifier circuit is formed by the smoothing capacitor Co.
As a result, a secondary-side DC output voltage Eo, which is a DC voltage at a level corresponding to the same voltage as the alternating voltage induced in each secondary winding, is obtained as the voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied as a main DC power source to a main load (not shown) and is also branched and input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 outputs a control signal as a voltage or current whose level is variable in accordance with the level of the secondary side DC output voltage Eo to the oscillation / drive circuit 2.
In the oscillation / drive circuit 2, the oscillation signal frequency generated by the oscillation circuit in the oscillation / drive circuit 2 is varied based on the control signal input from the control circuit 1. The frequency of the switching drive signal to be applied is changed. As a result, the switching frequency is varied. In this way, the switching frequency of the switching elements Q1, Q2 is variably controlled according to the level of the secondary side DC output voltage Eo, so that the resonance impedance of the primary side series resonance circuit changes and the primary side series resonance circuit is changed. The energy transmitted from the primary winding N1 to be formed to the secondary side is also varied, and the level of the secondary side DC output voltage Eo is also variably controlled. Thereby, constant voltage control of the secondary side DC output voltage Eo is achieved.
In the following, the constant voltage control method that achieves stabilization by variably controlling the switching frequency will be referred to as a “switching frequency control method”.

The waveform diagram of FIG. 12 shows the operation of the main part of the power supply circuit shown in FIG. In this figure, the operation when the load power Po = 200 W and the load power Po = 0 W in the circuit shown in FIG. 11 is shown. Note that the load power Po = 200 W is the maximum load power (Pomax) of the circuit shown in FIG. 11, and Po = 0 W is the minimum load power (Pomin).
In this figure, the input voltage condition is constant at AC input voltage VAC = 100V. Further, it is assumed that the secondary side DC output voltage Eo is 100 V or more (for example, Eo = 135 V in this case).

Further, corresponding to the above conditions of load power, input voltage, and secondary side DC output voltage level, the main part is selected as follows in the circuit of FIG.
・ Insulated converter transformer PIT ・ ・ ・ Gap length = 0.8mm, coupling coefficient k = 0.85 ・ Primary winding N1 = 20T
・ Secondary winding N2 = 50T (25T + 25T from the center tap)
・ Primary side series resonant capacitor C1 = 0.068μF
・ Partial resonance capacitor Cp = 1000pF

First, in FIG. 12, a rectangular wave voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2.
The period during which the voltage V1 is 0 level is the ON period in which the switching element Q2 is conductive. In this ON period, the switching current IQ2 having the waveform shown in FIG. Flowing. The period during which the voltage V1 is clamped at the level of the rectified and smoothed voltage Ei is a period during which the switching element Q2 is turned off, and the switching current IQ2 is at 0 level as shown in the figure.
Although not shown, the voltage across the other switching element Q1 and the switching current flowing through the switching circuit (Q1, DD1) are obtained as a waveform obtained by shifting the voltage V1 and the switching current IQ2 by 180 °. . That is, as described above, the switching element Q1 and the switching element Q2 perform the switching operation at the timing of turning on / off alternately.
The primary side series resonance current Io (not shown) flowing through the primary side series resonance circuit (C1-N1 (L1)) is the switching current flowing through these switching circuits (Q1, DD1) (Q2, DD2). It flows by the synthesized waveform.

Further, for example, as can be seen by comparing the waveform of the voltage V1 shown in this figure between when the load power Po = 200 W and when the load power Po = 0 W, as the switching frequency, the secondary side DC output voltage Eo is It can be seen that the primary-side switching frequency is controlled to be lower in the heavy load condition (Po = 200 W) than in the light load condition (Po = 0 W). That is, as the level of the secondary side DC output voltage Eo decreases due to a heavy load, the switching frequency is lowered and the level of the secondary side DC output voltage Eo increases due to a light load. In response to this, the switching frequency is increased. This indicates that a constant voltage control operation by upper side control is performed as a switching frequency control method.
In this case, as shown in the figure, the peak level of the switching current IQ2 when the load power Po = 200 W is 5.6 Ap, and the level of the switching current IQ2 when the load power Po = 0 W is 0.8 Ap.

Further, by obtaining the above-described primary side operation, an alternating voltage V2 having a waveform shown in the figure is induced in the secondary winding N2A of the insulating converter transformer PIT. During one half cycle in which the alternating voltage V2 is positive, the secondary side rectifier diode Do1 is made conductive. Further, the rectifier diode Do2 is turned on in the half cycle in which the alternating voltage V2 is negative (that is, in the half cycle in which the alternating voltage excited by the secondary winding portion N2B is positive). As a result, in the secondary-side double-wave rectifier circuit, as the rectified output current I2 flowing between the center tap of the secondary winding N2 and the secondary-side ground, the alternating voltage V2 has a positive and negative peak as shown in the figure. A waveform having a peak level with the same period as that of the level is obtained.
The peak level of the alternating voltage V2 is the level of the secondary side DC output voltage Eo. In this case, as the rectified output current I2, the peak level in each half cycle is different between 3Ap and 2Ap as shown in the figure, and this will be described later.

By the way, in the case of adopting a configuration as a resonance type converter that stabilizes the secondary side DC output voltage by the switching frequency control method as in the power supply circuit shown in FIG. 11, the switching frequency for stabilization is variable. The control range tends to be relatively wide.
This will be described with reference to FIG.
FIG. 13 shows the constant voltage control characteristics of a conventional power supply circuit configured to be stabilized by the switching frequency control method as in the power supply circuit shown in FIG. 11, with the switching frequency fs and the secondary side DC output voltage. This is shown by the relationship with the level of Eo.
In the description of this figure, it is assumed that the power supply circuit of FIG. 11 employs so-called upper side control as a switching frequency control method. Here, the upper side control means that the switching frequency is variably controlled in a frequency range higher than the resonance frequency fo of the primary side series resonance circuit, and the change of the resonance impedance generated thereby makes it possible to control the secondary side DC output voltage Eo. Control that controls the level.

As a general matter, the series resonant circuit has the smallest resonance impedance at the resonance frequency fo. Thereby, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs in the upper side control, the level of the secondary side DC output voltage Eo increases as the switching frequency fs approaches the resonance frequency fo. As it gets away from fo, it goes down.
Therefore, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs under the condition that the load power Po is constant, the switching frequency fs is the same as the resonance frequency fo of the primary side series resonance circuit as shown in FIG. It sometimes shows a peak and changes in a quadratic curve that decreases as the frequency moves away from the resonance frequency fo.

  Further, a characteristic is obtained that the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs is shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  Under such characteristics, when the secondary side DC output voltage Eo is stabilized by upper side control so that Eo = tg, the required switching frequency variable range (necessary control) Range) is a range indicated as Δfs.

In practice, the power supply circuit shown in FIG. 11 has an input fluctuation range of AC input voltage VAC = 85V to 120V as an AC 100V system, and maximum load power Pomax = 200 W of secondary side DC output voltage Eo as a main DC power source, minimum. Corresponding to the load condition of load power Pomin = 0 W (no load), constant voltage control is performed by the switching frequency control method so as to stabilize, for example, at the secondary side DC output voltage Eo = 135V.
When this condition is met, the variable range of the switching frequency fs necessary for constant voltage control in the conventional general power supply circuit is approximately fs = 80 kHz to 200 kHz or more, and Δfs is also 120 kHz. It will be a wide range accordingly.

Here, as a power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V is possible so as to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V region such as Europe. A so-called wide-range one is known.
Therefore, let us consider a configuration of the conventional power supply circuit that performs switching frequency control including the power supply circuit shown in FIG.
In the wide range correspondence, as described above, for example, it corresponds to the AC input voltage range of AC85V to 288V. Therefore, for example, the level fluctuation range of the secondary side DC output voltage Eo is larger than that corresponding to a single range of only the AC 100 V system or only the AC 200 V system. In order to perform constant voltage control on the secondary side DC output voltage Eo whose level fluctuation range is expanded corresponding to such an AC input voltage range, a wider switching frequency control range is required. For example, when the conventional switching frequency control range (fs = 80 kHz to 200 kHz) of the AC100V system as described above is used, the switching frequency variable range necessary for the wide range is about 80 kHz to 500 kHz. It will be necessary to expand to.

However, for an IC (oscillation / drive circuit 2) for driving a current switching element, the upper limit of the drive frequency that can be handled is about 200 kHz. Even if the switching drive IC capable of driving at a high frequency as described above is configured and mounted, the power conversion efficiency is significantly reduced when the switching element is driven at such a high frequency. Therefore, it is not practical as an actual power supply circuit.
For this reason, it has been considered extremely difficult to realize wide-range correspondence in the conventional power supply circuit only by the stabilization operation by the switching frequency control method.

  Further, in this case, the switching frequency control range is particularly wide when the double-wave rectifier circuit is configured as the secondary-side rectifier circuit as in the power supply circuit shown in FIG. Become.

First, in the case of a double-wave rectifier circuit, the secondary winding N2 is center-tapped to form two secondary winding portions (N2A, N2B). In these two secondary winding portions N2A and N2B, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding portion N2A → rectifier diode Do1 → smooth. It flows through the path of the capacitor Co → secondary winding portion N2A]. Further, in the other half cycle of the alternating voltage, the rectified current flows through [secondary winding portion N2B → rectifier diode Do2 → smoothing capacitor Co → secondary winding portion N2B].
That is, in the two-wave rectification, the two secondary winding portions are in a state in which current flows only in one half cycle and does not flow in the other half cycle.

According to such a two-wave rectification operation, there is a required capacitance between the secondary winding portion N2A and the secondary winding portion N2B wound around the bobbin of the insulating converter transformer PIT. Will be.
Since the inter-line capacitance is present in this way, the secondary winding is equivalently equivalent to the secondary winding as shown in FIG. 11 on the secondary side of the insulating converter transformer PIT in this case. The capacitor C2 is connected in parallel to the line N2.

Since the capacitor C2 is connected in parallel to the secondary winding N2, in this case also on the secondary side, the parallel is caused by the leakage inductance (L2) of the secondary winding N2 and the capacitance (C2) of the capacitor C2. Thus, a resonance circuit (partial resonance circuit) is formed, and the resonance operation similar to that of the primary side partial resonance circuit (L1 // Cp) can be obtained on the secondary side.
Incidentally, the capacitance of the capacitor C2 is determined by the number of litz wire bundles used as the secondary winding N2 and the window area of the bobbin around which the secondary winding N2 is wound. In the circuit of FIG. 11 under each condition described above, it is a very small amount of about 100 pF to 500 pF.

As a result of the parallel resonant circuit being formed on the secondary side as described above, the constant voltage characteristic of the secondary side DC output voltage Eo as shown in FIG. Actually, the characteristic is as shown in FIG.
In FIG. 14, first, a parallel resonance circuit is formed on the secondary side as described above, and when the resonance frequency of the series resonance circuit on the primary side is fo1, the parallel resonance circuit on the secondary side is formed. A resonance frequency fo2 exists.
In addition, since there are two different resonance points in this way, the characteristic curve particularly at Pomin corresponds to the peak and secondary resonance frequency fo2 according to the primary resonance frequency fo1. A bimodal curve having two peaks and a peak as shown in the figure is obtained.

  In this case, when the capacitance of the capacitor C2 is relatively small as described above, the level of the secondary side DC output voltage Eo tends to be relatively low under heavy load conditions. The resonance point on the secondary side does not become apparent (characteristic curve at Pomax). However, if the secondary-side DC output voltage Eo tends to rise sharply by approaching a no-load state due to a light load trend, does the secondary-side resonance point become obvious? As shown, a bimodal characteristic curve such as a characteristic curve at Po = 0 in the figure can be obtained.

When comparing the characteristic curve of this bimodal with the characteristic curve at the same Po = 0W in FIG. 13, the bimodal curve shown in FIG. 14 has no load compared to the case where it is a single-peaked curve. It can be seen that the switching frequency at the time tends to be higher.
And, according to this, as can be seen by comparing Δfs in each figure, the required control range Δfs of the switching frequency becomes wider in the bimodal in FIG.

FIG. 15 is a diagram showing fluctuation characteristics of the switching frequency fs with respect to load fluctuations in the circuit of FIG. 11 in which the secondary-side rectifier circuit is a double-wave rectifier circuit.
From this characteristic diagram, as described above, when the dual-wave rectifier circuit is used, when the load power Po is around 0 W, the secondary resonance point becomes obvious and the switching frequency tends to increase rapidly. Become.
According to experiments, when Pomax = 200 W, the switching frequency fs = 75.8 kHz, whereas when Pomin = 0 W, the switching frequency fs = 172.4 kHz.

  As described above, when the dual-wave rectifier circuit is configured on the secondary side as the configuration of the conventional power supply circuit, the necessary control range Δfs due to the presence of two resonance points by the primary side and secondary side resonance circuits. In addition, the necessary control range Δfs tends to expand.

  Here, since the necessary control range Δfs tends to expand as described above, in the conventional power supply circuit, in order to actually support a wide range, for example, the following configuration is used. It was supposed to be taken.

For one, a rectifier circuit system for generating a DC input voltage (Ei) by inputting a commercial AC power source, a voltage doubler rectifier circuit, a full-wave rectifier circuit, A function is given so that switching is performed with.
In this case, a commercial AC power supply level is detected, and a voltage doubler rectifier circuit or a full-wave rectifier circuit is formed according to the detected level, and a rectifier circuit is switched by a switch using an electromagnetic relay. The circuit is configured to switch the circuit connection in the system.

  However, such a rectifying circuit system switching configuration requires a required number of electromagnetic relays as described above. Further, it is necessary to provide at least two sets of smoothing capacitors in order to form a voltage doubler rectifier circuit. For this reason, the number of parts is increased and the cost is increased, and the mounting area of the power supply circuit board is increased and the size is increased. In particular, since these smoothing capacitors and electromagnetic relays are large among the components forming the power supply circuit, the substrate size becomes considerably large.

  Further, when the full-wave rectification operation and the voltage doubler rectification operation are switched, an instantaneous power failure occurs or the AC input voltage drops below the rating when an AC 200V commercial AC power supply is input. For example, when the level is lower than that corresponding to the AC200 system, it is assumed that a malfunction occurs in which the AC100V system is detected and switched to the voltage doubler rectifier circuit. When such a malfunction occurs, voltage doubler rectification is performed on the AC input voltage of AC200V system level, so that there is a possibility that the switching elements Q1, Q2, etc., for example, will be broken and destroyed.

  Therefore, as an actual circuit, in order to prevent the above-described malfunction, not only the DC input voltage of the main switching converter but also the DC input voltage of the converter circuit on the standby power supply side is detected. It is made to take. As a result, the addition of components for detecting the converter circuit on the standby power supply side further promotes the above-described cost increase and increase in circuit board size.

  In addition, the detection of the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunctions means that a wide range compatible power supply circuit having a circuit for switching the rectifying operation includes a standby power supply in addition to the main power supply. This means that it cannot actually be used unless it is an electronic device. That is, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and the use range is narrowed accordingly.

Also, as a configuration for wide range compatibility, the primary side current resonance type converter may be switched between half-bridge coupling and full-bridge coupling in accordance with AC 100 V / AC 200 V commercial AC power input. Are known.
With this configuration, even if the AC 200V system AC input voltage drops to the AC 100V system level due to, for example, the momentary power failure described above, the switching operation only changes from the half-bridge operation to the full-bridge operation. The switching element or the like does not exceed the breakdown voltage. For this reason, it is not necessary to detect the DC input voltage on the standby power supply side, and therefore, it can be adopted for an electronic device that does not have a standby power supply. In addition, since it is not switching in the commercial power supply line, switching of the circuit form by a semiconductor switch is possible, so that a large switch part such as an electromagnetic relay is not necessary.

However, in this configuration, it is necessary to provide at least four switching elements in order to form a full bridge coupling corresponding to the AC100V system. That is, it is necessary to add two switching elements as compared with the converter configuration using only the half-bridge coupling method that can be formed by two switching elements.
In this configuration, four stones perform switching operation in the full bridge operation, and three stone switching elements perform switching operation in the half bridge operation. Although the resonance type converter has low switching noise, as the number of switching elements that perform switching in this way increases, the switching noise becomes disadvantageous.

  In this way, in any of the configurations described above that are compatible with a wide range, when compared with a configuration compatible with a single range, an increase in circuit scale and cost increase due to an increase in the number of parts cannot be avoided. . In addition, there are new problems that did not occur in the configuration corresponding to the single range, such as limitation of the range of use to the device in the former configuration and increase in switching noise in the latter configuration.

In addition, if the control range of the switching frequency is correspondingly wide, there is a problem that the high-speed response characteristic for stabilization of the secondary side DC output voltage Eo is deteriorated.
Some electronic devices involve, for example, an operation that fluctuates so that the load condition is instantaneously switched between a state of maximum load and a state of almost no load. Such a load variation is also called a switching load. As a power supply circuit mounted on such a device, it is necessary to appropriately stabilize the secondary side DC output voltage in response to the load fluctuation that is the switching load.
However, when the switching frequency control range is wide as described with reference to FIGS. 13 and 14, the secondary side DC output voltage corresponds to the load fluctuation such as the switching load. It takes a relatively long time to vary the switching frequency to the required level. That is, an unsatisfactory result is obtained as a response characteristic of constant voltage control.
In particular, the power supply circuit shown in FIG. 11 has a large switching frequency in the load range from about load power Po = 25 W or less to 0 W as a switching frequency characteristic according to constant voltage control as shown in FIG. It can be seen that the constant voltage control response to the switching load as described above is disadvantageous.

Further, as a further problem, when the secondary side rectifier circuit is a double-wave rectifier as in the circuit shown in FIG. 11, there is a problem that a magnetic demagnetization occurs particularly in the insulating converter transformer PIT.
That is, as the secondary winding portion N2A and the secondary winding portion N2B, one winding portion is long and the other is short depending on which is wound first around the bobbin of the insulating converter transformer PIT. Due to the difference in winding length, a difference also occurs in the coupling coefficient between the primary winding N1 and the secondary winding portion N2A, and between the primary winding N1 and the secondary winding portion N2B.
11, the coupling coefficient between the primary winding N1 and the secondary winding part N2A is k = 0.86, and the coupling coefficient between the primary winding N1 and the secondary winding part N2B is k = 0. 85. This also causes a difference in the leakage inductance in each winding part. As a result, as shown in the waveform diagram of FIG. 12, the rectified output current I2 has a waveform with a different peak level in each half cycle. Will be.

Thus, the fact that the peak level of the rectified current is different in each half cycle means that the peak level of the current flowing through each secondary rectifier diode (Do1, Do2) is also different. As a result, as one rectifier diode, The withstand current level must be increased as compared with the case where a rectified current having an equal peak level flows through both diodes. Therefore, it is necessary to select a more expensive part with a higher withstand current level than when a uniform rectified current peak level is obtained, and this increases the manufacturing cost of the power supply circuit. Will be forced.
Further, when the peak level of the rectified current is made different as described above, there is a problem that the conduction loss in each of the rectifier diodes Do1 and Do2 is also biased.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, first, a switching means formed by including a switching element that performs switching by inputting a DC input voltage, and a switching drive means that switches the switching element are provided.
An insulating converter transformer formed by winding a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the primary winding; Prepare.
Also, the switching means is formed of at least a leakage inductance component of the primary winding of the insulating converter transformer and a capacitance of a primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type. A primary side series resonance circuit is provided.
Further, a rectifier circuit other than the double-wave rectifier circuit is provided to perform a rectification operation on the alternating voltage obtained in the secondary winding, and the rectified output is smoothed by a secondary side smoothing capacitor, and a secondary side DC Secondary side DC output voltage generation means for generating an output voltage is provided.
A constant voltage for controlling the secondary side DC output voltage by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. Control means are provided.
In addition, the insulating converter transformer has a gap length formed at a predetermined position of the core such that the coupling coefficient between the primary side and the secondary side is not more than a predetermined value.

The switching power supply circuit having the above configuration employs a switching converter configuration in which a primary side series resonance circuit having a primary side switching operation as a current resonance type is formed, and the secondary side rectifier circuit includes both waves. A rectifier circuit other than the rectifier circuit is provided. The stabilization of the secondary side DC output voltage is performed by a switching frequency control method that is performed by variably controlling the switching frequency of the primary side switching element.
In this way, when performing the stabilization operation by the switching frequency control method, the secondary side rectifier circuit is made other than the double-wave rectifier circuit, so that when the double-wave rectifier circuit is used, The inter-line capacitance that will exist can be eliminated, and parallel resonance operation will not be performed on the secondary side. First, in this respect, it is possible to reduce the variable control range (necessary control range) of the switching frequency necessary for stabilization.
In addition, in the present invention, the required control range can be further reduced by reducing the coupling coefficient between the primary side and the secondary side of the insulating converter transformer to a predetermined value or less.

As described above, according to the present invention, the variable control range (necessary control range) of the switching frequency necessary for the constant voltage control can be reduced as compared with the conventional case. Therefore, the wide range only by the stabilizing operation by the switching frequency control method. A corresponding power supply circuit can be easily realized.
In this way, wide-range support by switching frequency control is realized, for example, switching the rectifier circuit system according to the rated level of commercial AC power supply, or, for example, half-bridge coupling and full-bridge coupling It is no longer necessary to adopt a configuration for switching circuits between the two.
As a result, the circuit component parts and the board area can be reduced accordingly, and the application range of the power supply circuit to the electronic device can be expanded and switching noise can be advantageously obtained. .

  Further, in order to realize such a configuration of the present invention, basically, the gap formed in the core of the insulating converter transformer has only to be larger than the conventional one. A configuration corresponding to a wide range can be realized without additional parts from the configuration.

  If the required control range of the switching frequency is reduced as described above, for example, when the load power fluctuates at maximum / no load at high speed, the responsiveness of the constant voltage control is improved. In this respect, higher reliability can be obtained.

  FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit as the best mode for carrying out the present invention (hereinafter also referred to as an embodiment). The power supply circuit shown in this figure also employs a configuration in which a partial voltage resonance circuit is combined with a separately excited current resonance converter using a half-bridge coupling method as a basic configuration on the primary side.

First, in the power supply circuit shown in FIG. 1, a common mode noise filter including filter capacitors CL and CL and a common mode choke coil CMC is formed for the commercial AC power supply AC.
A full-wave rectifying / smoothing circuit including a bridge rectifier circuit Di and one smoothing capacitor Ci is connected to the commercial AC power supply AC that is a subsequent stage of the noise filter.
When this full-wave rectifying / smoothing circuit inputs a commercial AC power supply AC and performs a full-wave rectifying operation, a rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to the AC input voltage VAC.

  As shown in the figure, the current resonance type converter for switching (intermittently) by inputting the DC input voltage includes a switching circuit in which two switching elements Q1, Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A primary side partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the primary side partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In addition, an oscillation / drive circuit 2 is provided for switching the switching elements Q1, Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. In this case, for example, a general-purpose IC can be used. The oscillation circuit of the oscillation / drive circuit 2 generates an oscillation signal having a required frequency, and the drive circuit generates a switching drive signal that is a gate voltage for switching the MOS-FET by using the oscillation signal. The voltage is applied to the gates of the switching elements Q1, Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be continuously turned on / off at alternate timings according to the switching frequency according to the cycle of the switching drive signal.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the series connection of the primary side series resonant capacitor C1. As a result, the switching output is transmitted. The other end of the primary winding N1 is connected to the primary side ground.

Here, the insulating converter transformer PIT has a structure as shown in the cross-sectional view of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with resin etc. is provided by the shape divided | segmented so that it might mutually become independent about the primary side and the secondary side winding part. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding (N1) and the secondary side winding (N2) are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side Due to the different winding regions, the windings are wound around the inner magnetic legs of the EE type core. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G is formed as shown in the figure for the inner magnetic leg of the EE type core. In this case, as the gap G, for example, a gap length of about 2.0 mm or more is set, and as the coupling coefficient k between the primary side and the secondary side, for example, a loosely coupled state with k = 0.70 or less is obtained. ing. As an actual coupling coefficient k, a gap length = 2.8 mm was set and k = 0.625 was set. The gap G can be formed by making the inner magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

Incidentally, in the power supply circuit having the conventional current resonance type converter including the power supply circuit shown in FIG. 11, the gap formed in the core of the insulating converter transformer PIT is, for example, 1.0 mm as described above. By setting the degree, k = 0.8 to 0.9 was obtained as the coupling coefficient k.
In other words, in the present embodiment, the degree of coupling between the primary side and the secondary side of the insulating converter transformer PIT is set to be lower than in the prior art.

Returning to FIG.
The insulating converter transformer PIT generates a predetermined leakage inductance L1 in the primary winding N1 with the structure described with reference to FIG. Further, as described above, the primary winding N1 and the primary side series resonance capacitor C1 are connected in series. Accordingly, a series resonance circuit (primary side series resonance circuit) is formed by the leakage inductance L1 of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1.
In addition, the primary side series resonance circuit is connected to the switching output points of the switching elements Q1 and Q2, and therefore the switching outputs of the switching elements Q1 and Q2 are transmitted to the primary side series resonance circuit. Become. In the primary side series resonance circuit, the resonance operation is performed by the transmitted switching output, so that the operation of the primary side switching converter is a current resonance type.

By the way, according to the description so far, the primary side switching converter shown in this figure has an operation as a current resonance type by the primary side series resonance circuit (L1-C1) and the above-described primary side partial voltage resonance circuit (Cp / / L1) and partial voltage resonance operation.
That is, on the primary side of the power supply circuit shown in this figure, a configuration in which the resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit is adopted. Here, a switching converter in which two resonance circuits are combined in this way is referred to as a “composite resonance type converter”.

An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited (induced) in the secondary winding N2 of the insulating converter transformer PIT.
In this case, a full-wave rectifying / smoothing circuit including a bridge rectifying circuit formed by connecting rectifying diodes Do1 to Do4 as illustrated and a set of smoothing capacitors Co is provided for the secondary winding N2.
In this full-wave rectifying / smoothing circuit, in one half cycle of the alternating voltage excited by the secondary winding N2, the pair of rectifier diodes [Do1, Do4] of the bridge rectifier circuit is turned on, and is connected to the smoothing capacitor Co. The operation of charging the rectified current is obtained. Further, in the other half cycle of the alternating voltage excited in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining a set of rectifier diodes [Do2, Do3] conducting.
As a result, a voltage corresponding to the same multiple of the alternating voltage level excited in the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co (secondary side DC output voltage Eo).

  The secondary side DC output voltage Eo obtained in the smoothing capacitor Co in this way is supplied to a load (not shown) and is also branched and input as a detection voltage for the control circuit 1 described below.

The control circuit 1 is provided to stabilize the secondary side DC output voltage Eo by the switching frequency control method.
In this case, the control circuit 1 supplies the oscillation / drive circuit 2 with a detection output corresponding to the level change of the secondary side DC output voltage Eo as a detection input. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. For this purpose, the frequency of the oscillation signal generated by the internal oscillation circuit is varied.
By changing the switching frequency of the switching elements Q1 and Q2, the resonance impedance of the primary side series resonance circuit changes, and the amount of electric power transmitted from the primary winding N1 to the secondary winding N2 side of the insulating converter transformer PIT is reduced. Although it changes, this operates so as to stabilize the level of the secondary side DC output voltage Eo.

Here, in a general current resonance type converter, as a switching frequency control method, a frequency range higher than the resonance frequency fo of the primary side series resonance circuit is set as a variable range of the switching frequency. Take control method.
At this time, the series resonance circuit has the lowest resonance impedance at the resonance frequency. Therefore, when the upper side control method based on the resonance frequency of the series resonance circuit is employed as in the present embodiment, the resonance impedance is increased as the switching frequency fs increases. Become.
Therefore, for example, when the secondary side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. This lowers the resonance impedance and increases the amount of power transmitted from the primary side to the secondary side, so that the secondary side DC output voltage Eo rises.
On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the switching frequency is controlled to be increased. As a result, the resonance impedance is increased and the power transmission amount is reduced, so that the secondary side DC output voltage Eo is lowered. Thus, the secondary side DC output voltage Eo is stabilized by changing the switching frequency.

3 and 4 show operation waveforms of main parts of the power supply circuit shown in FIG.
Here, the power supply circuit shown in FIG. 1 adopts a configuration corresponding to a wide range as will be described later. Corresponding to this, FIG. 3 shows the operation waveforms when the AC input voltage VAC = 100V, and FIG. 4 shows the operation waveforms when the AC input voltage VAC = 230V. Further, as can be seen with reference to these drawings, in this case, each of the diagrams of FIGS. 3 and 4 further shows operation waveforms when the load power Po = 200 W and when the load power Po = 0 W, respectively. .
It is assumed that the load power Po = 200 W is the maximum load power (Pomax) of the power supply circuit in FIG. Further, the load power Po = 0W is the minimum load power (Pomin).

In order to obtain the experimental results shown in these figures, each part of the circuit of FIG. 1 was selected as follows.
Insulating converter transformer PIT: Gap G = 2.8 mm, primary winding N1 = 35T, secondary winding N2 = 25T, coupling coefficient k = 0.625,
・ Primary side series resonance capacitor C1 = 0.033μF
Note that, by setting the number of turns of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1, the resonance frequency fo of the primary side series resonance circuit in this case is set to about fo = 55 kHz.
Further, in this case, the induced voltage level per turn of the secondary winding N2 is set to 5 V / T by selecting each of the above parts.

First, in FIGS. 3 and 4, a rectangular wave voltage V1 is a voltage across the switching element Q2, and indicates the on / off timing of the switching element Q2. This voltage V1 has a waveform clamped at the level of the rectified and smoothed voltage Ei during an on period in which the switching element Q2 is conductive and turned on, and in an off period in which the switching element Q2 is nonconductive.
During the ON period of the switching element Q2, a switching current IQ2 having a waveform shown in the figure flows through the switching circuit system including the switching element Q2 and the clamp diode DD2. Further, the switching current IQ2 becomes 0 level during the OFF period of the switching element Q2.
Although not shown, as the voltage across the other switching element Q1 and the switching current flowing through the switching circuit (Q1, DD1), the voltage V1 and the switching current IQ2 have a waveform shifted by 180 ° phase shift. can get. That is, the switching element Q1 and the switching element Q2 perform the switching operation at the same cycle timing so as to be alternately turned on / off.
As the primary side series resonance current Io (not shown) flowing through the primary side series resonance circuit (L1-C1), switching currents flowing through these switching circuits (Q1, DD1) (Q2, DD2) are synthesized. It flows as an ingredient.

Note that, when the AC input voltage VAC = 100 V, the peak level of the switching current IQ2 is about 6.0 Ap when the load power Pomax = 200 W, as shown in FIG. In addition, when the load power Pomin = 0 W, about 4.0 Ap is obtained.
Further, as shown in FIG. 4, the peak level of the switching current IQ2 when the AC input voltage VAC = 230 V is about 5.0 Ap when the load power Pomax = 200 W, and about 3.4 Ap when the load power Pomin = 0 W. The result was obtained.

Then, by obtaining the primary side operation indicated by the voltage V1 and the switching current IQ2, the alternating voltage V2 having the waveform shown in the figure is excited in the secondary winding N2 of the insulating converter transformer PIT (each figure). (Refer to Pomax). This alternating voltage V2 has a waveform clamped at the level of the secondary side DC output voltage Eo.
Further, by obtaining the alternating voltage V2 having such a waveform, in the bridge rectifier circuit on the secondary side, the pair of rectifier diodes Do1 and Do4 are conducted and smoothed in the half cycle in which the alternating voltage V2 is positive. An operation of charging the capacitor Co with the rectified current is obtained. Further, in a half cycle in which the alternating voltage V2 is negative, the set of rectifier diodes Do2 and Do3 is turned on to charge the smoothing capacitor Co with a rectified current.
By performing such an operation, the secondary side rectified current I2 flowing through the secondary winding N2 shown in FIG. 1 is positive in the period in which the alternating voltage V2 is at the positive peak level as shown in the figure. In the period of negative polarity, a waveform that flows similarly due to negative polarity is obtained.

  Here, as can be seen with reference to the waveform of the secondary side rectified current I2, as the rectified current flowing on the secondary side of the circuit of FIG. 1, positive and negative peak levels can be obtained at the same level. . Specifically, as shown in the figure, when the AC input voltage VAC = 100 V and the load power Pomax, the positive and negative peak levels are equal to 3.0 Ap. Further, when the AC input voltage VAC = 230 V, the positive and negative peak levels at the load power Pomax are both 2.5 Ap.

In this way, the positive and negative peak levels of the secondary side rectified current are equal because the circuit of FIG. 1 is provided with a bridge rectifier circuit as the secondary side rectifier circuit and a rectifier circuit other than the double-wave rectifier circuit. by.
In other words, in the case of the circuit of FIG. 1, there is no occurrence of demagnetization in the secondary winding N2 of the insulating converter transformer PIT, unlike the case where the double-wave rectifier circuit of FIG. 11 is provided. This prevents a situation in which the peak level of the rectified current is different in each half cycle of the alternating voltage excited by.
According to this, the peak level of the rectified current flowing through each rectifier diode on the secondary side is not different, and therefore, the same current withstand level product can be selected. As a result, the circuit manufacturing cost can be reduced accordingly.
In addition, since the secondary side rectified current has the same peak level in each half cycle as described above, the problem that the conduction loss in each rectifier diode Do is biased is also solved.

In the circuit of FIG. 1, the secondary side rectifier circuit is a full-wave rectifier circuit using a bridge rectifier circuit, so that the number of turns of the secondary winding N2 for obtaining the same secondary side DC output voltage Eo level is It can be reduced to half of the case where both waves are rectified. Specifically, when the secondary side DC output voltage Eo = 135V is obtained, the secondary winding N2 = 50T (25T + 25T) in the conventional circuit of FIG. 11, whereas the circuit of FIG. The next winding N2 = 25T.
By reducing the number of turns of the secondary winding N2, it is possible to reduce the core size of the insulating converter transformer PIT, thereby reducing the size of the circuit.

Here, in the power supply circuit shown in FIG. 1, as described in FIG. 2, the coupling coefficient k in the insulating converter transformer PIT is set to a value lower than the conventional one. Further, as described above, the secondary side rectifier circuit includes a rectifier circuit other than the double-wave rectifier circuit.
In the embodiment, such a configuration realizes a so-called wide-range configuration that can operate in accordance with the input of the AC 100V system and the AC 200V system. This will be described with reference to FIG.

FIG. 5 shows the constant voltage control characteristics of the power supply circuit of FIG. 1 by the relationship between the switching frequency fs and the level of the secondary side DC output voltage Eo.
In this figure, at the same time, the constant voltage control characteristic of the power supply circuit when the coupling coefficient k in the insulating converter transformer PIT is set to the conventional value is indicated by a one-dot chain line.
For confirmation, it is assumed that the power supply circuit of FIG. 1 adopts so-called upper side control as a switching frequency control method.

As described above with reference to FIG. 13, the series resonant circuit has the smallest resonance impedance at the resonance frequency fo. Thereby, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs in the upper side control, the level of the secondary side DC output voltage Eo increases as the switching frequency fs approaches the resonance frequency fo. As it gets away from fo, it goes down.
Therefore, when the load power Po is constant, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs is as shown in the figure when the switching frequency fs is the same as the resonance frequency fo of the primary side series resonance circuit. It becomes a peak, and shows a quadratic curve change that decreases as it moves away from the resonance frequency fo.

  Further, a characteristic is obtained that the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs is shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  Here, the characteristic curve (characteristic curve 1) at the time of the maximum load power Pomax (Po = 200 W) in the case of the circuit of FIG. 1 shown in this figure and the characteristic curve (characteristic curve 2) at the time of Pomax in the conventional circuit. As can be seen from comparison with (2), approximately the same relatively steep quadratic curve is obtained as both characteristic curves at the maximum load power. On the other hand, when the minimum load power Pomin (Po = 0W) is compared, the curve of the circuit of FIG. 1 shown as the characteristic curve 3 and the conventional curve shown as the characteristic curve 4 are very gentle in the prior art. It can be understood that a steep characteristic is obtained in the circuit of FIG.

Then, under such a characteristic, when trying to stabilize the secondary side DC output voltage Eo by upper side control so that Eo = tg, it is necessary in the conventional power supply circuit. The variable range of the switching frequency (necessary control range Δfs) is a range indicated by a relatively wide ΔfsB as shown in the figure because the characteristic curve at the minimum load power Pomin is gentle as described above.
On the other hand, the necessary control range Δfs for stabilization in the power supply circuit shown in FIG. 1 is ΔfsA which is smaller than the above ΔfsB because the characteristic curve at the minimum load power Pomin is steeper than in the prior art. As a range.

In this way, in the embodiment in which the coupling coefficient k is set to a predetermined value lower than the conventional value, the increase in the switching frequency especially at the minimum load power Pomin is suppressed, so that it is necessary for stabilization. The control range Δfs is greatly reduced.
Here, in the characteristic diagram of FIG. 5, the necessary control range Δfs in the case of one single range of the AC100V system or the 200V system is shown, but in the other range, the necessary control range Δfs is also greatly increased. Reduction will be achieved. That is, similarly, the increase of the switching frequency at the time of the minimum load power Pomin is particularly suppressed, so that the required control range Δfs can be greatly reduced in comparison with the conventional one in both the AC100V system and the AC200V system. It is.
The reduction of the required control range Δfs in each single range in this way means that the required control range Δfs is also larger than the conventional control range Δfs even when it corresponds to the input from the AC 100V system to the AC 200V system. It can be greatly reduced.

  As described above, the necessary control range Δfs when the input from the AC 100V system to the AC 200V system is supported is greatly reduced, so that the power supply circuit of FIG. 1 can perform only the stabilization operation by the switching frequency control method. The configuration corresponding to the wide range can be realized more easily.

  By the way, as described above, as an embodiment, a full-wave rectification operation is performed by a bridge rectifier circuit as a secondary-side rectifier circuit, and a rectifier circuit other than a double-wave rectifier circuit such as the circuit of FIG. However, let us consider a case where a dual-wave rectifier circuit is provided as a secondary-side rectifier circuit under the setting of the coupling coefficient k of the insulating converter transformer PIT described in FIG.

First, since it is a double-wave rectifier circuit, even in this case, there is no change in the capacitance between the secondary windings divided by the center tap. A parallel resonant circuit is equivalently formed on the side, so that even in this case, the characteristic curve for the constant voltage control, particularly when the load power Po = 0 W, is the bimodal curve shown in FIG. Will be obtained.
In this way, the characteristic curve itself is not a single peak but a double peak characteristic. Therefore, in the case of a double-wave rectifier circuit, the required control range of the switching frequency is also set by the setting of the coupling coefficient k of the embodiment. Δfs cannot be reduced. This is because the characteristic curve is a double peak, and as a switching frequency in the vicinity of no load, the characteristic of rapidly increasing as shown in FIG. 15 is maintained. As a result, it is almost impossible to effectively reduce the necessary control range Δfs, and as a result, the necessary control range Δfs does not change even if the coupling coefficient k is set to a predetermined value or less.

On the other hand, if the parallel resonance circuit is not formed on the secondary side and the characteristic curve of constant voltage control is a single peak, as understood from the characteristic diagram of FIG. The switching frequency at no load (at Pomin) can be set to a lower value than when the conventional coupling coefficient is set. Therefore, the rapid increase characteristic of the switching frequency near the no load can be effectively improved. Figured.
For this reason, according to the embodiment in which the secondary side rectifier circuit is other than the double-wave rectifier circuit and the coupling coefficient k is reduced to a predetermined value or less, the required control range Δfs can be effectively reduced. This makes it possible to more easily realize a wide-range configuration with only a stabilizing operation by the switching frequency control method.

Next, FIG. 6 shows the result of an actual experiment on the switching frequency fs for the power supply circuit of FIG.
FIG. 6 shows a change characteristic of the switching frequency fs with respect to load fluctuation when the AC input voltage VAC is constant at 100 V and 230 V, respectively. In this case, the characteristics when the AC input voltage VAC = 100 V is indicated by a solid line, and the characteristics when VAC = 230 V are indicated by a broken line.
First, as shown in the figure, the switching frequency fs has a characteristic that increases according to a decrease in the load power Po at both the AC input voltage VAC = 100V and 230V. In particular, the characteristic when the AC input voltage VAC = 100 V is compared with the conventional characteristic when the AC input voltage VAC = 100 V as shown in FIG. 15, and the load power Po = 0 W (minimum load power). ) The sharp decline in the vicinity is greatly suppressed.

According to the experiment, the switching frequency fs with respect to the variation of the load power Po = 200 W to 0 W is fs = 78.1 kHz to 80.6 kHz when the AC input voltage VAC = 100 V, and fs = 100. A result of 0 kHz to 109.0 kHz was obtained.
According to this, the required control range Δfs in the AC100V system single range is about 2.5 kHz, and the required control range Δfs in the AC200V system single range is about 9.9 kHz.
The necessary control range Δfs for adapting to the wide range is approximately 78.1 kHz to 109.0 kHz, and Δfs = about 30 kHz.

  Such a required control range of the switching frequency is well within the frequency variable control range of the current switching drive IC (oscillation / drive circuit 2). From this, according to the power supply circuit of the embodiment, the switching frequency It can be understood that a configuration corresponding to a wide range by variable control can be realized at a practical level.

As described so far, the power supply circuit of the present embodiment shown in FIG. 1 is capable of handling a wide range only by a stabilizing operation by the switching frequency control method.
As a result, for example, when the wide range is supported, the rectification operation is switched for the rectifier circuit system for generating the DC input voltage (Ei) according to the rated level of the commercial AC power supply, or the half bridge coupling method and the full bridge There is no need to adopt a configuration for switching the type of the switching converter with the coupling method.
If such a circuit switching configuration is not necessary, for example, only one smoothing capacitor Ci can be used, and at least two switching elements required for half-bridge coupling can be used. Accordingly, it is possible to reduce the number of circuit components, the circuit scale, and the switching noise.
Further, if the circuit switching configuration is not required, it is not necessary to provide a special configuration for preventing malfunction due to switching, and in this respect also, the increase in the number of components and the suppression of the cost increase can be achieved. Furthermore, since a standby power supply is not essential to prevent malfunctions, the range of devices in which the power supply circuit can be used can be expanded.

  Further, in order to obtain the effect as such an embodiment, it is only necessary to change at least the gap length formed in the core of the insulating converter transformer PIT in the configuration of the conventional current resonance type converter. That is, it is possible to realize a wide range support without a special additional component compared to the configuration of the conventional current resonance type converter.

In addition, since the required control range Δfs of the switching frequency is greatly reduced as described above, the responsiveness of the constant voltage control is greatly increased regardless of whether it is compatible with a wide range or a single range. Will be improved.
That is, some electronic devices perform an operation of changing the load power Po so as to be switched (switched) at a relatively high speed between a maximum and no load, which is called a so-called switching load. An example of a device that performs such an operation as a switching load is a printer that is a peripheral device of a personal computer.
When a power supply circuit having a relatively wide necessary control range Δfs as shown in FIG. 11 is mounted on a device that operates as such a switching load, for example, as described above, the device has a steep Following the change of the load power, the switching frequency fs is variably controlled by a correspondingly large amount of change. For this reason, it has been difficult to obtain high-speed constant voltage control response.
On the other hand, in the present embodiment, as described above, the necessary control range Δfs is greatly reduced particularly in the region for each single range, so that the load power Po is steep between the maximum and no load. It is possible to stabilize the secondary side DC voltage Eo in response to fluctuations at a high speed. That is, the response performance of the constant voltage control with respect to the switching load can be greatly improved.

Next, FIG. 7 shows a configuration of a modification on the secondary side of the power supply circuit as the embodiment shown in FIG.
In FIG. 7, the configuration on the primary side is the same as that shown in FIG.
In the modification shown in FIG. 7, a voltage doubler half-wave rectifier circuit is configured as the secondary-side rectifier circuit.

First, the voltage doubler half-wave rectifier circuit includes two sets of smoothing capacitors Co, a rectifier diode Do1 and a rectifier diode Do2, and a smoothing capacitor Co1 and a smoothing capacitor Co2, as shown.
In this voltage doubler half-wave rectifier circuit, first, the anode of the rectifier diode Do1 is connected to one end portion (winding end portion) of the secondary winding N2. The cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co1.
The negative terminal of the smoothing capacitor Co1 is connected to the positive terminal of the other smoothing capacitor Co2, as shown, and the negative terminal of the smoothing capacitor Co2 is connected to the secondary side ground. In addition, the other end (winding start end) of the secondary winding N2 is connected to a connection point between the negative terminal of the smoothing capacitor Co1 and the positive terminal of the smoothing capacitor Co2.
Further, as shown in the figure, a rectifier diode Do2 is inserted between a connection point between one end of the secondary winding N2 and the anode of the rectifier diode Do1 and the secondary side ground. The rectifier diode Do2 is inserted so that the anode side is connected to the secondary side ground.

In the voltage doubler half-wave rectifier circuit having the above configuration, first, the rectifier diode Do1 is conducted in one half cycle of the alternating voltage excited in the secondary winding N2, and charges the rectified current to the smoothing capacitor Co1. As a result, a voltage is generated at both ends of the smoothing capacitor Co1 at a level corresponding to an equal voltage level of the alternating voltage excited by the secondary winding N2. In the other half cycle, the rectifier diode Do2 conducts to charge the smoothing capacitor Co2 with a rectified current, and both ends of the smoothing capacitor Co2 correspond to an equal voltage level of the alternating voltage excited by the secondary winding N2. Generate a voltage according to the specified level.
As a result, in one cycle of the alternating voltage excited in the secondary winding N2, there is a level corresponding to twice the alternating voltage level excited in the secondary winding N2 at both ends of the series connection of the smoothing capacitors Co1-Co2. Thus, the secondary side DC output voltage Eo is obtained.

In this way, depending on the configuration shown in FIG. 7, each of the smoothing capacitors Co is charged only in one half cycle of the alternating voltage excited in the secondary winding N2, and the smoothing capacitors Co (Co1− As a voltage level obtained for Co2), a double voltage half-wave rectification operation is obtained, which is a level corresponding to twice the alternating voltage level.
In this case, the secondary side DC output voltage Eo is also branched and supplied as a detection input of the control circuit 1 similar to that shown in FIG.

Here, for confirmation, in this case as well, since the secondary side rectifier circuit has a configuration other than the double-wave rectifier circuit, the same coupling coefficient k (gap length) as in the circuit of FIG. ), A configuration compatible with a wide range is realized.
Further, by adopting a configuration other than the double-wave rectifier circuit in this way, the current withstand level and conduction loss of each rectifier diode Do on the secondary side can be made equal.

In addition, when the secondary side rectifier circuit is a voltage doubler half-wave rectifier circuit as in the configuration shown in FIG. 7, in order to obtain the same secondary side DC output voltage Eo level, The number of turns can be further halved from the bridge rectifier circuit of FIG.
That is, according to this, compared with the conventional circuit of FIG. 11, the number of turns of the secondary winding N2 is reduced to ¼. Therefore, when the configuration on the secondary side of FIG. The insulation converter transformer PIT can be downsized.

In the following, an example of selection conditions for the main part when the power supply circuit of the embodiment in the case of the secondary side configuration in FIG. 7 is actually configured for the wide range will be shown.
-Insulating converter transformer PIT: ferrite core of EER-39, gap length = 2.4 mm, primary winding N1 = 37T, secondary winding N2 = 13T,
-Primary side series resonant capacitor C1 = 0.039μF
By selecting such a main part, it is possible to obtain substantially the same characteristics (FIGS. 5 and 6) as those described for the circuit of FIG.

FIG. 8 shows another modification of the configuration on the secondary side of the power supply circuit according to the embodiment.
In this figure as well, the configuration on the primary side is the same as that shown in FIG.
The modification shown in FIG. 8 includes a voltage doubler full wave rectifier circuit as a secondary side rectifier circuit.

First, in this case, the secondary winding N2 is divided into secondary winding portions N2A / N2B by applying a center tap. Here, a bridge rectifier circuit having the same connection form as that of the circuit of FIG. 1 is connected to the entire secondary winding N2.
Specifically, the end of the secondary winding N2 that is the winding end of the secondary winding N2 is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2. The Further, the end on the secondary winding portion N2B side, which is the winding start end of the secondary winding N2, is connected to a connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4.
The connection point between the cathode of the rectifier diode Do1 and the cathode of the rectifier diode Do3 is connected to the positive terminal of the smoothing capacitor Co. Also in this case, the negative terminal of the smoothing capacitor Co is connected to the secondary side ground.

In this case, the positive terminal of the capacitor Cc is connected to the center tap of the secondary winding N2. Further, the negative terminal of the capacitor Cc is connected to the connection point between the anodes of the rectifier diodes Do2 and Do4.
As shown in the figure, the connection point between the anodes of the rectifier diodes Do2 and Do4 is connected to the secondary side ground.

  In the voltage doubler full-wave rectifier circuit configured as described above, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding portion N2A → smoothing capacitor Cc → rectifier diode Do2 → secondary. It flows through the circulation path of the winding portion N2A]. Further, in the other half cycle of the alternating voltage excited in the secondary winding N2, the rectified current flows through a circulation path of [secondary winding portion N2B → smoothing capacitor Cc → rectifier diode Do4 → secondary winding portion N2B]. Flowing. In other words, in this case, for the smoothing capacitor Cc, a DC voltage is obtained at a level corresponding to the same multiple of the alternating voltage level excited in the secondary winding portion N2A and the secondary winding portion N2B in each half cycle. It is like that.

In addition, in the above half cycle of the alternating voltage excited in the secondary winding N2, the rectified current branches off from the circulation path [secondary winding portion N2B → rectifier diode Do3 → smoothing capacitor Co → It also flows through the path of the rectifier diode Do2].
As a result, in the half cycle, the smoothing capacitor Co is charged with a level in which the alternating voltage of the secondary winding N2B and the voltage across the smoothing capacitor Cc as described above are superimposed. That is, as the voltage across the smoothing capacitor Co, a level corresponding to twice the alternating voltage level obtained in the secondary winding is obtained.

  Further, also as the other half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is branched from the above circulation path [secondary winding portion N2A → rectifier diode Do1 → smoothing capacitor Co → rectifier diode. Do4] also flows, and in this case as well, the voltage across the smoothing capacitor Co is the alternating voltage level obtained in the secondary winding portion by the alternating voltage of the secondary winding portion N2A and the charge of the capacitor Cc. A level corresponding to twice this is obtained.

From such a rectifying operation, in the rectifying circuit in this case, the smoothing capacitor Co is charged in each half cycle of the alternating voltage obtained in the secondary winding N2. As the charging potential, a level corresponding to twice the alternating voltage induced in the secondary winding portion as described above is obtained.
From this, it can be understood that a double voltage full wave rectification operation is obtained by the secondary side rectifier circuit shown in FIG.

  Even when the voltage doubler full wave rectifier circuit is provided as the secondary side rectifier circuit in this way, in order to obtain the same secondary side DC output voltage Eo level, the number of turns of the entire secondary winding N2 is shown in FIG. It can be equivalent to the circuit shown in FIG. Therefore, even when the configuration of FIG. 8 is adopted, the size of the insulating converter transformer PIT similar to that of the circuit of FIG. 1 can be reduced.

For confirmation, the secondary side configuration shown in FIG. 8 is also provided with a center tap on the secondary winding N2 as in the case of the circuit of FIG. However, in the configuration of FIG. 8, the parallel resonance operation (partial voltage resonance operation) does not occur on the secondary side as in the case of the conventional double-wave rectifier circuit. In the configuration of FIG. 8, as understood from the above description, a rectification current flows in each secondary winding part in each half cycle of the alternating voltage, and unlike the case of the conventional double-wave rectification, This is because there is no line capacitance between the secondary winding parts because there is no period during which the rectified current does not flow in the winding part.
Since no parallel resonance operation occurs on the secondary side as described above, the required control range Δfs as in the case of FIG. 1 is reduced by setting the coupling coefficient k described in FIG. 2 even in the configuration of FIG. It is something that can be done.

  Further, in the case of the circuit of FIG. 11, the center tap is applied to the secondary winding N 2, so that the magnetic isolation is generated in the insulating converter transformer PIT. However, in the configuration of FIG. As can be understood from the explanation of the current path, since the rectified current flows through the entire secondary winding N2 in each half cycle, no demagnetization occurs. That is, from this, the configuration of FIG. 8 can also prevent the peak level of the rectified current from being biased in each half cycle as in the case of the circuit of FIG.

FIG. 9 shows still another modified example of the configuration on the secondary side of the power supply circuit according to the embodiment.
Also in FIG. 9, the configuration on the primary side is the same as in the case of FIG.
The modification of FIG. 9 includes a quadruple voltage rectifier circuit as a secondary side rectifier circuit.

As shown in the figure, the quadruple voltage rectifier circuit includes four rectifier diodes Do including rectifier diodes Do1 to Do4, capacitors Cc1 and Cc2, and smoothing capacitors Co1 and Co2.
In this case, a series connection of a capacitor Cc1 (negative terminal → positive terminal) → rectifier diode Do1 (anode → cathode) is connected to one end (end of winding) of the secondary winding N2 as shown in the figure. The positive terminal of the smoothing capacitor Co1 is connected through the via. The negative terminal of the smoothing capacitor Co1 is connected to the other end (winding end) of the secondary winding N2.
The positive terminal of the smoothing capacitor Co2 is connected to the connection point between the negative terminal of the smoothing capacitor Co1 and the other end of the secondary winding N2, and the negative terminal of the smoothing capacitor Co2 is connected to the secondary side ground. It is connected.

Further, a series connection circuit of a capacitor Cc2 (positive terminal → negative terminal) → rectifier diode Do4 (cathode → anode) is inserted between the one end of the secondary winding N2 and the secondary side ground. ing.
The anode of the rectifier diode Do3 is connected to the connection point between the capacitor Cc2 and the rectifier diode Do4 as shown. The cathode of the rectifier diode Do3 is connected to the connection point between the smoothing capacitors Co1 and Co2 and the other end of the secondary winding N2.
Further, the anode of the rectifier diode Do2 is connected to the connection point between the cathode of the rectifier diode Do3 and the other end of the secondary winding N2. The cathode of the rectifier diode Do2 is connected to the connection point between the capacitor Cc1 and the rectifier diode Do1.

In the quadruple voltage rectifier circuit having the above configuration, the rectified current is [secondary winding N2 → rectifier diode Do2 → capacitor Cc1 → secondary winding N2 in one half cycle of the alternating voltage excited in the secondary winding N2. ] By the circulation route. Similarly, also in the other half cycle of the alternating voltage, the rectified current flows through [secondary winding N2 → capacitor Cc2 → rectifier diode Do3 → secondary winding N2] through the circulation path.
In other words, even in this case, a DC voltage having a level corresponding to the same multiple of the alternating voltage level excited in the secondary winding N2 is obtained at both ends of the capacitors Cc1 and Cc2 in the corresponding half cycle. Become.

Even in this case, in each half cycle, the rectified current branches off from the circulation path and also flows through the following path.
First, in one half cycle of the alternating voltage described above, the rectified current branches and flows through the path [smoothing capacitor Co 2 → rectifier diode Do 4 → capacitor Cc 2 → secondary winding N 2]. At this time, charged charges are obtained at both ends of the capacitor Cc2 during this period by the previous circulation path. For this reason, depending on the rectification current path as described above, the smoothing capacitor Co2 is charged with a potential due to the alternating voltage obtained in the secondary winding N2 and the superimposed charge of the capacitor Cc2. Become.
That is, as a result, a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is generated in the smoothing capacitor Co2.

Further, in the other half cycle of the above alternating voltage, the rectified current branches and flows through the path [capacitor Cc1 → rectifier diode Do1 → smoothing capacitor Co1 → secondary winding N2]. The smoothing capacitor Co1 is charged with respect to the alternating voltage of the secondary winding N2 that has received the superimposed charge charge obtained by the capacitor Cc1.
In other words, even with the smoothing capacitor Co1, a DC voltage having a level corresponding to twice the alternating voltage level obtained at the secondary winding N2 is obtained as the voltage across the capacitor.

  In this way, a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is generated at both ends of the smoothing capacitor Co1 and the smoothing capacitor Co2. As a result, a secondary side DC output voltage Eo having a level corresponding to four times the alternating voltage level excited by the secondary winding N2 is obtained at both ends of the series connection of the smoothing capacitor Co1 and the smoothing capacitor Co2. It will be.

  When such a quadruple voltage rectifier circuit is employed, the number of turns of the secondary winding N2 can be reduced to about ¼ that in the case of FIG. 1, thereby further reducing the size of the insulating converter transformer PIT. It becomes possible.

Further, the circuit diagram of FIG. 10 shows a configuration of a modified example of the primary side of the power supply circuit of the embodiment.
As a modification of the primary side, the configuration of the switching converter is changed from the half bridge coupling method to the full bridge coupling method as shown in the figure.
In FIG. 10, the same parts as those already described in FIG.

In FIG. 10, as a full bridge coupling system, as shown in the figure, the half bridge connection of the switching elements Q3 and Q4 is connected in parallel to the half bridge connection of the switching elements Q1 and Q2.
As for the switching elements Q3 and Q4, similarly to the switching elements Q1 and Q2, a damper diode DD3 and a damper diode DD4, which are body diodes, are connected in parallel between the drain and the source, respectively.

In addition, in this case, a primary side series resonant circuit comprising a series connection of the primary winding N1 of the insulating converter transformer PIT and the primary side series resonant capacitor C1 is connected as follows.
First, one end (winding end) of the primary winding N1, which is one end of the primary side series resonance circuit, is connected to a connection point between the source of the switching element Q3 and the drain of the switching element Q4. The connection point between the source of the switching element Q3 and the drain of the switching element Q4 is one switching output point in the full-bridge coupling switching circuit system.
Further, with respect to the other end portion side of the primary side series resonance circuit, the other end (end of winding end) of the primary winding N1 is connected to the other switching output point via the series connection of the primary side series resonance capacitor C1. The connection is made to the connection point between the source of a certain switching element Q1 and the drain of the switching element Q2.

  In this case, the primary-side partial resonance capacitor Cp2 is connected in parallel with the source and drain of the switching element Q4. The primary side partial resonance capacitor Cp2 also forms a parallel resonance circuit (partial voltage resonance circuit) by its own capacitance and the leakage inductance L1 of the primary winding N1, and voltage resonates only when the switching elements Q3 and Q4 are turned off. Get voltage resonant operation.

  The oscillation / drive circuit 2 in this case is configured to drive four switching elements Q1 to Q4. Depending on the oscillation / drive circuit 2, switching driving is performed such that the group of switching elements [Q1, Q4] and the group of switching elements [Q3, Q4] are alternately turned on / off.

  Here, for example, as the load condition becomes a heavy load trend, the current flowing through the switching converter increases, the load on the circuit components increases, and the power loss also increases. Therefore, if the full-bridge coupling is used as described above, the necessary load current is provided by four switching elements. For example, each component is more than the case of the half-bridge coupling system including two switching elements. This reduces the burden on the power source and reduces power loss, which can be advantageous for heavy load conditions.

The present invention should not be limited to the embodiments described so far.
For example, the structure of the insulating converter transformer PIT may be changed as appropriate, including the core type.
In addition, the switching converter exemplified in the embodiment is based on a separately excited type current resonant converter, but may be configured to include, for example, a self excited type current resonant converter. In addition, as a switching element selected in the switching converter, for example, an element other than a MOS-FET such as a bipolar transistor or an IGBT (Insulated Gate Bipolar Transistor) may be employed.
Further, the constants of the component elements described above may be appropriately changed according to actual conditions and the like.
Further, as a configuration for handling heavy loads, a rectifying current circuit system for generating a rectified smoothing voltage Ei by inputting a commercial AC power supply AC (AC input voltage VAC) has a level corresponding to twice the AC input voltage VAC. A voltage doubler rectifier circuit that generates a rectified and smoothed voltage Ei can also be used. However, the configuration in which the rectifying and smoothing circuit system that generates the rectified and smoothing voltage Ei in this way is a voltage doubler rectifying circuit is a configuration that supports only a single range of the AC100V system.
Furthermore, the configuration of the rectifier circuit on the secondary side is not limited to the one exemplified in the embodiment, and a configuration other than the double-wave rectifier circuit that does not cause line capacitance in the secondary winding. If so, other rectifier circuit configurations may be employed.

It is a circuit diagram which shows the structure of the power supply circuit as embodiment in this invention. It is sectional drawing which shows the structural example of the insulation converter transformer with which the switching power supply circuit of embodiment is provided. It is a wave form diagram which shows the operation | movement waveform of the principal part at the time of AC100V in the power supply circuit of embodiment. It is a wave form diagram which shows the operation | movement waveform of the principal part at the time of AC230V in the power supply circuit of embodiment. It is a figure which shows the switching frequency control range (necessary control range) according to load fluctuation | variation as a constant voltage control characteristic of the power supply circuit of embodiment. It is a characteristic view shown about the change characteristic of the switching frequency with respect to the load fluctuation | variation about the power supply circuit of embodiment. It is the circuit diagram shown about the modification about the structure of the secondary side of the power supply circuit of embodiment. It is the circuit diagram shown about the other modification about the structure of the secondary side of the power supply circuit of embodiment. It is the circuit diagram shown about the other modification about the structure of the secondary side of the power supply circuit of embodiment. It is the circuit diagram shown about the modification about the structure of the primary side of the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art. FIG. 12 is a waveform diagram showing an operation of a main part in the power supply circuit shown in FIG. 11. It is the figure which showed the constant voltage control characteristic in the case of setting the coupling coefficient of a primary side and a secondary side as the conventional setting. It is the figure which showed the constant voltage control characteristic about the power supply circuit shown in FIG. 11 which made the secondary side rectifier circuit the double wave rectifier circuit. FIG. 12 is a characteristic diagram illustrating a change characteristic of a switching frequency with respect to a load change in the power supply circuit illustrated in FIG. 11.

Explanation of symbols

  1 Control circuit, 2 Oscillation / drive circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2, Q3, Q4 switching element, PIT isolation converter transformer, C1 primary side series resonant capacitor, Cp, Cp1, Cp2 primary side partial resonance Capacitor, N1 primary winding, N2 secondary winding, N2A, N2B secondary winding section, Do1 to Do4 (secondary side) rectifier diode, Co (secondary side) smoothing capacitor, Cc, Cc1, Cc2 capacitor

Claims (7)

Switching means formed with a switching element for switching by inputting a DC input voltage;
Switching driving means for switching and driving the switching element;
An insulating converter transformer formed by winding a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the primary winding;
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A rectifying circuit other than the double-wave rectifying circuit is provided, and the alternating voltage obtained in the secondary winding is rectified, and the rectified output is smoothed by the secondary side smoothing capacitor to obtain the secondary side DC output voltage. Secondary side DC output voltage generating means for generating
Constant voltage control means for performing constant voltage control on the secondary side DC output voltage by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. And
The insulating converter transformer has a gap length formed at a predetermined position of the core such that the coupling coefficient between the primary side and the secondary side is not more than a predetermined value.
A switching power supply circuit. The secondary side DC output voltage generating means is:
As the rectification operation, it is configured to perform full-wave rectification operation by a bridge rectification circuit,
The switching power supply circuit according to claim 1. The secondary side DC output voltage generating means is:
As the rectifying operation,
The secondary side smoothing capacitor is charged only during one half cycle of the alternating voltage excited by the secondary winding, and the secondary side DC output voltage is generated at a level corresponding to twice the alternating voltage level. Is configured to perform a double voltage half-wave rectification operation,
The switching power supply circuit according to claim 1. In the insulating converter transformer, a center tap is applied to the secondary winding to form a first secondary winding portion and a second secondary winding portion,
The secondary side DC output voltage generating means is configured as the rectifying operation.
The secondary side smoothing capacitor is charged in each half cycle of the alternating voltage excited by the secondary winding, and the first secondary winding unit and the first side are connected to both ends of the secondary side smoothing capacitor. Each of the two secondary windings is configured to perform a voltage doubler full-wave rectification operation that generates the secondary side DC output voltage at a level corresponding to twice the alternating voltage level obtained. ing,
The switching power supply circuit according to claim 1. The secondary side DC output voltage generating means is:
As the rectifying operation,
Configured to perform a quadruple voltage rectification operation adapted to generate the secondary side DC output voltage at a level corresponding to four times the alternating voltage level excited in the secondary winding;
The switching power supply circuit according to claim 1.   2. The switching power supply circuit according to claim 1, wherein the switching means has two switching elements connected by a half-bridge coupling method.   The switching power supply circuit according to claim 1, wherein the switching means includes four switching elements connected by a full bridge coupling method.

Images