JP2006262680A - Switching power circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain high efficiency, by realizing the tackling to the wide range of a voltage resonance type converter. <P>SOLUTION: A secondary parallel resonance circuit and a secondary series resonance circuit are combined with a voltage resonance type converter and are put in loose coupling of approximately 0.6 or smaller in integrated coupling coefficient kt of an insulated converter transformer PIT. As a result, its constant voltage control property is made sharp unimodal characteristics, thereby reducing the control range of switching frequency required for stabilization. Since the secondary series resonance circuit is combined, a satisfactory characteristics can be obtained for the power conversion efficiency. Moreover, further efficiency improvement is achieved by the setting of primary parallel resonance frequency fo1, the secondary frequency fo2, and the secondary series resonance frequency fo3. The peak level of the switching voltage is suppressed due to the drop in the integrated coupling coefficient kt. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit including a voltage resonance type converter.

As a so-called soft switching power supply of a resonance type, a current resonance type and a voltage resonance type are widely known. In the present situation, a current resonance type converter using a two-bridge switching element and a half-bridge coupling system is widely used on the background of practical application.
However, with the background of, for example, the improvement of the characteristics of high voltage switching elements, for example, the problem of withstand voltage in the practical use of voltage resonant converters has been cleared. In addition, a single-ended voltage resonant converter with a single switching element is advantageous in terms of input feedback noise and DC output voltage line noise components compared to a single-current resonant forward converter. It is also known that.

FIG. 12 shows a configuration example of a switching power supply circuit including a voltage resonance type converter by a single end system.
In the switching power supply circuit shown in this figure, the commercial AC power supply AC is rectified and smoothed by a rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci, and a rectified and smoothed voltage Ei is generated as a voltage across the smoothing capacitor Ci. Yes.
Note that the line of the commercial AC power supply AC is provided with a noise filter that includes a pair of common mode choke coils CMC and two across capacitors CL and removes common mode noise.

  The rectified and smoothed voltage Ei is input to the voltage resonant converter as a DC input voltage. As described above, this voltage resonance type converter adopts a single-end configuration including one switching element Q1. In this case, the voltage resonance type converter is a separately excited type, and the switching element Q1 of the MOS-FET is switched by the oscillation / drive circuit 2.

  A MOS-FET body diode DD is connected in parallel to the switching element Q1. A primary side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching element Q1.

  The primary side parallel resonant capacitor Cr forms a primary side parallel resonant circuit (voltage resonant circuit) with the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. The primary side parallel resonance circuit can obtain a voltage resonance type operation as the switching operation of the switching element Q1.

  The oscillation / drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1 in order to drive the switching element Q1. Thereby, the switching element Q1 performs a switching operation at a switching frequency corresponding to the cycle of the drive signal.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As the structure of the insulating converter transformer PIT, for example, as shown in FIG. And the bobbin B10 formed with resin etc. by the shape divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side is provided. The primary winding N1 is wound around one winding portion of the bobbin B10. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B10 on which the primary side winding and the secondary side winding are wound in this way to the EE type core (CR11, CR12), the primary side winding and the secondary side winding are different from each other. By the winding region, the EE type core is wound around the central magnetic leg. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, whereby k = 0.80 between the primary side and the secondary side. A coupling coefficient k of about ˜0.85 is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained.

  In FIG. 12, one end of the primary winding N1 of the insulating converter transformer PIT is inserted between the positive terminal of the switching element Q1 and the smoothing capacitor Ci, so that the switching output of the switching element Q1 is transmitted. It is like that. An alternating voltage induced by the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.

In this case, a secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. As a result, a secondary side parallel resonance circuit (voltage resonance circuit) is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2.
In addition, a half-wave rectifier circuit is formed by connecting a rectifier diode Do1 and a smoothing capacitor Co to the secondary side parallel resonant circuit as shown in the figure. This half-wave rectifier circuit uses, as a voltage across the smoothing capacitor Co, the secondary side DC output voltage Eo at a level corresponding to the same voltage as the alternating voltage V2 obtained in the secondary winding N2 (secondary side parallel resonant circuit). Generate. The secondary side DC output voltage Eo is supplied to the load and is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side DC output voltage Eo input as a detection voltage to the oscillation / drive control circuit 2.
The oscillation / drive circuit 2 adjusts the switching element Q1 so that the secondary side DC output voltage Eo becomes constant at a predetermined level in accordance with the level of the secondary side DC output voltage Eo indicated by the input detection output. Controls the switching operation. That is, a drive signal for obtaining a switching operation to be controlled is generated and output. Thereby, stabilization control of the secondary side DC output voltage Eo is performed.

14 and 15 show experimental results for the power supply circuit having the configuration shown in FIG. In the experiment, as a condition of VAC = 100 V corresponding to the AC 100 V system, the main part of the power supply circuit in FIG. 12 is set as follows.
For the isolated converter transformer PIT, EER-35 is selected for the core, and the gap length of the central magnetic leg is set to 1 mm. The number of turns T (number of turns) of the primary winding N1 and the secondary winding N2 was set to N1 = 43T and N2 = 43T, respectively. For the coupling coefficient k of the insulating converter transformer PIT, k = 0.81 was set.
Further, a primary side parallel resonant capacitor Cr = 6800 pF and a secondary side parallel resonant capacitor C2 = 0.01 μF were selected. Accordingly, the resonance frequency fo1 = 175 kHz of the primary side parallel resonance circuit and the resonance frequency fo2 = 164 kHz of the secondary side parallel resonance circuit are set.
The rated level of the secondary side DC output voltage Eo is 135V, and the corresponding load power is the maximum load power Pomax = 200 W to the minimum load power Pomin = 0 W.

First, FIG. 14 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG. 12 by the switching cycle of the switching element Q1, and FIG. 14 (a) shows the switching voltage at the maximum load power Pomax = 200W. V1, switching current IQ1, primary winding current I1, secondary winding voltage V2, secondary winding current I2, and secondary side rectified current ID1 are shown. FIG. 14B shows the switching voltage V1, switching current IQ1, primary winding current I1, secondary winding voltage V2, secondary winding current I2, and secondary side rectified current ID1 when the minimum load power Pomin = 0 W. It is shown.
The switching voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a zero level in the period TON in which the switching element Q1 is turned on and becomes a sinusoidal resonance pulse in the period TOFF in which the switching element Q1 is turned off. The resonance pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is a voltage resonance type.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD). The switching current IQ1 is zero level during the period TOFF, and has a negative polarity by flowing in the forward direction with respect to the body diode DD during the turn-on. Then, it is inverted and then obtained as a waveform that flows between the drain and source of the switching element Q1 and increases until it is turned off. For this reason, a peak level is obtained at the turn-off timing as the switching current IQ1.
The primary winding current I1 flowing in the primary winding N1 is obtained by combining the current component flowing as the switching current IQ1 in the period TON and the current flowing in the primary side parallel resonant capacitor Cr in the period TOFF. It becomes a waveform.

Further, as the operation of the secondary side rectifier circuit, the rectified current ID1 flowing through the rectifier diode Do1 is set to 0 level by the waveform shown after the peak level is obtained when the rectifier diode Do1 is turned on at the maximum load power Pomax = 200 W. The voltage decreases and flows with a waveform that becomes 0 level during the off period of the rectifier diode Do1. Note that, when the minimum load power Pomin = 0 W, the level is 0 even during the on period.
In this case, the secondary winding voltage V2 is a voltage obtained in the parallel circuit of the secondary winding N2 // secondary parallel resonant capacitor C2, and the secondary rectifier diode Do1 is conductive. Corresponding to the ON period, it is clamped by the level of the secondary side DC output voltage Eo, and a sine waveform in the negative polarity direction is obtained in the OFF period of the secondary side rectifier diode Do1. Further, the secondary winding current I2 flowing through the secondary winding N2 is a combination of the rectified current ID1 and the current flowing through the secondary parallel resonant circuit (N2 (L2) // C2). It flows by.

FIG. 15 shows the switching frequency fs with respect to the load fluctuation, the ON period TON, the OFF period TOFF, and the AC → DC power conversion efficiency (ηAC → DC) with respect to the load variation for the power supply circuit shown in FIG.
First, looking at AC → DC power conversion efficiency (ηAC → DC), it is 90% or more in the range of load power Po = 75 W to 200 W. As a voltage resonance type converter, it is known that a single-ended type in which the switching element Q1 is one stone can obtain a good result in terms of power conversion efficiency.

Further, depending on the switching frequency fs, the ON period TON, and the OFF period TOFF shown in FIG. 15, the switching operation as the constant voltage control characteristic with respect to the load fluctuation in the power supply circuit of FIG. 12 is shown. In this case, the switching frequency fs is controlled such that the switching frequency becomes higher as the load tends to be lighter. Further, with respect to the ON period TON and the OFF period TOFF, the OFF period TOFF is almost constant with respect to the load variation, whereas the ON period TON is shortened as the light load tends to be reduced. Yes. In other words, the power supply circuit shown in FIG. 12 variably controls the switching frequency so as to shorten the on-period TON in response to, for example, a light load tendency while keeping the off-period TOFF constant. become.
Thus, the inductive impedance obtained by providing the primary side parallel resonant circuit and the secondary side parallel resonant circuit is varied by variably controlling the switching frequency. Depending on the variable inductive impedance, the amount of power transmitted from the primary side to the secondary side and the amount of power transmitted from the secondary side parallel resonant circuit to the load change, and the secondary side DC output voltage Eo The level of is variable. As a result, the secondary side DC output voltage Eo is stabilized.

FIG. 16 schematically shows the constant voltage control characteristics of the power supply circuit shown in FIG. 12 by the relationship between the switching frequency fs (kHz) and the secondary side DC output voltage Eo.
Here, assuming that the resonance frequency of the primary side parallel resonance circuit is fo1 and the resonance frequency of the secondary side parallel resonance circuit is fo2, the circuit shown in FIG. The secondary parallel resonance frequency fo2 is low.
In addition, assuming constant voltage control characteristics with respect to the switching frequency fs under the condition of a certain AC input voltage VAC, as shown in the figure, the resonance impedance under the resonance impedance corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit is shown. The constant voltage control characteristics at the time of maximum load power Pomax / minimum load power Pomin are shown as characteristic curves A and B, respectively, and are maximum under the resonance impedance corresponding to the resonance frequency fo2 of the secondary side parallel resonance circuit. The constant voltage control characteristics at the time of load power Pomax / minimum load power Pomin are shown by characteristic curves C and D, respectively.
Furthermore, when the primary side parallel resonance frequency and the secondary side parallel resonance circuit are provided as in the circuit of FIG. 12, an intermediate resonance frequency fo exists between the resonance frequencies fo1 and fo2. The resonance impedance characteristic based on the relationship between the intermediate resonance frequency fo and the switching frequency fs is indicated by the characteristic curve E at the maximum load power Pomax, and is indicated by the characteristic curve F at the minimum load power Pomin.
In the voltage resonance type converter including the secondary side parallel resonance circuit, the level of the secondary side DC output voltage Eo is determined by the resonance impedance characteristic with respect to the switching frequency fs of the intermediate resonance frequency fo. Further, the voltage resonance type converter shown in FIG. 12 adopts a so-called lower side control method in which the switching frequency fs is variably controlled in a frequency region lower than the center resonance frequency fo.
Then, under the characteristics shown as characteristic curves E and F corresponding to the intermediate resonance frequency fo in FIG. 16, the rated level of the secondary side DC output voltage Eo (FIG. 12) is controlled by the switching frequency control corresponding to the lower side control. In the case of this circuit, when trying to achieve a constant voltage with 135V) as a target value, the variable range (necessary control range) of the switching frequency fs necessary for this is the section indicated by Δfs. In other words, the secondary side DC output voltage Eo becomes the rated level tg by changing the switching frequency so as to be a required value according to the load fluctuation in the frequency range corresponding to the section indicated by Δfs. It is controlled in this way.

JP 2000-152617 A

  By the way, against the background of diversification of various electronic devices, the power supply circuit is also required to be compatible with a so-called wide range that operates in accordance with any commercial AC power input of AC100V system and AC200V system. .

As described above, the power supply circuit having the configuration shown in FIG. 12 operates so as to stabilize the secondary side DC output voltage Eo by switching frequency control, and the switching frequency variable range (for this purpose) The necessary control range is indicated by Δfs described in FIG.
The power supply circuit shown in FIG. 12 is adapted to handle a relatively wide range of load fluctuations from 200 W to 0 W. In the power supply circuit of FIG. 12, the actual necessary control range corresponding to this load condition is fs = 117.6 kHz to 208.3 kHz and Δfs = 96.7 kHz, which is a relatively wide range.

Here, as a matter of course, the level of the secondary side DC output voltage Eo also varies as the level of the AC input voltage VAC changes. That is, the level of the secondary side DC output voltage Eo similarly increases and decreases according to the level increase and decrease of the AC input voltage VAC.
Therefore, in response to fluctuations in the AC input voltage in the wide range from the AC100 system to the AC200V system, for example, in the case of dealing with fluctuations in a single range of only the AC100 system or AC200V system, the secondary side It can be said that the level fluctuation of the DC output voltage Eo1 also increases. In order to perform the constant voltage control operation corresponding to the level fluctuation of the secondary side DC output voltage Eo1 expanded in this way, the above-mentioned range of 117.6 kHz to 208.3 kHz is set in the direction of higher frequency. An expanded, wider range of required control is required.

However, as an IC (oscillation / drive circuit 2) for driving a switching element at present, the upper limit of the drive frequency that can be handled is about 200 kHz. In addition, even if an IC capable of driving at a high frequency as described above is developed, the power conversion efficiency is remarkably lowered when the switching element is driven at a high frequency, so that it can be used as a power supply circuit. Practically impossible.
From this, it can be seen that, for example, it is very difficult to realize the wide range by the configuration of the power supply circuit shown in FIG.

  For this reason, as a switching power supply circuit equipped with a resonant converter, when an operation corresponding to a wide range is to be realized, for example, according to an AC100V system / 200V system commercial AC power supply input, a primary side switching converter The configuration is switched between half bridge and full bridge. Alternatively, the operation of the rectifier circuit that performs the rectification operation for the commercial AC power supply AC is switched by full-wave rectification / double voltage rectification according to the AC 100V / 200V commercial AC power input. .

However, when the circuit configuration is switched between the AC100V system and the AC200V system, the following problems occur.
For example, for switching according to such a commercial AC power supply level, a threshold value (for example, 150V) for the input voltage is set, and circuit switching corresponding to the AC 200V system is performed when the threshold is exceeded, and AC 100V system is coupled when the threshold is decreased. However, if only such switching is performed, the AC 100 system is also supported for a temporary decrease in AC input voltage due to, for example, a momentary power failure at the time of AC 200 V input. There is a risk of switching. In other words, for example, in the switching configuration of the rectifying operation, the AC voltage is switched to the voltage doubler rectifier circuit even though the input is AC200V, and the switching element is overvoltage-resistant. May be destroyed.

Therefore, in order to prevent the above-described malfunction, in practice, not only the DC input voltage of the main switching converter but also the DC input voltage of the converter circuit on the standby power supply side is detected. To be.
However, detecting the converter circuit on the standby power supply side in this way means mounting a comparator IC or the like for comparing the reference voltage and the input voltage, but this increases the number of components. Therefore, an increase in circuit manufacturing cost and an increase in circuit board size are promoted.

  In addition, detecting the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunction in this way means that it can only be actually used if it is an electronic device having a standby power supply in addition to the main power supply. become. That is, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and there is a problem that the use range is narrowed accordingly.

Further, in the configuration for switching between the half bridge and the full bridge, it is necessary to include at least four switching elements in order to enable the full bridge configuration. That is, if switching is not necessary, a half bridge that requires only two switching elements must be added in this case.
Further, even when the rectifying operation is switched, two smoothing capacitors Ci must be provided in order to obtain a double voltage rectifying operation. That is, one smoothing capacitor Ci has to be added as compared with a configuration in which only full-wave rectification is performed.
Also in these respects, the wide-range configuration with circuit switching as described above causes an increase in circuit manufacturing cost and an increase in the size of the power circuit board. In particular, in the configuration of switching the rectifying operation, the smoothing capacitor Ci and the like enter a large category among the components constituting the power supply circuit, so that the increase in the substrate size is further promoted.

Further, as described above, another problem caused by the wide control range of the switching frequency is that the high-speed response characteristic for stabilization of the secondary side DC output voltage Eo is deteriorated. Is mentioned.
Particularly in recent electronic devices, for example, the load power may be a so-called switching load condition in which the load power changes instantaneously between the maximum load and no load according to on / off of various driving units. In response to this, it is necessary for the power supply circuit side to perform constant voltage control of the secondary side DC output voltage Eo in accordance with the load power which fluctuates at such a high speed and widely.
However, if the switching frequency control range is wide as described above, it takes a lot of time to change to the switching frequency necessary for constant voltage control corresponding to the load changing between the maximum value and the minimum value. Will be required. That is, the responsiveness of constant voltage control becomes dull.

Further, the power supply circuit shown in FIG. 12 has a configuration including a primary side voltage resonance converter, and the power supply circuit having such a configuration has a characteristic that tends to be advantageous in terms of power conversion efficiency. As explained. However, in consideration of recent energy circumstances, environmental circumstances, and the like, electronic devices are required to have higher power conversion efficiency characteristics. In connection with this, the power supply circuit itself mounted in the electronic device is in a situation where further improvement in power conversion efficiency is required.
In particular, in the circuit shown in FIG. 12, as understood from the characteristics of FIG. 15, the power conversion efficiency is not improved under a light load condition, and the efficiency improvement in this region is the overall efficiency. It is important for improvement.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, a switching means formed by including a switching element that performs switching by inputting a DC input voltage, and a switching drive means that switches the switching element.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the switching output obtained in the primary winding are wound. Insulated converter transformer formed.
Further, a primary side parallel resonance circuit is provided which is formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of the primary side parallel resonance capacitor, and the operation of the switching means is a voltage resonance type.
Also, by connecting a secondary side parallel resonant capacitor in a parallel relationship with the secondary winding of the insulating converter transformer, a leakage inductance component including the secondary winding and the secondary side parallel resonant capacitor And a secondary side parallel resonant circuit formed by the capacitance of the second side.
Further, by connecting a secondary side series resonant capacitor in a series relationship with the secondary winding of the insulating converter transformer, a leakage inductance component including the secondary winding and the secondary side series resonant capacitor And a secondary side series resonance circuit formed by the capacitance of the second side.
A secondary-side DC output voltage generator configured to generate a secondary-side DC output voltage by performing a rectifying operation based on an alternating voltage induced in the secondary winding of the insulating converter transformer; Constant voltage control means for controlling the switching drive means according to the level of the secondary side DC output voltage and changing the switching frequency of the switching means to perform constant voltage control on the secondary side DC output voltage; Is provided.
In addition, the insulating converter transformer has an electromagnetic coupling type resonance circuit formed such that the coupling coefficient between the primary side and the secondary side includes at least the primary side parallel resonance circuit and the secondary side parallel resonance circuit. As for the output characteristics for the input of the frequency signal having the switching frequency, a single peak characteristic is obtained, and the ratio of the ON period of the switching element at the time of AC input voltage input of a predetermined level or more is less than a predetermined It is set to a predetermined value that is considered loosely coupled.
Then, the power conversion efficiency of a certain level or more is obtained at least under a predetermined load condition, the resonance frequency of the primary side parallel resonance circuit, the resonance frequency of the secondary side parallel resonance circuit, and the secondary side series resonance. The circuit is configured by setting the resonance frequency of the circuit.

The power supply circuit having the above configuration employs a configuration as a voltage resonance type converter including a primary side parallel resonance circuit on the primary side and a secondary side parallel resonance circuit and a secondary side series resonance circuit on the secondary side. By adopting a configuration in which the resonance circuits are combined on the primary side and the secondary side in this way, a coupled resonance circuit based on electromagnetic coupling of the insulating converter transformer is formed. In addition, by making the insulating converter transformer loosely coupled with a predetermined coupling coefficient, a steep single peak characteristic is obtained as an output characteristic for a frequency signal (switching output) of a switching frequency that is an input to the coupled resonance circuit. It becomes possible. As a result, the variable range (necessary control range) of the switching frequency required for stabilizing the secondary side DC output voltage can be reduced.
Further, although the secondary side series resonance circuit is combined, the combination of the secondary side series resonance circuit with respect to the primary side parallel resonance circuit can achieve high efficiency even under light load conditions. However, this combination cannot perform a stable ZVS (zero voltage switching) operation at an intermediate load, and an abnormal oscillation operation occurs, which is not a practical level. On the other hand, in the present invention, as described above, the coupling coefficient of the insulating converter transformer is reduced to a value that is a predetermined loose coupling, so that a stable ZVS operation can be realized and put into practical use. .
Further, by reducing the value of the coupling coefficient, the ratio of the ON period, particularly when an AC input voltage exceeding a predetermined level is input, becomes small as a characteristic of the OFF period and the ON period of the switching element with respect to the load fluctuation. Became clear. In other words, by setting the coupling coefficient to a value that is a predetermined loose coupling as described above, the ratio of the ON period when an AC input voltage of a predetermined level or more is input can be set to a predetermined value or less. As a result, the peak level of the switching voltage applied to the switching element can be suppressed.
In addition, as the present invention, by setting the resonance frequency of the primary side parallel resonance circuit, the resonance frequency of the secondary side parallel resonance circuit, and the resonance frequency of the secondary side series resonance circuit, under a predetermined load condition, A power conversion efficiency characteristic exceeding a certain level is obtained.

  In this way, the present invention provides a variable control range (necessary for switching frequency required for constant voltage control) of a voltage resonant converter that combines a primary side parallel resonant circuit, a secondary side parallel resonant circuit, and a secondary side series resonant circuit. The control range is reduced. As a result, the voltage resonance type switching converter can be easily realized in a wide range only by the switching frequency control.

  Moreover, although it is set as the voltage resonance type | mold converter provided with a secondary side series resonance circuit, the improvement of power conversion efficiency is achieved compared with the case where only a secondary side parallel resonance circuit is combined by this. Further, even when the secondary side series resonance circuit is provided as described above, the present invention can obtain a stable ZVS operation as described above, and can be put to practical use without any trouble.

  Furthermore, the ratio of the ON period of the switching element is reduced particularly when an AC input voltage of a predetermined level or higher is input by reducing the coupling coefficient to a value that is a predetermined loose coupling as described above. This suppresses the peak level of the switching voltage applied to the switching element.

  Further, according to the present invention, by setting the resonance frequency of the primary side parallel resonance circuit, the resonance frequency of the secondary side parallel resonance circuit, and the resonance frequency of the secondary side series resonance circuit, it is possible to cope with a load condition of a predetermined load power. Thus, power conversion efficiency exceeding a certain level is obtained. The voltage resonance type converter originally has high power conversion efficiency characteristics. However, according to the present invention, a power supply circuit including a voltage resonance type converter can be provided having better power conversion efficiency characteristics.

  As a basic configuration for realizing the wide range and high efficiency as described above, a required coupling coefficient is obtained for an insulating converter transformer with respect to a voltage resonant converter having at least a secondary side parallel resonant circuit. In addition, a secondary side series resonant capacitor for forming a secondary side series resonant circuit may be added. Therefore, the cost increases due to the increase in the number of parts, the circuit becomes larger, the weight increases, etc. There is also a merit that this can be realized without accompanying.

Hereinafter, the best mode (embodiment) for carrying out the present invention will be described.
The circuit diagram of FIG. 1 shows a configuration example of a switching power supply circuit as an embodiment of the present invention. The power supply circuit shown in this figure employs a basic configuration as a voltage resonance type switching converter by a single end system.
In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by the bridge rectifier circuit Di, and the rectified output is charged to the smoothing capacitor Ci. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter.

  In this figure, the switching converter that performs a switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage is formed as, for example, a single-ended voltage resonant converter including a single switching element Q1. In this case, a high breakdown voltage MOS-FET is selected as the switching element Q1. In this case, the driving method of the voltage resonance type converter is a separately excited type in which the switching element is switched by the oscillation / drive circuit 2.

A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.
The drain of the switching element Q1 is connected to the winding start end of a primary winding N1 of an insulating converter transformer PIT described later. The winding end end of the primary winding N1 is connected to the positive terminal of the smoothing capacitor Ei. Therefore, in this case, the DC input voltage (Ei) is supplied to the switching element Q1 through the serial connection of the primary winding N1. The source of the switching element Q1 is connected to the primary side ground.

  Since the MOS-FET is selected for the switching element Q1 in this case, a body diode DD is incorporated so as to be connected in parallel with the source and drain as shown in the figure. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
The primary side parallel resonance capacitor Cr has a primary side parallel resonance circuit (voltage resonance circuit) for the switching current flowing in the switching element Q1 by its own capacitance and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. Form. The primary side parallel resonance circuit performs a resonance operation, whereby a voltage resonance type operation is obtained as the switching operation of the switching element Q1. Accordingly, a sinusoidal resonance pulse waveform is obtained during the off period as the switching voltage V1, which is the voltage across the switching element Q1 (drain-source voltage).

  The oscillation / drive circuit 2 is a drive that is a gate voltage for switching the MOS-FET based on an oscillation circuit and an oscillation signal obtained by the oscillation circuit in order to drive the switching element Q1 by, for example, separate excitation. A signal is generated and applied to the gate of the switching element Q1. Thereby, the switching element Q1 continuously performs on / off operation according to the switching frequency corresponding to the cycle of the drive signal. That is, a switching operation is performed.

The insulating converter transformer PIT transmits the switching output of the primary side switching converter to the secondary side in a state where the primary side and the secondary side are galvanically insulated.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core in which E type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other. The E-type cores CR1 and CR2 are cores each having an E-shaped cross section as shown in the figure.
And the bobbin B formed with resin etc. is provided by the shape divided | segmented so that it might mutually become independent about the primary side and the secondary side winding part. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound.

  Furthermore, a gap G is formed as shown in the figure with respect to the central magnetic leg of the EE type core. The length of the gap G is an element in determining the coupling coefficient between the primary side and the secondary side. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

In addition, in the insulating converter transformer PIT in this case, the central portion of the two outer magnetic legs of the EE core, that is, the winding position of the primary winding N1 and the winding position of the secondary winding N2 The ferrite sheet cores FSC1 and FSC2 are provided so as to sandwich each of the portions corresponding to the boundary. These ferrite sheet cores FSC1 and FSC2 each have a predetermined thickness Ln2.
Further, as can be seen from the names, the ferrite sheet cores FSC1 and FSC2 in this case are made of the same ferrite as the E-type cores CR1 and CR2, and are provided between the outer magnetic legs. In addition, the ferrite sheet cores FSC1 and FSC2 further have a protruding portion (magnetic path generating portion) with a predetermined length Ln3 from the inner end surface portion of the outer magnetic leg to the inner magnetic leg side of the EE core. Provided. As a result, the end portions of the ferrite sheet cores FSC1 and FSC2 are inserted into the portion between the winding portions of the primary winding N1 and the secondary winding N2 in the bobbin B by a predetermined length. Become.

Here, for example, in the case of a simple EE type core structure in which the ferrite sheet cores FSC1 and FSC2 are not inserted as in the insulating converter transformer PIT shown in FIG. 13, the magnetic flux φ1 in FIG. A magnetic path is formed as shown by φ2. In FIG. 3, the bobbin B is omitted for easy understanding.
As can be seen from this figure, the magnetic paths of the magnetic fluxes φ1 and φ2 pass through the outer magnetic legs so as to straddle the primary winding N1 side and the secondary winding N2 side. For this reason, in the insulating converter transformer PIT shown in FIG. 13, the degree of coupling between the primary winding N1 and the secondary winding N2 is correspondingly high, and the coupling coefficient k set in the circuit of FIG. = 0.80 was obtained by forming a gap G having a gap length of about 1.0 mm on the inner magnetic leg.

On the other hand, when the ferrite sheet cores FSC1 and FSC2 are provided as shown in FIG. 2, the E-type cores CR1 and CR2 are ferrites of the same material as the ferrite sheet cores FSC1 and FSC2. Substantially, as shown in FIG. 3 (b), the shape of the EE core is assumed to have a shape in which the center part of the outer magnetic leg protrudes toward the center part side of the inner magnetic leg. Can do. The shape of such an EE type core is the same as that of the EE type core shape of FIG. 3A by the length Ln3 of the protruding portion (magnetic path generation site) and the central portion of the outer magnetic leg The spatial distance from the center part of the magnetic leg is shortened.
As these portions approach, the insulating converter transformer PIT shown in FIG. 3B generates magnetic fluxes indicated by φ11 and φ12 indicated by broken lines in FIG. 3B. The components of the magnetic fluxes φ11 and φ12 increase as the thickness Ln2 of the protruding portions of the ferrite sheet cores FSC1 and FSC2 increases and the amount of magnetic flux passing through the protruding portions of the ferrite sheet cores FSC1 and FSC2 increases. On the other hand, the component amount of the magnetic fluxes φ1 and φ2 decreases.

Here, the magnetic paths of the magnetic fluxes φ11 and φ12 are formed respectively corresponding to the primary winding N1 side and the secondary winding N2 side. Therefore, while the component amounts of the magnetic fluxes φ1 and φ2 described above decrease, the magnetic fluxes φ11 and φ12 tend to increase, which means that the degree of coupling between the primary winding N1 and the secondary winding N2 decreases. become.
That is, the coupling coefficient can be changed also by this ferrite sheet core (FSC), and the coupling coefficient can be set low according to the thickness Ln2 as described above.

In this way, the fact that the coupling coefficient is lowered by providing the ferrite sheet cores FSC1 and FSC2 means that according to the configuration of this example, a lower value is set as the coupling coefficient without increasing the gap G. be able to.
In the case of the present embodiment, specifically, the length of the gap G is set to about 1.0 mm as in the prior art, and the above-described thickness Ln2 of the ferrite sheet cores FSC1 and FSC2 is set to 1.5 mm. As a total coupling coefficient (hereinafter referred to as a total coupling coefficient kt) in the insulating converter transformer PIT, about kt = 0.6 is obtained.

As can be understood from the above description, the setting of the total coupling coefficient kt is such that a magnetic path generation site similar to the ferrite sheet core (FSC1, FSC2) is formed with respect to the basic shape of the EE type core. However, according to this, for example, an E-type core is molded so that an actual core shape as shown in FIG. 3B can be obtained without using a ferrite sheet core. Thus, by combining these, it is possible to set the total coupling coefficient kt similar to the case where the ferrite sheet core is used.
However, in the present embodiment, the ferrite sheet cores (FSC1, FSC2) are used as outer magnetic legs in consideration of the fact that the manufacturing process is simpler than molding the E-type core as described above. It is the form which is inserted.

In the case of the present embodiment, the insulating converter transformer PIT may have the structure shown in FIG.
In FIG. 4A, the insulating converter transformer PIT in this case is such that the magnetic legs of the E-type core CR3 and E-type core CR4 made of a ferrite material whose cross-sectional shape is E-shaped are opposed to each other. EE type core combined with
However, in this case, each of the two bobbins B5 for independently winding the primary winding (N1) and the secondary winding (N2) is formed on the EE core (CR3 / CR4). By attaching to each of the two outer magnetic legs, the primary side winding is wound around one outer magnetic leg and the secondary side winding is wound around the other outer magnetic leg as shown in the figure. It is supposed to be.
Also in this case, a gap G is formed with respect to the central magnetic leg of the EE core. Also in this case, the gap G can be formed by making the central magnetic legs of the E-type cores CR3 and CR4 shorter than the two outer magnetic legs.

In the insulating converter transformer PIT having the structure shown in FIG. 4A, a magnetic path as shown in FIG. 4B is formed. In FIG. 4B, the bobbin B5 is omitted for convenience of explanation.
First, the primary side winding and the secondary side winding are wound around different outer magnetic legs as described above, and as a magnetic path in this case, as shown in FIG. The magnetic path by the magnetic flux φ3 formed so as to straddle the two outer magnetic legs on the core CR3 side and the E-type core CR4 side, and the magnetic flux φ13 and the magnetic flux φ14 passing from the outer magnetic leg side to the inner magnetic leg side, respectively. To form a magnetic path.
As shown, the magnetic flux φ3 is formed so as to straddle the primary winding N1 side and the secondary winding N2 side.
One magnetic flux φ13, φ14 is formed corresponding to each of the primary winding N1 side and the secondary winding N2 side.

As described above, the magnetic fluxes φ13 and φ14 formed respectively corresponding to the primary winding N1 side and the secondary winding N2 side pass through the inner magnetic legs as described above. Therefore, in this case as well, it can be understood that the length of the gap G formed on the inner magnetic leg of the EE core is one element for setting the degree of coupling between the primary side and the secondary side.
However, according to the structure in this case, the degree of coupling between the primary side and the secondary side can be increased by decreasing the magnetic fluxes φ13 and φ14 by increasing the length of the gap G. On the contrary, by shortening the length of the gap G, the magnetic fluxes φ13 and φ14 are increased and lowered. In other words, in the case of the structure shown in FIG. 4, in reducing the coupling coefficient, the length of the gap G is made shorter. It is possible to reduce the value of the coupling coefficient without accompanying.

Thus, according to the structure of the insulating converter transformer PIT of the present embodiment, the coupling coefficient between the primary side and the secondary side can be reduced without increasing the gap G.
For example, when the coupling coefficient is decreased by increasing the gap G, the eddy current loss in the insulating converter transformer PIT increases. In this example, the gap G is increased when the coupling coefficient is decreased. By not having this, it is possible to suppress this eddy current loss and effectively suppress the decrease in efficiency.

  In the following, it is assumed that the configuration shown in FIG. 2 is adopted as the insulating converter transformer PIT, and the coupling coefficient on the primary side and the secondary side is indicated by the total coupling coefficient kt. For confirmation, the length of the gap G is set to be the same when setting the same coupling coefficient value regardless of the configuration of FIGS. 2 and 4. The effect of suppressing the eddy current loss can be obtained equally.

Returning to FIG.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the drain of the switching element Q1 as described above. As a result, the switching output of the switching element Q1 is transmitted to the primary winding N1, and an alternating voltage is generated in the primary winding N1.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2.
A secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. Thus, a secondary side parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2. This secondary side parallel resonance circuit performs a resonance operation in accordance with a rectification operation of a secondary side rectifier circuit described later. That is, the voltage resonance operation can be obtained on the secondary side as well as the primary side.

  Further, in the embodiment, the secondary side series resonant capacitor C3 is connected in series with the secondary winding N2. By connecting the secondary side series resonance capacitor C3, a series resonance operation (current resonance operation) is obtained together with the parallel resonance operation on the secondary side in this case.

In this case, the secondary side rectifier circuit is connected to the secondary winding N2 in which the secondary side parallel resonant capacitor C2 is connected in parallel and the secondary side series resonant capacitor C3 is connected in series as described above. , A rectifier diode Do1, a rectifier diode Do2, and a smoothing capacitor Co are connected as shown in FIG.
As a connection mode of the voltage doubler half-wave rectifier circuit, first, the anode of the rectifier diode Do1 is connected to the winding end end side of the secondary winding N2 through the series connection of the secondary side series resonant capacitor C3. The cathode of the rectifier diode Do2 is connected. Further, the cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The winding start end of the secondary winding N2, the anode of the rectifier diode Do2, and the negative terminal of the smoothing capacitor Co are connected to the secondary side ground.

The rectification operation of the double voltage half-wave rectifier circuit formed in this way is as follows.
First, in a half cycle corresponding to one polarity of the alternating voltage (V2) excited in the secondary winding N2, a forward voltage is applied to the rectifier diode Do2, so this period corresponds to this period. The rectifier diode Do2 becomes conductive, and an operation of charging the rectified current to the secondary side series resonant capacitor C3 is obtained. As a result, a voltage across the secondary side series resonant capacitor C3 having a level corresponding to the same multiple of the alternating voltage level induced in the secondary winding N2 is generated. In the next half cycle corresponding to the other polarity of the alternating voltage, a forward voltage is applied to the rectifier diode Do1 to conduct. At this time, the smoothing capacitor Co is charged with a potential obtained by superimposing the potential of the alternating voltage and the voltage across the secondary side series resonance capacitor C3.
As a result, the secondary side DC output voltage Eo having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co. In this rectification operation, the smoothing capacitor Co is charged only in correspondence with one half cycle of the alternating voltage excited in the secondary winding N2. That is, a rectifying operation as a double voltage half wave is obtained.
The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.
In such a voltage doubler half-wave rectifier circuit, the rectifier diode Do1 and the rectifier diode Do2 perform on / off operations at a relatively high frequency according to the switching frequency, so that a high-speed type (high-speed recovery type) is used. Is selected.

The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching element Q1 is driven such that the switching frequency is varied in accordance with the input detection output of the control circuit 1.
When the switching frequency of the switching element Q1 is variably controlled, the resonance impedance of the primary side and the secondary side in the power supply circuit changes and is transmitted from the primary winding N1 to the secondary winding N2 side of the insulating converter transformer PIT. And the amount of power to be supplied from the secondary side rectifier circuit to the load changes. As a result, an operation of controlling the level of the secondary side DC output voltage Eo so that the level fluctuation of the secondary side DC output voltage Eo is canceled is obtained. That is, the secondary side DC output voltage Eo is stabilized.

  Here, in the case of the present embodiment, the secondary winding N2 is connected to the secondary side series resonant capacitor C3 in addition to the secondary side parallel resonant capacitor C2, so that a series resonant operation is also obtained. Yes. The constant voltage control operation for the secondary side DC output voltage Eo in such a case includes the variable control of the switching frequency itself as described above and the ON period of the switching element Q1, as will be described later with reference to FIG. TON and variable control of the time length of the off period TOFF are performed in combination.

The actual configuration of the power supply circuit of the present example according to the circuit configuration shown in FIG. 1 is configured by setting the main part as follows.
First, for the insulating converter transformer PIT, EER-35 was selected for the core, and a gap length of 1.0 mm was set for the gap G. Further, N1 = 63T and N2 = 25T are selected for the number of turns (number of turns) T of the primary winding N1 and the secondary winding N2, and kt = 0.57 is set for the total coupling coefficient kt of the insulating converter transformer PIT. Is set. At this time, the leakage inductance L1 of the primary winding N1 = 403 μH and the leakage inductance L2 of the secondary winding N2 = 86 μH.
Further, Cr = 6800 pF was selected for the capacitance of the primary side parallel resonant capacitor Cr. The resonance frequency fo1 = 96 kHz of the primary side parallel resonance circuit is set by the capacitance setting for the primary side parallel resonance capacitor Cr and the leakage inductance L1 of the primary winding N1 obtained by the structure of the insulating converter transformer PIT. Also, C2 = 0.033 μF is selected as the capacitance of the secondary parallel resonant capacitor C2, and the capacitance setting and the leakage inductance L2 of the secondary winding N2 obtained by the structure of the insulating converter transformer PIT The secondary parallel resonance frequency fo2 = 94.5kHz is set. That is, it can be said that the relationship of fo1≈fo2 is obtained as the relationship between the primary side resonance frequency fo1 and the secondary side parallel resonance frequency fo2.
Further, for the secondary side series resonance capacitor C3, C3 = 0.1 μF was set, thereby setting the secondary side series resonance frequency fo3 = 54.3 kHz.
From these, the relationship between the resonance frequency fo1, the secondary side parallel resonance frequency fo2, and the secondary side series resonance frequency fo3 in this case is fo1≈fo2> fo3 as shown in FIG. It is what has been.
Further, in particular, in the comparison between the primary side resonance frequency fo1 and the secondary side series resonance frequency fo3, fo1≈fo3 × 0.56, and the resonance frequency fo3 is approximately half the resonance frequency fo1. It is supposed to be.
Further, the corresponding load power is the maximum load power Pomax = 200 W, the minimum load power Pomin = 0 W (no load), and the rated level of the secondary side DC output voltage Eo is 135V.

  The waveform diagram of FIG. 5 shows the operation of the main part of the power supply circuit of FIG. 1 configured as described above by the switching cycle of the switching element Q1, and FIG. 5 (a) shows the maximum load power Pomax = 200W. Switching voltage V1, switching current IQ1, primary winding current I1, alternating voltage V2, secondary winding current I2, secondary side rectified current ID1, secondary side rectified current ID2 when AC input voltage VAC = 100V is shown It is. Further, FIG. 5B shows the above waveforms when the AC input voltage VAC = 230 V under the condition of the maximum load power Pomax = 200 W.

In FIG. 5, the switching voltage V1 is a voltage between the drain and the source of the switching element Q1, and the switching current IQ1 is a current flowing from the drain side to the switching element Q1 (and the body diode DD). The on / off timing of the switching element Q1 is indicated by the switching voltage V1 and the switching current IQ1. One switching cycle is divided into a period TON in which the switching element Q1 is to be turned on and a period TOFF in which the switching element Q1 is to be turned off. It becomes. The resonance pulse of the switching voltage V1 is obtained as a sinusoidal resonance waveform because the operation of the primary side switching converter is a voltage resonance type.
In this case, when the AC input voltage VAC = 100 V shown in FIG. 5A, the peak level (V1-p) of the switching voltage V1 is 640 Vp. On the other hand, when the AC input voltage VAC = 230V shown in FIG. 5B, V1-p = 770Vp.

The switching current IQ1 is 0 level in the period TOFF. When the period TON ends and the period TON starts and reaches the turn-on timing, first, the switching current IQ1 has a negative waveform by flowing through the body diode DD. The waveform reverses to the positive polarity by flowing from the drain to the source.
The waveform of the switching current IQ1 shown in this figure indicates that ZVS is properly performed.
When the AC input voltage VAC = 100V in FIG. 5A, the peak level of the switching current IO1 is 3.4 Ap, and when the AC input voltage VAC = 230 V shown in FIG. 5B, IQ1 = 4.0 Ap. Become.

  The primary winding current I1 is a current that flows through the primary winding N1, and is a combination of the current component that flows through the switching current IQ1 and the current that flows through the primary side parallel resonant capacitor Cr. The waveform of the period TOFF in the primary winding current I1 corresponds to the current waveform flowing in the primary side parallel resonant capacitor Cr.

The operation of the secondary side rectifier circuit is indicated by the alternating voltage V2, the secondary winding current I2, and the secondary side rectified currents ID1 and ID2.
Depending on the alternating voltage V2 induced in the secondary winding N2, as described above, the rectifier diodes Do1 and Do2 are turned on alternately corresponding to each half-cycle period of the alternating voltage V2.
The secondary side rectified currents ID1 and ID2 flow through the smoothing capacitor Co in an alternating manner by a half-wave sine wave shape as shown in the figure. The secondary winding current I2 flowing through the secondary winding N2 is obtained by synthesizing the secondary side rectified currents ID1 and ID2, and has a sine wave shape as shown in the figure. The sinusoidal shape of the secondary winding current I2 is obtained by the influence of the resonance operation of the series resonance circuit formed on the secondary side.
In the case of the present embodiment, the alternating voltage V2 is a voltage at which the induced voltage level of the secondary winding N2 is higher than the secondary DC output voltage Eo as shown in the figure at the maximum load power Pomax. Thus, a waveform clamped by a level of ½ of the secondary side DC output voltage Eo is obtained corresponding to a period in which the rectifier diode Do1 and the rectifier diode Do2 are turned on.
In this case, the peak levels of the secondary side rectified currents ID1 and ID2 at the AC input voltage VAC = 100 V shown in FIG. 5A are 5.0 Ap and 6.0 Ap, respectively, as shown in FIG. 5B. The peak levels of the secondary side rectified currents ID1 and ID2 when the AC input voltage VAC = 230 V are 6.0 Ap and 7.0 Ap, respectively.

FIG. 6 shows, as experimental results for the power supply circuit shown in FIG. 1, AC → DC power conversion efficiency (ηAC → DC) with respect to load fluctuation, switching frequency fs, time length of period TON and period TOFF, and switching voltage V1. The change characteristic of the peak level V1-p is shown.
As will be described later, according to the configuration of the power supply circuit of the present embodiment described with reference to FIG. 1, a wide-range configuration capable of operating in accordance with both the AC100V system and AC200V system inputs is realized. Is done. Accordingly, in FIG. 6, each characteristic when the AC input voltage VAC = 100 V is indicated by a solid line, and each characteristic when the AC input voltage VAC = 230 V is indicated by a broken line.

In FIG. 6, first, the switching frequency fs changes with a tendency to increase in accordance with a light load tendency. Further, when the AC input voltage VAC is 100V and 230V, the value is higher when VAC = 230V.
In these cases, the constant voltage control operation in this case is a control to increase the switching frequency fs as the secondary side DC output voltage Eo increases in accordance with the light load tendency and the AC input voltage increasing tendency. It is shown that.

Further, in FIG. 6, the time length of the period TON decreases as the light load tendency is increased, and conversely, the period TOFF increases as the light load tendency is increased. In addition, comparing the AC input voltage VAC = 100V and VAC = 230V, the period TON is shorter when the AC input voltage VAC = 230V, and the period TOFF is the AC input voltage VAC = 230V. It can be seen that is controlled so that the time length becomes longer.
Therefore, depending on the variable control of the time length of the period TON / period TOFF, the period TON is shortened as the secondary side DC output voltage Eo increases according to the light load tendency and the AC input voltage increasing tendency. , Indicating that the control for extending the period TOFF is being performed, and as a result, the control of the tendency to stabilize the level of the secondary side DC output voltage Eo is performed, similarly to the variable control of the switching frequency fs itself. I understand.
As described above, in the circuit of FIG. 1, the switching frequency variable control and the variable control of the time length of the period TON / TOFF are combined to stabilize the secondary side DC output voltage Eo.

In addition, according to the characteristic of the switching frequency fs in this case, it turns out that the switching frequency fs is continuously variably controlled with respect to the load fluctuation.
Here, if a configuration in which only the series resonant circuit is combined with the secondary side is assumed, the switching frequency fs is not continuously variably controlled with respect to the load variation, and the intermediate load region There is a region where the switching frequency fs is constant (for example, in a range of about Po = 120 W to 50 W).
As in the power supply circuit shown in FIG. 1, the constant voltage control operation in which the switching frequency fs is continuously variably controlled is compared with the characteristic diagram shown in FIG. It can be seen that this is closer to the case where only the parallel resonant circuit is combined. Therefore, the stabilizing operation of the secondary side DC output voltage Eo is not a combination of the primary side parallel resonant circuit and the secondary side series resonant circuit, but a combination of the primary side and secondary side parallel resonant circuits. It can be considered that the action by is dominant.

  In FIG. 6, the peak level V1-p of the switching voltage V1 tends to increase as the load of the AC input voltage VAC increases at 100V and 230V as the load increases. It can also be seen that under the same load power condition, the value is higher when the AC input voltage VAC = 230V.

Here, in each characteristic shown in FIG. 6, the specific value of the switching frequency fs corresponds to the range of the maximum load power Pomax = 200 W to the minimum load power Pomin = 0 W when the AC input voltage VAC = 100 V. When fs = 83.3kHz to 137.0kHz, Δfs = 53.7kHz, and the period TON / TOFF corresponding to the change in switching frequency is TON = 8.0μs to 3.6μs, TOFF = 4.0μs to 4.8μs. It was.
Also, when AC input voltage VAC = 230V, corresponding to the range of maximum load power Pomax = 200W to minimum load power Pomin = 0W, Δfs = 27kHz from fs = 126kHz to 153kHz, corresponding to this change in switching frequency In the period TON / TOFF to be performed, results are obtained that TON = 3.7 μs to 1.5 μs and TOFF = 4.2 μs to 5.0 μs.

The AC → DC power conversion efficiency (ηAC → DC) is as follows. When AC input voltage VAC = 100V, maximum load power Pomax = 200W, ηAC → DC = 92.1%, AC input voltage VAC = 230V A measurement result of ηAC → DC = 91.0% was obtained.
It should be noted that, as shown in FIG. 6, the AC → DC power conversion efficiency in this case is from maximum load power (Po = 200 W) to 25% load power (Po = 50 W vicinity). It is a characteristic that it gradually rises with respect to the load fluctuations and gradually declines as the load power drops thereafter.
This is a characteristic opposite to the characteristic that the efficiency improves as the load becomes heavy as in the case of the circuit of FIG. 12 in which only the parallel resonant circuit is formed on the secondary side (see FIG. 15). Thus, it can be seen that the efficiency at a load of 25% or less is clearly improved as compared with the case of the circuit of FIG.
Specifically, in the present embodiment, particularly when the AC input voltage VAC = 100 V, the characteristic is high efficiency that becomes ηAC → DC = 90% or more with respect to the variation range of the load power Po = 200 W to 25 W. ing.

Here, regarding the characteristics of the power supply circuit of FIG. 1 shown in FIG. 6, first, the characteristics regarding the switching frequency fs are compared with those of the power supply circuit of FIG.
In the power supply circuit of FIG. 12, with an input of AC input voltage VAC = 100 V, fs = 117.6 kHz to 208.3 kHz and Δfs = 96.7 kHz with respect to fluctuations of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W. It had been.
On the other hand, in the power supply circuit of FIG. 1, with an input of AC input voltage VAC = 100 V, fs = 83.3 kHz to 137.0 kHz and Δfs = 8 with respect to fluctuations of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W. It is 53.7 kHz, and it can be seen that the required control range is greatly shortened compared with the characteristics of the power supply circuit of FIG. Further, in the power supply circuit of FIG. 1, with an input of AC input voltage VAC = 230 V, Δfs = 27 kHz at fs = 126 kHz to 153 kHz with respect to fluctuations of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W. Even under this condition, the required control range is greatly shortened compared to the characteristics of the power supply circuit of FIG.

The characteristics of the switching frequency fs of the power supply circuit of FIG. 1 are as follows: AC 100 V system to AC 200 V system range (for example, VAC = 85 V to under the condition of corresponding load power of maximum load power Pomax = 200 W to minimum load power Pomin = 0 W). H.264V) that can be stabilized in response to commercial AC power input, so-called wide-range compatibility is realized.
Note that the wide-range support in the circuit of FIG. 1 will be described later.

  Further, regarding the AC → DC power conversion efficiency, the power conversion efficiency decreases as the light load condition is reached in the circuit of FIG. 12, so that the high efficiency range of ηAC → DC = 90% or more is the load power Po = 200 W. Although it was set to a range of ˜75 W, as described above, in this example, the range is expanded to a range of Po = 200 W to 25 W.

In the present embodiment, the main factor for improving the power conversion efficiency in this way is that a series resonant circuit is formed on the secondary side.
That is, it is known that the configuration in which the secondary side series resonant circuit is combined with the primary side parallel resonant circuit as the voltage resonant converter as in this embodiment is inherently advantageous in terms of power conversion efficiency. Yes. This is because the energy for generating the secondary side DC output voltage Eo can be covered by the resonance energy obtained by the secondary side series resonance circuit. That is, in this respect, the energy to be transmitted to the secondary winding N2 can be reduced as compared with the case where the secondary series resonance circuit is not provided, and the power conversion efficiency is improved accordingly.
This configuration has a characteristic property that the power conversion efficiency increases as the load tends to increase from the maximum load power. As compared with the configuration of the combination of parallel resonant circuits (FIG. 12) that tends to decrease, a very good characteristic can be obtained as a power conversion efficiency characteristic with respect to load fluctuation.

Moreover, the improvement of power conversion efficiency is also achieved by setting each resonance frequency.
That is, as the power conversion efficiency characteristic (ηAC → DC) of the present embodiment, ηAC → DC = 90% or more is obtained with respect to the load fluctuation range from the load power Po = 200 W to 25 W as described above. However, the power conversion efficiency characteristics with respect to such a load condition are finally obtained by adjusting the resonance frequencies fo1, fo2, and fo3. In other words, various settings were made for the resonance frequencies fo1, fo2, and fo3, and experiments were performed. The above-described fo1 = 96 kHz, fo2 = 94.5 kHz, fo3 = 54.3 kHz were set, and a relationship that was regarded as fo1≈fo2> fo3 was set. This is the characteristic finally obtained.

Note that such improvement in power conversion efficiency by setting the resonance frequency is also shown by the waveform of the switching current IQ1 shown in FIG.
That is, as can be seen by comparing the switching current IQ1 in FIG. 5A and the previous FIG. 14A, the waveform of the switching current IQ1 in FIG. The waveform is such that a peak level of 3.4 Ap is obtained at the timing before the turn-off when the on-period TON of Q1 ends and transitions to the off-period TOFF. And when the turn-off timing is reached, the level is further lowered.
The waveform of the switching current IQ1 is influenced by the waveform of the secondary winding current I2. That is, it has a waveform component corresponding to the current flowing in the secondary resonance circuit in which the parallel resonance circuit and the series resonance circuit are combined. The waveform of the secondary winding current I2 is determined by the setting of the resonance frequency fo2 and the resonance frequency fo3 with respect to the resonance frequency fo1.
From this, the waveform of the switching current IQ1 shown in FIG. 5A depends on the appropriate setting of the resonance frequencies fo1, fo2, and fo3 of the primary side parallel resonance circuit, the secondary parallel resonance circuit, and the secondary side series resonance circuit. It means that it is obtained.
The waveform of the switching current IQ1 in FIG. 5A means that the level of the switching current IQ1 at the time of turn-off is suppressed. If the level of the switching current IQ1 at the time of turn-off is suppressed, the switching loss and the conduction loss at the time of turn-off are reduced correspondingly.
Such a reduction in switching loss and conduction loss of the switching element is one factor that high power conversion efficiency characteristics are obtained for the power supply circuit of the present embodiment.

Here, in the circuit of FIG. 1 as described above, the power conversion efficiency can be improved even if the series resonant circuit is formed on the secondary side. However, the secondary side series resonant circuit is simply provided. In such a case, an abnormal operation occurs during an intermediate load, making it difficult to put it to practical use.
When the secondary side series resonant circuit is combined with the primary side parallel resonant circuit as in the circuit of FIG. 1, the primary side switching current IQ1 is referred to as a so-called biting current, which is due to the positive polarity at the timing when the switching element Q1 is turned off. Current flows.
This biting current increases the switching loss and makes it impossible to obtain a stable ZVS (zero voltage switching) operation. In any case, when such an abnormal operation occurs, for example, a phase-gain characteristic of the constant voltage control circuit system is shifted, and a switching operation is performed in an abnormal oscillation state. And since these abnormal operation | movement arises, the practical use was made difficult.
Such an abnormal operation at the time of intermediate load is caused by an interaction caused by simultaneous operation of the primary side parallel resonance circuit forming the voltage resonance type converter and the secondary side series resonance circuit.

Therefore, in the present embodiment, the coupling coefficient (total coupling coefficient kt) of the insulating converter transformer PIT is set to a predetermined value lower than that in the prior art.
By setting the total coupling coefficient kt to be lower in this way, the interaction between the primary side parallel resonance circuit and the secondary side series resonance circuit is diluted, and abnormal operation at the time of intermediate load is eliminated. It will follow. Specifically, the biting current of the switching current IQ1 is not observed, and a waveform corresponding to normal ZVS is obtained. This also eliminates the abnormal oscillation operation described above.
In this way, in the configuration of the present embodiment shown in FIG. 1, a configuration in which the secondary side series resonant circuit is combined with the primary side parallel resonant circuit can be realized as a practically usable one.

Note that the loose coupling state up to the total coupling coefficient kt included in the insulating converter transformer PIT of the present embodiment is due to an increase in power transmission loss from the primary side to the secondary side in the conventional voltage resonance type converter. There is a background that has not been done so far because it leads to a decrease in power conversion efficiency.
However, in the present embodiment, as shown as the experimental result of FIG. 6, very good power conversion efficiency characteristics can be obtained over almost the entire range of the corresponding load power.
This is because a resonance circuit (secondary side parallel resonance circuit and secondary side series resonance circuit) is also formed on the secondary side. That is, by providing the secondary side resonance circuit, as described above, it is possible to supply power as the secondary side DC output voltage Eo including the increase in energy obtained by the resonance operation. Thus, a decrease in power conversion efficiency due to the loose coupling is compensated.

Moreover, the experimental result that the raise of the peak level of switching voltage V1 was suppressed by setting to a lower value about the total coupling coefficient kt as mentioned above was obtained. This will be described below.
First, the peak level V1-p of the switching voltage V1 is
V1−p = Ei {(1 + π / 2) × (TON / TOFF)}
It is represented by
From this equation, it can be understood that the peak level V1-p increases as the rectified smoothing voltage (DC input voltage) Ei increases (that is, in this case, the AC input voltage VAC increases). In addition, it can be understood that the value increases as the value of “TON / TOFF” increases, that is, as the ratio of the period TON in one switching cycle increases.

  From the characteristic diagram of FIG. 6, it can be seen that the value of “TON / TOFF” increases as the heavy load condition is reached, and becomes maximum at the maximum load power Pomax = 200 W. Since the value of the peak level V1-p increases as the AC input voltage VAC rises as described above, the value of the peak level V1-p depends on the AC input voltage VAC and the load power Po. It can be understood that the maximum value is obtained when becomes the maximum.

Here, if attention is paid to the characteristic that the peak level V1-p increases according to the load power as described above, the characteristic of the period TON with respect to the period TOFF is shown as the characteristic of the period TON and the period TOFF with respect to the load fluctuation shown in FIG. If the time length is further reduced, the above-described “TON / TOFF” value is decreased, and as a result, an increase in the peak level V1-p can be suppressed.
In particular, since the peak level V1-p has a maximum value at the maximum AC input voltage, in order to suppress the peak value V1-p on the maximum value side, on the maximum AC input voltage side as indicated by a broken line in FIG. As a characteristic of the period TON and the period TOFF, it is only necessary to obtain a characteristic in which the ratio of the period TON becomes smaller.

As a result of the experiment, when the value of the total coupling coefficient kt of the insulating converter transformer PIT in the configuration of FIG. In the characteristics obtained on the side (in this case, the AC 200V system is input because of the wide range configuration), the change in the time ratio between the period TOFF and the period TON becomes large, and as a result, the period TON within one switching cycle The result was that the proportion of the share decreased. That is, by setting the value of the total coupling coefficient kt lower in this way, the result that the value of “TON / TOFF” as described above can be further reduced particularly on the maximum AC input voltage side is obtained. It is.
In this way, since the characteristic that the value of “TON / TOFF” becomes smaller especially on the maximum AC input voltage side is obtained, the value of “TON / TOFF” when the maximum AC input voltage is input is also smaller. To be. As the value of “TON / TOFF” at the time of inputting the maximum AC input voltage becomes smaller in this way, the maximum value of the peak level V1-p of the switching voltage V1 is also lowered accordingly. Become.

As the value of the coupling coefficient kt is lowered in this way, the maximum value of the peak level V1-p is suppressed.
If the maximum value of the peak level V1-p of the switching voltage V1 is suppressed, the voltage level applied to the switching element Q1 is also reduced accordingly, so that a part having a lower withstand voltage is selected as the switching element Q1. can do.
If a component with a lower breakdown voltage can be selected, the element size can be reduced, and the circuit area can be reduced. Further, since the component cost is lower, the circuit manufacturing cost can be reduced accordingly.

Here, in the present embodiment, kt = 0.57 is set as the total coupling coefficient kt. However, according to the setting of the total coupling coefficient kt, in this case, the maximum peak level V1-p of the switching voltage V1 is set. The value (at the time of maximum AC input voltage and maximum load power: in this case VAC = 264V · Po = 200 W) can be suppressed to 820V.
With the maximum value of the peak level V1-p, a 900V withstand voltage product can be used as the switching element Q1 in this case.

Next, the wide range correspondence realized by the configuration of FIG. 1 will be described.
First, the power supply circuit shown in FIG. 1 adopts a configuration as a voltage resonance type converter including a parallel resonance circuit together with the series resonance circuit as a secondary side resonance circuit. That is, the power supply circuit shown in FIG. 1 is also provided with a parallel resonant circuit on each of the primary side and the secondary side, as in the conventional configuration shown in FIG.
Here, from the viewpoint of wide range compatibility, the interaction between the parallel resonance circuits provided on the primary side and the secondary side is dominant. From the characteristics of the respective switching frequencies fs shown in FIG. 6 and FIG. 15, the constant voltage control operation in this case is a combination of the primary side parallel resonance circuit and the secondary side parallel resonance circuit. It can be understood from the fact that the switching frequency fs is continuously variably controlled as in the case of FIG.
Therefore, hereinafter, the wide-range support in the configuration of FIG. 1 will be described based on the constant voltage control characteristic when only the secondary side parallel resonant circuit is combined with the primary side parallel resonant circuit.

The combination of parallel resonant circuits on the primary side and the secondary side as described above means that these parallel resonant circuits are formed via electromagnetic induction of the insulating converter transformer PIT. If such a configuration is seen from the relationship between the primary side parallel resonance circuit and the secondary side parallel resonance circuit, a coupled resonance circuit by electromagnetic coupling is formed, to which a frequency signal corresponding to the switching frequency fs is input. Can be viewed as equivalent.
The constant voltage control characteristic for the secondary side DC output voltage Eo of the power supply circuit shown in FIG. Depending on it. This point will be described with reference to FIG.

FIG. 7 shows the output characteristics with respect to the input (switching frequency signal) for the above-described electromagnetically coupled resonance circuit. That is, the control characteristic for the secondary side DC output voltage Eo is shown by the relationship with the switching frequency fs. In this figure, the horizontal axis represents the switching frequency, and the vertical axis represents the level of the secondary side DC output voltage Eo.
As described in FIG. 1, in this embodiment, the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side parallel resonance circuit are set to be approximately equal. Actually, the resonance frequency fo1 is slightly higher than the resonance frequency fo1. In FIG. 7, the horizontal axis indicating the switching frequency fs is shown corresponding to the resonance frequencies fo1 and fo2. In this case, the resonance frequency fo1 This is shown as being higher than the resonance frequency fo2.

  Here, it is assumed that the coupling degree of the insulating converter transformer PIT is set to a tight coupling state where the total coupling coefficient kt = 1. In this case, the leakage inductance L1 of the primary winding N1 and the leakage inductance L2 of the secondary winding N2 are 0 respectively.

で表され、
周波数ｆ２は、

で表される。
また、上記（数１）（数２）における項の１つであるｆｏは、一次側直列共振回路の共振周波数ｆｏ１と、二次側並列共振回路の共振周波数ｆｏ２との中間に存在する中間共振周波数であり、１次側のインピーダンスと２次側のインピーダンスと、一次側と二次側とで共通となるインピーダンス（相互結合インダクタンスＭ）により決定される周波数である。
なお、相互結合インダクタンスＭについては、

により表される。 As described above, as a constant voltage control characteristic in a state where the primary side and the secondary side of the insulating converter transformer PIT are tightly coupled, as shown as the characteristic curve 1 in FIG. The resonance frequency fo1 and the resonance frequency fo2 of the secondary side parallel resonance circuit have so-called bimodal characteristics in which the secondary side DC output voltage Eo peaks at frequencies f1 and f2.
Here, the frequency f1 is

Represented by
The frequency f2 is

It is represented by
Further, fo, which is one of the terms in the above (Equation 1) and (Equation 2), is an intermediate resonance that exists between the resonance frequency fo1 of the primary side series resonance circuit and the resonance frequency fo2 of the secondary side parallel resonance circuit. It is a frequency determined by the impedance on the primary side and the impedance on the secondary side, and the impedance (mutual coupling inductance M) that is common to the primary side and the secondary side.
For the mutual coupling inductance M,

It is represented by

  Further, assuming that the total coupling coefficient kt is gradually decreased from the state of kt = 1, that is, when the degree of loose coupling is gradually increased from the tightly coupled state, it is shown in FIG. The characteristic curve 1 shows a change in which the bimodal tendency gradually diminishes and becomes flat near the intermediate resonance frequency fo. Then, when the total coupling coefficient kt is reduced to a certain level, a so-called critical coupling state is obtained. In this critical coupling state, as shown by the characteristic curve 2, there is no tendency as a bimodal characteristic, and the curve shape becomes flat with the intermediate resonance frequency fo as the center.

Further, if the overall coupling coefficient kt is reduced from the critical coupling state and the loose coupling state is strengthened, as shown by the characteristic curve 3 in FIG. A unimodal characteristic is obtained. Further, when comparing the characteristic curve 3 with the characteristic curves 1 and 2, the characteristic curve 3 has a peak level itself lower than that of the characteristic curves 1 and 2, but as a quadratic curve shape thereof, It can be seen that it has a steeper slope.
Insulating converter transformer PIT of the present embodiment is set to a loosely coupled state in which total coupling coefficient kt≈0.6 or less. In the setting of the total coupling coefficient kt, the operation is based on the single peak characteristic shown as the characteristic curve 3.

  When the unimodal characteristic shown in FIG. 7 is compared with the constant voltage control characteristic of the conventional power supply circuit (FIG. 12) shown in FIG. 16, the characteristic shown in FIG. For the characteristics, the slope is considerably gentler in terms of a quadratic function.

  Since the characteristic shown in FIG. 16 is moderately curved as described above, the necessary control range of the switching frequency for performing constant voltage control on the secondary side DC output voltage Eo is, for example, AC input voltage VAC = 100V. Even under the condition corresponding to a single range by the input of fs = 117.6 kHz to 208.3 kHz, Δfs = 96.7 kHz. For this reason, as described above, it is very difficult to achieve a wide range only by constant voltage control by switching frequency control.

On the other hand, the constant voltage control characteristic of the present embodiment is a single peak characteristic shown by the characteristic curve 3 in FIG. 7, and the constant voltage control operation is, for example, as shown in FIG. Become.
In FIG. 8, the characteristic curves A and B at the maximum load power Pomax and the minimum load power Pomin when the AC input voltage VAC = 100 V (AC 100 V system) for the power supply circuit of the present embodiment shown in FIG. Four characteristic curves are shown, which are characteristic curves C and D at the maximum load power Pomax and at the minimum load power Pomin when the AC input voltage VAC = 230 V (AC 200 V system).

As can be seen from FIG. 8, first, when the AC input voltage VAC = 100 V corresponding to the AC 100 V system input, the switching required for making the secondary side DC output voltage Eo constant at the required rated level tg. The frequency variable control range (necessary control range) is indicated by Δfs1. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve A to the switching frequency fs at level tg in the characteristic curve B.
In addition, when the AC input voltage VAC = 230 V corresponding to the AC 200 V system input, the variable control range (necessary control) of the switching frequency necessary for making the secondary side DC output voltage Eo constant at the required rated level tg. (Range) is indicated by Δfs2. That is, the frequency range is from the switching frequency fs at level tg in the characteristic curve C to the switching frequency fs at level tg in the characteristic curve D.

As described above, the unimodal characteristic that is the control characteristic of the secondary side DC output voltage Eo in the present embodiment is considerably steep in a quadratic function curve as compared with the control characteristic shown in FIG. It is.
For this reason, Δfs1 and Δfs2 which are the respective required control ranges when the AC input voltage VAC = 100V and VAC = 230V are considerably reduced as compared with Δfs shown in FIG.
This also changes the frequency variable range (ΔfsA) from the lowest switching frequency at Δfs1 (switching frequency fs at level tg in characteristic curve A) to the highest switching frequency at Δfs2 (switching frequency fs at level tg in characteristic curve A). ) Will be correspondingly narrow.
For confirmation, the actual Δfs1, Δfs2, and ΔfsA measured in the power supply circuit of FIG.
Δfs1 = 53.7kHz (= 137.0kHz−83.3kHz)
Δfs2 = 43.5kHz (= 153kHz-126kHz)
ΔfsA ≒ 70 kHz (= 153kHz−83.3kHz)
It becomes.

The frequency variable range ΔfsA is sufficiently within the variable range of the switching frequency corresponding to the current switching drive IC (oscillation / drive circuit 2). That is, in the power supply circuit shown in FIG. 1, the switching frequency can be actually variably controlled within the frequency variable range ΔfsA.
In this way, the power supply circuit of the present embodiment shown in FIG. 1 is suitable for any AC 100 V system or AC 200 V system commercial AC power input, and the secondary side DC output voltage Eo that is a proper main DC power supply. Can be stabilized. In other words, wide range support is possible only by switching frequency control.

  Incidentally, a coupled resonance circuit using electromagnetic coupling is already known as a technique for expanding the amplification bandwidth of an amplifier circuit using a transistor, for example, as an intermediate frequency transformer amplifier. However, in such a field, the bimodal characteristic in the tight coupling or the flat characteristic in the critical coupling is used, and the single peak characteristic in the loose coupling is not used. In the present embodiment, in such a coupled resonant circuit technology using electromagnetic coupling, a single-peak characteristic with loose coupling that has not been employed in the field of communication technology is actively used in the field of resonant switching converters. It can be said that. This reduces the variable range (necessary control range) of the switching frequency necessary to stabilize the secondary side DC output voltage Eo as described above, and supports a wide range only by constant voltage control in switching frequency control. Is possible.

In this way, the necessary control range (Δfs) of the switching frequency fs for constant voltage control is reduced under the conditions of AC 100 V system and AC 200 V system commercial AC power supply input. Responsiveness and control sensitivity will be greatly improved.
Some electronic devices perform an operation of changing the load power Po so as to be switched (switched) at a relatively high speed between a maximum and no load, which is a so-called switching load. An example of a device that performs such an operation as a switching load is a printer that is a peripheral device of a personal computer.
For a device that operates as such a switching load, for example, when a power supply circuit having a relatively wide necessary control range Δfs as shown in FIG. Following the change in the load power, the switching frequency fs is variably controlled by a correspondingly large amount of change. For this reason, it has been difficult to obtain high-speed constant voltage control response.
On the other hand, in the present embodiment, since the necessary control range Δfs is greatly reduced particularly in the area for each single range, it is possible to quickly cope with a steep fluctuation between the maximum load power Po and no load. In response, the secondary side DC voltage Eo can be stabilized. That is, the response performance of the constant voltage control with respect to the switching load is greatly improved.

As described above, in the switching power supply circuit of the present embodiment, the insulating converter transformer PIT is used as a voltage resonance type converter in which the secondary side parallel resonance circuit and the secondary side series resonance circuit are combined with the primary side parallel resonance circuit. By reducing the total coupling coefficient kt to a predetermined value that is regarded as loosely coupled, it is possible to achieve a wide range only by variable control of the switching frequency.
Further, since the secondary side series resonance circuit is combined as described above, the power conversion efficiency is improved as compared with the conventional case due to the characteristic that increases as the light load condition shown in FIG. Can be planned. At this time, since the total coupling coefficient kt is set to a predetermined value or less as described above, it is possible to prevent abnormal operation when the secondary side series resonance circuit is combined, and this is practical. Can be realized.
Furthermore, the power conversion efficiency is further improved by setting the resonance frequencies fo1, fo2, and fo3 of the primary side parallel resonance circuit, the secondary side parallel resonance circuit, and the secondary side series resonance circuit.

  Further, in the present embodiment, as the total coupling coefficient kt is reduced to a predetermined value that is loosely coupled as described above, the ratio of the ON period TON particularly on the maximum AC input voltage side is reduced. As a result, the peak level V1-p of the switching voltage V1 is suppressed from increasing, and the breakdown voltage level of the switching element Q1 is thereby reduced. At this time, by reducing the total coupling coefficient kt to kt = 0.57 described above, the maximum value of the peak level V1-p is suppressed to about 820V, and as a result, a 900V withstand voltage product is selected as the switching element Q1. Can do.

  As a basic configuration for realizing such wide-range compatibility, high efficiency, and the effect of suppressing the peak level V1-p of the switching voltage V1, a voltage resonance type converter including a secondary side parallel resonance circuit ( For example, compared to the conventional configuration shown in FIG. 12, at least the insulating converter transformer PIT has a structure capable of obtaining a total coupling coefficient kt below the required level, and a secondary series resonance capacitor C3 is added to form a secondary series resonance circuit. Therefore, there is also an advantage that this can be realized without increasing the cost due to an increase in the number of parts, increasing the size of the circuit, and increasing the weight.

  In order to reduce the coupling coefficient to a relatively low value as described above, in this embodiment, the insulating converter transformer PIT has the structure shown in FIG. This can be achieved. In other words, it is not necessary to enlarge the gap G when setting a coupling coefficient lower than that in the prior art, and it is possible to effectively prevent a decrease in efficiency due to eddy current loss.

Subsequently, as variations of the power supply circuit of the present embodiment, variations of the secondary side rectifier circuit are shown in FIGS.
FIG. 9 shows the configuration of a power supply circuit as a first modification of the present invention.
In addition, in each figure from this FIG. 9 to FIG. 11, the same part as the part already demonstrated in FIG. 1 is attached | subjected, and description is abbreviate | omitted. In particular, the configuration on the primary side is the same as in FIG.

The power supply circuit shown in FIG. 9 includes a voltage doubler full wave rectifier circuit as a secondary side rectifier circuit.
First, also in this case, a secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2.
In addition, as the voltage doubler full-wave rectifier circuit, first, the secondary winding N2 is provided with a center tap as shown in the figure, and the secondary winding portions N2A and N2B are connected to the secondary winding portions N2A and N2B with the center tap as a boundary. To divide. The same predetermined number of turns (number of turns) is set in the secondary winding portions N2A and N2B.
Further, a secondary side series resonant capacitor C3A is connected in series to the end of the secondary winding N2 on the secondary winding portion N2A side, and the secondary winding N2 has a secondary winding portion N2B side end. A secondary side series resonant capacitor C3B is connected in series to the end. Thus, on the secondary side in this case, the secondary side resonance circuit by the connection of the parallel resonance capacitor C2 and the leakage inductance component of the secondary winding N2A and the capacitance of the secondary side series resonance capacitor C3A are formed. A first secondary side series resonance circuit and a second secondary side series resonance circuit composed of the leakage inductance component of the secondary winding N2B and the capacitance of the secondary side series resonance capacitor C3B are formed. .

The end of the secondary winding N2 on the secondary winding N2A side is connected to the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 via the series connection of the secondary side series resonant capacitor C3A. Connect. Further, the end of the secondary winding N2 on the secondary winding N2B side is connected to the connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4 via the series connection of the secondary side series resonant capacitor C3B. Connect.
The cathodes of the rectifier diodes Do1 and Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground.
Further, the connection points of the anodes of the rectifier diodes Do2 and Do4 and the center tap of the secondary winding N2 are also connected to the secondary side ground.

In the above connection form, the first voltage-doubling half including the first secondary-side series resonance circuit including the secondary winding N2A, the secondary-side series resonance capacitor C3A, the rectifier diodes Do1 and Do2, and the smoothing capacitor Co. A second voltage-doubling half comprising a wave rectifier circuit and a second secondary side series resonant circuit comprising a secondary winding portion N2B, a secondary side series resonant capacitor C3B, rectifier diodes Do1, Do2 and a smoothing capacitor Co. A wave rectifier circuit is formed.
In the first voltage-doubling half-wave rectifier circuit, during the half-cycle period of one polarity of the alternating voltage induced in the secondary winding N2, [secondary winding portion N2A → rectifier diode Do2 → secondary series The rectification operation is performed by the rectification current path of the resonance capacitor C3A → secondary winding portion N2A], and the secondary side series resonance capacitor C3A is charged by the potential of the alternating voltage (V2) of the secondary winding portion N2A. In the period of the other half cycle, the rectification operation is performed by the rectification current path of [secondary winding part N2A → secondary series resonance capacitor C3A → rectifier diode Do1 → smoothing capacitor Co → secondary winding part N2A]. Thus, the smoothing capacitor Co is charged by the superimposed potential of the both-ends voltage of the secondary side series resonance capacitor C3A and the alternating voltage of the secondary winding N2A.
In addition, the second voltage-doubling half-wave rectifier circuit [secondary winding portion N2B → rectifier diode Do4 → second] in the half cycle period of the other polarity of the alternating voltage induced in the secondary winding N2. The secondary side series resonant capacitor C3B is rectified by the rectified current path of the secondary side series resonant capacitor C3B → secondary winding part N2B], and the secondary side series resonant capacitor C3B is driven by the alternating voltage (equivalent to V2) of the secondary winding part N2A. Rectified current path of [secondary winding portion N2B → secondary side series resonance capacitor C3B → rectifier diode Do3 → smoothing capacitor Co → secondary winding portion N2B] during the half cycle period of the one polarity. The smoothing capacitor Co is charged by the superimposed potential of the voltage across the secondary side series resonant capacitor C3B and the alternating voltage of the secondary winding N2B.

  According to the above rectifying operation, for the smoothing capacitor Co, the induced voltage of the secondary winding N2B and the secondary voltage corresponding to the half cycle of one polarity of the alternating voltage of the secondary winding N2 are applied. The rectified current is charged by the superimposed potential of the voltage across the side series resonant capacitor C3B, and the induced voltage of the secondary winding N2A and the secondary side series resonant capacitor C3A The rectified current is charged by the superimposed potential of the both-end voltage. As a result, a level corresponding to twice the induced voltage level (V2) of the secondary winding portions N2A and N2B is obtained as the secondary side DC output voltage Eo which is the voltage across the smoothing capacitor Co. That is, a voltage doubler full wave rectifier circuit is obtained.

  Here, for convenience of explanation, the rectification current path has been described assuming that the secondary side parallel resonance capacitor C2 is not provided. However, as an actual rectification operation, a parallel resonance circuit is provided by providing the secondary side parallel resonance capacitor C2. It is performed under the influence of the operation. As a result of the rectification operation actually performed in this way, a level corresponding to twice the alternating voltage level excited in the secondary winding N2 is obtained as the generated secondary side DC output voltage Eo. Also in this case, charging to the smoothing capacitor Co is performed corresponding to each half cycle, so that the double voltage full wave rectification operation can be obtained as the rectification operation.

Here, also in the power supply circuit of FIG. 9 provided with the voltage doubler full wave rectifier circuit for the secondary side rectifier circuit as described above, kt≈0.6 is set as the total coupling coefficient kt for the isolated converter transformer PIT, With respect to the parallel resonance frequency fo1, the secondary side parallel resonance frequency fo2, and the secondary side series resonance frequency fo3, the respective resonance frequencies are set so that the relationship of fo1≈fo2> fo3 is obtained.
As a result, the power supply circuit as the first modification can be adapted to a wide range only by the switching frequency control, and the responsiveness of the constant voltage control operation to the switching load can be improved. Further, abnormal oscillation due to the combination of the secondary side series common circuit is also prevented, and the AC → DC power conversion efficiency is improved as in the case of FIG.
Further, as in the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

FIG. 10 shows a configuration of a power supply circuit as a second modification.
In the second modification, a bridge full-wave rectifier circuit including a bridge rectifier circuit including four rectifier diodes Do1, Do2, Do3, Do4 is provided as a secondary-side rectifier smoothing circuit.
Also in this case, first, a parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. Further, a secondary side series resonant capacitor C3 is connected in series with the secondary winding N2.

In the bridge rectifier circuit, the connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 is a positive input terminal, and the connection point of the cathode of the rectifier diode Do1 and the cathode of the rectifier diode Do3 is a positive output terminal. The connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4 is a negative input terminal, and the connection point between the anode of the rectifier diode Do2 and the anode of the rectifier diode Do4 is a negative output terminal.
The positive input terminal of the bridge rectifier circuit is connected to the winding end of the secondary winding N2 via the secondary side series resonant capacitor C3, and the positive output terminal is connected to the positive terminal of the smoothing capacitor Co. The negative input terminal is connected to the winding start end of the secondary winding N2, and the negative output terminal is connected to the secondary side ground. The negative terminal of the smoothing capacitor Co is also connected to the secondary side ground.

  In the secondary side rectifier circuit formed in this way, the rectifier diode Do1 and the rectifier diode Do4 conduct and perform rectification corresponding to the half cycle of one polarity of the alternating voltage V2 of the secondary winding N2. Thus, the smoothing capacitor Co is charged by the rectified current ID1. In correspondence with the other half cycle, the rectifier diode Do2 and the rectifier diode Do3 conduct and rectify, and the rectification current ID2 charges the smoothing capacitor Co. As a result, a secondary side DC output voltage Eo of a level corresponding to the same voltage as the induced voltage (V2) of the secondary winding N2 is generated as the voltage across the smoothing capacitor Co.

Also in the power supply circuit of FIG. 10 having such a circuit configuration, kt≈0.6 is set as the total coupling coefficient kt for the insulating converter transformer PIT, and the primary side parallel resonance frequency fo1, the secondary side parallel resonance frequency fo2, the secondary Regarding the side series resonance frequency fo3, each resonance frequency is set so that the relationship of fo1≈fo2> fo3 can be obtained even in this case.
As a result, the power supply circuit of the second modified example shown in FIG. 10 can be adapted to a wide range only by switching frequency control, and can improve the responsiveness of the constant voltage control operation to the switching load. Further, abnormal oscillation due to the combination of the secondary side series common circuit is also prevented, and the AC → DC power conversion efficiency is improved as in the case of FIG.
Further, as in the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

FIG. 11 shows a configuration of a power supply circuit as a third modification.
In the third modification, the secondary side rectifier circuit is a quadruple voltage rectifier circuit.
First, also in this case, a parallel resonant capacitor C2 is connected in parallel to the secondary winding N2.
The quadruple voltage rectifier circuit includes four rectifier diodes including rectifier diodes Do1 to Do4, a secondary side series resonance capacitor C3A, a secondary side series resonance capacitor C3B, and smoothing capacitors Co1 and Co2, as shown in the figure. Formed in preparation.
In this case, one end (end of winding) of the secondary winding N2 is connected via a series connection of a secondary side series resonant capacitor C3A → rectifier diode Do1 (anode → cathode) as shown in the figure. The positive terminal of the smoothing capacitor Co1 is connected. The negative terminal of the smoothing capacitor Co1 is connected to the other end (winding end) of the secondary winding N2.
Further, the positive terminal of the smoothing capacitor Co2 is connected to the connection point between the negative terminal of the smoothing capacitor Co1 and the other end of the secondary winding N2, and the negative terminal of the smoothing capacitor Co2 is connected to the secondary side ground. It is connected.

Further, a series connection circuit of a secondary side series resonance capacitor C3B → rectifier diode Do4 (cathode → anode) is inserted between the one end of the secondary winding N2 and the secondary side ground. .
The anode of the rectifier diode Do3 is connected to the connection point between the secondary side series resonant capacitor C3B and the rectifier diode Do4 as shown in the figure. The cathode of the rectifier diode Do3 is connected to the connection point between the connection point of the smoothing capacitors Co1 and Co2 and the other end of the secondary winding N2.
Further, the anode of the rectifier diode Do2 is connected to the connection point between the cathode of the rectifier diode Do3 and the other end of the secondary winding N2. The cathode of the rectifier diode Do2 is connected to the connection point between the secondary side series resonant capacitor C3A and the rectifier diode Do1.

In the quadruple voltage rectifier circuit having the above configuration, in one half cycle of the alternating voltage excited in the secondary winding N2, the rectified current is [secondary winding N2 → rectifier diode Do2 → secondary series resonant capacitor C3A → It flows through the circulation path of the secondary winding N2]. Similarly, in the other half cycle of the alternating voltage, the rectified current flows through [secondary winding N2 → secondary series resonance capacitor C3B → rectifier diode Do3 → secondary winding N2] through the circulation path.
In other words, even in this case, the DC voltage at a level corresponding to the same voltage as the alternating voltage level excited in the secondary winding N2 is provided at both ends of the secondary side series resonant capacitors C3A and C3B in the corresponding half cycle. Will be obtained.

Even in this case, the rectified current branches from the circulation path and flows through the following paths in each half cycle.
First, in one half cycle of the alternating voltage described above, the rectified current branches and also flows through the path [smoothing capacitor Co2 → rectifier diode Do4 → secondary series resonant capacitor C3B → secondary winding N2]. At this time, the secondary side series resonance capacitor C3B is charged during this period by the previous circulation path. For this reason, depending on the rectification current path as described above, the smoothing capacitor Co2 is charged with a voltage level due to the superposition of the alternating voltage obtained at the secondary winding N2 and the voltage across the secondary side series resonance capacitor C3B. Will be done.
That is, a smoothing capacitor Co2 generates a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2.

Further, in the other half cycle of the alternating voltage, the rectified current branches and flows through a path of [secondary side series resonant capacitor C3A → rectifier diode Do1 → smoothing capacitor Co1 → secondary winding N2]. The smoothing capacitor Co1 is charged according to the voltage level of the alternating voltage of the secondary winding N2 that has received the superimposed voltage across the secondary side series resonant capacitor C3A obtained by the previous circulation path.
In other words, even with the smoothing capacitor Co1, a voltage corresponding to twice the alternating voltage level obtained at the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co1.

  In this manner, a DC voltage having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is generated at both ends of the smoothing capacitor Co1 and the smoothing capacitor Co2. As a result, a secondary side DC output voltage Eo having a level corresponding to four times the alternating voltage level excited in the secondary winding N2 is obtained at both ends of the series connection of the smoothing capacitor Co1 and the smoothing capacitor Co2. It will be.

  In this case as well, for convenience of explanation, the rectification current path has been described assuming that there is no secondary side parallel resonance capacitor C2. However, as an actual rectification operation, the secondary side parallel resonance capacitor C2 is also provided in this case. This is performed under the influence of the operation as a parallel resonant circuit. As a result of the rectification operation actually performed as described above, the level corresponding to four times the alternating voltage level excited in the secondary winding N2 is obtained as the generated secondary side DC output voltage Eo. As an operation, a quadruple voltage rectification operation can be obtained.

Also in the power supply circuit of FIG. 11 adopting the above circuit configuration, kt≈0.6 is set as the total coupling coefficient kt for the insulating converter transformer PIT, and the primary side parallel resonance frequency fo1, the secondary side parallel resonance frequency fo2, the secondary side series With respect to the resonance frequency fo3, each resonance frequency is set so that the relationship of fo1≈fo2> fo3 is obtained even in this case.
As a result, the power supply circuit as the third modified example of FIG. 11 can be adapted to the wide range only by the switching frequency control, and the response of the constant voltage control operation to the switching load can be improved. Further, abnormal oscillation due to the combination of the secondary side series common circuit is also prevented, and the AC → DC power conversion efficiency is improved as in the case of FIG.
Further, as in the case of FIG. 1, the maximum value of the peak level V1-p is similarly suppressed by reducing the total coupling coefficient kt to a predetermined value.

In addition, as a variation of the secondary side rectifying / smoothing circuit, a configuration other than those shown in FIGS. 9 to 11 can be used.
However, in the present embodiment, a series resonance circuit is combined with the secondary side, and therefore, the secondary winding current in each half cycle of the entire secondary winding N2 so as to obtain a series resonance operation. Need to be washed away. For this reason, the half-wave rectifier circuit and the double-wave rectifier circuit in which the secondary winding current flows only in a half cycle are excluded.

  The present invention is not limited to the configurations shown as the above embodiments. For example, other detailed circuit configurations of the primary side voltage resonance type converter are conceivable. As the switching element, it may be possible to select an element other than the MOS-FET. In each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

It is a circuit diagram which shows the structural example of the power supply circuit as embodiment of this invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a figure for demonstrating the magnetic path formed in the insulated converter transformer by the structure shown in FIG. It is a figure shown about the other structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped, and its magnetic path. It is a wave form diagram which shows operation | movement of the principal part of the power supply circuit of embodiment by a switching period. It is a figure which shows the fluctuation | variation characteristics of the AC-> DC power conversion efficiency with respect to load fluctuation | variation, switching frequency, the ON / OFF period of a switching element, and the peak level of a switching voltage about the power supply circuit of embodiment. It is a figure which shows the constant voltage control characteristic about the power supply circuit of embodiment. It is a figure which shows the switching frequency control range (required control range) according to alternating current input voltage conditions and load fluctuation | variation as constant voltage control operation | movement of the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as a 1st modification. It is a circuit diagram which shows the structural example of the power supply circuit as a 2nd modification. It is a circuit diagram which shows the structural example of the power supply circuit as a 3rd modification. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art example. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit shown in FIG. 12 is provided. FIG. 13 is a waveform diagram showing an operation of a main part of the power supply circuit shown in FIG. 12. FIG. 13 is a diagram showing fluctuation characteristics of AC → DC power conversion efficiency, switching frequency, and switching element on / off period with respect to load fluctuations for the power supply circuit shown in FIG. 12. It is a figure which shows notionally the constant voltage control characteristic about the conventional power supply circuit.

Explanation of symbols

  1. Control circuit, 2. Oscillation / drive circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT insulation converter transformer, CR1, CR2, CR3, CR4 E type core, FSC1, FSC2 ferrite sheet core, Cr primary side parallel Resonant capacitor, N1 primary winding, N2 secondary winding, C2 secondary side parallel resonant capacitor, C3, C3A, C3B secondary side series resonant capacitor, Do1, Do2, Do3, Do4 rectifier diode, Co (secondary side) Smoothing capacitor, N2A, N2B secondary winding

Claims (3)

Switching means formed with a switching element for switching by inputting a DC input voltage;
Switching driving means for switching and driving the switching element;
Formed by winding at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is induced by the switching output obtained in the primary winding An isolated converter transformer,
A primary side parallel resonant circuit formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of the primary side parallel resonant capacitor, and the operation of the switching means is a voltage resonant type;
By connecting a secondary side parallel resonant capacitor in parallel with the secondary winding of the insulating converter transformer, a leakage inductance component including the secondary winding and a capacitance of the secondary side parallel resonant capacitor A secondary parallel resonant circuit formed by
By connecting a secondary side series resonant capacitor in series with the secondary winding of the insulating converter transformer, a leakage inductance component including the secondary winding and a capacitance of the secondary side series resonant capacitor A secondary side series resonant circuit formed by
A secondary side DC output voltage generating means configured to perform a rectifying operation based on an alternating voltage induced in the secondary winding of the insulating converter transformer to generate a secondary side DC output voltage;
Constant voltage control means for performing constant voltage control on the secondary side DC output voltage by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. And
The insulation converter transformer has a coupling coefficient between the primary side and the secondary side,
For an electromagnetically coupled resonant circuit formed with at least the primary side parallel resonant circuit and the secondary side parallel resonant circuit, a unimodal characteristic is obtained as an output characteristic for the input of a frequency signal having the switching frequency, And the ratio of the ON period of the switching element when an AC input voltage of a predetermined level or higher is input is set to a predetermined value that is regarded as loosely coupled,
Further, the power conversion efficiency of a certain level or more is obtained at least under a predetermined load condition, the resonance frequency of the primary side parallel resonance circuit, the resonance frequency of the secondary side parallel resonance circuit, and the secondary side series resonance. The resonant frequency of the circuit is set,
A switching power supply circuit. The insulation converter transformer is
The primary winding and the secondary winding are wound around the central magnetic leg of the EE type core, a gap is formed with respect to the central magnetic leg, and the outer magnet of the EE type core is formed. A ferrite sheet core is provided so as to protrude toward the central magnetic leg at a predetermined position of the leg,
The switching power supply circuit according to claim 1. The insulation converter transformer is
The primary winding is wound around one outer magnetic leg of the EE core, and the secondary winding is wound around the other outer magnetic leg, and a gap is formed with respect to the central magnetic leg of the EE core. Formed,
The switching power supply circuit according to claim 1.

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