JP2006296054A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To put into practical use a voltage resonance type converter that performs ZVS in a totally corresponding load region. <P>SOLUTION: A primary-side switching converter constitutes a DC input voltage as an E-class resonance type that is obtained by rectifying and smoothing commercial AC power by a rectifying circuit (including a smoothing capacitor Ci), and a switching power supply circuit that uses a choke coil of a primary-side series resonance circuit of the E-class resonance type converter as a primary winding N1 of an insulated converter transformer PIT is constituted. A general coupling coefficient between a primary side and a secondary side is set to be 0.65 more or less by a coupling coefficient of the insulated converter transformer PIT itself, and equivalent parallel connection between the choke coil 10 and the primary winding N1. Each of a resonance frequency fo1 of the primary-side series resonance circuit and a resonance frequency fo2 of a secondary-side parallel resonance circuit is set equal at a level of 60kHz. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit including a voltage resonance type converter.

As a so-called soft switching power supply of a resonance type, a current resonance type and a voltage resonance type are widely known. At present, half bridge coupling type current resonance type converters with two stone switching elements are widely used due to the ease of practical use.
However, at present, for example, the characteristics of high-voltage switching elements have been improved, and the problem of withstand voltage when the voltage resonance converter is put into practical use has been cleared. In addition, a single-ended voltage resonant converter with a single switching element is advantageous in terms of input feedback noise and DC output voltage line noise components compared to a single-current resonant forward converter. It is also known that.

  FIG. 10 shows a configuration example of a switching power supply circuit including a voltage resonance type converter by a single end system.

In the switching power supply circuit shown in this figure, the commercial AC power supply AC is rectified and smoothed by a rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci, thereby generating a rectified and smoothed voltage Ei as a voltage across the smoothing capacitor Ci. is doing.
Note that the line of the commercial AC power supply AC is provided with a noise filter that includes a pair of common mode choke coils CMC and two across capacitors CL and removes common mode noise.

  The rectified and smoothed voltage Ei is input to the voltage resonant converter as a DC input voltage. As described above, this voltage resonance type converter employs a single-ended configuration including one switching element Q. In this case, the voltage resonance type converter is a separately excited type, and the switching element Q1 of the MOS-FET is switched by the oscillation / drive circuit 2.

  A MOS-FET body diode DD is connected in parallel to the switching element Q1. A primary side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching element Q1.

  The primary side parallel resonant capacitor Cr forms a primary side parallel resonant circuit (voltage resonant circuit) with the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. The primary side parallel resonance circuit can obtain a voltage resonance type operation as the switching operation of the switching element Q1.

  The oscillation / drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1 in order to drive the switching element Q1. Thereby, the switching element Q1 performs a switching operation at a switching frequency corresponding to the cycle of the drive signal.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As the structure of the insulating converter transformer PIT, for example, an EE type core in which an E type core made of a ferrite material is combined is provided. Then, after dividing the winding part on the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the central magnetic leg of the EE core.
In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, whereby k = 0.80 between the primary side and the secondary side. A coupling coefficient k of about ˜0.85 is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained. Further, the value of the coupling coefficient k becomes a setting element of the leakage inductance (L1).

  One end of the primary winding N1 of the insulating converter transformer PIT is inserted between the positive terminal of the switching element Q1 and the smoothing capacitor Ci, so that the switching output of the switching element Q1 is transmitted. Yes. An alternating voltage induced by the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.

In this case, the secondary side series resonant capacitor C2 is connected in series to one end of the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C2 Thus, a secondary side series resonance circuit (current resonance circuit) is formed.
In addition, a double voltage half-wave rectifier circuit is formed by connecting rectifier diodes Do1 and Do2 and a smoothing capacitor Co to the secondary side series resonant circuit as shown in the figure. This voltage doubler half-wave rectifier circuit generates a secondary side DC output voltage Eo at a level corresponding to twice the alternating voltage V2 induced in the secondary winding N2 as a voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to the load and is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side DC output voltage Eo input as a detection voltage to the oscillation / drive circuit 2.
The oscillation / drive circuit 2 adjusts the switching element Q1 so that the secondary side DC output voltage Eo becomes constant at a predetermined level in accordance with the level of the secondary side DC output voltage Eo indicated by the input detection output. Controls the switching operation. That is, a drive signal for obtaining a switching operation to be controlled is generated and output. Thereby, stabilization control of the secondary side DC output voltage Eo is performed.

11 and 12 show experimental results for the power supply circuit having the configuration shown in FIG. In the experiment, the main part of the power supply circuit of FIG. 10 is set as follows.
For the isolated converter transformer PIT, EER-35 is selected for the core, and the gap length of the central magnetic leg is set to 1 mm. The number of turns T (number of turns) of the primary winding N1 and the secondary winding N2 is N1 = 39T and N2 = 23T, respectively, and the induced voltage level per turn (T) of the secondary winding N2 is as follows. 3V / T was set. For the coupling coefficient k of the insulating converter transformer PIT, k = 0.81 was set.
Further, Cr = 3900 pF was selected for the capacitance of the primary side parallel resonant capacitor Cr, and C2 = 0.1 μF was selected for the capacitance of the secondary side series resonant capacitor C2. Accordingly, the resonance frequency fo1 = 230 kHz of the primary side parallel resonance circuit and the resonance frequency fo2 = 82 kHz of the secondary side series resonance circuit are set. In this case, the relative relationship between the resonance frequencies fo1 and fo2 can be expressed as fo1≈2.8 × fo2.
The rated level of the secondary side DC output voltage Eo is 135V, and the corresponding load power is maximum load power Pomax = 200 W to minimum load power Pomin = 0 W.

FIG. 11 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG. 10 by the switching cycle of the switching element Q1, and FIG. 11A shows the voltage V1 when the maximum load power Pomax = 200 W, the switching A current IQ1, a primary winding current I1, a secondary winding current I2, and a secondary side rectified current ID1, ID2 are shown. FIG. 11B shows the voltage V1, the switching current IQ1, the primary winding current I1, and the secondary winding current I2 when the intermediate load power Po = 120 W. FIG. 11C shows the voltage V1 and the switching current IQ1 when the minimum load power Pomin = 0W.
The voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a zero level in the period TON in which the switching element Q1 is turned on and becomes a sinusoidal resonance pulse in the period TOFF in which the switching element Q1 is turned off. The resonance pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is a voltage resonance type.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD), and flows as a waveform shown in the figure in the period TON, and is obtained as a waveform that becomes 0 level in the period TOFF.
The primary winding current I1 flowing through the primary winding N1 is a combination of the current component flowing as the switching current IQ1 during the period TON and the current flowing through the primary parallel resonant capacitor Cr during the period TOFF.

Although only shown in FIG. 11 (a), as the operation of the secondary side rectifier circuit, the rectified currents ID1 and ID2 flowing through the rectifier diodes Do1 and Do2 flow in a sine wave shape as shown in the figure. . In this case, the waveform of the rectified current ID1 is such that the resonance operation of the secondary side series resonant circuit appears more dominantly than the rectified current ID2.
The secondary winding current I2 flowing through the secondary winding N2 is obtained as a waveform obtained by combining the rectified currents ID1 and ID2.

FIG. 12 shows the switching frequency fs with respect to load fluctuation, the ON period TON, the OFF period TOFF, and the AC → DC power conversion efficiency ηAC → DC for the power supply circuit shown in FIG.
First, looking at AC → DC power conversion efficiency ηAC → DC, it can be seen that a high efficiency of 90% or more is obtained in a wide range from load power Po = 50 W to 200 W. The inventors of the present application have previously confirmed through experiments that such characteristics are obtained when a secondary-side series resonant circuit is combined with a single-ended voltage resonant converter.

  Further, depending on the switching frequency fs, the ON period TON, and the OFF period TOFF in FIG. 12, a switching operation as a constant voltage control characteristic with respect to load fluctuations in the power supply circuit shown in FIG. 10 is shown. In this case, the switching frequency fs is substantially constant with respect to the load fluctuation. On the other hand, the ON period TON and the OFF period TOFF are linearly changed so as to be opposite to each other as illustrated. This means that the switching operation is controlled by changing the time ratio between the on period and the off period while the switching frequency (switching period) is made substantially constant with respect to the fluctuation of the secondary side DC output voltage Eo. It shows that you are doing. Such control can be regarded as PWM (Pulse Width Modulation) control that varies the on / off period within one cycle. By the PWM control, the power supply circuit shown in FIG. 10 stabilizes the secondary side DC output voltage Eo.

FIG. 13 schematically shows the constant voltage control characteristics of the power supply circuit shown in FIG. 10 by the relationship between the switching frequency fs (kHz) and the secondary side DC output voltage Eo.
Since the power supply circuit shown in FIG. 10 includes a primary side parallel resonance circuit and a secondary side series resonance circuit, the resonance impedance characteristic according to the resonance frequency fo1 of the primary side parallel resonance circuit and the resonance of the secondary side series resonance circuit. It has two resonance impedance characteristics in combination with the resonance impedance characteristics corresponding to the frequency fo2. Further, since the power supply circuit shown in FIG. 10 has a relationship of fo1≈2.8 × fo2, as shown in FIG. 13, the secondary side series is connected to the primary side parallel resonance frequency fo1. The resonance frequency fo2 is low.
In addition, assuming constant voltage control characteristics with respect to the switching frequency fs under the condition of a certain AC input voltage VAC, as shown in the figure, the resonance impedance under the resonance impedance corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit is shown. The constant voltage control characteristics at the time of maximum load power Pomax / minimum load power Pomin are shown as characteristic curves A and B, respectively, and are maximum under the resonance impedance corresponding to the resonance frequency fo2 of the secondary side series resonance circuit. The constant voltage control characteristics at the time of load power Pomax / minimum load power Pomin are shown by characteristic curves C and D, respectively. If constant voltage control is attempted with the characteristic shown in FIG. 13 and the rated level tg of the secondary side DC output voltage Eo, the variable range (necessary control range) of the switching frequency fs required for that purpose is , Δfs can be expressed as a section.

The necessary control range Δfs shown in FIG. 13 is the minimum load power corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit from the characteristic curve C at the maximum load power Pomax corresponding to the resonance frequency fo2 of the secondary side series resonance circuit. It reaches the characteristic curve A at Pomin. During that time, it corresponds to the characteristic curve D at the minimum load Pomin corresponding to the resonance frequency fo2 of the secondary side series resonance circuit and the resonance frequency fo1 of the primary side parallel resonance circuit. It crosses the characteristic curve B at the maximum load power Pomax.
For this reason, as a constant voltage control operation of the power supply circuit shown in FIG. 10, the switching frequency fs is substantially fixed and the PWM control state for changing the time ratio of the period TON / TOFF in one switching cycle is as follows. Switching drive control is performed. It should be noted that this means that the maximum load power Pomax = 200 W, load power Po = 100 W, and minimum load power Pomin = 0 W shown in FIGS. 11 (a), 11 (b), and 11 (c) This is also indicated by the fact that the period length is substantially constant and the widths of the periods TOFF and TON are changed.
Such an operation is a state where the resonance impedance (capacitive impedance) of the resonance frequency fo1 of the primary side parallel resonance circuit becomes dominant as the resonance impedance characteristic according to the load fluctuation in the power supply circuit, and the secondary side series resonance circuit It is assumed that the transition to the state where the resonance frequency fo2 (inductive impedance) becomes dominant is performed under a narrow switching frequency variable range (Δfs).

JP 2000-134925 A

The power supply circuit shown in FIG. 10 has the following problems.
In the waveform diagram of FIG. 11 described above, the switching current IQ1 at the time of the maximum load power Pomax shown in FIG. 11A is 0 level until the end point of the off period TOFF which is the turn-on timing, and the on period When TON is reached, first, a negative current flows through the body diode DD, and then reverses to flow through the drain-source of the switching element Q1. This operation shows a state where ZVS (Zero Voltage Switching) is properly performed.
On the other hand, the switching current IQ1 at Po = 120W corresponding to the intermediate load shown in FIG. 11B is the timing before the end of the turn-off timing OFF period TOFF. The movement which flows automatically is obtained. This operation is an abnormal operation in which ZVS is not properly performed.

That is, as shown in FIG. 10, it is known that the voltage resonance type converter including the secondary side series resonance circuit becomes an abnormal operation in which ZVS is not properly executed at the intermediate load. As an actual example of the power supply circuit of FIG. 10, it has been confirmed that such an abnormal operation occurs in the region of the load fluctuation range as section A shown in FIG.
As described above, the voltage resonance type converter including the secondary side series resonance circuit originally has a characteristic that high efficiency can be satisfactorily maintained with respect to load fluctuation. ), A corresponding peak current flows when the switching element Q1 is turned on. This causes an increase in switching loss and causes a decrease in power conversion efficiency.
In any case, when the abnormal operation as described above occurs, for example, a shift occurs in the phase-gain characteristic of the constant voltage control circuit system, and the switching operation is performed in the abnormal oscillation state. For this reason, in the present situation, it is strongly recognized that practical application is difficult.

In view of the above problems, the present invention is configured as a switching power supply circuit as follows.
In other words, a rectifying / smoothing circuit that is formed with at least a rectifying element and a smoothing capacitor, generates a rectified smoothing voltage as a voltage across the smoothing capacitor by inputting a commercial AC power supply and rectifying and smoothing, and a rectified smoothing voltage as a DC input A switching element that performs switching by inputting the voltage; and a switching driving unit that performs switching driving of the switching element.
The first inductor inserted in series with respect to the path through which the rectified and smoothed voltage is input to the switching element is connected in parallel with the switching element, and at least the inductance of the first inductor and its own A primary side parallel resonance capacitor that forms a primary side parallel resonance circuit in which a predetermined first resonance frequency is set by the capacitance.
Further, the second inductor is connected in series with the second inductor, so that at least the predetermined inductance that is regarded as equivalent to the first resonance frequency by the inductance of the second inductor and its own capacitance. A primary side series resonance circuit in which the second resonance frequency is set is formed, and a primary connection circuit is provided so that the series connection circuit of the second inductor and itself is connected in parallel with the switching element. A side series resonant capacitor.
In addition, the second inductor is wound as a primary winding, and a secondary winding in which an alternating voltage is induced by the switching output obtained by the primary winding is wound, and is regarded as loosely coupled. Rectified by inputting the converter transformer in which its own coupling coefficient is set and the alternating voltage induced in the secondary winding of the converter transformer so that the required total coupling coefficient between the primary side and the secondary side can be obtained. A secondary-side DC output voltage generating unit configured to operate and generate a secondary-side DC output voltage is provided.

  In the present invention, the “coupling coefficient” indicates the degree of electromagnetic coupling. A numerical value of 1 indicates the highest degree of coupling, and a numerical value of 0 indicates the lowest degree of coupling (coupling) Not). In addition, the term “coupling coefficient” is generally used as a general term regardless of the form of configuration. However, in order to distinguish it from the coupling coefficient of the converter transformer itself, an electromagnetic wave between the entire primary side and the entire secondary side is used. The degree of effective coupling is called “total coupling coefficient”. For example, when no other inductance component is added to the converter transformer, the value of the coupling coefficient matches the value of the total coupling coefficient.

The power supply circuit having the above configuration forms a circuit form as a class E switching converter on the primary side. The class E switching converter is a type of soft switching converter called a composite resonance type including a parallel resonance circuit (primary side parallel resonance circuit) and a series resonance circuit (primary side series resonance circuit). Further, in the present invention, the resonance frequencies of the primary side parallel resonance circuit and the primary side series resonance circuit are set to be substantially equal. In addition, a power supply circuit is configured with an inductor (second inductor) forming a series resonance circuit (primary side series resonance circuit) in a class E switching converter as a primary winding of a converter transformer. In this way, the primary side switching converter is a class E switching converter, so that, for example, ZVS (Zero Voltage) under a load condition that is an intermediate load, such as an interaction between the primary side parallel resonant circuit and the secondary side series resonant circuit. Switching: Zero voltage switching) This eliminates the factors that prevent operation.
In such a configuration, the total coupling coefficient between the primary side and the secondary side of the converter transformer includes the coupling coefficient of the converter transformer itself, the first inductor and the primary winding (second inductor) of the converter transformer. It is determined by the leakage inductance on the primary side obtained by the equivalent parallel circuit. In consideration of the fact that the total coupling coefficient is set in this way, the present invention sets the coupling coefficient of the converter transformer itself so as to obtain a required total coupling coefficient regarded as loose coupling. ing. This is also a factor for avoiding a state in which the ZVS operation cannot be obtained under a load condition that is an intermediate load.

  In the present invention, the primary class E switching converter performs switching by inputting a rectified and smoothed voltage that is a voltage across a smoothing capacitor that forms a rectifying and smoothing circuit that rectifies and smoothes a commercial AC power supply. ing. At this time, the current flowing from the smoothing capacitor to the class E switching converter becomes a direct current by flowing to the switching element side via the first inductor forming the primary side parallel resonant circuit.

In this way, the present invention provides an abnormal operation in which a ZVS (Zero Voltage Switching) operation cannot be obtained under a load condition range of an intermediate load as a switching power supply circuit having a parallel resonant circuit on the primary side. It will be resolved. As a result, the practical application of the voltage resonance type converter including the secondary side series resonance circuit is easily realized.
Moreover, since the current flowing into the switching converter from the smoothing capacitor of the rectifying and smoothing circuit that generates the rectified and smoothed voltage (DC input voltage) from the commercial AC power supply becomes DC, the capacitance of the component element as the smoothing capacitor is reduced. It is possible to select a general-purpose product, and for example, effects such as cost reduction and downsizing of the smoothing capacitor can be obtained.

Prior to describing the best mode for carrying out the present invention (hereinafter referred to as an embodiment), a switching converter (hereinafter referred to as a class E) that performs switching operation by a class E resonance type, which is the background art of the present embodiment. A basic configuration of the switching converter) will be described with reference to FIGS.
FIG. 8 shows a basic configuration as a class E switching converter. The class E switching converter shown in this figure employs a configuration as a DC / AC inverter that operates in a class E resonance type.
The class E switching converter shown in this figure includes one switching element Q1. In this case, the switching element Q1 is a MOS-FET. The switching element Q1 as the MOS-FET is formed such that the body diode DD is connected in parallel with the drain-source in such a manner that the direction opposite to the drain → source is the forward direction. .
Similarly, a parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.

  The drain of the switching element Q1 is connected to the positive electrode of the DC input voltage Ein through a series connection of the choke coil L10. The source of the switching element Q1 is connected to the negative electrode of the DC input voltage Ein.

  Further, one end of the choke coil L11 is connected to the drain of the switching element Q1, and a series resonant capacitor C11 is connected in series to the other end. An impedance Z serving as a load is inserted between the series resonant capacitor C11 and the negative electrode of the DC input voltage Ein. Specific examples of the impedance Z here include a piezoelectric transformer and a high-frequency fluorescent lamp.

  The class E switching converter having such a configuration is formed by a parallel resonance circuit formed by the inductance of the choke coil L10 and the capacitance of the parallel resonance capacitor Cr, an inductance of the choke coil L11, and a capacitance of the series resonance capacitor C11. It can be regarded as a form of a composite resonance type converter including a series resonance circuit.

FIG. 9 shows the operation of the main part of the class E switching converter having the configuration shown in FIG.
The switching voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a 0 level in the period TON in which the switching element Q1 is on and becomes a sine wave pulse in the period TOFF in which the switching element Q1 is off. This switching pulse waveform is obtained by the resonance operation (voltage resonance operation) of the parallel resonance circuit.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD), and is 0 level in the period TOFF, and in the period TON, first, it flows negatively by flowing through the body diode DD for a certain period from the start time. Thereafter, it is inverted to become positive polarity, and flows from the drain to the source of the switching element Q1.
As the output of the class E switching converter, the current I2 that flows through the series resonant circuit includes the switching current IQ1 that flows through the switching element Q1 (and the body diode DD), and the current that flows through the parallel resonant capacitor Cr. The result is a synthesized waveform that includes a sine wave component.

  Further, depending on the relationship between the switching current IQ1 and the switching voltage V1, it is also shown that the ZVS operation is obtained at the turn-off timing of the switching element Q1, and the ZVS and ZCS operations are obtained at the turn-on timing.

  Further, the current I1 flowing into the class E switching converter from the positive terminal of the DC input voltage Ein through the choke coil L10 sets the relationship of L10> L11 for the inductances of the choke coils L10 and L11. As shown in the figure, a pulsating flow waveform having a predetermined average level is obtained. Such a pulsating waveform can be viewed as an approximate direct current.

In this embodiment, a class E switching converter based on the above basic configuration is applied to a power supply circuit. First, a configuration example of the power supply circuit according to the first embodiment is shown in the circuit diagram of FIG.
In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by the bridge rectifier circuit Di, and the rectified output is charged to the smoothing capacitor Ci. That is, the commercial AC power supply is rectified and smoothed by the rectifying and smoothing circuit including the bridge rectifying circuit Di and the smoothing capacitor Ci. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter.

In this figure, the switching converter that performs the switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage is formed as a class E switching converter based on the basic configuration of FIG.
In this case, a high breakdown voltage MOS-FET is selected as the switching element Q1. In this case, the driving method of the class E switching converter is a separately excited type in which the switching element is switched by the oscillation / drive circuit 2.

The drain of the switching element Q1 is connected to the positive terminal of the smoothing capacitor Ci through a series connection of the choke coil L10. Accordingly, in this case, the DC input voltage (Ei) is supplied to the switching element Q1 through the series connection of the choke coil L10. The source of the switching element Q1 is connected to the primary side ground.
A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.

  Since the MOS-FET is selected for the switching element Q1 in this case, a body diode DD is incorporated so as to be connected in parallel with the source and drain as shown in the figure. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
The primary side parallel resonance capacitor Cr has a primary side parallel resonance circuit (voltage resonance circuit) for the switching current flowing in the switching element Q1 by its own capacitance and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. Form. When the primary side parallel resonant circuit performs a resonance operation, a voltage resonance type operation is obtained as one of the switching operations of the switching element Q1. Accordingly, a sinusoidal resonance pulse waveform is obtained as an end-to-end voltage (drain-source voltage) V1 of the switching element Q1 during the off period.

  Further, a series connection circuit composed of a primary winding N1 of an insulating converter transformer PIT described later and a primary side series resonance capacitor C11 is connected in parallel between the drain and source of the switching element Q1. In this case, the winding end end of the primary winding N1 is connected to the drain of the switching element Q1, and the winding start end is connected to the primary side series resonance capacitor C11. The pole terminal on the side not connected to the primary winding N1 of the primary side series resonance capacitor C11 is connected to the source of the switching element Q1 at the primary side ground potential.

  The oscillation / drive circuit 2 is a gate voltage for switching the MOS-FET based on the oscillation circuit and the oscillation signal obtained by the oscillation circuit in order to drive the switching element Q1 by, for example, separate excitation. A drive signal is generated and applied to the gate of the switching element Q1. Thereby, the switching element Q1 continuously performs on / off operation according to the drive signal waveform. That is, a switching operation is performed.

The insulating converter transformer PIT transmits the switching output of the primary side switching converter to the secondary side in a state where the primary side and the secondary side are galvanically insulated.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. In this embodiment, two sets of secondary windings N2A and N2B are wound as secondary windings as will be described later, but these secondary windings N2A and N2B are particularly distinguished. In addition, when handling collectively, it may be described as secondary winding N2.
By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G having a gap length of about 2 mm or more is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with about k≈0.8 or less is obtained. That is, it is in a more loosely coupled state than the insulating converter transformer PIT of the power supply circuit shown in FIG. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  The primary winding N1 of the insulating converter transformer PIT is an element for forming a primary series resonance circuit in a class E switching converter formed on the primary side, as will be described later, and depends on the switching output of the switching element Q1. Alternate output is obtained.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. The secondary winding N2 in this case is divided into secondary winding portions N2A and N2B with the center tap as a boundary by applying a center tap. The center tap is connected to the secondary side ground.
In addition, in this case, two rectifier diodes Do1 and Do2 and one smoothing capacitor Co are provided as component elements forming the secondary side rectifier circuit. The anode of the rectifier diode Do1 is connected to the end of the secondary winding N2 on the secondary winding portion N2A side, and the anode of the rectifier diode Do2 is connected to the end of the secondary winding N2 on the secondary winding portion N2B side To do. The cathodes of the rectifier diodes Do1 and Do2 are both connected to the positive terminal of the smoothing capacitor Co, and the negative terminal of the smoothing capacitor Co is connected to the center tap of the secondary winding N2 at the secondary side ground. With such a connection mode, a double-wave rectifier circuit is formed as the secondary-side rectifier circuit.

  In the secondary-side double-wave rectifier circuit formed in this way, the secondary winding portion NA corresponds to the half cycle of one polarity of the secondary winding voltage V3 induced in the secondary winding N2. The rectification current ID1 flows through the path of the rectifier diode Do1 → smoothing capacitor Co to charge the smoothing capacitor Co. Further, in response to the other half cycle of the polarity of the secondary winding V3, the rectification current ID2 flows through the path of the secondary winding NB → rectifier diode Do2 → smoothing capacitor Co to charge the smoothing capacitor Co. . In this manner, the double-wave rectification operation is performed in which the smoothing capacitor Co is charged with the rectified current corresponding to the positive and negative half-cycle periods of the secondary winding voltage V3. As a result, as the voltage across the smoothing capacitor Co, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the induced voltage level of the secondary winding N2 is obtained. The secondary side DC output voltage Eo is supplied as electric power to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching element Q 1 is driven mainly by changing the switching frequency according to the input detection output of the control circuit 1.

  As described above, the switching frequency of the switching element Q1 is variably controlled, whereby the primary and secondary resonance impedances in the power supply circuit change, and the primary winding N1 to the secondary winding N2 from the primary winding N1 of the insulating converter transformer PIT change. The amount of power transmitted to the side and the amount of power to be supplied to the load from the secondary side rectifier circuit will change. As a result, an operation for controlling the level of the secondary side DC output voltage Eo is obtained such that the level fluctuation of the secondary side DC output voltage Eo is canceled. That is, the secondary side DC output voltage Eo is stabilized.

The switching converter (Q1, Cr, L10, N1, C11) formed on the primary side of the power supply circuit of the present embodiment formed as described above, and the circuit as the class E converter previously shown in FIG. Comparing the configuration, the switching converter of the present embodiment omits the impedance Z as a load from the circuit of FIG. 8, and replaces the choke coil L11 with the primary winding N1 (leakage inductance L1) of the insulating converter transformer PIT. Can be seen as a replacement. Further, as described above, in the primary side switching converter of the present embodiment, a parallel resonant circuit is formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonant circuit, and the primary winding N1 of the insulating converter transformer PIT is formed. The leakage inductance L1 and the secondary winding N2 form a primary side series resonance circuit.
From this, it can be said that the primary side switching converter of this embodiment is formed as a class E switching converter that performs a class E resonance type switching operation. Then, the switching output (alternate output) obtained by the switching operation of the primary side switching converter is passed through the magnetic coupling in the insulating converter transformer PIT, and the primary winding N1 corresponding to the choke coil L11 to the secondary winding N2. And rectified on the secondary side to obtain a DC output voltage Eo.
Further, the class E switching converter formed as in the present embodiment has a choke coil L10 and a switching element Q1 (and body diode DD) that forms a voltage resonance type converter together with a primary side parallel resonance capacitor Cr. It can also be seen that the configuration is a composite resonance type converter and a soft switching power supply in which a series connection circuit of a primary winding N1 and a primary side series resonance capacitor C11 forming a primary side series resonance circuit are connected in parallel.

  In the configuration of the present embodiment, the primary side series resonant capacitor C11 is connected in series to the primary winding N1 of the insulating converter transformer PIT. A rectifying operation in which secondary-side rectified currents ID1 and ID2 are caused to flow through the smoothing capacitor Co by the secondary-side double-wave rectifier circuit in a state in which no demagnetization is caused by the action of DC blocking by the primary-side series resonance capacitor C11. Obtained properly.

Here, as a practical example of the power supply circuit of the circuit configuration shown in FIG. 1, the main part is configured by setting as follows.
First, for the insulating converter transformer PIT, EER-35 was selected for the core, and a gap length of 1.6 mm was set for the gap G. As for the number of turns (number of turns) T of the primary winding N1 and the secondary winding N2, N1 = 59T, N2A = 30T, and N2B = 20T were selected. Thereby, k = 0.75 is set for the coupling coefficient k of the insulating converter transformer PIT.
As is well known, the EER core is one of the types and standards of the core as a product, and it is also known that this type has EE. In the present application, in the case of the EE type, both types of EER and EE are handled as EE type cores in accordance with the EE-shaped cross section.
For choke coil L10, the core of EER-28 was selected, the gap length formed in the central magnetic leg was set to 1.2mm, and the inductance (leakage inductance) was set to 1.05mH. .
Further, Cr = 6800 pF was selected for the capacitance of the primary side parallel resonant capacitor Cr. The resonance frequency fo1 = 59.6 kHz of the primary side parallel resonance circuit is set by the capacitance setting for the primary side parallel resonance capacitor Cr and the inductance of the choke coil L10.
Further, the primary side series resonance capacitor C11 is selected to be 0.015 μF, and by this capacitance setting and the leakage inductance on the primary side corresponding to the total coupling coefficient kt on the primary side of the insulating converter transformer PIT described below, The resonance frequency fo2 = 60.3 kHz of the primary side series resonance circuit is set. Relatively, it can be said that fo1≈fo2 is set and almost the same value is set.
The corresponding load power is the maximum load power Pomax = 300 W, the minimum load power Pomin = 0 W (no load), and the rated level of the secondary side DC output voltage Eo is 175V.

Here, when the primary winding N1 and the choke coil L10 of the insulating converter transformer PIT in the power supply circuit shown in FIG. 1 are viewed in the switching cycle, it can be regarded as equivalent to being in a parallel connection relationship. In this case, the magnetic flux generated from the choke coil L10 is not coupled to the secondary winding N2 of the insulating converter transformer PIT. From this, it can be considered that the leakage inductance component on the primary side of the insulating converter transformer PIT increases.
For this reason, as described above, for example, k = 0.75 is set as the coupling coefficient k as the insulating converter transformer PIT itself. However, as described above, the leakage on the primary side of the insulating converter transformer PIT is set. As the inductance increases, a value lower than 0.75 can be obtained as a total coupling coefficient of the insulating converter transformer PIT in the power supply circuit. That is, as a power supply circuit, the overall coupling degree between the primary side and the secondary side of the insulating converter transformer PIT is set lower than the coupling coefficient k due to the structure of the insulating converter transformer PIT itself. become. Here, the numerical value regarding the degree of coupling is treated as the total coupling coefficient kt.
As the present embodiment, for example, by setting the predetermined inductance value described above for the choke coil L10, the overall coupling coefficient kt is set to a value that is regarded as loose coupling of about 0.7 or less. In practice, kt = 0.65 is set. In this case, the setting factors for the total coupling coefficient kt are the coupling coefficient k due to the structure of the insulating converter transformer PIT itself and the inductance of the choke coil L10. As described above, for the primary side series resonant circuit formed by connecting the primary winding N1 and the primary side series resonant capacitor C11 in series, the leakage inductance on the primary side according to the total coupling coefficient kt, It is set by the capacitance of the primary side series resonance capacitor C11.

  The waveform diagram of FIG. 3 shows the operation of the main part of the power supply circuit shown in FIG. 1 by the switching period of the switching element Q1, and FIG. I1, switching voltage V1, primary winding current I2, switching current IQ1, series resonance voltage V2, secondary winding voltage V3, and secondary side rectified currents ID1 and ID2 are shown. FIG. 3B shows an input current I1, a switching voltage V1, a primary winding current I2, and a switching current IQ1 when the load power Po, which is an intermediate load, is 225 W. FIG. 3C shows the input current I1, the switching voltage V1, the primary winding current I2, and the switching current IQ1 when the minimum load power Pomin = 0W. The AC input voltage condition is VAC = 100V.

The basic operation of the power supply circuit of FIG. 1 will be described with reference to a waveform diagram when the maximum load power Pomax = 300 W shown in FIG.
The input current I1 is a current that tends to flow from the smoothing capacitor Ci into the primary side switching converter. An inductance larger than the leakage inductance L1 of the primary winding N1 is set in a line between the positive terminal of the smoothing capacitor Co and the drain side of the switching element Q1, which is a path through which the input current I1 flows into the switching element Q1. By inserting the choke coil L10, the input current I1 flows through the choke coil L10. For this reason, the input current I1 becomes a pulsating flow having an average value I0. The input current I1 having such a waveform can be viewed as a direct current. That is, in the present embodiment, the current flowing from the smoothing capacitor Ci into the switching converter is a direct current. The input current I1 flowing through the choke coil L10 is shunted to the series circuit of the primary winding N1 and the primary side series resonance capacitor C11, the switching element Q1 (and the body diode DD), and the primary side parallel resonance capacitor Cr. Flow.

The switching element Q1 performs a switching operation by inputting the voltage (Ei) across the smoothing capacitor Ci as a DC input voltage. The switching voltage V1 is a voltage between the drain and the source of the switching element Q1, and the switching current IQ1 is a current that flows from the drain side to the switching element Q1 (and the body diode DD). The on / off timing of the switching element Q1 is indicated by the switching voltage V1 and the switching current IQ1. One switching cycle is divided into a period TON in which the switching element Q1 is to be turned on and a period TOFF in which the switching element Q1 is to be turned off. It becomes. The resonance pulse of the switching voltage V1 is obtained as a sinusoidal resonance waveform by the resonance operation of the primary side parallel resonance circuit.
The switching current IQ1 is 0 level in the period TOFF. When the period TON ends and the period TON starts and reaches the turn-on timing, first, the switching current IQ1 has a negative waveform by flowing through the body diode DD. By inverting and flowing from the drain to the source, a waveform with positive polarity is obtained.

  The primary winding current I2 is a current that flows through the primary winding N1 in accordance with the switching operation of the switching element Q1, and the switching current IQ1 and the current that flows through the primary parallel resonant capacitor Cr are synthesized with the polarity shown in this figure. It can be seen as a thing. This primary winding current I2 can be viewed as the output current of the primary side series resonant circuit. That is, when the switching element Q1 performs an on / off operation, the voltage resonance pulse, which is the switching voltage V1 in the period TOFF, forms the primary side series resonance circuit, and the primary side series resonance capacitor C11 series connection circuit. To be applied. As a result, the primary side series resonance circuit performs a resonance operation, and the primary winding current I2 becomes a sinusoidal alternating waveform corresponding to the switching period. The series resonance voltage V2 is a voltage across the primary side series resonance capacitor C11. This series resonance voltage V2 also becomes a sinusoidal alternating waveform corresponding to the switching period as shown in the figure.

At the timing when the switching element Q1 is turned off when the period TON ends and the switching element Q1 is turned off, an operation of flowing the positive primary winding current I2 is started so as to charge the primary side parallel resonance capacitor Cr. The switching voltage V1 starts to rise in a sine wave form from the 0 level, and the voltage resonance pulse rises. When the primary winding current I2 is reversed to the negative polarity, the primary side parallel resonant capacitor Cr shifts from charging to discharging, and the voltage resonance pulse decreases in a sinusoidal form from the peak level.
If the voltage resonance pulse waveform as the switching voltage V1 falls to the 0 level, first, the body diode DD is turned on and the negative primary winding current I2 flows. At this time, the switching voltage V1 is at the 0 level, and when the primary winding current I2 flows through the body diode DD for a fixed period, the drain-source of the switching element Q1 is turned on, and the positive primary winding current I2 is Shed. In this way, during the period TON, the primary winding current I2 flows through the switching element Q1 (and the body diode DD), whereby the waveform of the switching current IQ1 is obtained. Such an operation indicates that the ZVS operation by the primary side parallel resonance circuit and the ZCS operation by the primary side series resonance circuit are obtained when the switching element Q1 is turned on and off.

Further, the operation of the secondary side rectifier circuit is shown by the secondary winding voltage V3 and the secondary side rectified currents ID1 and ID2.
Depending on the secondary winding voltage V3, which is an alternating voltage induced in the secondary winding N2, as described above, the rectifier diodes Do1 and Do2 are provided every half cycle of the secondary winding voltage V3. Conducts alternately. Thereby, the secondary winding voltage V3 becomes an alternating waveform clamped by the absolute value level corresponding to the secondary side DC output voltage Eo according to the conduction period of the rectifier diodes Do1 and Do2.
The secondary side rectified currents ID1 and ID2 flow through the smoothing capacitor Co in an alternating manner by a half-wave sine wave shape as shown in the figure. The secondary winding current I2 flowing through the secondary winding N2 is obtained by synthesizing the secondary side rectified currents ID1 and ID2, and has a sine wave shape as shown in the figure. Further, the secondary side rectified currents ID1 and ID2 in this case flow at the same peak level due to substantially the same conduction angle.

  Based on the operation of each unit shown in FIG. 3 (a), when the intermediate load power Po = 225W shown in FIG. 3 (b) and the minimum load power Pomin = 0W shown in FIG. 3 (c). Referring to the waveform, as the operation of the primary side switching converter, the period length of one switching cycle (TOFF + TON) is shortened as the load tends to no load. This indicates that the switching frequency is appropriately changed as a constant voltage control operation according to the load fluctuation in the range of the maximum load power Pomax to the minimum load power Pomin, as will be described later. Further, focusing on the time ratio between the period TOFF and the period TON, the period TOFF tends to increase and the period TON tends to decrease as the light load tends to no load. This indicates that a change in the time ratio between the period TOFF and the period TON by PWM control is also given as a constant voltage control operation according to the load fluctuation in the range of the maximum load power Pomax to the minimum load power Pomin.

  Further, according to the waveform of the switching current IQ1 when the intermediate load power Po = 225 W shown in FIG. 3B, as in the case of FIG. 3A, the body diode due to the negative polarity at the timing when the period TON is started. It can be seen that the operation is flowing through DD. That is, ZVS is appropriately obtained. The same applies to the switching current IQ1 when the minimum load power Pomin = 0 W shown in FIG. This indicates that the power supply circuit shown in FIG. 1 guarantees the ZVS operation in the entire range of the corresponding load power.

FIG. 4 shows experimental results of the power supply circuit shown in FIG. 1 with respect to AC → DC power conversion efficiency (ηAC → DC) with respect to load fluctuation, switching frequency fs, time lengths of periods TON and TOFF, and switching current IQ1. The change characteristic is shown.
As shown in this figure, the switching frequency fs has a substantially constant slope according to the tendency of light load in the range from the maximum load power Pomax = 300 W to the minimum load power Pomin = 0 W (no load). It is changing with the tendency to become higher.
Further, the time length of the period TON during which the switching element Q1 is turned on has a substantially constant slope as the load becomes heavy in the range from the minimum load power Pomin = 0W to the maximum load power Pomax = 300W. It tends to increase. On the other hand, the time length of the period TOFF in which the switching element Q1 is turned off is almost equal to the tendency of heavy load in the range from the minimum load power Pomin = 0W to the maximum load power Pomax = 300W. It tends to decrease with a certain inclination. The degree of inclination is greater in the period TON than in the period TOFF.

Such characteristics of the switching frequency fs and the periods TON and TOFF indicate that switching frequency control for changing the switching frequency fs is first performed as constant voltage control. The change of the switching frequency fs means that the time length of one switching cycle represented by TON + TOFF changes, but in this embodiment, the time of one switching cycle changes as the switching frequency fs increases. When the length (TON + TOFF) becomes shorter, the period TON becomes shorter at a constant rate, whereas the period TOFF becomes longer at a constant rate. Such changes in the periods TON and TOFF within one switching cycle can be regarded as the PWM control for controlling the conduction period of the switching element Q1.
From this, it can be said that the present embodiment is a composite constant voltage control operation in which the switching frequency control and the PWM control are performed simultaneously. Such composite control has high control sensitivity.

Regarding the power supply circuit of FIG. 1, the actual measurement result regarding the constant voltage control is as follows.
First, the variable range (Δfs) of the switching frequency fs required for stabilizing the secondary side DC output voltage Eo at 175 V against load fluctuations of the maximum load power Pomax = 300 W to the minimum load power Pomin = 0 W is as follows. 14.9 KHz, and the fluctuation range (ΔTON) of the period TON is 3.5 μs, and the fluctuation range (ΔTOFF) of the period TOFF is 1.5 μs.

  Further, the AC → DC power conversion efficiency (ηAC → DC) in this case is a characteristic that increases as the light load tends to occur in the range from the maximum load power Pomax = 300 W to the load power Po = 100 W. When the load power Po is about 100 W, a maximum value of 93% or more is obtained. Further, it is a good characteristic that 90% or more is obtained in the range from the maximum load power Pomax = 300 W to the load power Po = 25 W.

  In addition, the switching current IQ1 has a characteristic that becomes higher with a substantially constant slope in accordance with the tendency of heavy load in the range from the minimum load power Pomin = 0 W (no load) to the maximum load power Pomax = 300. is there.

Here, when the power supply circuit of the present embodiment shown in FIG. 1 is compared with the power supply circuit shown in FIG. 10 as a conventional example, the following can be said.
In general, a power supply circuit having a voltage resonance type converter on the primary side has a narrow control range of load power and cannot maintain ZVS at light load, so it is considered impossible to put it to practical use as it is. ing. Therefore, as shown in FIG. 10, a secondary side series resonant circuit is provided for the primary side voltage resonant converter, and a power supply circuit in which a voltage doubler half-wave rectifier circuit is formed as a secondary side rectifier circuit is configured. As a result of experiments conducted by the inventor, it was confirmed that characteristics closer to realization can be obtained than a power supply circuit having a voltage resonant converter so far.
However, in the power supply circuit of FIG. 10, as described with reference to FIG. 11, the switching element Q1 has a positive direction (in this case, drain → source direction) before the OFF period (TOFF) of the switching element Q1 does not end at an intermediate load. ) Causes an abnormal operation in which the ZVS operation cannot be obtained. For this reason, even with the configuration of the power supply circuit of FIG.
On the other hand, in the power supply circuit according to the embodiment shown in FIG. 1, the ZVS operation is obtained over the entire range of the corresponding load power as described with reference to the waveform diagram of FIG. That is, the abnormal operation at the intermediate load is eliminated. As a result, a single-ended voltage resonant converter including a secondary side series resonant circuit can be easily put into practical use.

It has been confirmed that the abnormal operation at the time of intermediate load of the power supply circuit shown in FIG. 10 is likely to occur when a complex resonance type converter having a secondary resonance circuit in the voltage resonance type converter is configured. This is mainly caused by the interaction between the primary side parallel resonant circuit forming the voltage resonant converter and the secondary side series resonant circuit (rectifier circuit) operating simultaneously.
Therefore, in the case of the present embodiment, the abnormal operation at the time of the intermediate load described above is due to the fact that the circuit configuration is a combination of the primary side voltage resonance type converter and the secondary side series resonance circuit. First, the primary side switching converter has a configuration in which a class E switching converter is applied instead of the voltage resonance type converter.
Further, the overall coupling coefficient kt = 0.65 was set, and the degree of coupling between the primary side and the secondary side of the insulating converter transformer PIT was lowered than before. Thus, the interaction between the operation of the primary side switching converter and the rectification operation (switching operation) of the secondary side rectifier circuit is dilute to obtain an operation of suppressing the abnormal operation at the time of intermediate load.
In addition, since the primary side switching converter is a class E switching converter, there is no need to provide a series resonance circuit on the secondary side, so in this embodiment, the secondary side resonance circuit is omitted. ing. When the secondary side resonance circuit is omitted, a half-wave rectification circuit or a double-wave rectification circuit that does not generate a rectification operation in one half cycle of the alternating voltage of the secondary winding can be adopted as the secondary-side rectification circuit. . The power supply circuit of FIG. 1 shows a configuration provided with a double-wave rectifier circuit as a secondary side rectifier circuit.

In the power supply circuit shown in FIG. 10, the current flowing from the smoothing capacitor Ci into the switching converter flows into the switching element Q1 and the primary side parallel resonant capacitor Cr via the primary winding N1 of the insulating converter transformer PIT. In this case, the current flowing from the smoothing capacitor Ci into the switching converter is the primary winding current I1, and has a relatively high frequency due to the switching period. That is, a charging / discharging current flows to the smoothing capacitor Ci at a high frequency with respect to the commercial AC power supply cycle.
An aluminum electrolytic capacitor is often employed depending on the fact that a high breakdown voltage is required for the component element as the smoothing capacitor Ci. Compared to other types of capacitors and the like, aluminum electrolytic capacitors have the property that when operated at a high frequency, the capacitance decreases and the loss angle tangent tends to increase. For this reason, it is necessary to select a special product having a low ESR (equivalent series resistance) and a large allowable ripple current as the aluminum electrolytic capacitor used for the smoothing capacitor Ci. It is also necessary to select a correspondingly large value for the capacitance of the element as the smoothing capacitor Ci. For example, in the configuration of the power supply circuit of FIG. 10, about 1000 μF is selected when the maximum load power Pomax = 300 W equivalent to the present embodiment is supported. Such an aluminum electrolytic capacitor is more expensive than a general-purpose aluminum electrolytic capacitor, and is disadvantageous in terms of cost, including an increase in component price in accordance with an increase in capacitance.

  On the other hand, in the power supply circuit of the present embodiment shown in FIG. 1, the current flowing from the smoothing capacitor Ci to the switching converter is switched via the choke coil L10 to the switching element Q1, the primary side parallel resonant capacitor Cr, the primary side series. It flows through the resonance circuit (N1-C11). For this reason, the current flowing from the smoothing capacitor Co into the switching converter becomes a direct current as shown also as the input current I1 in FIG. In this way, since the current flowing from the smoothing capacitor Ci into the switching converter becomes a direct current, in the present embodiment, the above-described problems of a decrease in capacitance and an increase in tangent of the loss angle do not occur. Therefore, a general-purpose aluminum electrolytic capacitor can be selected as the smoothing capacitor Ci. Also, a lower value than the case of the circuit of FIG. 10 can be selected for the capacitance of the element as the smoothing capacitor Ci. As an actual circuit of FIG. 1, 680 μF can be selected. In this way, in the present embodiment, it is possible to reduce the cost of the smoothing capacitor Ci.

In the power supply circuit according to the present embodiment, the total coupling coefficient kt between the primary side and the secondary side of the insulating converter transformer PIT is set to about 0.65. The total coupling coefficient kt is an apparent increase in the leakage inductance of the primary winding N1 due to the equivalent coupling coefficient k of the insulating converter transformer PIT and the primary winding N1 and the choke coil L10 connected in parallel. It is obtained by.
For example, if an overall coupling coefficient k of about 0.65 is obtained under the configuration of the power supply circuit shown in FIG. 10, about 0.65 is set for the coupling coefficient kt of the insulating converter transformer PIT itself. become. For this purpose, for example, when the insulating converter transformer PIT adopts the structure shown in FIG. 2, the gap G may be enlarged to 2 mm or more to form a core. However, since increasing the gap length causes an increase in eddy current loss (eddy current loss), it is not desirable to increase the gap length unnecessarily. There is a possibility that the loss may not be ignored.
In the case of the present embodiment, the coupling coefficient of the insulating converter transformer PIT itself is obtained by obtaining the value of the total coupling coefficient kt of about 0.65 by the increase in the leakage inductance of the primary winding N1 as described above. k can be suppressed by a decrease to about 0.75. Accordingly, a gap length of about 1.6 mm can be set for the gap G formed in the core of the insulating converter transformer PIT. With such a gap length, there is no need to consider eddy current loss.

Hereinafter, as other embodiments of the present invention, variations of the secondary side rectifier circuit are shown in FIGS.
The circuit diagram of FIG. 5 shows a configuration example of a power supply circuit as the second embodiment. In this figure, the same parts as those in FIG. Also in the power supply circuit shown in this figure, the insulation converter transformer PIT has the same structure as that shown in FIG. 2, and the coupling coefficient k of the insulation converter transformer PIT itself is set to about 0.75. The total coupling coefficient kt between the primary side and the secondary side of the insulating converter transformer PIT is set to about 0.65 by equivalently connecting the primary winding N1 and the choke coil L10 in parallel. This also applies to the power supply circuits shown in FIGS.

The secondary side rectifier circuit of the power supply circuit shown in this figure is a full-wave rectifier circuit including a bridge rectifier circuit.
The secondary winding N2 of the insulating converter transformer PIT in this case corresponds to the secondary side rectifier circuit being a bridge full-wave rectifier circuit, so that one set of secondary winding N2 is not subjected to a center tap and is predetermined. It is provided by being wound by the number of turns. A bridge rectifier circuit composed of four rectifier diodes Do1, Do2, Do3, Do4 and one smoothing capacitor Co are connected to the secondary winding N2 as follows.
The winding end end of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2. Further, the winding start end of the secondary winding N2 is connected to a connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4. The cathode of the rectifier diode Do1 and the cathode of the rectifier diode Do3 are connected to the positive terminal of the smoothing capacitor Co. The negative terminal of the smoothing capacitor Co is connected to the secondary side ground. The anode of the rectifier diode Do2 and the anode of the rectifier diode Do4 are also connected to the secondary side ground.

Depending on the full-wave rectifier circuit formed as described above, a pair of rectifier diodes [Do1, Do4] of the bridge rectifier circuit is formed in one half cycle of the alternating voltage induced (excited) in the secondary winding N2. An operation of conducting and charging the rectified current to the smoothing capacitor Co is obtained. Further, in the other half cycle of the alternating voltage induced in the secondary winding N2, the operation of charging the rectified current to the smoothing capacitor Co by obtaining a set of rectifier diodes [Do2, Do3] conductive.
As a result, a secondary side DC output voltage Eo having a level corresponding to the same level as the level of the alternating voltage induced in the secondary winding N2 is generated as the voltage across the smoothing capacitor Co.

The circuit diagram of FIG. 6 shows a configuration example of a power supply circuit as the third embodiment. In this figure, the same parts as those in FIG. 1 and FIG.
The secondary side rectifier circuit of the power supply circuit shown in this figure is formed as a voltage doubler full wave rectifier circuit.
The secondary winding N2 in this case is divided into secondary winding portions N2A and N2B with the center tap as a boundary by applying a center tap. The number of turns of the secondary windings N2A and N2B is the same. Then, rectifier diodes Do1, Do2, Do3, Do4, capacitor CoA, and smoothing capacitor Co are connected to secondary winding N2 as follows.

The end of the secondary winding N2 on the secondary winding portion N2A side is connected to a connection point between the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2. The end of the secondary winding portion N2B side is connected to a connection point between the anode of the rectifier diode Do3 and the cathode of the rectifier diode Do4.
The center tap of the secondary winding N2 is connected to the positive terminal of the capacitor CoA, and the negative terminal of the capacitor CoA is connected to the secondary side ground together with the connection points of the anodes of the rectifier diodes Do2 and Do4. The cathodes of the rectifier diodes Do1 and Do3 are connected to the positive terminal of the smoothing capacitor Co, and the negative terminal of the smoothing capacitor Co is connected to the secondary side ground.

  The voltage doubler full-wave rectifier circuit thus formed includes a first voltage doubler half-wave rectifier circuit including a secondary winding N2A, rectifier diodes Do1 and Do2, a capacitor CoA, and a smoothing capacitor Co. It can be considered that the second voltage doubler half-wave rectifier circuit including the winding portion N2B, the rectifier diodes Do3 and Do4, the capacitor CoA, and the smoothing capacitor Co is combined. In this case, the first voltage half-wave rectifier circuit and the second voltage half-wave rectifier circuit share the capacitor CoA and the smoothing capacitor Co.

The first voltage doubling half-wave rectifier circuit has a secondary winding portion N2A → capacitor CoA during the half-wave period in which the end of the secondary winding portion N2A is negative with respect to the induced voltage of the secondary winding N2. → Rectifier diode Do2 → Secondary winding portion N2A is passed through a rectified current to charge capacitor CoA, so that the voltage across capacitor CoA corresponds to the same level as the induced voltage level of secondary winding portion N2A. Generate a potential. The secondary winding N2A → rectifier diode Do1 → smoothing capacitor Co → in the half-wave period in which the end on the secondary winding N2A side is positive with respect to the induced voltage V2 of the next secondary winding N2. A rectified current is caused to flow through the path from the capacitor CoA to the secondary winding portion N2B. At this time, the smoothing capacitor Co is charged with a potential obtained by superimposing the voltage across the capacitor CoA on the potential (V2) of the secondary winding portion N2A.
The second voltage doubler half-wave rectifier circuit charges the capacitor CoA by passing a rectified current through the path of the secondary winding N2B → capacitor CoA → rectifier diode Do4 → secondary winding N2B. The induced voltage of the line N2 is executed in a half-wave period in which the end on the secondary winding part N2B side is negative, and the half-wave period in which the end on the next secondary winding part N2B side is positive , A rectification current flows through the path of secondary winding N2B → rectifier diode Do3 → smoothing capacitor Co → capacitor CoA → secondary winding N2B, and the potential of the secondary winding N2B with respect to the smoothing capacitor Co. Charging is performed with a potential obtained by superimposing the voltage across the capacitor CoA on the same as the potential of the secondary winding N2A.
By such an operation, for the smoothing capacitor Co, charging by the superimposed potential of the secondary winding portion N2A and the capacitor CoA and the secondary winding are performed every positive and negative half cycles of the induced voltage of the secondary winding N2. The charging by the superimposed potential of the part N2B and the capacitor CoA is performed alternately. As a result, the secondary side DC output voltage Eo is obtained as the rectified and smoothed voltage at a level corresponding to twice the induced voltage level of the secondary winding N2A or N2B as the voltage across the smoothing capacitor Co.

The circuit diagram of FIG. 7 shows a configuration example of a power supply circuit according to the fourth embodiment. In this figure, the same parts as those in FIG. 1, FIG. 5, and FIG.
The power supply circuit shown in this figure includes a voltage doubler rectifier circuit as a secondary side rectifier circuit. This voltage doubler rectifier circuit is formed by connecting two rectifier diodes Do1 and Do2 and two smoothing capacitors Co1 and Co2 as shown in the figure to a set of secondary winding N2.
In this case, the anode of the rectifier diode Do1 and the cathode of the rectifier diode Do2 are connected to the winding end of the secondary winding N2. The cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co1.
The smoothing capacitors Co1 and Co2 are connected in series so that the negative electrode terminal of the smoothing capacitor Co1 and the positive electrode terminal of the smoothing capacitor Co2 are connected, and the winding of the secondary winding N2 is connected to the connection point of the smoothing capacitors Co1 and Co2. Connect the start end.
The negative terminal of the smoothing capacitor Co2 and the anode of the rectifier diode Do2 are connected to the secondary side ground.

In the secondary side voltage doubler rectifier circuit formed in this way, in the half cycle of one polarity of the induced voltage of the secondary winding N2, the path of the secondary winding N2 → rectifier diode Do1 → smoothing capacitor Co1. The rectified current flows through and the smoothing capacitor Co1 is charged. Further, in the half cycle of the other polarity of the induced voltage of the secondary winding N2, a rectification current flows through the path of the secondary winding N2, rectifier diode Do2, and smoothing capacitor Co2, and the smoothing capacitor Co2 is charged. In this way, the charging to the smoothing capacitor Co1 and the charging to the smoothing capacitor Co2 are alternately performed every positive and negative half cycles of the induced voltage of the secondary winding N2, and each of the smoothing capacitors Co1 and Co2 has two A potential corresponding to an equal magnification of the induced voltage level of the next winding N2 is obtained. As a result, the secondary side DC output voltage Eo having a level corresponding to the same voltage as the induced voltage level of the secondary winding N2 is obtained as the voltage across the series connection circuit of the smoothing capacitors Co1-Co2.
For example, if the secondary side DC output voltage Eo of the same level as the power supply circuit shown in FIG. 1 is obtained, the number of turns (number of turns) of the secondary winding N2 is the number of turns of the secondary winding N2 of FIG. About ½ of that. Thereby, the winding process of the secondary winding N2 can be simplified.

In addition, as this invention, it is not limited to the structure shown as said each embodiment. For example, a detailed circuit configuration of the primary class E switching converter, a configuration of the secondary rectifier circuit, and the like are also conceivable.
As the main switching element (and auxiliary switching element), it is also conceivable to select an element other than the MOS-FET, such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor. In each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

1 is a circuit diagram illustrating a configuration example of a power supply circuit according to a first embodiment of the present invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a wave form diagram which shows operation | movement of the principal part of the power supply circuit of 1st Embodiment with a switching period. It is a figure which shows the fluctuation | variation characteristic of AC-> DC power conversion efficiency with respect to load fluctuation | variation, switching frequency, the ON period of a switching element, an OFF period, and a switching current about the power supply circuit of 1st Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as 2nd Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as 3rd Embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as 4th Embodiment. It is a circuit diagram which shows the basic structural example of a class E switching converter. FIG. 9 is a waveform diagram showing an operation of the class E switching converter shown in FIG. 8. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art example. It is a wave form diagram which shows the operation | movement of the principal part of the power supply circuit shown in FIG. It is a figure which shows the fluctuation characteristic of the ON-period of the AC-> DC power conversion efficiency with respect to load fluctuation | variation, switching frequency, and a switching element about the power supply circuit shown in FIG. It is a figure which shows notionally the constant voltage control characteristic about the conventional power supply circuit.

Explanation of symbols

  1. Control circuit, 2. Oscillation / drive circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, Cr primary side parallel resonance capacitor, N1 primary winding, N2 (N2A, N2B) Secondary winding , C11 Primary side series resonant capacitor, Do1, Do2, Do3, Do4 rectifier diode, Co (secondary side) smoothing capacitor

Claims (1)

A rectifying / smoothing circuit that is formed with at least a rectifying element and a smoothing capacitor, and generates a rectified smoothing voltage as a voltage across the smoothing capacitor by inputting a commercial AC power supply and rectifying and smoothing;
A switching element that performs switching by inputting the rectified and smoothed voltage as a DC voltage;
Switching driving means for switching and driving the switching element;
A first inductor inserted in series with respect to a path through which the rectified and smoothed voltage is input to the switching element;
A primary side that is connected in parallel with the switching element and forms a primary side parallel resonant circuit in which a predetermined first resonance frequency is set by at least the inductance of the first inductor and its own capacitance. A parallel resonant capacitor;
A second inductor;
A predetermined second resonance frequency that is considered to be equivalent to the first resonance frequency by at least the inductance of the second inductor and its own capacitance by being connected in series with the second inductor. Forming a primary side series resonant circuit in which a series connection circuit of the second inductor and itself is connected in parallel with the switching element. When,
The second inductor is wound as a primary winding, and is formed by winding a secondary winding in which an alternating voltage is induced by the switching output obtained in the primary winding. A converter transformer in which its own coupling coefficient is set so that a total coupling coefficient between the primary side and the secondary side can be obtained;
A secondary side DC output voltage generating means configured to input an alternating voltage induced in the secondary winding of the converter transformer and perform a rectifying operation to generate a secondary side DC output voltage;
A switching power supply circuit.

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