JP2007068380A - Switching power circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a single-end resonant converter which becomes ZVS (zero voltage switching) in all corresponding load ranges. <P>SOLUTION: A primary-side switching converter is constituted as an E-class resonance; a first primary-side series resonance circuit of the E-class resonant converter is formed of a leakage inductance generated in a primary winding N1 of an insulation converter transformer PIT and a series resonance capacitor C11; a second primary-side series resonance circuit is formed of an inductance generated in a choke coil winding N10 and a series resonance capacitor C11; and a primary-side parallel resonance circuit is formed of a leakage inductance generated in the primary winding N1 of the insulation converter transformer PIT, an inductance generated in the choke coil winding N10, and a parallel resonance capacitor Cr. Further, a winding ratio between a first secondary winding N2 and a second secondary winding N2' is determined as a secondary winding of the insulation converter transformer PIT so that the size of currents flowing through a rectifier diode D01 and a rectifier diode D02, which are generated due to magnetic flux of different polarity can become the same. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit.

As a so-called soft switching power supply of a resonance type, a current resonance type and a voltage resonance type are widely known. At present, half bridge coupling type current resonance type converters with two stone switching elements are widely used due to the ease of practical use.
However, at present, for example, the characteristics of high-voltage switching elements have been improved, and the problem of withstand voltage in the practical application of a voltage resonance type converter has been cleared. In addition, a single-ended voltage resonant converter with a single switching element is advantageous in terms of input feedback noise and DC output voltage line noise components compared to a single-current resonant forward converter. It is also known that.

FIG. 9 shows a configuration example of a switching power supply circuit including a voltage resonance type converter by a single end system.
In the switching power supply circuit shown in this figure, the commercial AC power supply AC is rectified and smoothed by a rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci, thereby generating a rectified and smoothed voltage Ei as a voltage across the smoothing capacitor Ci. is doing.
Note that the line of the commercial AC power supply AC is provided with a noise filter that includes a pair of common mode choke coils CMC and two across capacitors CL and removes common mode noise.

  The rectified and smoothed voltage Ei is input to the voltage resonant converter as a DC input voltage. As described above, this voltage resonance type converter adopts a single-end configuration including one switching element Q1. In this case, the voltage resonance type converter is a separately excited type, and the switching element Q1 of the MOS-FET is switched by the oscillation / drive circuit 2.

  A MOS-FET body diode DD is connected in parallel to the switching element Q1. A primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.

  The primary side parallel resonant capacitor Cr forms a primary side parallel resonant circuit (voltage resonant circuit) with the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. The primary side parallel resonance circuit can obtain a voltage resonance type operation as the switching operation of the switching element Q1.

  The oscillation / drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1 in order to drive the switching element Q1. Thereby, the switching element Q1 performs a switching operation at a switching frequency corresponding to the cycle of the drive signal.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As the structure of the insulating converter transformer PIT, for example, an EE-shaped core that is a combination of E-shaped cores made of a ferrite material is provided. Then, after dividing the winding part on the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the central magnetic leg of the EE-shaped core.
In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE-shaped core of the insulating converter transformer PIT, whereby k = 0.80 between the primary side and the secondary side. A coupling coefficient k of about ˜0.85 is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained. Further, the value of the coupling coefficient k becomes a setting element of the leakage inductance (L1).

  One end of the primary winding N1 of the insulating converter transformer PIT is inserted between the positive terminal of the switching element Q1 and the smoothing capacitor Ci, so that the switching output of the switching element Q1 is transmitted. Yes. An alternating voltage induced by the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.

In this case, the secondary side series resonant capacitor C2 is connected in series to one end of the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C2 As a result, a secondary side series resonance circuit (current resonance circuit) is formed.
In addition, a double voltage half-wave rectifier circuit is formed by connecting rectifier diodes Do1 and Do2 and a smoothing capacitor Co to the secondary side series resonant circuit as shown in the figure. This voltage doubler half-wave rectifier circuit generates a secondary side DC output voltage Eo at a level corresponding to twice the alternating voltage V2 induced in the secondary winding N2 as a voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied to the load and is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side DC output voltage Eo input as a detection voltage to the oscillation / drive circuit 2.
The oscillation / drive circuit 2 outputs a drive signal having a variable frequency according to the level of the secondary side DC output voltage Eo indicated by the input detection output, so that the secondary side DC output voltage Eo is at a predetermined level. The switching operation of the switching element Q1 is controlled so as to be constant. Thereby, stabilization control of the secondary side DC output voltage Eo is performed.

10 and 11 show experimental results for the power supply circuit having the configuration shown in FIG. In the experiment, the main part of the power supply circuit of FIG. 9 is set as follows.
For the isolated converter transformer PIT, EER-35 is selected for the core, and the gap length of the central magnetic leg is set to 1 mm. The number of turns T (number of turns) of the primary winding N1 and the secondary winding N2 is N1 = 39T and N2 = 23T, respectively, and the induced voltage level per turn (T) of the secondary winding N2 is as follows. 3V / T was set. For the coupling coefficient k of the insulating converter transformer PIT, k = 0.81 was set.
Further, Cr = 3900 pF was selected for the capacitance of the primary side parallel resonant capacitor Cr, and C2 = 0.1 μF was selected for the capacitance of the secondary side series resonant capacitor C2. Accordingly, the resonance frequency fo1 = 230 kHz of the primary side parallel resonance circuit and the resonance frequency fo2 = 82 kHz of the secondary side series resonance circuit are set. In this case, the relative relationship between the resonance frequencies fo1 and fo2 can be expressed as fo1≈2.8 × fo2.
The rated level of the secondary side DC output voltage Eo is 135V, and the corresponding load power is maximum load power Pomax = 200 W to minimum load power Pomin = 0 W.

FIG. 10 is a waveform diagram showing the operation of the main part in the power supply circuit shown in FIG. 9 by the switching cycle of the switching element Q1, and FIG. 10A shows the voltage V1 when the maximum load power Pomax = 200 W, the switching A current IQ1, a primary winding current I1, a secondary winding current I2, and a secondary side rectified current ID1, ID2 are shown. FIG. 10B shows the voltage V1, the switching current IQ1, the primary winding current I1, and the secondary winding current I2 when the intermediate load power Po = 120 W. FIG. 10C shows the voltage V1 and the switching current IQ1 when the minimum load power Pomin = 0 W.
The voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a zero level in the period TON in which the switching element Q1 is on and becomes a sinusoidal resonance pulse in the period TOFF in which the switching element Q1 is off. The resonance pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is a voltage resonance type.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD), and flows as a waveform shown in the figure in the period TON, and is obtained as a waveform that becomes 0 level in the period TOFF.
The primary winding current I1 flowing through the primary winding N1 is a combination of the current component flowing as the switching current IQ1 during the period TON and the current flowing through the primary parallel resonant capacitor Cr during the period TOFF.

Although only shown in FIG. 10A, as the operation of the secondary side rectifier circuit, the rectified currents ID1 and ID2 flowing in the rectifier diodes Do1 and Do2 flow in a sine wave shape as shown in the figure. . In this case, the waveform of the rectified current ID1 is such that the resonance operation of the secondary side series resonant circuit appears more dominantly than the rectified current ID2.
The secondary winding current I2 flowing through the secondary winding N2 is obtained as a waveform obtained by combining the rectified currents ID1 and ID2.

FIG. 11 shows the switching frequency fs with respect to the load fluctuation, the ON period TON, the OFF period TOFF, and the AC → DC power conversion efficiency (ηAC → DC) with respect to the load variation for the power supply circuit shown in FIG.
First, looking at the AC → DC power conversion efficiency (ηAC → DC), it can be seen that a high efficiency of 90% or more is obtained in a wide range from the load power Po = 50 W to 200 W. The inventors of the present application have previously confirmed through experiments that such characteristics are obtained when a secondary-side series resonant circuit is combined with a single-ended voltage resonant converter.

  Further, depending on the switching frequency fs, the ON period TON, and the OFF period TOFF in FIG. 11, a switching operation as a constant voltage control characteristic with respect to load fluctuations in the power supply circuit shown in FIG. 9 is shown. In this case, the switching frequency fs is substantially constant with respect to the load fluctuation. On the other hand, the ON period TON and the OFF period TOFF are linearly changed so as to be opposite to each other as illustrated. This means that the switching operation is controlled by changing the time ratio between the on period and the off period while the switching frequency (switching period) is made substantially constant with respect to the fluctuation of the secondary side DC output voltage Eo. It shows that you are doing. Such control can be regarded as PWM (Pulse Width Modulation) control that varies the on / off period within one cycle. By the PWM control, the power supply circuit shown in FIG. 9 stabilizes the secondary side DC output voltage Eo.

FIG. 12 schematically shows the constant voltage control characteristics of the power supply circuit shown in FIG. 9 by the relationship between the switching frequency fs (kHz) and the secondary side DC output voltage Eo.
Since the power supply circuit shown in FIG. 9 includes a primary side parallel resonance circuit and a secondary side series resonance circuit, the resonance impedance characteristic according to the resonance frequency fo1 of the primary side parallel resonance circuit and the resonance of the secondary side series resonance circuit. It has two resonance impedance characteristics in combination with the resonance impedance characteristics corresponding to the frequency fo2. Further, since the power supply circuit shown in FIG. 9 has a relationship of fo1≈2.8 × fo2, as shown in FIG. 12, the secondary side series is connected to the primary side parallel resonance frequency fo1. The resonance frequency fo2 is low.
In addition, assuming constant voltage control characteristics with respect to the switching frequency fs under the condition of a certain AC input voltage VAC, as shown in the figure, the resonance impedance under the resonance impedance corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit is shown. The constant voltage control characteristics at the time of maximum load power Pomax / minimum load power Pomin are shown as characteristic curves A and B, respectively, and are maximum under the resonance impedance corresponding to the resonance frequency fo2 of the secondary side series resonance circuit. The constant voltage control characteristics at the time of load power Pomax / minimum load power Pomin are shown by characteristic curves C and D, respectively. Then, under the characteristics shown in FIG. 12, when the constant voltage control is attempted with the rated level tg of the secondary side DC output voltage Eo, the variable range (necessary control range) of the switching frequency fs necessary for that purpose is , Δfs can be expressed as a section.

The necessary control range Δfs shown in FIG. 12 is the minimum load power corresponding to the resonance frequency fo1 of the primary side parallel resonance circuit from the characteristic curve C at the maximum load power Pomax corresponding to the resonance frequency fo2 of the secondary side series resonance circuit. It reaches the characteristic curve A at Pomin. During that time, it corresponds to the characteristic curve D at the minimum load Pomin corresponding to the resonance frequency fo2 of the secondary side series resonance circuit and the resonance frequency fo1 of the primary side parallel resonance circuit. It crosses the characteristic curve B at the maximum load power Pomax.
For this reason, as a constant voltage control operation of the power supply circuit shown in FIG. 9, the switching frequency fs is substantially fixed, and the PWM control state for changing the time ratio of the period TON / TOFF in one switching cycle is as follows. Switching drive control is performed. Note that this indicates that one switching cycle (TOFF + TOzN) shown in FIGS. 10A, 10B, and 10C when the maximum load power Pomax = 200 W, the load power Po = 100 W, and the minimum load power Pomin = 0 W is shown. This is also indicated by the fact that the period length is substantially constant and the widths of the periods TOFF and TON are changed.
Such an operation is a state where the resonance impedance (capacitive impedance) of the resonance frequency fo1 of the primary side parallel resonance circuit becomes dominant as the resonance impedance characteristic according to the load fluctuation in the power supply circuit, and the secondary side series resonance circuit It is assumed that the transition to the state where the resonance frequency fo2 (inductive impedance) becomes dominant is performed under a narrow switching frequency variable range (Δfs).

JP 2000-134925 A

The power supply circuit shown in FIG. 9 has the following problems.
In the waveform diagram of FIG. 10 described above, the switching current IQ1 at the maximum load power Pomax shown in FIG. 10A is 0 level until the end of the off period TOFF that is the turn-on timing, and the on period When TON is reached, first, a negative current flows through the body diode DD, and then reverses to flow through the drain-source of the switching element Q1. This operation shows a state where ZVS (Zero Voltage Switching) is properly performed.
On the other hand, the switching current IQ1 at Po = 120W corresponding to the intermediate load shown in FIG. 10B is the timing before the end of the turn-off timing OFF period TOFF, and the switching current IQ1 is noise. The movement which flows automatically is obtained. This operation is an abnormal operation in which ZVS is not properly performed.

In other words, as shown in FIG. 9, it is known that the voltage resonance type converter including the secondary side series resonance circuit has an abnormal operation in which ZVS is not properly executed at an intermediate load. In practice, the power supply circuit of FIG. 9 has been confirmed to have such an abnormal operation in the region of the load fluctuation range as section A shown in FIG.
As described above, the voltage resonance type converter including the secondary side series resonance circuit originally has the characteristic that high efficiency can be satisfactorily maintained with respect to the load fluctuation. ), A corresponding peak current flows when the switching element Q1 is turned on. This causes an increase in switching loss and causes a decrease in power conversion efficiency.
In any case, when the abnormal operation as described above occurs, for example, a shift occurs in the phase-gain characteristic of the constant voltage control circuit system, and the switching operation is performed in the abnormal oscillation state. For this reason, in the present situation, it is strongly recognized that practical application is difficult.

In view of the above problems, the present invention is configured as a switching power supply circuit as follows.
A primary-side rectifying and smoothing circuit having a primary-side rectifying element and a primary-side smoothing capacitor that rectifies and smoothes a commercial AC power supply to generate a rectified and smoothed voltage; a switching element that switches the rectified and smoothed voltage to an AC voltage; and A converter transformer that inputs an AC voltage to the primary winding and generates an AC voltage in the secondary winding; a secondary-side rectifying element that generates an output DC voltage by rectifying and smoothing the AC voltage generated in the secondary winding; In a switching power supply circuit comprising a secondary side rectifying and smoothing circuit having a secondary side smoothing capacitor, and switching element control means for controlling the switching element based on the output DC voltage,
The rectified and smoothed voltage is supplied to one winding end of the primary winding of the converter transformer via a choke coil connected to the output end of the primary side rectifying and smoothing circuit, and the other primary winding of the converter transformer is supplied. The switching element is connected to a winding end to generate the AC voltage, and a series resonance capacitor is connected to a connection point between one winding end of the primary winding of the converter transformer and the choke coil. The resonance frequency is dominated by the first series resonance circuit whose resonance frequency is governed by the inductance of the choke coil and the series resonance capacitor, the leakage inductance generated in the primary winding of the converter transformer, and the series resonance capacitor. Forming a second series resonant circuit that receives the resonance frequency of the first series resonant circuit and the above The resonance frequency of the two series resonance circuits is set to be approximately equal, and the resonance frequency is determined by the primary side parallel resonance capacitor connected in parallel to the switching element, the inductance of the choke coil, and the leakage inductance generated in the primary winding. Forming a parallel resonance circuit to be controlled, and setting the resonance frequency of the parallel resonance circuit to a frequency higher than the resonance frequency of the first series resonance circuit and the resonance frequency of the second series resonance circuit; The secondary winding includes a first secondary winding and a second secondary winding, and the secondary-side rectifying element includes the first secondary winding and the second secondary winding. Currents having different polarities from each other are taken out by the first rectifier diode and the second rectifier diode from each of the windings. And wherein the determining the winding ratio of the secondary winding the diameter so as to be substantially equal to the first secondary winding the second current of.

The power supply circuit having the above configuration includes a primary side rectifying and smoothing circuit having a primary side rectifying element and a primary side smoothing capacitor for rectifying and smoothing commercial AC power to generate a rectified and smoothed voltage, and switching the rectified and smoothed voltage to an AC voltage. A switching element for conversion, a converter transformer that inputs the AC voltage to the primary winding and generates an AC voltage in the secondary winding, and an output DC voltage by rectifying and smoothing the AC voltage generated in the secondary winding A secondary-side rectifying / smoothing circuit having a secondary-side rectifying element and a secondary-side smoothing capacitor, and switching element control means for controlling the switching element based on the output DC voltage. Therefore, a predetermined output DC voltage can be obtained from the commercial AC power supply.
And it has the following features. The rectified and smoothed voltage is supplied to one winding end of the primary winding of the converter transformer via a choke coil connected to the output end of the primary side rectifying and smoothing circuit, and the other primary winding of the converter transformer is supplied. The AC voltage is generated by connecting the switching element to a winding end. Therefore, it becomes close to the so-called class E operation, and the current supplied from the smoothing rectifier circuit has a pulsating waveform close to direct current.
In addition, by connecting a series resonant capacitor to the connection point between one winding end of the primary winding of the converter transformer and the choke coil, the resonance frequency is governed by the inductance of the choke coil and the series resonant capacitor. And a second series resonance circuit whose resonance frequency is governed by the leakage inductance generated in the primary winding of the converter transformer and the series resonance capacitor. The resonance frequency of the first series resonance circuit and the resonance frequency of the second series resonance circuit are set to be substantially equal, and the primary side parallel resonance capacitor connected in parallel to the switching element, the inductance of the choke coil, and the primary winding A parallel circuit whose resonance frequency is dominated by the leakage inductance generated in the wire. To form a circuit to set the resonance frequency of the parallel resonant circuit to a frequency higher than the resonant frequency of the resonant frequency and the second series resonant circuit of the first series resonant circuit. Therefore, a so-called current / voltage resonance type converter is configured, and the output DC voltage can be controlled by controlling the frequency of the AC voltage within a narrow range.
The secondary winding includes a first secondary winding and a second secondary winding, and the secondary-side rectifying element includes the first secondary winding and the second secondary winding. Current from each of the two secondary windings having different polarities is taken out by the first rectifier diode and the second rectifier diode, and the magnitudes of the currents having different polarities from each other are obtained. The winding ratio between the first secondary winding and the second secondary winding is determined so that the two are substantially equal. Therefore, current can be efficiently extracted from the secondary winding.

In this way, the present invention provides an abnormal operation in which a ZVS (Zero Voltage Switching) operation cannot be obtained under a load condition range of an intermediate load as a switching power supply circuit having a parallel resonant circuit on the primary side. It will be resolved.
Moreover, since the current flowing into the switching converter from the smoothing capacitor of the rectifying and smoothing circuit that generates the rectified and smoothed voltage (DC input voltage) from the commercial AC power supply becomes DC, the capacitance of the component element as the smoothing capacitor is reduced. It is possible to select a general-purpose product, and for example, effects such as cost reduction and downsizing of the smoothing capacitor can be obtained.
Furthermore, as described above, the power loss is reduced according to the reduction in the amount of current flowing in the power supply circuit, so that the overall power conversion efficiency characteristic is greatly improved.

Prior to describing the best mode for carrying out the present invention (hereinafter referred to as an embodiment), a switching converter (hereinafter referred to as a class E) that performs switching operation by a class E resonance type, which is the background art of the present embodiment. A basic configuration of the switching converter) will be described with reference to FIGS.
FIG. 6 shows a basic configuration as a class E switching converter. The class E switching converter shown in this figure employs a configuration as a DC-AC inverter that operates in a class E resonance type.
The class E switching converter shown in this figure includes one switching element Q1. In this case, the switching element Q1 is a MOS-FET. In the switching element Q1 as the MOS-FET, a body diode DD is formed so as to be connected in parallel between the drain and the source. The forward direction of the body diode DD in this case is along the direction from the source to the drain.
Similarly, a primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.

  The drain of the switching element Q1 is connected to the positive electrode of the DC input voltage Ein through a series connection of the choke coil L10. The source of the switching element Q1 is connected to the negative electrode of the DC input voltage Ein.

  Further, one end of the choke coil L11 is connected to the drain of the switching element Q1, and a series resonant capacitor C11 is connected in series to the other end. An impedance Z serving as a load is inserted between the series resonant capacitor C11 and the negative electrode of the DC input voltage Ein. Specific examples of the impedance Z here include a piezoelectric transformer and a high-frequency fluorescent lamp.

  The class E switching converter having such a configuration is formed by a parallel resonance circuit formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonance capacitor Cr, an inductance of the choke coil L11, and a capacitance of the series resonance capacitor C11. It can be considered that this is a form of a composite resonance type converter including a series resonance circuit. In addition, it can be said that it is the same as a single-ended voltage resonant converter in that it is formed with only one switching element.

FIG. 7 shows the operation of the main part of the class E switching converter having the configuration shown in FIG.
The switching voltage V1 is a voltage obtained at both ends of the switching element Q1, and has a waveform that is a 0 level in the period TON in which the switching element Q1 is on and becomes a sine wave pulse in the period TOFF in which the switching element Q1 is off. This switching pulse waveform is obtained by the resonance operation (voltage resonance operation) of the parallel resonance circuit.

The switching current IQ1 is a current that flows through the switching element Q1 (and the body diode DD), and is 0 level in the period TOFF, and in the period TON, first, it flows negatively by flowing through the body diode DD for a certain period from the start time. Thereafter, it is inverted to become positive polarity, and flows from the drain to the source of the switching element Q1.
As the output of the class E switching converter, the current I2 that flows through the series resonant circuit includes the switching current IQ1 that flows through the switching element Q1 (and the body diode DD) and the current that flows through the primary side parallel resonant capacitor Cr. It becomes a composite and becomes a waveform including a sine wave component.

  Further, depending on the relationship between the switching current IQ1 and the switching voltage V1, it is also shown that the ZVS operation is obtained at the turn-off timing of the switching element Q1, and the ZVS and ZCS operations are obtained at the turn-on timing.

  The current I1 flowing into the class E switching converter from the positive terminal of the DC input voltage Ein through the choke coil L10 sets the relationship of L10> L11 for the inductances of the choke coils L10 and L11. As shown in the figure, a pulsating flow waveform having a predetermined average level is obtained. Such a pulsating waveform can be viewed as an approximate direct current.

  The inventor of the present application applied a class E switching converter based on the above basic configuration to configure a power supply circuit, and conducted an experiment on this power supply circuit. A configuration example of this power supply circuit is shown in a circuit diagram of FIG.

  In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by the bridge rectifier circuit Di, and the rectified output is charged to the smoothing capacitor Ci. That is, the commercial AC power supply is rectified and smoothed by the rectifying and smoothing circuit including the bridge rectifying circuit Di and the smoothing capacitor Ci. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter.

In this figure, the switching converter that performs the switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage is formed as a class E switching converter based on the basic configuration of FIG.
In this case, a high breakdown voltage MOS-FET is selected as the switching element Q1. In this case, the driving method of the class E switching converter is a separately excited type in which the switching element is switched by the oscillation / drive circuit 2.

The drain of the switching element Q1 is connected to the positive terminal of the smoothing capacitor Ci through a series connection of the choke coil L10. Accordingly, in this case, the DC input voltage (Ei) is supplied to the switching element Q1 through the series connection of the choke coil L10. The source of the switching element Q1 is connected to the primary side ground. The inductor (first inductor) as the choke coil L10 is a functional part corresponding to the choke coil L10 in the class E switching converter shown in FIG.
A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.

  Since the MOS-FET is selected for the switching element Q1 in this case, a body diode DD is incorporated so as to be connected in parallel with the source and drain as shown in the figure. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side parallel resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
The primary side parallel resonance capacitor Cr has a primary side parallel resonance circuit (voltage resonance circuit) for the switching current flowing in the switching element Q1 by its own capacitance and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. Form. When the primary side parallel resonant circuit performs a resonance operation, a voltage resonance type operation is obtained as one of the switching operations of the switching element Q1. Accordingly, a sinusoidal resonance pulse waveform is obtained as an end-to-end voltage (drain-source voltage) V1 of the switching element Q1 during the off period.

  Further, a series connection circuit composed of a primary winding N1 of an insulating converter transformer PIT described later and a primary side series resonance capacitor C11 is connected in parallel between the drain and source of the switching element Q1. In this case, the winding end end of the primary winding N1 is connected to the drain of the switching element Q1, and the winding start end is connected to the primary side series resonance capacitor C11. The pole terminal on the side not connected to the primary winding N1 of the primary side series resonance capacitor C11 is connected to the source of the switching element Q1 at the primary side ground potential.

  The oscillation / drive circuit 2 is a gate voltage for switching the MOS-FET based on the oscillation circuit and the oscillation signal obtained by the oscillation circuit in order to drive the switching element Q1 by, for example, separate excitation. A drive signal is generated and applied to the gate of the switching element Q1. Thereby, the switching element Q1 continuously performs on / off operation according to the drive signal waveform. That is, a switching operation is performed.

  The insulating converter transformer PIT transmits the switching output of the primary side switching converter to the secondary side in a state where the primary side and the secondary side are galvanically isolated. For this purpose, the primary winding N1 and the secondary side Winding N2 is wound.

As an example of the structure of the insulating converter transformer PIT in this case, an EE-shaped core obtained by combining an E-shaped core made of a ferrite material is provided. Then, after dividing the winding part on the primary side and the secondary side, the primary winding N1 and the secondary winding N2 are wound around the central magnetic leg of the EE-shaped core.
In addition, a gap of about 1.6 mm is formed with respect to the central magnetic leg of the EE-shaped core of the insulating converter transformer PIT, whereby k = 0.75 between the primary side and the secondary side. A degree of coupling coefficient k is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained.

  The primary winding N1 of the insulating converter transformer PIT is an element for forming a primary series resonance circuit in a class E switching converter formed on the primary side, as will be described later, and depends on the switching output of the switching element Q1. Alternate output is obtained.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2.
A secondary side series resonance capacitor C2 is connected to the secondary winding N2 by a serial connection relationship. Thus, a secondary side series resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side series resonance capacitor C2. This secondary side series resonance circuit performs a resonance operation in accordance with a rectification operation of a secondary side rectifier circuit described later, and thereby the secondary winding current flowing in the secondary winding N2 becomes a sine wave. That is, a current resonance operation is obtained on the secondary side.

  The secondary side rectifier circuit in this case has two rectifier diodes Do1 and Do2 and one smoothing with respect to the secondary winding N2 to which the secondary side series resonant capacitor C2 is connected in series as described above. By connecting the capacitor Co, a double voltage half-wave rectifier circuit is formed. As a connection mode of the voltage doubler half-wave rectifier circuit, first, the anode of the rectifier diode Do1 and the rectifier diode Do2 are connected to the winding end end side of the secondary winding N2 via the secondary side series resonant capacitor C2. Connect the cathode. Further, the cathode of the rectifier diode Do1 is connected to the positive terminal of the smoothing capacitor Co. The winding start end of the secondary winding N2 and the anode of the rectifier diode Do2 are connected to the negative terminal of the smoothing capacitor Co at the secondary side ground potential.

The rectification operation of the voltage doubler half-wave rectifier circuit thus formed is as follows.
First, in the half cycle corresponding to one polarity of the voltage across the secondary winding N2 (secondary winding voltage), which is an alternating voltage induced in the secondary winding N2, a forward voltage is applied to the rectifier diode Do2. As a result, the rectifier diode Do2 becomes conductive, and an operation of charging the rectified current to the secondary side series resonant capacitor C2 is obtained. As a result, a voltage across the level corresponding to the same multiple of the alternating voltage level induced in the secondary winding N2 is generated in the secondary side series resonance capacitor C2. In the next half cycle corresponding to the other polarity of the secondary winding voltage V2, a forward voltage is applied to the rectifier diode Do2 to conduct. At this time, the smoothing capacitor Co is charged with a potential obtained by superimposing the potential of the secondary winding voltage V1 and the voltage across the secondary side series resonance capacitor C2.
As a result, the secondary side DC output voltage Eo having a level corresponding to twice the alternating voltage level excited by the secondary winding N2 is obtained as the voltage across the smoothing capacitor Co.
In this rectification operation, the smoothing capacitor Co is charged only in one half cycle of the alternating voltage excited by the secondary winding N2. That is, a rectification operation as a double voltage half wave is obtained. Further, in such a rectifying operation, it may be considered that the rectifying operation is performed on the resonance output of the secondary side series resonance circuit formed by the series connection of the secondary winding N2 and the secondary side series resonance capacitor C2. it can.
The secondary side DC output voltage Eo thus generated is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, the switching frequency is varied in accordance with the input detection output of the control circuit 1, and the time ratio (conduction angle) between the ON period TON and the OFF period TOFF in one switching cycle is accordingly accompanied. Is switched so as to drive the switching element Q1. This operation is a constant voltage control operation for the secondary side DC output voltage.

  By variably controlling the switching frequency and conduction angle of the switching element Q1 as described above, the primary side and secondary side resonance impedances and the power transmission effective period in the power supply circuit are changed, and the primary winding of the insulating converter transformer PIT is changed. The amount of power transmitted from the line N1 to the secondary winding N2 side and the amount of power to be supplied from the secondary side rectifier circuit to the load will change. As a result, an operation for controlling the level of the secondary side DC output voltage Eo is obtained such that the level fluctuation of the secondary side DC output voltage Eo is canceled. That is, the secondary side DC output voltage Eo is stabilized.

The switching converter (Q1, Cr, L10, N1, C11) formed on the primary side of the power supply circuit of FIG. 8 formed as described above, and the circuit configuration as the class E converter previously shown in FIG. 8, the switching converter in FIG. 8 omits the load impedance Z from the circuit of FIG. 6, and the winding of the choke coil L11 is replaced with the primary winding N1 (leakage inductance L1) of the insulating converter transformer PIT. It can be seen as a replacement. Further, in the primary side switching converter in FIG. 8, a primary side parallel resonance circuit is formed by the inductance of the choke coil L10 and the capacitance of the primary side parallel resonance capacitor Cr, and the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT A primary side series resonance circuit is formed by the capacitance of the primary side series resonance capacitor C11.
From this, it can be said that the primary side switching converter of FIG. 8 is formed as a class E switching converter that performs a class E resonance type switching operation. Then, the switching output (alternate output) obtained by the switching operation of the primary side switching converter is passed through the magnetic coupling in the insulating converter transformer PIT, and the primary winding N1 corresponding to the choke coil L11 to the secondary winding N2. And rectified on the secondary side to obtain a DC output voltage Eo. That is, the power supply circuit shown in FIG. 8 is configured as a DC-DC converter including a class E switching converter on the primary side.
In addition, the primary class E switching converter formed in this way has a choke coil L10 and a primary side parallel resonant capacitor Cr together with the switching element Q1 (and body diode DD) forming a voltage resonant converter. It can also be regarded as a configuration of a composite resonance type converter and a soft switching power source in which a series connection circuit of a primary winding N1 and a primary side series resonance capacitor C11 forming a primary side series resonance circuit are connected in parallel.

By the way, in general, a power supply circuit having a voltage resonance type converter on the primary side has a narrow control range of load power and cannot maintain ZVS at light load, so that it cannot be put into practical use as it is. It is considered. Therefore, as shown in FIG. 9 as a prior art example, a power supply circuit in which a secondary side series resonant circuit is provided for a primary side voltage resonant converter and a voltage doubler half-wave rectifier circuit is formed as a secondary side rectifier circuit. As a result of an experiment conducted by the inventor of the present invention, it was confirmed that characteristics closer to realization can be obtained than a power supply circuit having a voltage resonant converter so far.
However, in the power supply circuit of FIG. 9, as described with reference to FIG. 10, in the intermediate load, the switching element Q <b> 1 is in the positive direction (in this case, the drain → source direction) before the switching element Q <b> 1 is off. ) Causes an abnormal operation in which the ZVS operation cannot be obtained. For this reason, even with the configuration of the power supply circuit of FIG. 9, it was still difficult to put into practical use.

As described above, the power supply circuit described with reference to FIG. 8 is a composite resonance type switching converter having a circuit form of a voltage resonance type converter on the primary side. It can be said that it has a common configuration.
However, when the power supply circuit of FIG. 8 was tested, it was confirmed that the abnormal operation in which ZVS could not be obtained at the intermediate load was eliminated, and normal switching operation was obtained over the entire range of the predetermined corresponding load power. It was.

It has been confirmed that the abnormal operation at the intermediate load of the power supply circuit shown in FIG. 9 is likely to occur when a complex resonant converter of the type provided with a secondary series resonant circuit is configured in the voltage resonant converter. This is mainly caused by the interaction between the primary side parallel resonant circuit forming the voltage resonant converter and the secondary side series resonant circuit (rectifier circuit) operating simultaneously. That is, it can be understood that the abnormal operation at the time of the intermediate load described above is mainly due to the circuit configuration in which the primary side voltage resonance type converter and the secondary side series resonance circuit are combined. Based on this, first, the power supply circuit shown in FIG. 8 has a configuration in which a class E switching converter is applied as the primary side switching converter instead of the voltage resonance type converter.
Due to such a configuration, the power supply circuit of FIG. 8 has an abnormal operation in which ZVS cannot be obtained at an intermediate load regardless of whether or not a series resonance circuit is provided on the secondary side. It was canceled.

In this manner, in the power supply circuit of FIG. 8, the abnormal operation at the intermediate load, which is a problem in the power supply circuit of FIG. 9 as the conventional example, is eliminated.
However, in the power supply circuit of FIG. 8, the choke coil L10 that is not provided in the power supply circuit of FIG. 9 is inserted into the line through which the DC input voltage flows into the switching converter. The choke coil L10 is, for example, about 1 mH, and has a considerably large inductance as compared with the primary winding N1 of the insulating converter transformer PIT corresponding to the choke coil L11. For this reason, the power loss due to iron loss, copper loss, etc. in the choke coil L10 is correspondingly large, and as a result, the reduction in power conversion efficiency of the entire power supply circuit becomes correspondingly significant. For example, in comparison with the power supply circuit of FIG. 9, it has been confirmed by experiments that the value of AC → DC power conversion efficiency (ηAC → DC) of the power supply circuit of FIG. 8 is reduced by about 1%.
Further, in the power supply circuit shown in FIG. 8, since it is necessary to set an inductance that is considerably large for the choke coil L10 as described above, relatively large components such as a core for forming the choke coil L10 are used. Will be selected. This becomes a hindrance to promoting cost reduction and circuit miniaturization.

Therefore, in the present embodiment, the power supply circuit shown in FIG. 8 is further advanced, and the class E switching converter is applied as the power supply circuit to eliminate the abnormal operation at the intermediate load. The present invention proposes a configuration for improving the power conversion efficiency by compensating for the decrease in the power conversion efficiency, and reducing the size of the parts corresponding to the choke coil L10.
As such a power supply circuit according to this embodiment, a configuration example of the power supply circuit according to the first embodiment is shown in FIG. In this figure, the same parts as those in FIG.

First, in the power supply circuit shown in FIG. 1, one end of the choke coil winding N10 is connected to the positive terminal of the smoothing capacitor Ci, and the other end of the choke coil winding N10 is connected to one of the primary windings N1. Connect to the end. In this case, the primary side series resonant capacitor C11 has one end connected to a connection point between one end of the primary winding N1 and the other end of the choke coil winding N10. The end of the other pole is connected to the connection point between one pole of the primary side parallel resonant capacitor Cr and the source of the switching element Q1 at the primary side ground potential, so that the primary side series resonant capacitor C11 and the primary side are connected. A series connection relationship with the winding N1 is obtained.
Also in this case, the primary side parallel resonant capacitor Cr is connected in parallel with the source-drain of the switching element Q1.

  In the present embodiment, the choke coil winding N10 inserted as described above corresponds to the winding as the choke coil L10 in FIG. 6 or FIG. In the present embodiment, the choke coil winding N10 is wound around a core of a predetermined shape size, thereby constituting a component element as the choke coil PCC.

In the circuit configuration as described above, the primary side parallel resonant circuit includes a series connection circuit including the choke coil winding N10 and the primary winding N1, and a primary side parallel resonance capacitor Cr connected in parallel to the series connection circuit. Is formed by the combined inductance obtained by the inductance L10 of the choke coil winding N10 (choke coil PCC), the leakage inductance L1 of the primary winding N1, and the capacitance of the primary parallel resonant capacitor Cr. Become.
The primary side series resonant circuit is formed by the capacitance of the primary side series resonant capacitor C11 and the leakage inductance L1 of the primary winding N1, based on the series connection of the primary side series resonant capacitor C11 and the primary winding N1. The first primary side series resonant circuit is provided. Further, based on the series connection of the choke coil winding N10 and the primary side series resonance capacitor C11, a second primary side series formed by the inductance L10 of the choke coil winding N10 and the capacitance of the primary side series resonance capacitor C11. A resonance circuit is provided.

In response to the switching operation of the switching element Q1, a charge / discharge current is supplied to the primary side parallel resonance capacitor Cr during the period when the switching element Q1 is turned off by the voltage resonance operation of the primary side parallel resonance circuit. This charging / discharging current generates a substantially half-wave sinusoidal resonant pulse voltage as the voltage across the primary parallel resonant capacitor Cr. In the circuit of FIG. 1, since the primary winding N1 is inserted in the primary side parallel resonance circuit, the resonance pulse voltage is superimposed on the alternating voltage generated according to the switching current in the primary winding N1. Occurs.
The first primary-side series resonance circuit performs a resonance operation so that a resonance current flows through the path of the primary-side series resonance capacitor C11-primary winding N1-switching element Q1 when the switching element Q1 is on.
Further, the second primary side series resonance circuit performs a resonance operation such that a resonance current flows through the path of the primary side series resonance capacitor C11-choke coil winding N10-smoothing capacitor Ci according to the switching operation of the switching element Q1. I do.
In this way, the first primary side series resonant circuit and the second primary side series resonant circuit operate in combination, so that, for example, a series resonant current that should flow through the primary winding N1 is applied to the choke coil winding N10. However, it will flow separately.
Also, the primary side series resonance circuit corresponding to the first primary side series resonance circuit corresponding to the first primary side series resonance capacitor C11−the primary winding N1−the switching element Q1 and the second primary side series resonance circuit. When the current path of the capacitor C11, the choke coil winding N10, and the smoothing capacitor Ci is viewed in an alternating manner with a switching cycle, the current paths of both are in a parallel relationship with the primary side series resonance capacitor C11 in common. It can be seen as being.

Since the resonance current is shunted in this way, the inductance (L1) obtained at both ends of the primary side winding N1 and the inductance (L10) obtained at both ends of the choke coil winding N10 can be made equal. The size of the choke coil PCC provided with the coil winding N10 can be reduced. The resonance frequency of the first primary side series resonance circuit is governed by the value of the leakage inductance (L1) generated in the primary winding N1 and the value of the primary side series resonance capacitor C11, and the second primary side series resonance circuit. The inductance (L1) obtained at both ends of the primary side winding N1 and the choke coil winding are controlled by the value of the inductance (L10) obtained at the choke coil winding N10 and the value of the primary side series resonance capacitor C11. By making the inductance (L10) obtained at both ends of N10 equal, the resonance frequency of the first primary side series resonance circuit is made equal to the resonance frequency of the second primary side series resonance circuit, and the switching element Q1 On the other hand, good control characteristics can be obtained even if the variable frequency of the AC signal supplied from the transmission / drive circuit 2 forming part of the control means is narrow.
The resonance frequency is “dominated” means that the value of the resonance frequency of the first primary side series resonance circuit is equal to the value of the leakage inductance (L1) generated in the primary winding N1 and the value of the primary side series resonance capacitor C11. Greatly depends on the value of the inductance (L10) obtained in the choke coil winding N10 and the value of the primary side series resonance capacitor C11, and greatly depends on the value of the resonance frequency of the second primary side series resonance circuit. For example, the smoothing capacitor Ci also forms part of the second primary side series resonance circuit, but the value of the smoothing capacitor Ci is smaller than the value of the primary side series resonance capacitor C11. It is so large that it does not dominate the resonant frequency.

On the secondary side, an intermediate tap is applied to the secondary winding so that the currents flowing through the rectifier diode D01 and the rectifier diode D02 are equal. If the currents flowing through the rectifier diode D01 and the rectifier diode D02 are set to be equal in this way, the load current flows uniformly over the entire period, and the efficiency of the switching power supply circuit can be improved.
Further, since the secondary side partial voltage resonance capacitor C3 is provided, it is possible to prevent partial voltage resonance and switching loss from occurring at the on / off switching point of the rectifier diode D01 and the rectifier diode D02. Can be improved.

Hereinafter, further details of the switching power supply circuit shown in FIG. 1 will be described. FIG. 5 shows a structural example of the insulating converter transformer PIT provided in the power supply circuit of FIG. 1 having the above configuration.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-shaped cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 and the secondary winding N2 ′ are wound around the other winding portion.
By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE-shaped core (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE-shaped core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained.

  In addition, a gap G having a gap length of about 1.6 mm or more is formed on the central magnetic leg of the EE-shaped core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with about k≈0.75 is obtained. That is, as a conventional technique, the state is more loosely coupled than the insulating converter transformer PIT of the power supply circuit shown in FIG. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs. In the present embodiment, EER-35 is used as the core material, the gap G is 2.2 mm, the number of turns of the primary winding N1 is 50T, the number of turns of the secondary winding N2 is 35T, and the secondary winding N2. The number of turns of 'was set to 45T. For the coupling coefficient k between the primary side and the secondary side in the insulating converter transformer PIT itself, a value that is smaller than, for example, k = 0.7 is set.

  The choke coil PCC can also be configured by winding the EE-shaped core having a predetermined shape size. In the present embodiment, ER-28 is used as the core material, the gap G is 1.2 mm, the number of turns of the choke coil winding N10 is 45T, and the inductance value of the inductor L10 is 303 μH.

And about the power supply circuit of the circuit form shown in FIG. 1, when obtaining the experimental result mentioned later, the principal part was selected as follows.
Regarding the respective capacitances of the primary side parallel resonant capacitor Cr, the primary side series resonant capacitor C11, and the secondary side partial voltage resonant capacitor C3,
Cr = 5600pF
C11 = 0.027μF
C3 = 220pF
Was selected.
The primary side parallel resonance frequency f01 is 85.8 kHz, the primary side series resonance frequency f02 composed of the primary side series resonance capacitor C11 and the inductor L1 is 54.8 kHz, and the primary side composed of the primary side series resonance capacitor C11 and the inductor L10. The series resonance frequency f02 ′ is 55.7 kHz, the values of the two primary side series resonance frequencies are substantially equal, and the frequencies of the primary side series resonance frequency f02 and the primary side series resonance frequency f02 ′ are compared to the primary side parallel resonance frequency f01. Set low.
Further, the corresponding load power is the maximum load power Pomax = 300 W, the minimum load power Pomin = 0 W (no load), and the rated level of the secondary side DC output voltage Eo is 175V.

As an experimental result of the power supply circuit of FIG. 1, waveform diagrams of FIGS. 2 (a) and 2 (b) are given. In FIG. 2A, switching voltage V1, switching current IQ1, primary winding voltage V2, choke coil current I1, primary winding current I2 under the conditions of maximum load power Pomax = 300 W and AC input voltage VAC = 100V. , Secondary side alternating voltage V3, and secondary winding current I3.
In FIG. 2B, switching voltage V1, switching current IQ1, primary winding voltage V2, choke coil current I1, primary winding current under the conditions of minimum load power Pomax = 0W and AC input voltage VAC = 100V. I2, secondary side alternating voltage V3, and secondary winding current I3 are shown.

The basic operation of the power supply circuit of FIG. 1 will be described with reference to the waveform diagram shown in FIG.
The input current I1 is a current that tends to flow from the smoothing capacitor Ci into the primary side switching converter. The input current I1 flows through the combined inductance of the inductance L10 of the choke coil winding N10 and the leakage inductance L1 of the primary winding N1. For this reason, the current flowing from the smoothing capacitor Ci into the switching converter becomes a pulsating flow.

The switching element Q1 performs a switching operation by inputting the voltage (Ei) across the smoothing capacitor Ci as a DC input voltage. The switching voltage V1 is a voltage between the drain and source of the switching element Q1.
The switching current IQ1 is a current that flows from the drain side to the switching element Q1 (and the body diode DD). The switching cycle is divided into a period TON in which the switching element Q1 is to be turned on and a period TOFF in which the switching element Q1 is to be turned off, and the switching voltage V1 has a waveform in which a resonance pulse is obtained in the period TOFF with a zero level. Become. The voltage resonance pulse of the switching voltage V1 is obtained as a sinusoidal resonance waveform by the resonance operation of the primary side parallel resonance circuit.
The switching current IQ1 is 0 level in the period TOFF. When the period TON ends and the period TON starts and reaches the turn-on timing, first, the switching current IQ1 has a negative waveform by flowing through the body diode DD. By inverting and flowing from the drain to the source, a waveform with positive polarity is obtained.

  The primary winding current I2 is a current that flows through the primary winding N1 in accordance with the switching operation of the switching element Q1, and in this case, a waveform obtained by almost synthesizing the switching current IQ1 and the capacitor current Icr. Become. When the switching element Q1 is turned on / off, the resonance pulse voltage, which is the switching voltage V1 in the period TOFF, is connected in series to the primary winding N1 and the primary side series resonance capacitor C11 that form the first primary side series resonance circuit. Applied to the circuit. As a result, the primary side series resonance circuit performs a resonance operation, and the primary winding current I2 becomes an alternating waveform corresponding to the switching cycle by the sine wave component. The primary winding voltage V2 is a voltage across the primary winding N1. This primary winding voltage V2 also has an alternating waveform corresponding to the switching cycle by a sine wave as shown in the figure.

At the timing when the switching element Q1 is turned off after the period TON ends and the switching element Q1 is turned off, the primary winding current I2 flows as a capacitor current Icr with a positive polarity with respect to the primary side parallel resonance capacitor Cr, thereby causing the primary side parallel. The operation of charging the resonant capacitor Cr is started, and in response, the switching voltage V1 starts to rise in a sine wave form from the 0 level, and the voltage resonant pulse rises. When the capacitor current Icr is inverted to a negative polarity, the primary side parallel resonant capacitor Cr shifts from charging to discharging, and the voltage resonance pulse decreases in a sinusoidal form from the peak level.
Then, if the voltage resonance pulse waveform as the switching voltage V1 falls to the 0 level, a period TON in which the switching element Q1 (and the body diode DD) is turned on is started. When this period TON is reached, first, the body diode DD is turned on, and the negative primary winding current I2 flows. At this time, the switching voltage V1 is at the 0 level, and when the primary winding current I2 flows through the body diode DD for a fixed period, the drain-source of the switching element Q1 is turned on, and the positive primary winding current I2 is Shed. In this way, during the period TON, the primary winding current I2 flows through the switching element Q1 (and the body diode DD), whereby the waveform of the switching current IQ1 is obtained. Such an operation indicates that the ZVS operation by the primary side parallel resonance circuit and the ZCS operation by the primary side series resonance circuit are obtained when the switching element Q1 is turned on and off.

Further, depending on the secondary side alternating voltage V3, the operation of the secondary side rectifier circuit is indicated.
The secondary side alternating voltage V3 is a voltage across the connection circuit of the secondary winding N2 and secondary winding N2 'and the secondary side partial voltage resonance capacitor C3, and is input to the secondary side rectifier circuit. As the alternating voltage V3, a forward voltage is applied to the rectifier diodes Do1 and Do2 every half period of the secondary alternating voltage V3, and the rectifier diodes Do1 and Do2 are alternately turned on in response thereto. As a result, the secondary side alternating voltage V3 is clamped at an absolute value level corresponding to the secondary side DC output voltage Eo depending on the conduction period of the rectifier diode Do1 and the rectifier diode Do2.
Here, since the insulating converter transformer PIT cannot transmit direct current, if the waveform on the primary side has a distortion deviating from the sine wave, the positive voltage generated in the secondary winding N2 and the secondary winding N2 ′ is positive. The zero level of the voltage is determined so that the time integral value of the negative voltage and the time integral value of the negative voltage are equal, and the number of turns of the secondary winding N2 and the secondary winding N2 'is equal. , The magnitudes of the currents flowing through the rectifier diode Do1 and the rectifier diode Do2 are different.
Therefore, the number of turns of the secondary winding N2 and the secondary winding N2 ′ is set to be different for 35T and 45T, respectively, and thereby the magnitude of the current flowing through the rectifier diode Do1 and the rectifier diode Do2. Are made equal.
Further, a partial voltage resonance capacitor C3 is provided, and the efficiency is improved by passing a current at a voltage switching point through the partial voltage resonance capacitor C3. The value of the partial voltage resonance capacitor C3 was able to obtain a good efficiency improvement effect in the range of about 200 pF to 470 pF.

In the combination of the primary side voltage resonance converter, the secondary side voltage doubler half-wave rectification current resonance circuit and the loosely coupled insulating converter transformer PIT having a coupling coefficient of 0.7 or less shown in FIG. 8, the constant input voltage VAC = AC = The experimental result has been obtained that the variable range Δfs of the switching frequency fs with respect to the load fluctuation of the maximum load power Pomax = 300 W to the minimum load power Pomin = 0 W under the input condition of 100 V is Δfs = 20.8 kHz. Moreover, the AC → DC power conversion efficiency (ηAC → DC) is ηAC → DC = 91.7% when the maximum load power Pomax = 300 W, and ηAC → DC = 93.8% when the load power Po = 75 W. It has been.
Compared with this, the AC → DC power conversion efficiency (ηAC → DC) of the modified class E switching operation multiple resonance converter of the present embodiment shown in FIG. 1 is 91.3% and 100 W when the maximum load power Pomax = 300 W. The time is 91.7%. Further, an experimental result that Δfs = 11.0 kHz is obtained for the variable range Δfs of the switching frequency fs, which is about half that of the circuit shown in FIG.

  As described above, in the circuit of FIG. 1 or the circuit of FIG. 8 in which the class E switching converter is applied to the primary side switching converter, the abnormality at the intermediate load regardless of the presence or absence of the secondary side series resonance circuit. Operation is canceled.

Further, in the power supply circuit shown in FIG. 9 as the conventional example, the current flowing from the smoothing capacitor Ci into the switching converter passes through the primary winding N1 of the insulating converter transformer PIT, the switching element Q1, and the primary side parallel resonant capacitor. Flows into Cr. In this case, the current flowing from the smoothing capacitor Ci into the switching converter is the primary winding current I1, and has a relatively high frequency due to the switching period. That is, a charging / discharging current flows to the smoothing capacitor Ci at a high frequency with respect to the commercial AC power supply cycle.
An aluminum electrolytic capacitor is often employed depending on the fact that a high breakdown voltage is required for the component element as the smoothing capacitor Ci. Compared to other types of capacitors and the like, aluminum electrolytic capacitors have the property that when operated at a high frequency, the capacitance decreases and the loss angle tangent tends to increase. For this reason, it is necessary to select a special product having a low ESR (equivalent series resistance) and a large allowable ripple current as the aluminum electrolytic capacitor used for the smoothing capacitor Ci. It is also necessary to select a correspondingly large value for the capacitance of the element as the smoothing capacitor Ci. For example, in the configuration of the power supply circuit of FIG. 9, when the maximum load power Pomax = 300 W equivalent to the present embodiment is to be supported, about 1000 μF is selected. Such an aluminum electrolytic capacitor is more expensive than a general-purpose aluminum electrolytic capacitor, and is disadvantageous in terms of cost, including an increase in component price in accordance with an increase in capacitance.

  On the other hand, in the power supply circuit of the present embodiment shown in FIG. 1, the current flowing from the smoothing capacitor Ci to the switching converter is switched to the switching element Q1 side through the series connection of the choke coil winding N10 and the primary winding N1. It is supposed to flow through. For this reason, the current flowing from the smoothing capacitor Co into the switching converter becomes a direct current as shown also as the input current I1 in FIG. In this manner, since the current flowing from the smoothing capacitor Ci into the switching converter becomes a direct current, in the present embodiment, the above-described problems of decrease in capacitance and increase in tangent of loss angle do not occur. Along with this, the ripple of the commercial AC power supply period in the DC input voltage Ei is also reduced. For example, the ripple is 7.5 Vp-p in the configuration of the power supply circuit of FIG. 9, whereas it is 5 Vp-p in the power supply circuit of FIG. For this reason, as this embodiment, a general-purpose aluminum electrolytic capacitor can be selected as the smoothing capacitor Ci. Further, since the ripple voltage is small for the capacitance of the element as the smoothing capacitor Ci, a value lower than that in the case of the circuit of FIG. 9 can be selected. As an actual circuit of FIG. 1, 680 μF can be selected. In this way, in the present embodiment, it is possible to reduce the cost of the smoothing capacitor Ci. Further, the waveform of the input current I1 has a sine wave shape, which can also reduce the high frequency noise.

  Further, as described above, the abnormal operation at the time of intermediate load is eliminated, and an appropriate ZVS operation is obtained. As a phenomenon of the abnormal operation, as shown in FIG. 10B, the switching element Q1 is turned on at a timing before the turn-on (start of the period TON), and the positive switching current IQ1 is generated between the source and the drain. However, depending on the operation of the switching current IQ1, the switching loss is increased. In the present embodiment, since the operation of the switching current IQ1 corresponding to the abnormal operation does not occur, the increase of the switching loss due to this is eliminated, and this is one of the factors for improving the power conversion efficiency. .

Further, as can be seen by comparing the switching current IQ1 of FIGS. 2A and 10A, the waveform of the switching current IQ1 of FIG. 2A corresponding to the present embodiment is at the end of the period TON. The waveform has a peak at the previous timing. The waveform of the switching current IQ1 shown in FIG. 2 means that the level of the switching current IQ1 at the time of turn-off is suppressed. If the level of the switching current IQ1 at the time of turn-off is suppressed, the switching loss at the time of turn-off is reduced correspondingly, and the power conversion efficiency is improved.
Such a waveform of the switching current IQ1 is obtained by performing the class E switching operation for the primary side switching converter.

Further, depending on the connection mode on the primary side of the power supply circuit of FIG. 8, the first primary side series resonant circuit in the present embodiment is formed, but the second primary side series resonant circuit is not formed. In this case, the series resonance current that flows to the primary side flows as the primary winding current I2 as it is.
On the other hand, in the present embodiment, as described above, the primary side series resonance circuit includes two sets of the first primary side series resonance circuit and the second primary side series resonance circuit. The series resonance current that should flow through is divided into the primary winding N1 and the choke coil winding N10. This reduces the magnitude of the primary winding current I2 and improves efficiency.
Further, in the present embodiment, the waveform of the input current I1 is a sine wave, but the effect of reducing high-frequency noise can be obtained.

  On the secondary side, efficiency is improved by making the number of turns of the secondary winding N2 and the secondary winding N2 'different and making the current of both positive and negative polarity generated according to the polarity of the magnetic flux uniform. In addition, by providing a secondary side partial voltage resonance circuit, a partial voltage resonance operation is performed at the timing when the secondary side rectifier diode is turned on / off, and a current that is about to flow to the secondary side rectifier diode is It flows to the partial voltage resonance capacitor, and conduction loss and switching loss in the rectifier diode are reduced.

  Further, the present invention is not limited to the configurations shown as the above embodiments. For example, for the main switching element (and auxiliary switching element), it is also conceivable to select an element other than the MOS-FET, such as an IGBT (Insulated Gate Bipolar Transistor) or a bipolar transistor. In each of the above embodiments, a separately excited switching converter is cited, but the present invention can also be applied to a case where it is configured as a self-excited type.

It is a circuit diagram which shows the structural example of the power supply circuit as embodiment of this invention. It is a wave form diagram which shows operation | movement of the principal part in the power supply circuit as embodiment by a switching period. It is a figure which shows the fluctuation | variation characteristic of AC-> DC power conversion efficiency with respect to load fluctuation | variation, and the switching frequency about the power supply circuit as embodiment. It is a figure which shows the fluctuation | variation characteristic of AC-> DC power conversion efficiency with respect to load fluctuation | variation, and the switching frequency about the power supply circuit as embodiment. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a circuit diagram which shows the basic structural example of a class E switching converter. FIG. 7 is a waveform diagram showing an operation of the class E switching converter shown in FIG. 6. It is a circuit diagram which shows the structural example of the switching power supply circuit to which the class E switching converter shown in FIG. 6 is applied. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art example. FIG. 10 is a waveform diagram showing an operation of a main part of the power supply circuit shown in FIG. 9. FIG. 10 is a diagram showing AC-to-DC power conversion efficiency with respect to load fluctuation, switching frequency, and on-period fluctuation characteristics of the switching element for the power supply circuit shown in FIG. 9. It is a figure which shows notionally the constant voltage control characteristic about the power supply circuit shown in FIG.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, PCC choke coil N10 choke coil winding, Cr primary side parallel resonant capacitor, N1 primary winding, N2 , N2 'secondary winding, C11 primary side series resonant capacitor, Do1, Do2 (secondary side) rectifier diode, Co (secondary side) smoothing capacitor,

Claims (2)

A primary-side rectifying and smoothing circuit having a primary-side rectifying element and a primary-side smoothing capacitor that rectifies and smoothes a commercial AC power supply to generate a rectified and smoothed voltage;
A switching element that switches the rectified and smoothed voltage to an alternating voltage;
A converter transformer that inputs the AC voltage to the primary winding and generates an AC voltage in the secondary winding;
A secondary-side rectifying / smoothing circuit having a secondary-side rectifying element and a secondary-side smoothing capacitor that rectifies and smoothes the AC voltage generated in the secondary winding to generate an output DC voltage;
Switching element control means for controlling the switching element based on the output DC voltage;
In a switching power supply circuit comprising:
The rectified and smoothed voltage is supplied to one winding end of the primary winding of the converter transformer via a choke coil connected to the output end of the primary side rectifying and smoothing circuit, and the other primary winding of the converter transformer is supplied. Connect the switching element to the winding end to generate the AC voltage,
By connecting a series resonant capacitor to the connection point between one winding end of the primary winding of the converter transformer and the choke coil, the leakage inductance generated in the primary winding of the converter transformer and the series resonant capacitor Forming a first series resonant circuit governed by a resonant frequency, and a second series resonant circuit governed by a resonant frequency by the inductance of the choke coil and the series resonant capacitor, and The resonance frequency of the series resonance circuit and the resonance frequency of the second series resonance circuit are set substantially equal,
A parallel resonance circuit in which a resonance frequency is governed by a primary side parallel resonance capacitor connected in parallel to the switching element, an inductance of the choke coil, and a leakage inductance generated in the primary winding is formed, and the parallel resonance circuit Is set to a frequency higher than the resonance frequency of the first series resonance circuit and the resonance frequency of the second series resonance circuit,
The secondary winding includes a first secondary winding and a second secondary winding, and the secondary-side rectifying element includes the first secondary winding and the second secondary winding. Currents in directions different in polarity from each of the secondary windings are taken out by the first rectifier diode and the second rectifier diode, and the magnitudes of the currents in directions different in polarity from each other are obtained. A switching power supply circuit characterized in that a winding ratio between the first secondary winding and the second secondary winding is determined to be substantially equal. One winding end of the first secondary winding and one winding end of the second secondary winding are connected so that a voltage is added to the first secondary winding. The first secondary side rectifying element is connected to the other winding end of the second secondary winding, and the second secondary side rectifying element is connected to the other winding end of the second secondary winding, A partial voltage resonance capacitor for causing partial voltage resonance is connected between the other winding end of the first secondary winding and the other winding end of the second secondary winding. The switching power supply circuit according to claim 1.

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