JP2008079375A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply circuit by which a stable output DC voltage can be obtained without employing a magnetic amplifier control method requiring many count of components and which makes less noises. <P>SOLUTION: In a push-pull system converter wherein DC voltage is supplied to the center tap of a converter transformer PIT via a choke coil Lo1, the primary winding and the secondary winding of the converter transformer are made to be not larger than 0.8 in a coupling factor to cause a leakage inductor, forming a resonance circuit on the primary side and the secondary side respectively. A time ratio (TON/TOFF) between switching elements Q1 and Q2 is set to be 1 to roughly 2 and the resonance frequency on the secondary side is set to be roughly not larger than 2/3 of the resonance frequency of the primary side voltage resonance circuit to carry out ZVS operation. In this case, the choke coil Lo1 and the converter transformer PIT are formed as a decrypting component sharing a core material to be reduced in size of the device. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

  In recent years, most power supply circuits that rectify a commercial power supply to obtain a desired DC voltage are switching power supply circuits. A switching power supply circuit is used as a power source for various electronic devices as a high-power DC-DC converter while miniaturizing a transformer and other devices by increasing a switching frequency.

  As a DC-DC converter system, various systems such as a system with one switching element, a half-bridge system with two switching elements, a full-bridge system with four switching elements, etc. are often used. Yes. Furthermore, a push-pull type DC-DC converter (hereinafter abbreviated as a push-pull converter) having two switching elements is also used (for example, see Non-Patent Document 1).

  A circuit example of the push-pull converter is shown in FIG. In FIG. 23, only the power part of the push-pull converter is described, and the control circuit that controls the gates of the switching element Q1 and the switching element Q2, which are MOS-FETs, constituting the push-pull converter is not described. The switching element Q1 and the switching element Q2 are complementarily controlled by a control circuit (not shown) (the switching element Q1 and the switching element Q2 are turned on (conducted) and off (disconnected). In other words, the switching element Q1 includes a body diode DD1, and the switching element Q2 includes a body diode DD2.

  The converter transformer PIT is formed by winding a primary winding N1, a primary winding N1 ′, a secondary winding N2 and a secondary winding N2 ′ around a ferrite core (magnetic core). The coupling coefficient between the line N1 and the primary winding N1 ′, and the secondary winding N2 and the secondary winding N2 ′ is close to 1. The primary winding N1 and the primary winding N1 ′ have the same number of turns, and the connection point between the primary winding N1 and the primary winding N1 ′ is the winding start of one winding and the winding of the other winding. The center tap is connected to the end. Here, the inductor L1 is a leakage inductor generated in the primary winding N1, the inductor L1 ′ is a leakage inductor generated in the primary winding N1 ′, the inductor L2 is a leakage inductor generated in the secondary winding N2, and the inductor L2 ′ is the primary. When the coupling coefficient is close to 1, each of the inductor L1, the inductor L1 ′, and the inductor L2 is an operation of the circuit. It is a minute thing that does not have a big influence on

  A choke coil Lo1 is connected to the center tap, and a DC voltage Ei is supplied via the choke coil Lo1, and the primary winding N1 and the primary winding N1 ′, the secondary winding N2 and the secondary winding are supplied. Due to the imbalance of the coupling coefficient with the line N2 ′, the occurrence of bias is suppressed. Further, a primary side voltage resonance capacitor C1 is connected to both ends of the primary winding N1 and the primary winding N1 '.

  For the secondary side, the high speed switching diode Do1 is connected to the secondary winding N2, the high speed switching diode Do2 is connected to the secondary winding N2 ′, and the choke coil Lo20 and the secondary are connected to the output side of the rectifier circuit. The output DC voltage Eo is obtained by arranging a side smoothing capacitor Co to constitute a double-wave rectifier circuit having a choke input smoothing circuit. The conduction angle of the high-speed switching diode Do1 and the high-speed switching diode Do2 is expanded by having a smoothing circuit for choke input.

  Further, the gates of the switching elements Q1 and Q2 are controlled by a control circuit (not shown), but the time ratio (TON / TOFF) of each of the switching elements Q1 and Q2 is 1, that is, the time TON. Length = time TOFF length = time TS length / 2. Here, the time TON is the time when the switching element Q1 or the switching element Q2 is turned on, the time TOFF is the time when the switching element Q1 or the switching element Q2 is disconnected, and the time TS is the time of one cycle of the switching period. is there. In this way, the waveform of the voltage V4 generated between the secondary winding N2 and the secondary winding N2 'is a sine wave.

  FIG. 24 shows the waveform of each part of the circuit shown in FIG. In order from the top of FIG. 24, voltage V3 (see FIG. 23), voltage V1 (see FIG. 23), current IQ1 (see FIG. 23), voltage V2 (see FIG. 23), and current IQ2 (see FIG. 23). And voltage V4 (see FIG. 23). As shown by the current IQ1 and the current IQ2, the body diode DD1 and the body diode DD2 do not conduct, and a ZVS (Zero Voltage Switching) operation is performed as shown by the voltage V1 and the voltage V2.

  In this operation mode, when the frequency fs, which is the switching frequency between the switching element Q1 and the switching element Q2, is increased, the switching element Q1 and the switching element Q2 become conductive during the time TOFF, and the switching loss increases. When the switching loss exceeds a predetermined value, the so-called thermal runaway state may occur and the switching element Q1 and the switching element Q2 may be destroyed.

  Therefore, when controlling the output DC voltage Eo, as shown in FIG. 25, the operational amplifier OP, the saturable magnetic core SR-1 and the saturable magnetic core SR-2 on the secondary side, the diode Dr3 functioning as a reset diode, the diode A magnetic amplifier composed of Dr4 and a diode Dr5 is added, and the conduction angle of the high-speed switching diode Do1 and the high-speed switching diode Do2 is controlled by the action of this magnetic amplifier, and the output DC voltage Eo is controlled with the frequency fs fixed. A magamp control system has been proposed.

  In this case, the voltage appearing at both ends of the saturable magnetic core SR-1 and the saturable magnetic core SR-2 becomes a waveform including harmonics and the switching noise also increases. In addition, saturable magnetic core SR-1 and saturable magnetic core SR-2 having a square hysteresis characteristic, diode Dr3, diode Dr4, and diode Dr5 are required as additional components.

  Therefore, in order to eliminate the need for the saturable magnetic core SR-2, the diode Dr3, the diode Dr4, and the diode Dr5, the primary winding N1 and the primary winding N1 ′ and the secondary winding N2 are not known in the art. It can be considered that the coupling coefficient with the secondary winding N2 ′ is loosely coupled, and the values of the inductor L1, the inductor L1 ′, and the inductor L2 are increased, and the voltage is controlled using this value.

  For example, FIG. 26 shows an example of the push-pull converter shown in FIG. 1 in which the values of the inductor L1, the inductor L1 ′, and the inductor L2 are increased to decrease the value of the primary side voltage resonant capacitor C1, and the time ratio (TON FIG. 6 is a waveform diagram of voltage V1, current IQ1, and voltage V2 when the value of / TOFF is 2. When the value of the time ratio is 2, the value of the output DC voltage Eo that is the output voltage when the frequency fs that is the switching frequency (the reciprocal of the time TS that is the switching period) is changed is shown in FIG. As shown in The graph indicated by Po = 0 is a graph when the power supplied to a load (not shown) is 0 W (watts), and the graph indicated by Pomax is a graph when the power supplied to the load is maximum.

  As can be seen from FIG. 27, the value of the output DC voltage Eo is the lowest at the frequency fo1, the case where the value from the frequency fo1 to the frequency fs is high, and the case where the value from the frequency fo1 to the frequency fs is low. The value of the output DC voltage Eo increases. That is, this graph shows a primary side parallel resonance curve formed by the choke coil Lo1, the inductor L1, the primary side voltage resonance capacitor C1, the choke coil Lo1, the inductor L1 ′, and the primary side voltage resonance capacitor C1. It is. From FIG. 27, the inventors of the present application have found that the output DC voltage Eo can be controlled by controlling the frequency fs. This technique is neither a known technique nor a public technique.

  The variable frequency range for setting the value of the output DC voltage Eo to a constant value from when the power supplied to the load is 0 W to when the power supplied to the load is maximum is given by the variable frequency range Δfs. The variable range of the switching frequency is, for example, 100 kHz or more and is large. For this reason, the maximum load power within the range in which the ZVS operation is possible is limited to 100 W or less.

  In recent years, there has been an active movement to use such a DC-DC converter in a device having a low-level signal system such as an audio device. When the DC-DC converter is used in an audio device, ZVS operation or ZCS (Zero) It is stated that it is preferable to perform a current switching operation (see, for example, Non-Patent Document 2).

However, when the load power is 200 W or more, there are very few examples of practical use of a soft switching power supply by a push-pull converter based on voltage resonance that operates with a commercial AC power supply, and there are few academic papers and there are few known techniques in this field.
Ninomiya, et al. “Operational Characteristics of Voltage Resonant Push-Pull Converters Using Magnetic Control System” Transactions of the Institute of Electronics, Information and Communication Engineers B Vol. J70-B No. 11 pp 1307-1315, November 1987 Shin Nakagawa, et al. “Switching power supply with income characteristics suitable for audio” IEICE Technical Report EE2006-9 May 2006

  In the above-described push-pull converter, the configuration of means for performing constant voltage control with respect to a wide range of load power fluctuations is complicated, and the number of components greatly increases. In order to reduce noise, it is desirable to use a secondary voltage waveform as a sine wave for use in an audio device. However, in order to use a secondary voltage waveform as a sine wave, the switching on the primary side is preferable. It is necessary to set the time ratio (TON / TOFF) of the element to 1. When a magnetic amplifier is provided in order to keep the voltage supplied from the secondary side to the load constant over a wide range of load fluctuations while keeping the duty ratio (TON / TOFF) of the switching element at 1. There is a problem that the waveform of the high frequency voltage on the secondary side is not a sine wave and noise increases.

  Further, without setting the time ratio (TON / TOFF) of the switching element on the primary side to 1, for example, the switching frequency is controlled by setting the time ratio to 2, and the voltage supplied from the secondary side to the load is kept constant. In this case, the time TOFF is increased at the time of light load, and it becomes difficult to perform the ZVS operation. In such a case, the maximum load value is limited to, for example, 100 W or less, and the variable range of the switching frequency is, for example, 100 kHz or more as described above.

  In addition, the push-pull converter shown in the background art described above has a problem that a choke coil is required on the secondary side because the choke input method is adopted, which increases the size of the power supply device and increases the cost of the power supply device. is doing.

  The present invention provides a switching power supply circuit that solves the above-described problems and obtains a stable output DC voltage without adopting a mag-amplifier control method with a large number of parts, generates less noise, and reduces the size of the device. The purpose is to provide.

The switching power supply circuit of the present invention is a switching power supply circuit that converts primary side DC power into AC power and further converts it into secondary side DC power, and the primary side DC power is supplied to one terminal. A primary winding formed by separating a first primary winding and a second primary winding by a choke coil and a center tap connected to the other terminal of the choke coil; And a converter transformer having a secondary winding magnetically loosely coupled to each of the primary winding and the second primary winding, and connected to an end of the first primary winding First switching element to be operated and a second switching element connected to an end of the second primary winding, and oscillation for driving on and off the first switching element and the second switching element .Drive circuit and the cho A primary-side voltage resonant capacitor that forms a primary-side voltage resonant circuit together with an inductor generated in each of the coil, the first primary winding, and the second primary winding; and an inductor generated in the secondary winding And a secondary side voltage resonance capacitor connected in parallel to the secondary winding so as to form a secondary side voltage resonance circuit,
A secondary side rectifying element whose input side is connected to the secondary side resonance circuit, and a secondary side smoothing capacitor connected to the output side of the secondary side rectifying element to obtain a smoothed output DC voltage; And a control circuit for supplying a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit, the first switching element and the second The time ratio, which is the ratio between the on and off states of the switching elements, is set to 1 or more, and the converter transformer includes a first core material having a first gap and forming a first magnetic path. Each of the first primary winding, the second primary winding, and the secondary winding is wound so that a magnetic flux passing through the first magnetic path is linked, and the choke The coil includes the first core material as a part of a magnetic path component. The choke coil winding includes a second core material that forms a second magnetic path having a second gap so that a magnetic flux passing through the first magnetic path does not pass. The magnetic flux passing through the second magnetic path is wound so as to interlink, and the converter transformer and the choke coil are integrally formed as a composite part.

  In this switching power supply circuit, the primary side DC power is supplied to the center tap of the primary winding of the converter transformer via the choke coil. The converter and the lance are formed by loosely coupling the primary winding and the secondary winding, thereby generating a leakage inductance in each of the primary winding and the secondary winding. Each of the first switching element and the second switching element is connected to each end of the primary winding, whereby AC power is supplied to the primary winding. The oscillation / drive circuit turns on / off each of the first switching element and the second switching element based on a signal from the control circuit. The primary side voltage resonance capacitor, the primary choke coil, and the inductor generated in the first primary winding and the second primary winding form a primary side voltage resonance circuit. Further, a secondary side voltage resonance circuit is formed by an inductor generated in the secondary winding and a secondary side voltage resonance capacitor connected in parallel to the secondary winding. The secondary side rectifying element is connected to the input side of the secondary side rectifying element in the secondary side resonance circuit, and the secondary side smoothing capacitor is connected to the output side of the secondary side rectifying element so as to obtain a smoothed output DC voltage. The control circuit supplies a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit. The ZVS operation is obtained by setting the time ratio, which is the ratio of ON and OFF of the first switching element and the second switching element, to 1 or more, respectively. The converter transformer includes a first core material that has a first gap and forms a first magnetic path, and includes a first primary winding, a second primary winding, and a secondary winding. Each of the wires is wound so that the magnetic flux passing through the first magnetic path is linked, and the choke coil includes the first core material as a part of the component of the magnetic path, and the first magnetic path A choke coil winding passes through the second magnetic path, the second core material forming a second magnetic path having a second gap so that the magnetic flux passing through the second magnetic path does not pass. The converter transformer and the choke coil are integrally formed as a composite part that is not affected magnetically by winding the magnetic flux so as to interlink, and the apparatus can be miniaturized.

Another switching power supply circuit of the present invention is a switching power supply circuit that converts primary side DC power into AC power and further converts it into secondary side DC power, and the primary side DC power is supplied to one terminal. The first primary winding and the first primary winding by the center tap to which the primary DC power is supplied by the choke coil that is connected to the other terminal of the choke coil. A primary winding formed by separating a second primary winding that is loosely coupled, and each of the first primary winding and the second primary winding are magnetically loose. A converter transformer having a secondary winding to be coupled; a first switching element connected to an end of the first primary winding; and an end of the second primary winding A second switching element to be operated;
Oscillation / drive circuit for driving on / off of the first switching element and the second switching element, and the choke coil, the first primary winding, and the second primary winding, respectively. A primary side voltage resonance capacitor that forms a primary side voltage resonance circuit together with an inductor, and a parallel connection to the secondary winding so as to form a secondary side voltage resonance circuit together with the inductor generated in the secondary winding; A secondary side voltage resonance capacitor, a secondary side rectifying element whose input side is connected to the secondary side resonance circuit, and a smoothed output DC voltage connected to the output side of the secondary side rectifying element A secondary side smoothing capacitor, and a control circuit for supplying a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit, Duty ratio 1 is the ratio of the on and off switching element and the second switching elements are respectively set to one or more.

  In this switching power supply circuit, the primary primary DC power is separated from the first primary winding and the second primary winding, which is loosely coupled to the first primary winding, via the choke coil. It is supplied to the center tap of the primary winding of the converter transformer to be formed. As a result, leakage inductance is generated in each of the first primary winding and the second primary winding. In addition, the primary winding and the secondary winding of the converter transformer are formed as magnetically loose coupling, which further causes a leakage inductance in each of the primary winding and the secondary winding. Each of the first switching element and the second switching element is connected to each end of the primary winding, whereby AC power is supplied to the primary winding. The oscillation / drive circuit turns on / off each of the first switching element and the second switching element based on a signal from the control circuit. The primary side voltage resonance capacitor, the primary choke coil, and the inductor generated in the first primary winding and the second primary winding form a primary side voltage resonance circuit. Further, a secondary side voltage resonance circuit is formed by an inductor generated in the secondary winding and a secondary side voltage resonance capacitor connected in parallel to the secondary winding. The secondary side rectifying element is connected to the input side of the secondary side rectifying element in the secondary side resonance circuit, and the secondary side smoothing capacitor is connected to the output side of the secondary side rectifying element so as to obtain a smoothed output DC voltage. The control circuit supplies a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit. The ZVS operation is obtained by setting the time ratio, which is the ratio of ON and OFF of the first switching element and the second switching element, to 1 or more, respectively.

  Still another switching power supply circuit according to the present invention is a switching power supply circuit that converts primary side DC power into AC power and further converts it into secondary side DC power, and is a center to which the primary side DC power is supplied. A primary winding formed by separating a first primary winding and a second primary winding loosely coupled to the first primary winding by a tap; and the first 1 A converter transformer having a secondary winding and a secondary winding that is magnetically loosely coupled to each of the second winding and the second primary winding, and is connected to an end of the first primary winding A first switching element and a second switching element connected to an end of the second primary winding, and an oscillation drive that drives the first switching element and the second switching element on and off A circuit, the first primary winding and the second primary A primary-side voltage resonant capacitor that forms a primary-side voltage resonant circuit with an inductor that occurs in each of the lines; and a secondary-side voltage resonant circuit that forms a secondary-side voltage resonant circuit with an inductor that occurs in the secondary winding. A secondary-side voltage resonant capacitor connected in parallel, a secondary-side rectifying element whose input side is connected to the secondary-side resonant circuit, and an output direct current connected to the output side of the secondary-side rectifying element and smoothed A secondary smoothing capacitor configured to obtain a voltage, and a control circuit configured to supply a control signal whose frequency is variable so that the value of the output DC voltage is set to a predetermined value to the oscillation / drive circuit. And the time ratio, which is the ratio of ON to OFF of the first switching element and the second switching element, is set to 1 or more.

  In this switching power supply circuit, a converter transformer formed by separating a primary DC power from a first primary winding and a second primary winding that is loosely coupled to the first primary winding. To the center tap of the primary winding. As a result, leakage inductance is generated in each of the first primary winding and the second primary winding, which is equivalent to the case where the primary DC power is supplied to the center tap via the inductor. In addition, the primary winding and the secondary winding of the converter transformer are formed as magnetically loose coupling, which further causes a leakage inductance in each of the primary winding and the secondary winding. Each of the first switching element and the second switching element is connected to each end of the primary winding, whereby AC power is supplied to the primary winding. The oscillation / drive circuit turns on / off each of the first switching element and the second switching element based on a signal from the control circuit. The primary side voltage resonance capacitor, the primary choke coil, and the inductor generated in the first primary winding and the second primary winding form a primary side voltage resonance circuit. Further, a secondary side voltage resonance circuit is formed by an inductor generated in the secondary winding and a secondary side voltage resonance capacitor connected in parallel to the secondary winding. The secondary side rectifying element is connected to the input side of the secondary side rectifying element in the secondary side resonance circuit, and the secondary side smoothing capacitor is connected to the output side of the secondary side rectifying element so as to obtain a smoothed output DC voltage. The control circuit supplies a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit. The ZVS operation is obtained by setting the time ratio, which is the ratio of ON and OFF of the first switching element and the second switching element, to 1 or more, respectively.

  According to the present invention, it is possible to provide a switching power supply circuit that can obtain a stable output DC voltage without adopting a mag-amplifier control method with a large number of parts, and that can reduce noise and reduce the size of the device.

  The best mode for carrying out the present invention (hereinafter referred to as an embodiment) will be described. The switching power supply circuit of the embodiment is a push-pull method as a switching method. That is, the primary winding of the converter transformer includes a center tap, and the primary winding is formed by separating the first primary winding and the second primary winding by the center tap. Then, DC power is supplied through a choke coil connected to the center tap. A switching element is connected to each end of the primary winding. Then, the primary winding and the secondary winding of the converter transformer are loosely coupled to generate a leakage inductor, and have a resonance circuit on each of the primary side and the secondary side. A constant output DC voltage is generated on the secondary side while the ZVS operation is performed by the action of the resonance circuit. Here, the primary side resonance circuit is a voltage resonance circuit, and the time ratio (TON / TOFF) is 1 or more. The primary-side resonance circuit is mainly formed by a choke coil connected to the center tap of the primary winding, an inductor generated in the primary winding, and a primary voltage resonance capacitor. The secondary resonance circuit is mainly formed by an inductor generated in the secondary winding and a secondary resonance capacitor. When the secondary side resonance circuit is a voltage resonance circuit and the secondary side resonance frequency is set to about 2/3 or less of the resonance frequency of the primary side voltage resonance circuit, ZVS operation can be obtained in a wide load power range. . In addition, a stable output DC voltage can be obtained, and noise generation can be reduced.

  Here, the converter transformer and the choke coil may be combined as an integrated composite part to reduce the size of the switching power supply circuit, and the first primary winding and the second 1 By making the secondary winding loosely coupled, thereby making the inductance value of the leakage inductor generated in each of the first primary winding and the second primary winding larger, the choke coil can be further downsized, The apparatus is reduced in size, and the first primary winding and the second primary winding are further loosely coupled, whereby the first primary winding and the second primary winding. By further increasing the inductance value of the leakage inductor generated in each of the wires, the choke coil can be omitted and the apparatus can be downsized.

  The switching power supply circuit according to the first embodiment will be described with reference to FIGS.

  The switching power supply circuit according to the first embodiment includes a choke coil Lo1 having a choke coil winding No to which primary side DC power is supplied to one terminal, and a center tap connected to the other terminal of the choke coil Lo1. A primary winding formed by separating the primary winding N1 and the primary winding N1 ′, and each of the primary winding N1 and the primary winding N1 ′ are magnetically loosely coupled, for example, a coupling coefficient is A converter transformer PIT having a secondary winding N2 of 0.8 or less, and a switching element Q1 connected to the end of the primary winding N1 and an end of the primary winding N1 ′ Along with the switching element Q2, the oscillation / drive circuit 2 for driving the switching element Q1 and the switching element Q2 on and off, and the inductor generated in the choke coil Lo1 and the primary winding N1 The second primary side voltage together with the primary voltage resonant capacitor C1, the choke coil Lo1, and the inductor generated in the primary winding N1 ′ connected in parallel to the switching element Q1 so as to form one primary side voltage resonant circuit. The secondary winding N2 (or the secondary winding N2 (or so as to form a secondary resonance circuit together with the primary voltage resonance capacitor C2 connected in parallel to the switching element Q2 so as to form a resonance circuit and the inductor generated in the secondary winding N2). A secondary side voltage resonance capacitor C3 connected in parallel to the secondary winding N2 and the secondary winding N2 ′) and a secondary side rectifier Do (or a high speed) whose input side is connected to the secondary side resonance circuit. Switching diodes Do1 to high-speed switching diodes Do4) and a smoothed output DC voltage Eo connected to the output side of the secondary side rectifying element Do, etc. Control signal whose frequency is variable so that the value of the output DC voltage Eo is a predetermined value, and the secondary side smoothing capacitor Co (or the secondary side smoothing capacitor Co1 and the secondary side smoothing capacitor Co2). And a control circuit 1 for supplying the oscillation / drive circuit 2 to the oscillation / drive circuit 2, each of which is set to have a time ratio of 1 or more, which is a ratio of ON and OFF of the switching element Q1 and the switching element Q2.

  Here, in the present embodiment, the primary side voltage resonant capacitor C1 is an embodiment of the first primary side voltage resonant capacitor, and the primary side voltage resonant capacitor C2 is the second primary side voltage resonant capacitor. It is one embodiment. Both forms a primary-side voltage resonant capacitor, which is an embodiment of the primary-side voltage resonant capacitor. Further, as another embodiment of the primary side voltage resonance capacitor, the primary side voltage resonance capacitor C1 and the primary side voltage resonance capacitor C2 are replaced with both ends of the primary winding N1 and the primary winding N1 ′. Only the primary side voltage resonance capacitor C1 may be provided.

  Further, the converter transformer PIT and the inductor Lo1 are configured as a converter transformer / inductor PIT / PCC which is an integral part by combining both. That is, the converter transformer PIT includes an E-type core CR1 and an E-type core CR2 that are first core materials having a gap G1 (see FIG. 2) as a first gap and forming a first magnetic path. Each of the first primary winding N1, the second primary winding N1 ′, and the secondary winding N2 has a first magnetic path (the center magnetic leg of the E-type core CR1 shown in FIG. 2). The magnetic flux passing through the central magnetic leg of the E-type core CR2 and the gap G2) is wound so as to be linked. The choke coil Lo1 includes a first core material as a part of a magnetic path component, and has a second gap G2 so that a magnetic flux passing through the first magnetic path does not pass. An E-type core CR3 which is a second core material forming the second magnetic path is provided, and the magnetic flux generated by the choke coil winding No is wound so as to pass through the second magnetic path.

  The sequence from the AC power supply AC side to the secondary side of the switching power supply circuit shown in FIG.

  The switching power supply circuit has a common mode filter having an across line capacitor CL1, a common mode choke coil CMC, and an across line capacitor CL2. For example, a plug (not shown) is connected to the across-line capacitor CL1 on the input side of the common mode filter via a power switch and a fuse, and the AC power is obtained by inserting the plug into an outlet connected to the AC power supply AC. Is supplied to the switching power supply circuit. Here, the value of the input AC voltage VAC, which is the voltage of the AC power supply AC, is in the range of 85V to 144V (effective value).

  The output side of the common mode filter (the side of the across-line capacitor CL2) is the connection point between the anode of the diode Di1 and the cathode of the diode Di2 that are the input side of the primary side rectifier element Di, Connected to the connection point.

  The positive side terminal of the primary side smoothing capacitor Ci is connected to the connection point between the cathode of the diode Di1 which is the output side of the primary side rectifying element Di and the cathode of the diode Di3, and the anode of the diode Di2 and the anode of the diode Di4 Is connected to the negative terminal of the primary smoothing capacitor Ci. In this way, the primary side rectifying and smoothing circuit is formed, and a DC voltage is supplied to the push-pull converter from both ends of the primary side smoothing capacitor.

  The push-pull converter includes a switching element Q1 and a switching element Q2, a choke coil Lo1, a primary side voltage resonance capacitor C1, a primary side voltage resonance capacitor C2, a control circuit 1, an oscillation / drive circuit 2, a converter transformer PIT, and a secondary side. A voltage resonance capacitor C3, a secondary side rectifying element Do, a secondary side smoothing capacitor Co, and a secondary side inductor Lo2 are provided.

  Switching element Q1 and switching element Q2 are MOS-FETs, and body diode DD1 and body diode DD2 are diodes formed in the process of manufacturing a MOS-FET. Here, the specification of the MOS-FET is 5A / 900V.

  The converter transformer PIT has functions of insulating the primary side and the secondary side and performing voltage conversion, but also functions as an inductor L1, an inductor L1 ', and an inductor L2. Here, each inductor is a leakage inductor formed by loosely coupling a primary winding composed of a primary winding N1 and a primary winding N1 ′ of the converter transformer PIT and a secondary winding N2. It is. How to form such a leakage inductor will be specifically described with reference to a cross-sectional view of the converter transformer / inductor PIT / PCC shown in FIG.

  In the converter transformer / inductor PIT / PCC shown in FIG. 2, the converter transformer PIT and the inductor Lo1 are integrally configured as a composite part. The converter transformer PIT includes an EE type core in which an E type core CR1 and an E type core CR2 made of a ferrite material as a first core material are combined so that their magnetic legs face each other. The primary side and secondary side winding portions are divided so as to be independent from each other, and provided with a bobbin B formed of, for example, resin. Then, the bobbin B around which the primary winding N1 and the primary winding N1 ′ as the primary winding portion and the secondary winding N2 as the secondary winding portion are wound is attached to the EE core. The primary winding N1 and the primary winding N1 ′ are wound in one region, and the secondary winding N2 is separated into a winding region different from this one region, and the central magnetic leg of the EE core It will be in the state wound by. In this way, the overall structure of the converter transformer PIT is obtained.

  Here, as shown in FIG. 2, the primary winding N1 ′ is wound outside the primary winding N1, and the primary winding N1 and the primary winding N1 ′ are tightly coupled. . The primary winding N1 and the secondary winding N2 are loosely coupled due to the existence of the gap G1 as the first gap. Similarly, the primary winding N1 ′ and the secondary winding N2 are gap G1. Because of the existence of loose coupling. The primary winding N1 and the primary winding N1 ′ have the same number of turns, but the length of the primary winding N1 ′ is longer than the length of the primary winding N1, so that the secondary side The coupling coefficient for the winding N2 is larger in the primary winding N1 ′ than in the primary winding N1.

  A gap G1 of 1.6 mm (millimeter) is formed with respect to the central magnetic leg of the EE type core. As a result, the value of the coupling coefficient k is made smaller than 1 as the value of the coupling coefficient k between the primary side and the secondary side. That is, by using loose coupling, a part of the magnetic flux generated by the primary winding N1 and the primary winding N1 ′ and passing through the central magnetic leg, which is the first magnetic path, is coupled to the secondary winding N2. A part of the magnetic flux generated in the secondary winding N2 is not interlinked with the primary winding N1 and the primary winding N1 ′, and the effect of the non-interlinking magnetic flux causes the inductor L1, the inductor L1 ′, The inductor L2 is formed. As shown in FIG. 2, the gap G1 is formed by making the central magnetic legs of the E-type core CR1 and the E-type core CR2 shorter than the two outer magnetic legs. The core material was EER-35 (core material name).

  In the first embodiment, the number of turns of the primary winding N1 and the number of turns of the primary winding N1 ′ are 40T (turns), respectively, and the number of turns of the secondary winding N2 is 35T. The value of the coupling coefficient k between the line N1 and the secondary winding N2 is 0.715, and the value of the coupling coefficient k between the primary winding N1 ′ and the secondary winding N2 is 0.735. The value of the output DC voltage Eo at this time is 175 V (volt).

  As shown in FIG. 2, the choke coil Lo1 includes an E-type core CR3 joined to the side surface of the converter transformer PIT, and is included as a part of the E-type core CR3 as a second core material and a magnetic path component. The E-type core CR1 and the E-type core CR2 as one core material form a magnetic path of the choke coil Lo1. As is apparent from FIG. 2, the gap G2 as the second gap is provided so that the magnetic flux passing through the first magnetic path (the magnetic flux generated by the converter transformer PIT) does not pass through the second magnetic path. is doing. That is, the magnetic flux generated by the converter transformer PIT cannot pass through the second magnetic path (the magnetic path through which the magnetic flux generated by the choke coil Lo1 passes) having a high magnetic resistance value due to the gap G2. Similarly, the magnetic flux passing through the second magnetic path cannot pass through the first magnetic path due to the action of the gap G1. Thus, since the magnetic flux passing through the first magnetic path and the magnetic flux passing through the second magnetic path do not pass through each other magnetic path, the converter transformer PIT and the choke coil Lo1 are here. Is a composite part but functions independently.

  The choke coil winding No is wound around the central magnetic leg of the E-type core CR3 so that the magnetic flux passing through the second magnetic path is linked. The gap G2 was 0.8 mm which is half of the gap G1, the number of turns of the choke coil winding No was 43T, and the inductance value of the choke coil Lo1 was 400 μH (micro henry). The value of 400 μH is larger than the inductors L1 and L1 ′. The choke coil winding No, the primary winding N1, the primary winding N1 ', and the secondary winding N2 are litz wires.

  The connection mode on the primary side will be described. The choke coil Lo1 is connected between the primary side smoothing capacitor Ci, the primary winding N1 of the converter transformer PIT, and the center tap of the primary winding N1 ′, and the end of the primary winding N1 of the converter transformer PIT. Is connected to the drain of the switching element Q1, and the end of the primary winding N1 ′ of the converter transformer PIT is connected to the drain of the switching element Q2. The sources of the switching element Q1 and the switching element Q2 are grounded to the primary side ground. The primary side voltage resonance capacitor C1 is connected in parallel between the drain and source of the switching element Q1, and the primary side voltage resonance capacitor C2 is connected in parallel between the drain and source of the switching element Q2.

  The output of the oscillation / drive circuit 2 is connected to the gate of the switching element Q1 and the gate of the switching element Q1. The oscillation / drive circuit 2 drives the gate of the switching element Q1 and the gate of the switching element Q2 according to the output from the control circuit 1.

  The connection mode on the secondary side will be described. A secondary side rectifying element Do is connected to the secondary winding N2 of the converter transformer PIT. The secondary side rectifying element Do is configured by bridge-connecting high speed switching diodes Do1 to Do4, and a secondary side smoothing capacitor Co is connected to the output side of the secondary side rectifying element Do. An output DC voltage Eo that is a voltage supplied to the load from both ends of the secondary side smoothing capacitor Co is obtained. Here, the value of the output DC voltage Eo was set to 175 V as described above.

  The control circuit 1 has its input side connected to the secondary side smoothing capacitor Co, and oscillates a detection output (error voltage) corresponding to the difference between the input output DC voltage Eo and a predetermined reference voltage value. Supply to the drive circuit 2

  In such a push-pull converter, a first primary voltage resonance circuit is mainly formed by the choke coil Lo1, the inductor L1, and the primary voltage resonance capacitor C1. Here, mainly means the resonance frequency of the first primary side voltage resonance circuit by the characteristics of these components, that is, the inductance of the choke coil Lo1, the inductance of the inductor L1, and the capacitance of the primary side voltage resonance capacitor C1. It is said that is generally determined. For example, the capacitance of the primary side smoothing capacitor Ci influences the resonance frequency, but the influence is negligible. Therefore, it is not used as an element for forming the first primary side voltage resonance circuit.

  Further, a second primary voltage resonance circuit is mainly formed by the choke coil Lo1 and the inductor L1 'and the primary voltage resonance capacitor C2. That is, since the resonance frequency of the second primary side voltage resonance circuit is substantially determined by the inductance of the choke coil Lo1, the inductance of the inductor L1 ', and the capacitance of the primary side voltage resonance capacitor C2, the primary side smoothing capacitor Ci, etc. Is not an element forming the second primary side voltage resonance circuit. Each is set to the frequency fo1 so that the resonance frequency of the first primary side voltage resonance circuit and the resonance frequency of the second primary side voltage resonance circuit are substantially equal. Here, the resonance frequency of the first primary side voltage resonance circuit and the second primary side voltage resonance circuit do not work independently, depending on which of the switching element Q1 and the switching element Q2 is turned off. Thus, both operate in cooperation to form a primary side voltage resonance circuit.

  Also, a secondary side voltage resonance circuit is formed mainly by the inductance of the inductor L2 and the capacitance of the secondary side voltage resonance capacitor C3, and the resonance frequency is set to the frequency fo2. Here, mainly as in the case described above, for example, the secondary side inductor Lo2 and the like also affect the resonance frequency, but the effect is negligible, so that as an element forming the secondary side voltage resonance circuit No. The relationship between the frequency fo1 and the frequency fo2 is set to approximately 2/3 or less of the frequency fo1 as shown in FIG. Here, the value of the secondary inductor Lo2 is several μH or less, for example, 3.3 μH, which is very small compared to the inductance value of the inductor L2. Therefore, although the secondary side inductor Lo2 becomes an element constituting the secondary side voltage resonance circuit, it hardly affects the resonance frequency of the secondary side voltage resonance circuit.

  The operation performed by the secondary inductor Lo2 will be described. When the secondary inductor Lo2 is not provided, that is, when a conductive wire is used instead of the secondary inductor Lo2, the current I4 (see FIG. 1) and the current I3 (see FIG. 1) are set to, for example, 1 Ringing of .5 MHz (mega hertz) occurs. Since this ringing has a high frequency, it is harmful as a noise component. For example, this noise leaks to the AC power supply AC side, which adversely affects the operation of other devices. In addition, when only the converter unit that does not include the primary side rectifier circuit is used as a DC-DC converter, this ringing has an adverse effect on the input DC line side.

  Such ringing is caused by an equivalent inductance (ESL) existing in the secondary side smoothing capacitor Co. In the absence of the secondary side inductor Lo2, a resonance circuit having a resonance frequency near 1.5 MHz formed by the capacitance of the secondary side voltage resonance capacitor C3 and this equivalent inductance is formed. This is because the resonance circuit is excited by operation. Further, ringing of the same frequency is also generated in the current I4 (see FIG. 1) that is a current flowing on the input side of the secondary side rectifying element Do. On the other hand, when the secondary inductor Lo2 is present, the ringing generated in the currents I3 and I4 described above disappears, as shown in FIGS.

  Here, the capacitance value of the primary side voltage resonance capacitor C1 and the capacitance value of the primary side voltage resonance capacitor C2 are 4700 pF (Pico Farad), and the capacitance value of the secondary side voltage resonance capacitor C3 is 0.047 μF. (Micro Farad). At this time, the value of the frequency fo1 was 90.1 kHz (kilohertz), and the value of the frequency fo2 was 53.5 kHz (kilohertz).

  Next, the operation of the switching power supply circuit having the above-described configuration will be described. The basic operation principle of such a push-pull converter operates as a multi-resonance converter, which is a known technique, and has the following characteristics.

  The duty ratio between the switching element Q1 and the switching element Q2 is set to 1 or more, and is selected to be 2 or more at the maximum load, for example. With this setting, the choke coil Lo20 required in FIG. 23 can be deleted. That is, even when the choke coil Lo20 is not provided, the current I2, which is the resonance current flowing through the secondary winding N2, becomes a sine wave, and the voltage V4 generated at both ends of the secondary side voltage resonance capacitor C3 also becomes a sine wave. The conduction angle of the switching diode Do1 to the high-speed switching diode Do4 increases. Further, since the secondary side inductor Lo2 is for eliminating ringing, the inductance value of the secondary side inductor Lo2 can be made smaller than the inductance value of the choke coil Lo20. As a result, the size of the apparatus can be reduced and the cost of the apparatus can be reduced.

  Also, each of the switching element Q1 and the switching element Q2 is turned off by connecting the primary side voltage resonant capacitor C1 in parallel to the switching element Q1 and connecting the primary side voltage resonant capacitor C2 in parallel to the switching element Q2. The phase of the primary side resonant current C1 that flows through the primary side voltage resonant capacitor C2 and the current I4 that is the current that flows through the secondary side voltage resonant capacitor C3 at the time TOFF that is And the conduction angle of the body diode DD1 of the switching element Q1 and the body diode DD2 of the switching element Q2 is enlarged, and the ZVS operation is more reliable.

  As shown in FIG. 3, when the output DC voltage Eo is 175 V, the variable frequency range Δfs, which is the range between the value of the frequency fs corresponding to heavy load and the value of the frequency fs corresponding to zero load. The value of can be made very small.

  FIG. 4 shows each of the power conversion efficiency ηAC → DC, frequency fs, time TON, and time TOFF with respect to load fluctuations when the value of the load power Po when the input AC voltage VAC is 100 V is 0 W (no load) to 300 W. Show. In this switching power supply circuit, the value of the output DC voltage Eo is 175V.

  FIG. 5 shows an operation waveform of each part in the switching period. The load power at this time is 300 W, and the input AC voltage is 100V. In order from the top of FIG. 5, voltage V1 (see FIG. 1) that is the drain voltage of switching element Q1, current IQ1 (see FIG. 1) that is the drain current of switching element Q1, and voltage that is the drain voltage of switching element Q2 V2 (refer to FIG. 1), current IQ2 (refer to FIG. 1) which is the drain current of the switching element Q2, current I2 (refer to FIG. 1) which flows through the secondary winding N2, secondary side voltage resonance capacitor A current I3 (refer to FIG. 1) which flows through C3, a voltage V3 (refer to FIG. 1) which is a voltage across the secondary winding N2, and a current which flows into the input side of the secondary rectifier Do. Each of I4 (refer FIG. 1) is shown.

  FIG. 6 shows an operation waveform of each part in the switching period. The load power at this time is 0 W, and the input AC voltage is 100V. The voltage V1, current IQ1, voltage V2, current IQ2, current I2, current I3, voltage V3, and current I4 are shown in order from the top of FIG.

  From the experimental data shown in FIGS. 4 to 6, the following becomes clear about the switching power supply circuit shown in FIG. The ZVS operation is performed when the value of the load power Po is in the range of 0 W to 300 W. The value of power conversion efficiency ηAC → DC when the load power Po is 300 W is 91.8%.

  The switching power supply circuit shown in FIG. 1 has the following advantages.

  In the fs control of the voltage resonance push-pull converter shown in the background art, the load power Po is limited to 100 W or less. On the other hand, the switching power supply circuit shown in FIG. 1 uses a voltage resonance push-pull converter, but the converter transformer PIT is loosely coupled, and the voltage resonance circuit is combined on the secondary side to constitute a multiple resonance type converter. It is possible to perform fs control (control to change the output DC voltage Eo by changing the frequency fs, which is a switching frequency) in a range from a heavy load of 300 W or more to a no load.

  By using the converter transformer / inductor PIT / PCC as a composite part in which the functions of the converter transformer PIT and the inductor Lo1 are integrated, it is possible to reduce the size of the apparatus and to improve the power conversion efficiency ηAC → DC. And can. The reason why the power conversion efficiency ηAC → DC can be improved is that the converter transformer, the inductor PIT, and the PCC can be reduced in size, while the cross-sectional area of the E-type core CR3 that forms the magnetic path of the inductor Lo1 can be increased. This is because copper loss and iron loss are reduced. That is, when the inductor Lo1 is formed as a single component having the same inductance value as that in the converter transformer / inductor PIT / PCC, in consideration of cost, a smaller core material EE-25 is used, and the litz wire The number of turns is also large. For this reason, the copper loss becomes larger. On the other hand, when the core material used for the converter transformer PIT is also used as the choke coil in this way, the core CR3, which is a larger core, can be used for the same cost. The improvement of power conversion efficiency ηAC → DC when using the converter transformer / inductor PIT / PCC was improved by 0.3% at the maximum load power of 300 W, and the core temperature was also reduced by about 5 ° C.

  By providing the secondary side inductor Lo2, the ringing current that has been flowing in the secondary side voltage resonant capacitor C3 no longer flows, the operation waveform becomes a smooth sine wave shape, and the generation of noise is reduced.

  By adding a voltage resonance circuit to the secondary side, a sinusoidal voltage can be obtained from the voltage resonance circuit on the secondary side even if the choke coil Lo20 used in the background art is deleted. Here, the inductance of the choke coil Lo20 used in the background art is set to a large value in order to obtain a large inductance value, and the apparatus is enlarged and the apparatus cost is high. Then, by not using the choke coil Lo20, the size of the device can be reduced and the cost of the device can be reduced.

  FIG. 7 is a modification of the switching power supply circuit shown in FIG.

  The primary side of the switching power supply circuit of FIG. 7 is replaced with the primary side voltage resonance capacitor C1 and the primary side voltage resonance capacitor C2, and the primary side voltage resonance capacitor is used as the primary side voltage resonance capacitor C1. The first and second primary windings N1 and N1 ′ are connected in parallel to both ends of the primary winding N1 and the second primary winding N1 ′.

  The secondary side of the switching power supply circuit of FIG. 7 has a connection point between the secondary winding N2 and the secondary winding N2 ′ as a center tap and a secondary at the end of the secondary winding N2 and the secondary winding N2 ′. The secondary side is configured as a double-wave rectifier circuit having a high-speed switching diode Do1 and a high-speed switching diode Do2 for both-wave rectification connected to the side voltage resonant capacitor C3. Further, since the secondary winding N2 and the secondary winding N2 ′ are wound so as to have a positive polarity, the secondary winding N2 and the secondary winding N2 ′ forming the secondary side voltage resonance circuit The value of the inductance constituting the secondary side voltage resonance circuit generated at the end of the capacitor becomes larger than the value of the inductance in the circuit shown in FIG. 1, and the capacitance of the secondary side voltage resonance capacitor C3 can be reduced by this amount. Here, the secondary-side inductor Lo2 is connected between the connection point between the cathode of the high-speed switching diode Do1 and the cathode of the high-speed switching diode Do2, and the secondary-side smoothing capacitor Co to eliminate ringing.

  The switching power supply circuit shown in FIG. 8 is a switching power supply circuit obtained by modifying the secondary side of the switching power supply circuit shown in FIG. The secondary side constitutes a voltage doubler full wave rectifier circuit. That is, a DC voltage is held in the secondary side smoothing capacitor Co1, and this voltage is added to the voltage of the secondary side smoothing capacitor Co2, thereby being obtained from both ends of the secondary winding N2 or the secondary winding N2 ′. A voltage twice the voltage is obtained as the output DC voltage Eo. Here, the secondary-side inductor Lo2 is connected between the connection point between the cathode of the high-speed switching diode Do1 and the cathode of the high-speed switching diode Do2, and the secondary-side smoothing capacitor Co to eliminate ringing.

  FIG. 9 shows a modification of the secondary side of the switching power supply circuit shown in FIG. 1 using a voltage doubler half-wave rectifier circuit. Half of the voltage doubler full wave rectifier circuit shown in FIG. 9 is used. Thus, even when the voltage doubler half-wave rectifier circuit is used, the basic advantage of the switching power supply circuit shown in FIG. 1 can be maintained as it is. Here, the secondary-side inductor Lo2 is connected between the connection point between the cathode of the high-speed switching diode Do1 and the cathode of the high-speed switching diode Do2, and the secondary-side smoothing capacitor Co to eliminate ringing.

  A switching power supply circuit according to the second embodiment will be described with reference to FIGS. In the description of the switching power supply circuit of the second embodiment, the same components as those in the switching power supply circuit of the first embodiment are denoted by the same reference numerals, and a part of the description is omitted.

  The switching power supply circuit of the second embodiment is a switching power supply circuit that converts primary side DC power into AC power and further converts it into secondary side DC power, and the primary side DC power is supplied to one terminal. Choke coil PCC and a primary winding that is loosely coupled to primary winding N1 and primary winding N1 by a center tap connected to the other terminal of choke coil PCC and supplied with primary DC power. A converter transformer having a primary winding formed separately from N1 ′ and a secondary winding that is magnetically loosely coupled to each of the primary winding N1 and the second primary winding PIT, switching element Q1 connected to the end of primary winding N1, switching element Q2 connected to the end of primary winding N1 ′, and switching element Q1 and switching element Q2 are driven on / off. The primary side voltage resonance capacitor C1 and the inductor generated in the choke coil PCC and the primary winding N1 ′ together with the vibration / drive circuit 2, the choke coil PCC and the inductor generated in the primary winding N1 form a primary side voltage resonance circuit. The secondary winding N2 (or the secondary winding so as to form a secondary side voltage resonance circuit together with a primary side voltage resonance capacitor C2 that forms a primary side voltage resonance circuit together with an inductor generated in the secondary winding N2. A secondary side voltage resonance capacitor C3 connected in parallel to the line N2 and the secondary winding N2 ′), and a secondary side rectifier element Do (or high-speed switching diode Do1 to D2) whose input side is connected to the secondary side resonance circuit. It is connected to the output side of the high-speed switching diode Do4) and the secondary side rectifying element Do, etc. so as to obtain a smoothed output DC voltage Eo. A secondary-side smoothing capacitor Co (or secondary-side smoothing capacitor Co1 and secondary-side smoothing capacitor Co2) and a control signal whose frequency is variable so that the value of the output DC voltage Eo is a predetermined value are oscillated. And a control circuit 1 for supplying to the drive circuit 2, each of which is set to have a time ratio of 1 or more, which is a ratio of ON and OFF of the switching element Q1 and the switching element Q2.

  Here, in the present embodiment, the primary side voltage resonant capacitor C1 is an embodiment of the first primary side voltage resonant capacitor, and the primary side voltage resonant capacitor C2 is the second primary side voltage resonant capacitor. It is one embodiment. Both forms a primary-side voltage resonant capacitor, which is an embodiment of the primary-side voltage capacitor. As another embodiment of the primary side voltage resonance capacitor, instead of the primary side voltage resonance capacitor C1 and the primary side voltage resonance capacitor C2, both ends of the primary winding N1 and the primary winding N1 ′ are provided. Only the primary side voltage resonance capacitor C1 may be provided.

  The sequence from the AC power supply AC side to the secondary side of the switching power supply circuit shown in FIG. 10 will be described in order.

  The switching power supply circuit has a common mode filter having an across line capacitor CL1, a common mode choke coil CMC, and an across line capacitor CL2. For example, a plug (not shown) is connected to the across-line capacitor CL1 on the input side of the common mode filter via a power switch and a fuse, and the AC power is obtained by inserting the plug into an outlet connected to the AC power supply AC. Is supplied to the switching power supply circuit. Here, the value of the input AC voltage VAC, which is the voltage of the AC power supply AC, is in the range of 85V to 144V.

  The output side of the common mode filter (the side of the across-line capacitor CL2) is the connection point between the anode of the diode Di1 and the cathode of the diode Di2 that are the input side of the primary side rectifier element Di, Connected to the connection point.

  The positive side terminal of the primary side smoothing capacitor Ci is connected to the connection point between the cathode of the diode Di1 which is the output side of the primary side rectifying element Di and the cathode of the diode Di3, and the anode of the diode Di2 and the anode of the diode Di4 Is connected to the negative terminal of the primary smoothing capacitor Ci. In this way, the primary side rectifying and smoothing circuit is formed, and a DC voltage is supplied to the push-pull converter from both ends of the primary side smoothing capacitor.

  The push-pull converter includes a switching element Q1 and a switching element Q2, a choke coil Lo1, a primary side voltage resonance capacitor C1, a primary side voltage resonance capacitor C2, a control circuit 1, an oscillation / drive circuit 2, a converter transformer PIT, and a secondary side. A voltage resonance capacitor C3, a secondary side rectifying element Do, a secondary side smoothing capacitor Co, and a secondary side inductor Lo2 are provided.

  Switching element Q1 and switching element Q2 are MOS-FETs, and body diode DD1 and body diode DD2 are diodes formed in the process of manufacturing a MOS-FET. Here, the specification of the MOS-FET is 5A / 900V.

  The converter transformer PIT has functions of insulating the primary side and the secondary side and performing voltage conversion, but also functions as an inductor L1, an inductor L1 ', and an inductor L2. Here, each inductor is a leakage inductor formed by the converter transformer PIT. How to form such a leakage inductor will be specifically described with reference to a sectional view of the converter transformer PIT shown in FIG.

  The converter transformer PIT includes an EE type core in which an E type core CR1 and an E type core CR2 made of a ferrite material are combined so that their magnetic legs face each other. The winding portions of the primary winding N1, the primary winding N1 ', and the secondary winding N2 are divided so as to be independent from each other, and provided with a bobbin B formed of, for example, resin. The bobbin B around which the primary winding N1 and the primary winding N1 ′ as the primary winding portion and the secondary winding N2 as the secondary winding portion are wound is attached to the EE core. ing.

  With such a configuration, the magnetic flux passing through the magnetic path formed by the center magnetic leg of the EE core and the gap G is linked to the primary winding N1, the primary winding N1 ′, and the secondary winding N2. Will be. However, since this magnetic path has a gap G, a part of the magnetic flux leaks from the gap G, so that the degree of mutual magnetic coupling of the windings wound around this magnetic path is small (loose coupling). ). Since the secondary winding N2 is sandwiched between the primary winding N1 and the primary winding N1 ′ and each winding is wound around the core material without passing through other windings, the primary winding is particularly The degree of coupling between the line N1 and the primary winding N1 ′ is small, the degree of coupling between the primary winding N1 and the secondary winding N2, and between the primary winding N1 ′ and the secondary winding N2. The degree of coupling is also small. That is, the degree of coupling between the windings is small. In this way, the overall structure of the converter transformer PIT is obtained.

  A gap G of 1.6 mm (millimeters) is formed with respect to the central magnetic leg of the EE type core. Accordingly, as described above, the respective windings are loosely coupled to each other, and a part of the magnetic flux generated in the primary winding N1, the primary winding N1 ′, and the secondary winding N2 is made to other windings. The inductor L1, the inductor L1 ′, and the inductor L2 are formed by the effect of the magnetic flux not interlinked with the line. The gap G is formed by making the central magnetic legs of the E-type core CR1 and the E-type core CR2 shorter than the two outer magnetic legs. The core material was EER-35 (core material name). The number of turns of the primary winding N1 and the number of turns of the primary winding N1 ′ are 60T (turns), and the number of turns of the secondary winding N2 is 35T. The value of the coupling coefficient k with the winding N1 ′ is 0.60, the value of the coupling coefficient k between the primary winding N1 and the secondary winding N2, and the primary winding N1 ′ and the secondary winding. The value of the coupling coefficient k with N2 was 0.80. As a result, the value of the inductor L1 and the value of the inductor L1 ′ are 580 μH due to the loose coupling between the primary winding N1 and the secondary winding N2, and the primary winding N1 ′ and the secondary winding N2. Yes, each of the value of the inductor L1l and the value of the inductor L1l ′ generated by the loose coupling between the primary winding N1 and the primary winding N1 ′ was 200 μH. In the converter transformer PIT in FIG. 23 shown as the background art, the coupling coefficient between the primary winding N1 and the primary winding N1 ′ is 0.98. In this case, the value of the inductor L1l and the value of the inductor L1l ′ Each is a small one.

  In the second embodiment, the choke coil Lo1 in the first embodiment is configured as a choke coil PCC as an independent component, and the choke coil PCC has a structure similar to the converter transformer PIT. FIG. 12 shows a cross-sectional view of the choke coil PCC. The bobbin B has only one winding region and is provided with a winding No. Also, cores CR4 and CR5, which are EE-25 (model number) smaller than EER-35, are adopted as the core size, the gap G is 0.8 mm, and the inductance value of the choke coil PCC is 200 μH. It has gained. The value of 200 μH is set to have a size approximately equal to the values of the inductor L1l and the inductor L1l ′.

  The connection mode on the primary side will be described. The choke coil Lo1 is connected between the primary side smoothing capacitor Ci, the primary winding N1 of the converter transformer PIT, and the center tap of the primary winding N1 ′, and the end of the primary winding N1 of the converter transformer PIT. Is connected to the drain of the switching element Q1, and the end of the primary winding N1 ′ of the converter transformer PIT is connected to the drain of the switching element Q2. The sources of the switching element Q1 and the switching element Q2 are grounded to the primary side ground. The primary side voltage resonance capacitor C1 is connected in parallel between the drain and source of the switching element Q1, and the primary side voltage resonance capacitor C2 is connected in parallel between the drain and source of the switching element Q2.

  The output of the oscillation / drive circuit 2 is connected to the gate of the switching element Q1 and the gate of the switching element Q1. The oscillation / drive circuit 2 drives the gate of the switching element Q1 and the gate of the switching element Q2 according to the output from the control circuit 1.

  The connection mode on the secondary side will be described. A secondary side rectifying element Do is connected to the secondary winding N2 of the converter transformer PIT. The secondary side rectifying element Do is configured by bridge-connecting high speed switching diodes Do1 to Do4, and a secondary side smoothing capacitor Co is connected to the output side of the secondary side rectifying element Do. An output DC voltage Eo that is a voltage supplied to the load from both ends of the secondary side smoothing capacitor Co is obtained. Here, the value of the output DC voltage Eo was 175 V (volts).

  The control circuit 1 has its input side connected to the secondary side smoothing capacitor Co, and oscillates a detection output (error voltage) corresponding to the difference between the input output DC voltage Eo and a predetermined reference voltage value. Supply to the drive circuit 2

  In such a push-pull converter, a first primary voltage resonance circuit is mainly formed by the choke coil PCC, the inductor L1l, the inductor L1, and the primary voltage resonance capacitor C1. Here, mainly means the resonance frequency of the first primary side voltage resonance circuit by the characteristics of these components, that is, the inductance of the choke coil PCC and the inductance of the inductor L1 and the capacitance of the primary side voltage resonance capacitor C1. It is said that is generally determined. For example, the capacitance of the primary side smoothing capacitor Ci influences the resonance frequency, but the influence is negligible. Therefore, it is not used as an element for forming the first primary side voltage resonance circuit.

  Further, a second primary side voltage resonance circuit is mainly formed by the choke coil PCC, the inductor L1l ', the inductor L1', and the primary side voltage resonance capacitor C2. That is, the resonance frequency of the second primary side voltage resonance circuit is generally determined by the inductance of the choke coil PCC, the inductance of the inductor L1l, the inductance of the inductor L1 ′, and the capacitance of the primary side voltage resonance capacitor C2. The side smoothing capacitor Ci or the like is not an element forming the second primary side voltage resonance circuit. Each is set to the frequency fo1 so that the resonance frequency of the first primary side voltage resonance circuit and the resonance frequency of the second primary side voltage resonance circuit are substantially equal.

  Also, a secondary side voltage resonance circuit is formed mainly by the inductance of the inductor L2 and the capacitance of the secondary side voltage resonance capacitor C3, and the resonance frequency is set to the frequency fo2. Here, mainly as in the case described above, for example, the secondary side inductor Lo2 and the like also affect the resonance frequency, but the effect is negligible, so that as an element forming the secondary side voltage resonance circuit No. As shown in FIG. 3, the relationship between the frequency fo1 and the frequency fo2 is such that the value of the frequency fo2 is set to approximately 2/3 or less of the frequency fo1. Here, the value of the secondary inductor Lo2 is set to several μH or less, for example, 3.3 μH, which is very small as compared with the inductance value of the inductor L2. Therefore, although the secondary side inductor Lo2 becomes an element constituting the secondary side voltage resonance circuit, it hardly affects the resonance frequency of the secondary side voltage resonance circuit.

  The effect of the secondary inductor Lo2 is exactly the same as that shown in the first embodiment. When the secondary inductor Lo2 is not provided, that is, when a conductor is used instead of the secondary inductor Lo2, For example, ringing of 1.5 MHz (mega hertz) occurs in I3 (see FIG. 10) and current I4 (see FIG. 10). Since this ringing has a high frequency, it is harmful as a noise component. For example, when a switching power supply circuit is configured as an AC power supply AC side or a DC-DC converter, this noise leaks to the input DC side, which adversely affects the operation of other devices. As described above, the cause of such ringing is the equivalent inductance (ESL) existing in the secondary side smoothing capacitor Co. By connecting the transmission inductance and the secondary side inductor Lo2 in series, this ringing occurs. The 1.5 Mhz ringing disappears.

  Here, the capacitance value of the primary side voltage resonance capacitor C1 and the capacitance value of the primary side voltage resonance capacitor C2 are 3300 pF (Pico Farad), and the capacitance value of the secondary side voltage resonance capacitor C3 is 0.068 μF. (Micro Farad). At this time, the value of the frequency fo1 was 99.3 kHz, and the value of the frequency fo2 was 52.7 kHz.

  Next, the operation of the switching power supply circuit having the above-described configuration will be described. The basic operation principle of such a push-pull converter operates as a multi-resonance converter, which is a known technique, and has the following characteristics.

  The duty ratio between the switching element Q1 and the switching element Q2 is set to 1 or more, and is selected to be 2 or more at the maximum load, for example. With this setting, the choke coil Lo20 required in FIG. 23 can be deleted. That is, even when the choke coil Lo20 is not provided, the current I2, which is the resonance current flowing through the secondary winding N2, becomes a sine wave, and the voltage V4 generated at both ends of the secondary side voltage resonance capacitor C3 also becomes a sine wave. The conduction angle of the switching diode Do1 to the high-speed switching diode Do4 increases. Further, since the secondary side inductor Lo2 is for eliminating ringing, the inductance value of the secondary side inductor Lo2 can be made smaller than the inductance value of the choke coil Lo20, and the apparatus can be miniaturized.

  Further, by making the primary winding N1 and the primary winding N1 ′ loosely coupled, an inductor L1l and an inductor L1 ′ are obtained, and a part of the inductance value required for the choke coil PCC is obtained by the inductor L1l and the inductor L1l. The choke coil PCC can be reduced in size by compensating for the inductance value generated by the inductor L1 ′.

  Also, each of the switching element Q1 and the switching element Q2 is turned off by connecting the primary side voltage resonant capacitor C1 in parallel to the switching element Q1 and connecting the primary side voltage resonant capacitor C2 in parallel to the switching element Q2. The phase of the primary side resonant current C1 that flows through the primary side voltage resonant capacitor C2 and the current I4 that is the current that flows through the secondary side voltage resonant capacitor C3 at the time TOFF that is And the conduction angle of the body diode DD1 of the switching element Q1 and the body diode DD2 of the switching element Q2 is enlarged, and the ZVS operation is more reliable.

  As shown in FIG. 13, when the output DC voltage Eo is set to 175 V, the variable frequency range Δfs, which is a range between the value of the frequency fs corresponding to heavy load and the value of the frequency fs corresponding to zero load. The value of can be made very small.

  FIG. 13 shows the values of power conversion efficiency ηAC → DC, frequency fs, time TON, and time TOFF for load fluctuations when the input AC voltage VAC is 100 V and the load power Po is 0 W (no load) to 300 W. Show.

  FIG. 14 shows an operation waveform of each part in the switching period. The load power at this time is 300 W, and the input AC voltage is 100V. In order from the top of FIG. 14, voltage V1 (see FIG. 10) that is the drain voltage of switching element Q1, current IQ1 (see FIG. 10) that is the drain current of switching element Q1, and voltage that is the drain voltage of switching element Q2 V2 (see FIG. 10), current IQ2 (see FIG. 10) which is the drain current of the switching element Q2, current I2 (see FIG. 10) which is the current flowing in the secondary winding N2, secondary side voltage resonance capacitor A current I3 that flows through C3 (see FIG. 10), a voltage V3 that operates across the secondary winding N2 (see FIG. 10), and a current that flows through the input side of the secondary rectifier Do. Each of I4 (refer FIG. 10) is shown.

  FIG. 15 shows an operation waveform of each part in the switching period. The load power at this time is 0 W, and the input AC voltage is 100V. Each of the voltage V1, the current IQ1, the voltage V2, the current IQ2, the current I2, the current I3, the voltage V3, and the current I4 is shown in order from the top of FIG.

  The following becomes clear about the switching power supply circuit shown in FIG. The ZVS operation is performed when the value of the load power Po is in the range of 0 W to 300 W. Further, the frequency fs has an inflection point at an intermediate load in which the value of the load power Po is around 100 W. The value of power conversion efficiency ηAC → DC when the load power Po is 300 W is 91.8%. Here, the efficiency is improved by 0.4% compared with the value of power conversion efficiency ηAC → DC at 300 W when the choke coil PCC having an inductance value of 400 μH is used using the core material of EE-25. This is obtained as a result of reducing the copper loss and the iron loss by reducing the number of turns of the winding No. composed of the litz wire because the inductance value required for the choke coil PCC may be small. It is an effect. As described above, the inductance value of the choke coil PCC can be set to 200 μH because the primary winding N1 and the primary winding N1 ′ are loosely coupled, and the inductor L1l and the inductor L1l ′ formed as leakage inductors. This is because each inductance value is set to 200 μH.

  The switching power supply circuit shown in FIG. 10 has the following advantages.

  In the fs control of the voltage resonance push-pull converter shown in the background art, the load power Po is limited to 100 W or less. On the other hand, the switching power supply circuit shown in FIG. 10 uses a voltage resonance push-pull converter, but is configured as a multiple resonance type converter by combining the converter transformer PIT with a loose coupling and a voltage resonance circuit on the secondary side. It is possible to perform fs control (control to change the output DC voltage Eo by changing the frequency fs, which is a switching frequency) in a range from a heavy load of 300 W or more to a no load.

  Loss in the choke coil PCC is reduced by loosely coupling the primary winding N1 and the primary winding N1 ′ and reducing the inductance value of the choke coil PCC by the inductor L1l and the inductor L1l ′ formed as leakage inductors. . Further, as shown in FIG. 11, the primary winding N1, the primary winding N2, and the primary winding N1 ′ are arranged in a row so that the primary winding N1 and the primary winding N1 ′ Assuming that each length is equal, the operation of the first primary side resonance circuit and the operation of the second primary side resonance circuit can be well balanced.

  By providing the secondary side inductor Lo2, the ringing current that has been flowing in the secondary side voltage resonant capacitor C3 no longer flows, the operation waveform becomes a smooth sine wave shape, and the generation of noise is reduced.

  By adding a voltage resonance circuit to the secondary side, a sine wave voltage can be obtained from the voltage resonance circuit on the secondary side even if the choke coil Lo20 used in the background art is deleted. Here, the secondary inductor Lo2 is for preventing ringing, and its inductance value is, for example, 1/100 of the inductance value of the choke coil Lo20 as compared to the choke coil Lo20 used in the background art. The shape and cost are greatly different, and the secondary inductor Lo2 is smaller and less expensive than the choke coil Lo20.

  FIG. 16 is a modification of the switching power supply circuit shown in FIG.

  The primary side of the switching power supply circuit of FIG. 16 is replaced with the primary side voltage resonance capacitor C1 and the primary side voltage resonance capacitor C2, and the primary side voltage resonance capacitor is used as the primary side voltage resonance capacitor C1. The first and second primary windings N1 and N1 ′ are connected in parallel to both ends of the primary winding N1 and the second primary winding N1 ′.

  The secondary side of the switching power supply circuit of FIG. 16 has a connection point between the secondary winding N2 and the secondary winding N2 ′ as a center tap and a secondary at the end of the secondary winding N2 and the secondary winding N2 ′. The secondary side is configured as a double-wave rectifier circuit having a high-speed switching diode Do1 and a high-speed switching diode Do2 for both-wave rectification connected to the side voltage resonant capacitor C3. Further, since the secondary winding N2 and the secondary winding N2 ′ are wound so as to have a positive polarity, the secondary winding N2 and the secondary winding N2 ′ forming the secondary side voltage resonance circuit The value of the inductance constituting the secondary side voltage resonance circuit generated at the end of the capacitor becomes larger than the inductance value in the circuit shown in FIG. 10, and the capacitance of the secondary side voltage resonance capacitor C3 can be reduced by this amount. The secondary-side inductor Lo2 is connected to the output-side connection point between the high-speed switching diode Do1 and the high-speed diode Do2, thereby eliminating the ringing.

  As the switching power supply circuit shown in FIG. 10 or FIG. 16, a secondary side circuit similar to that shown in FIG. 8 in the description of the first embodiment can be used. In FIG. 8, the secondary side forms a voltage doubler full wave rectifier circuit. That is, a DC voltage is held in the secondary side smoothing capacitor Co1, and this voltage is added to the voltage of the secondary side smoothing capacitor Co2, thereby being obtained from both ends of the secondary winding N2 or the secondary winding N2 ′. A voltage twice the voltage is obtained as the output DC voltage Eo. The secondary-side inductor Lo2 is connected to the output-side connection point between the high-speed switching diode Do1 and the high-speed diode Do2, thereby eliminating the ringing.

  The switching power supply circuit shown in FIG. 10 or FIG. 16 can also use a secondary side circuit similar to that shown in FIG. 9 in the description of the first embodiment. In FIG. 9, the secondary side uses a voltage doubler half-wave rectifier circuit. Half of the voltage doubler full wave rectifier circuit shown in FIG. 8 is used. Thus, even when the voltage doubler half-wave rectifier circuit is used, the basic advantage of the switching power supply circuit shown in FIG. 10 can be maintained as it is. The secondary-side inductor Lo2 is connected to the high-speed switching diode Do1, and the secondary-side inductor Lo2 'is connected to the high-speed diode Do2, thereby eliminating the ringing.

  A switching power supply circuit according to the third embodiment will be described with reference to FIGS.

  The switching power supply circuit according to the third embodiment is a switching power supply circuit that converts primary side DC power into AC power and further converts it into secondary side DC power, and is provided by a center tap to which primary side DC power is supplied. A primary winding formed by separating a primary winding N1 and a primary winding N1 ′ that is loosely coupled to the primary winding N1, and a primary winding N1 and a primary winding N1 ′ A converter transformer having a secondary winding N2 that is magnetically loosely coupled to each other, and a switching element Q1 connected to an end of the primary winding N1 and an end of the primary winding N1 ′ Primary side voltage resonance together with the switching element Q2, the oscillation / drive circuit 2 for driving the switching element Q1 and the switching element Q2 on and off, and the inductors generated in the primary winding N1 and the primary winding N1 ′, respectively. Circuit Primary side voltage resonance capacitor C1 connected in parallel to switching element Q1 and primary side voltage resonance capacitor C2 connected in parallel to switching element Q2 (or connected to the drain of switching element Q1 and the drain of switching element Q2) Primary side voltage resonance capacitor C1) and secondary side voltage resonance capacitor C3 connected in parallel to the secondary winding N2 so as to form a secondary side voltage resonance circuit together with an inductor generated in the secondary winding N2. The secondary side rectifying element Do (or the high speed switching diode Do1 to the high speed switching diode Do4) whose input side is connected to the secondary side resonance circuit and the output side of the secondary side rectifying element Do are smoothed. The secondary side smoothing capacitor Co (or the secondary side smoothing capacitor) adapted to obtain the output DC voltage Eo. And a control circuit 1 for supplying a control signal whose frequency is variable so that the value of the output DC voltage Eo is a predetermined value to the oscillation / drive circuit 2. In addition, each of the time ratios, which is the ratio of ON and OFF of the switching element Q1 and the switching element Q2, is set to 1 or more. In the following, the details will be described, but parts having the same configuration as in the first embodiment and having the same functions are denoted by the same reference numerals as in the first embodiment, and a part of the description is omitted. .

  The steps from the AC power supply AC side to the secondary side of the switching power supply circuit shown in FIG.

  The switching power supply circuit has a common mode filter having an across line capacitor CL1, a common mode choke coil CMC, and an across line capacitor CL2. For example, a plug (not shown) is connected to the across-line capacitor CL1 on the input side of the common mode filter via a power switch and a fuse, and the AC power is obtained by inserting the plug into an outlet connected to the AC power supply AC. Is supplied to the switching power supply circuit. Here, the value of the input AC voltage VAC, which is the voltage of the AC power supply AC, is in the range of 85V to 144V.

  The output side of the common mode filter (the side of the across-line capacitor CL2) is the connection point between the anode of the diode Di1 and the cathode of the diode Di2 that are the input side of the primary side rectifier element Di, Connected to the connection point.

  The positive side terminal of the primary side smoothing capacitor Ci is connected to the connection point between the cathode of the diode Di1 which is the output side of the primary side rectifying element Di and the cathode of the diode Di3, and the anode of the diode Di2 and the anode of the diode Di4 Is connected to the negative terminal of the primary smoothing capacitor Ci. In this way, the primary side rectifying and smoothing circuit is formed, and a DC voltage is supplied to the push-pull converter from both ends of the primary side smoothing capacitor.

  The push-pull converter includes a switching element Q1 and a switching element Q2, a primary side voltage resonance capacitor C1, a primary side voltage resonance capacitor C2, a control circuit 1, an oscillation / drive circuit 2, a converter transformer PIT, and a secondary side voltage resonance capacitor C3. A secondary side rectifying element Do and a secondary side smoothing capacitor Co are provided.

  Switching element Q1 and switching element Q2 are MOS-FETs, and body diode DD1 and body diode DD2 are diodes formed in the process of manufacturing a MOS-FET. Here, the specification of the MOS-FET is 5A / 900V.

  The converter transformer PIT has a function of insulating the primary side and the secondary side and performing voltage conversion, and further includes an inductor L1 and an inductor L1 ′ that are leakage inductors generated on the primary side and the secondary side. It also functions as an inductor L2 and also functions as an inductor L1l and an inductor L1l ′, which are leakage inductances generated in the primary winding N1 and the primary winding N1 ′. Here, each inductor is a leakage inductor formed by the converter transformer PIT.

  As in FIG. 2, the converter transformer PIT used in the second embodiment shown in FIG. 18 is an EE type core in which an E type core CR1 and an E type core CR2 made of a ferrite material are combined so that their magnetic legs face each other. Is provided. The EE type core includes two outer magnetic legs formed by the E type core CR1 and the E type core CR2 and a central magnetic leg having a gap G shorter than the outer magnetic legs. . The secondary winding N2 is wound around the central magnetic leg, the primary winding N1 is wound around one of the external magnetic legs, and the primary winding N1 'is wound around the other external magnetic leg.

  By winding the primary winding N1 around one of the external magnetic legs and winding the primary winding N1 'around the other of the external magnetic legs, a part of the magnetic flux generated at one of the external magnetic legs is part of the other external magnetic leg. The central magnetic leg is short-circuited without passing through the magnetic leg. As a result, the magnetic coupling coefficient between the primary winding N1 and the primary winding N1 'is a loose coupling of about 0.3. A gap G of 1.6 mm (millimeters) is formed with respect to the central magnetic leg of the EE type core. As a result, the value of the coupling coefficient k between the primary side and the secondary side is a loose coupling of about 0.41. The gap G is formed by making the central magnetic legs of the E-type core CR1 and the E-type core CR2 shorter than the two outer magnetic legs. The core material was EER-35 (core material name). The number of turns of the primary winding N1 and the number of turns of the primary winding N1 'are respectively the same number of turns of 75T (turns), and the number of turns of the secondary winding N2 is 42T.

  As a result, the value of the inductor L1 and the value of the inductor L1 ′ are 545 μH due to the loose coupling between the primary winding N1 and the secondary winding N2, and the primary winding N1 ′ and the secondary winding N2. Yes, each of the value of the inductor L1l and the value of the inductor L1l ′ due to the loose coupling of the primary winding N1 and the primary winding N1 ′ is 455 μH, the value of the inductor L2 is 233 μH, and 1 The value of the inductor L2l generated in the secondary winding generated by the loose coupling between the secondary winding N1 and the primary winding N1 ′ was 195 μH. In the converter transformer PIT shown in FIG. 23 as the background art as a comparative example, the coupling coefficient between the primary winding N1 and the primary winding N1 ′ is 0.98. In this case, the value of the inductor L1l, the inductor Each value of L1l ′ is small.

  The connection mode on the primary side will be described. The primary side smoothing capacitor Ci is connected between the primary winding N1 of the converter transformer PIT and the center tap of the primary winding N1 ′, and the end of the primary winding N1 of the converter transformer PIT is connected to the switching element Q1. The end of the primary winding N1 ′ of the converter transformer PIT is connected to the drain of the switching element Q2. The sources of the switching element Q1 and the switching element Q2 are grounded to the primary side ground. The primary side voltage resonance capacitor C1 is connected in parallel between the drain and source of the switching element Q1, and the primary side voltage resonance capacitor C2 is connected in parallel between the drain and source of the switching element Q2.

  The output of the oscillation / drive circuit 2 is connected to the gate of the switching element Q1 and the gate of the switching element Q1. The oscillation / drive circuit 2 drives the gate of the switching element Q1 and the gate of the switching element Q2 according to the output from the control circuit 1.

  The connection mode on the secondary side will be described. A secondary side rectifying element Do is connected to the secondary winding N2 of the converter transformer PIT. The secondary side rectifying element Do is configured by bridge-connecting high speed switching diodes Do1 to Do4, and a secondary side smoothing capacitor Co is connected to the output side of the secondary side rectifying element Do. An output DC voltage Eo that is a voltage supplied to the load from both ends of the secondary side smoothing capacitor Co is obtained. Here, the value of the output DC voltage Eo was 175 V (volts).

  The control circuit 1 has its input side connected to the secondary side smoothing capacitor Co, and oscillates a detection output (error voltage) corresponding to the difference between the input output DC voltage Eo and a predetermined reference voltage value. Supply to the drive circuit 2

  In such a push-pull converter, a first primary side voltage resonance circuit is formed by the inductor L1l, the inductor L1, and the primary side voltage resonance capacitor C1. Here, mainly means that the resonance frequency of the first primary-side voltage resonance circuit depends on the characteristics of these components, that is, the inductance of the inductor L1l, the inductance of the inductor L1, and the capacitance of the primary-side voltage resonance capacitor C1. It is to say that it is generally determined. For example, the capacitance of the primary side smoothing capacitor Ci influences the resonance frequency, but the influence is negligible. Therefore, it is not used as an element for forming the first primary side voltage resonance circuit.

  Further, a second primary side voltage resonance circuit is mainly formed by the inductor L1l ', the inductor L1' and the primary side voltage resonance capacitor C2. That is, since the resonance frequency of the second primary voltage resonance circuit is substantially determined by the inductance of the inductor L1l ′, the inductance of the inductor L1 ′, and the capacitance of the primary voltage resonance capacitor C2, the primary smoothing capacitor Ci and the like Is not an element forming the second primary side voltage resonance circuit. Each is set to the frequency fo1 so that the resonance frequency of the first primary side voltage resonance circuit and the resonance frequency of the second primary side voltage resonance circuit are substantially equal.

  Further, a secondary side voltage resonance circuit is mainly formed by the inductance of the inductor L21, the inductance of the inductor L2, and the capacitance of the secondary side voltage resonance capacitor C3, and the resonance frequency is set to the frequency fo2. Here, mainly, as described above, for example, the secondary side smoothing capacitor Co or the like also affects the resonance frequency, but the influence is negligible. Not as. As shown in FIG. 4, the relationship between the frequency fo1 and the frequency fo2 is such that the value of the frequency fo2 is set to approximately 2/3 or less of the frequency fo1.

  Here, the capacitance value of the primary side voltage resonance capacitor C1 and the capacitance value of the primary side voltage resonance capacitor C2 were 4700 pF, and the capacitance value of the secondary side voltage resonance capacitor C3 was 0.033 μF. At this time, the value of the frequency fo1 was 99.5 kHz, and the value of the frequency fo2 was 59.5 kHz.

  Next, the operation of the switching power supply circuit having the above-described configuration will be described. The basic operation principle of such a push-pull converter operates as a multi-resonance converter, which is a known technique, and has the following characteristics.

  The duty ratio between the switching element Q1 and the switching element Q2 is set to 1 or more, and is selected to be 2 or more at the maximum load, for example. With this setting, the choke coil Lo20 required in FIG. 23 can be deleted. That is, even when the choke coil Lo20 is not provided, the current I2, which is the resonance current flowing through the secondary winding N2, becomes a sine wave, and the voltage V4 generated at both ends of the secondary side voltage resonance capacitor C3 also becomes a sine wave. The conduction angle of the switching diode Do1 to the high-speed switching diode Do4 increases.

  Also, each of the switching element Q1 and the switching element Q2 is turned off by connecting the primary side voltage resonant capacitor C1 in parallel to the switching element Q1 and connecting the primary side voltage resonant capacitor C2 in parallel to the switching element Q2. The phase of the primary side resonant current C1 that flows through the primary side voltage resonant capacitor C2 and the current I4 that is the current that flows through the secondary side voltage resonant capacitor C3 at the time TOFF that is And the conduction angle of the body diode DD1 of the switching element Q1 and the body diode DD2 of the switching element Q2 is enlarged, and the ZVS operation is more reliable.

  Conventionally, the choke coil required between the primary side smoothing capacitor Ci and the primary winding is not provided.

  Further, as shown in FIG. 4, when the output DC voltage Eo is 175 V, the variable frequency range Δfs, which is a range between the value of the frequency fs corresponding to heavy load and the value of the frequency fs corresponding to zero load. The value of can be made very small.

  FIG. 19 shows the values of power conversion efficiency ηAC → DC, frequency fs, time TON, and time TOFF for load fluctuations when the value of the load power Po when the input AC voltage VAC is 100 V is 0 W (no load) to 300 W. Show. In addition, about this switching power supply circuit, the value of the output DC voltage Eo was set to 175V.

  FIG. 20 shows an operation waveform of each part in the switching period. The load power at this time is 300 W, and the input AC voltage is 100V. In order from the top of FIG. 20, a voltage V1 (see FIG. 17) that is the drain voltage of the switching element Q1, a current IQ1 (see FIG. 17) that is the drain current of the switching element Q1, and a voltage that is the drain voltage of the switching element Q2. V2 (see FIG. 17), current IQ2 (see FIG. 17) which is the drain current of the switching element Q2, current I2 (see FIG. 17) which is the current flowing in the secondary winding N2, secondary side voltage resonance capacitor A current I3 (refer to FIG. 17) that flows in C3, a voltage V3 (refer to FIG. 17) that is a voltage across the secondary winding N2, and a current that flows to the input side of the secondary rectifier Do. Each of I4 (refer FIG. 17) is shown.

  FIG. 21 shows an operation waveform of each part in the switching period. The load power at this time is 0 W, and the input AC voltage is 100V. In the order from the top of FIG. 19, each of voltage V1, current IQ1, voltage V2, current IQ2, current I2, current I3, voltage V3, and current I4 is shown.

  From the experimental data shown in FIGS. 19 to 21, the following becomes clear about the switching power supply circuit shown in FIG. The ZVS operation is performed when the value of the load power Po is in the range of 0 W to 300 W.

  The switching power supply circuit shown in FIG. 17 has the following advantages.

  In the fs control of the voltage resonance push-pull converter shown in the background art, the load power Po is limited to 100 W or less. On the other hand, the switching power supply circuit shown in FIG. 17 uses a voltage resonance push-pull converter, but the converter transformer PIT is loosely coupled, and the voltage resonance circuit is combined on the secondary side to constitute a multiple resonance type converter. It is possible to perform fs control (control to change the output DC voltage Eo by changing the frequency fs, which is a switching frequency) in a range from a heavy load of 300 W or more to a no load.

  By adding a voltage resonance circuit on the secondary side, a sine wave voltage can be obtained from the voltage resonance circuit on the secondary side even if the choke coil Lo20 used in the background art is deleted, and noise can be reduced. It was.

  Since the primary winding N1 and the primary winding N1 ′ are loosely coupled with a coupling coefficient k of 0.3, conventionally, it is necessary between the primary-side smoothing capacitor Ci and the primary winding. Therefore, the choke coil is not required, and the apparatus can be miniaturized.

  FIG. 22 shows a modification of the switching power supply circuit shown in FIG.

  The primary side of the switching power supply circuit of FIG. 22 is replaced with the primary side voltage resonance capacitor C1 and the primary side voltage resonance capacitor C2, and the primary side voltage resonance capacitor is used as the primary side voltage resonance capacitor C1. The first and second primary windings N1 and N1 ′ are connected in parallel to both ends of the primary winding N1 and the second primary winding N1 ′.

  The secondary side of the switching power supply circuit of FIG. 22 has a connection point between the secondary winding N2 and the secondary winding N2 ′ as a center tap, and the secondary side at the end of the secondary winding N2 and the tertiary winding N3. The secondary side is configured as a double-wave rectifier circuit having a high-speed switching diode Do1 and a high-speed switching diode Do2 connected to the voltage resonance capacitor C3 and having a double-wave rectification. Further, since the secondary winding N2, the secondary winding N2 ′, and the tertiary winding N3 are wound so as to have a positive polarity, the secondary windings N2 and 2 that form the secondary side voltage resonance circuit The inductance value constituting the secondary side voltage resonance circuit generated at the end of the secondary winding N2 ′ is larger than the inductance value in the circuit shown in FIG. 16, and the capacitance of the secondary side voltage resonance capacitor C3 is increased by this amount. Can be small. The secondary-side inductor Lo2 is connected to the output-side connection point between the high-speed switching diode Do1 and the high-speed diode Do2, thereby eliminating the ringing.

  The switching power supply circuit shown in FIG. 17 or 22 can also use a secondary side circuit similar to that shown in FIG. 8 in the description of the first embodiment. In FIG. 8, the secondary side forms a voltage doubler full wave rectifier circuit. That is, a DC voltage is held in the secondary side smoothing capacitor Co1, and this voltage is added to the voltage of the secondary side smoothing capacitor Co2, thereby being obtained from both ends of the secondary winding N2 or the secondary winding N2 ′. A voltage twice the voltage is obtained as the output DC voltage Eo. The secondary-side inductor Lo2 is connected to the output-side connection point between the high-speed switching diode Do1 and the high-speed diode Do2, thereby eliminating the ringing.

  The switching power supply circuit shown in FIG. 17 or 22 can also use a secondary side circuit similar to that shown in FIG. 9 in the description of the first embodiment. In FIG. 9, the secondary side uses a voltage doubler half-wave rectifier circuit. Half of the voltage doubler full wave rectifier circuit shown in FIG. 8 is used. Thus, even when the voltage doubler half-wave rectifier circuit is used, the basic advantage of the switching power supply circuit shown in FIG. 17 can be maintained as it is. The secondary-side inductor Lo2 is connected to the high-speed switching diode Do1, and the secondary-side inductor Lo2 'is connected to the high-speed diode Do2, thereby eliminating the ringing.

  In addition, this invention is not limited to embodiment mentioned above, Embodiment can be changed as needed.

It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is sectional drawing of the converter transformer inductor of embodiment. It is a figure which shows the control characteristic of the switching power supply circuit of embodiment. It is a figure which shows the characteristic of the power supply efficiency with respect to the load power of the switching power supply circuit of embodiment, a switching frequency, TON, and TOFF. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram showing a part of an example of composition of a switching power supply circuit of an embodiment. It is a circuit diagram showing a part of an example of composition of a switching power supply circuit of an embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is sectional drawing of the converter transformer of embodiment. It is sectional drawing of the choke coil of embodiment. It is a figure which shows the characteristic of the power supply efficiency with respect to the load power of the switching power supply circuit of embodiment, a switching frequency, TON, and TOFF. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is sectional drawing of the converter transformer of embodiment. It is a figure which shows the characteristic of the power supply efficiency with respect to the load power of the switching power supply circuit of embodiment, a switching frequency, TON, and TOFF. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a figure explaining background art. It is a figure explaining background art. It is a figure explaining background art. It is a figure explaining background art. It is a figure explaining background art.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Control circuit, 2 Oscillation drive circuit, B bobbin, C1 primary side voltage resonance capacitor, C2 primary side voltage resonance capacitor, C3 secondary side voltage resonance capacitor, Ci primary side smoothing capacitor, CL1, CL2 Cross line capacitor , CMC common mode choke coil, Co, Co1, Co2 secondary side smoothing capacitor, CR1, CR2, CR3 E type core, DD1, DD2 body diode, Di primary side rectifier, Di1, Di2, Di3, Di4, Do1, Do2, Do3, Do4 High-speed switching diode, Do secondary side rectifier, Eo output DC voltage, G, G1, G2 gap, L1, L1l, L2l, L1l inductor, Lo1, PCC choke coil (inductor), Lo2 secondary side Inductor, N1, N1 'primary winding, N2 secondary winding, No choke coil winding, PIT converter transformer, PIT / PCC converter transformer / inductor, Q1, Q2 switching element

Claims (14)

A switching power supply circuit that converts primary side DC power to AC power and further converts to secondary side DC power,
A choke coil to which the primary DC power is supplied to one terminal;
A primary winding formed by separating a first primary winding and a second primary winding by a center tap connected to the other terminal of the choke coil; and the first primary winding. A converter transformer having a wire and a secondary winding that is magnetically loosely coupled to each of the second primary windings;
A first switching element connected to an end of the first primary winding and a second switching element connected to an end of the second primary winding;
An oscillation / drive circuit for driving on and off the first switching element and the second switching element;
A primary-side voltage resonance capacitor that forms a primary-side voltage resonance circuit together with inductors generated in each of the choke coil, the first primary winding, and the second primary winding;
A secondary-side voltage resonant capacitor connected in parallel to the secondary winding to form a secondary-side voltage resonant circuit with an inductor generated in the secondary winding;
A secondary side rectifying element having an input side connected to the secondary side resonance circuit;
A secondary-side smoothing capacitor connected to the output side of the secondary-side rectifying element to obtain a smoothed output DC voltage;
A control circuit for supplying a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit,
Each of the time ratios, which is a ratio of on and off of the first switching element and the second switching element, is set to 1 or more,
The converter transformer is
A first core material having a first gap and forming a first magnetic path, wherein the first primary winding, the second primary winding, and the secondary winding; Each is wound so that the magnetic flux passing through the first magnetic path is linked,
The choke coil is
The first core material is included as a part of a magnetic path component, and a second magnetic path having a second gap is formed so that magnetic flux passing through the first magnetic path does not pass. Comprising the second core material, the choke coil winding is wound so that the magnetic flux passing through the second magnetic path is interlinked,
The switching power supply circuit according to claim 1, wherein the converter transformer and the choke coil are integrally formed as a composite part. A switching power supply circuit that converts primary side DC power to AC power and further converts to secondary side DC power,
A choke coil to which the primary DC power is supplied to one terminal;
A first tap connected to the other terminal of the choke coil is loosely coupled to the first primary winding and the first primary winding by a center tap to which the primary DC power is supplied. A primary winding formed by separating two primary windings, and a secondary magnetically loosely coupled to each of the first primary winding and the second primary winding. A converter transformer having windings;
A first switching element connected to an end of the first primary winding and a second switching element connected to an end of the second primary winding;
An oscillation / drive circuit for driving on and off the first switching element and the second switching element;
A primary-side voltage resonance capacitor that forms a primary-side voltage resonance circuit together with inductors generated in each of the choke coil, the first primary winding, and the second primary winding;
A secondary-side voltage resonant capacitor connected in parallel to the secondary winding to form a secondary-side voltage resonant circuit with an inductor generated in the secondary winding;
A secondary side rectifying element having an input side connected to the secondary side resonance circuit;
A secondary-side smoothing capacitor connected to the output side of the secondary-side rectifying element to obtain a smoothed output DC voltage;
A control circuit for supplying a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit,
A switching power supply circuit, wherein a time ratio, which is a ratio of ON to OFF of the first switching element and the second switching element, is set to 1 or more. A switching power supply circuit that converts primary side DC power to AC power and further converts to secondary side DC power,
The first primary winding and the second primary winding that is loosely coupled to the first primary winding are separated from each other by a center tap to which the primary side DC power is supplied. A converter transformer having a primary winding, and a secondary winding that is magnetically loosely coupled to each of the first primary winding and the second primary winding;
A first switching element connected to an end of the first primary winding and a second switching element connected to an end of the second primary winding;
An oscillation / drive circuit for driving on and off the first switching element and the second switching element;
A primary-side voltage resonant capacitor that forms a primary-side voltage resonant circuit together with an inductor generated in each of the first primary winding and the second primary winding;
A secondary-side voltage resonant capacitor connected in parallel to the secondary winding to form a secondary-side voltage resonant circuit with an inductor generated in the secondary winding;
A secondary side rectifying element having an input side connected to the secondary side resonance circuit;
A secondary-side smoothing capacitor connected to the output side of the secondary-side rectifying element to obtain a smoothed output DC voltage;
A control circuit for supplying a control signal whose frequency is variable so that the value of the output DC voltage is a predetermined value to the oscillation / drive circuit,
A switching power supply circuit, wherein a time ratio, which is a ratio of ON to OFF of the first switching element and the second switching element, is set to 1 or more. The converter transformer is
Comprising a core material forming a magnetic path with a gap;
The secondary winding is sandwiched between the first primary winding and the second primary winding, and each of the windings is wound around the core material without passing through another winding, The switching power supply circuit according to claim 2, wherein the first primary winding, the second primary winding, and the secondary winding are loosely coupled to each other. The converter transformer is
Comprising an EE-type core material comprising two outer magnetic legs and a central magnetic leg having a shorter length than the outer magnetic legs and having a gap;
Winding the secondary winding around the central magnetic leg;
Winding the first primary winding around one of the external magnetic legs;
Winding the second primary winding around the other of the external magnetic legs;
4. The switching power supply circuit according to claim 3, wherein the first primary winding, the second primary winding, and the secondary winding are loosely coupled to each other. A secondary inductor is provided between the output side of the secondary rectifier and the secondary smoothing capacitor;
The inductance value of the secondary inductor is set to a magnitude that prevents ringing that occurs on the secondary side and does not affect the resonance frequency of the secondary voltage resonance circuit. The switching power supply circuit according to any one of claims 1 to 3. The primary side voltage resonant capacitor includes a first primary side voltage resonant capacitor connected in parallel to the first switching element and a second primary side voltage connected in parallel to the second switching element. The switching power supply circuit according to claim 1, comprising a resonance capacitor. 4. The primary voltage resonant capacitor is connected in parallel to both ends of a series connection of the first primary winding and the second primary winding. The switching power supply circuit according to 1 above. The resonance frequency of the secondary side resonance circuit is set to be lower than about 2/3 of the resonance frequency of the resonance frequency of the primary side voltage resonance circuit. The switching power supply circuit according to 1. 4. The switching power supply circuit according to claim 1, wherein the secondary side rectifier circuit is formed as a full-wave rectifier circuit. 5. 4. The switching power supply circuit according to claim 1, wherein the secondary side rectifier circuit is formed as a double-wave rectifier circuit. 5. 4. The switching power supply circuit according to claim 1, wherein the secondary side rectifier circuit is formed as a voltage doubler full wave rectifier circuit. 5. 4. The switching power supply circuit according to claim 1, wherein the secondary side rectifier circuit is formed as a voltage doubler half-wave rectifier circuit. 5. 4. The switching power supply circuit according to claim 1, wherein the duty ratio is approximately 2 when the load power is maximized.

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