JP2008154390A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply circuit where the occurrence of noise is reduced, a value of the fluctuation range of frequency is narrowed, and a device is miniaturized using a single piece of a switching element. <P>SOLUTION: This switching power supply circuit is configured as a class-E multiplex resonant converter having one piece of the switching element Q1. This multiplex resonant converter is provided with the primary side resonance circuit and the secondary side resonance circuit. The primary-side resonance circuit includes a primary-side parallel resonance circuit, in which a resonance frequency fop1 is controlled by a primary-side parallel resonance capacitor C1 and an inductor Lp of a choke coil PCC, and a primary-side series resonance circuit, in which a resonance frequency fos1 is controlled by a primary-side series resonance capacitor C2 and an inductor L1 which occurs in the primary winding N1. The secondary-side resonance circuit includes a secondary-side parallel resonance circuit, in which a resonance frequency fop2 is controlled by a secondary side parallel resonance capacitor C3 and an inductor L2 which occurs in the secondary winding N2. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

  In recent years, most power supply circuits that rectify a commercial power supply to obtain a desired DC voltage are switching power supply circuits. A switching power supply circuit is used as a power source for various electronic devices as a high-power DC-DC converter while miniaturizing a transformer and other devices by increasing a switching frequency.

  There are various DC-DC converter systems, for example, a half-bridge system with two switching elements, a full-bridge system with four switching elements, and a push-pull DC-DC system with two switching elements. A DC converter (hereinafter abbreviated as a push-pull converter) is also used (see, for example, Non-Patent Document 1). Furthermore, a DC-DC converter (hereinafter abbreviated as a single-ended converter) using a single switching element is also used (see, for example, Patent Document 1). Hereinafter, as a background art, each of the push-pull converter and the single-ended converter will be described in order.

  An example of a push-pull converter circuit is shown in FIG. In FIG. 20, only the power part of the push-pull converter is described, and the control circuit for controlling the gates of the switching element Q11 and the switching element Q12, which are MOS-FETs, constituting the push-pull converter is not described. On (conduction) and off (disconnection) of switching element Q11 and switching element Q12 are controlled by a control circuit (not shown). Here, as a general configuration of the MOS-FET, the switching element Q11 includes a body diode DD11, and the switching element Q12 includes a body diode DD12.

  The converter transformer PIT is formed by winding a primary winding N1, a primary winding N1 ′, a secondary winding N2 and a secondary winding N2 ′ around a ferrite core (magnetic core). The coupling coefficient between the line N1 and the primary winding N1 ′, and the secondary winding N2 and the secondary winding N2 ′ is close to 1. The primary winding N1 and the primary winding N1 ′ have the same number of turns, and the connection point between the primary winding N1 and the primary winding N1 ′ is the winding start of one winding and the winding of the other winding. The center tap is connected to the end. Here, the inductor L1 is a leakage inductor generated in the primary winding N1, the inductor L1 ′ is a leakage inductor generated in the primary winding N1 ′, the inductor L2 is a leakage inductor generated in the secondary winding N2, and the inductor L2 ′ is the primary. A leakage inductor generated in the winding N2 ′, formed by the primary winding formed by the primary winding N1 and the primary winding N1 ′, and the secondary winding N2 and the secondary winding N2 ′. Inductors generated in the respective windings when the coupling coefficient of the windings on the secondary side is not 1.

  A choke coil Lo1 is connected to the center tap, and a voltage Ei is supplied via the choke coil Lo1, and the primary winding N1 and the primary winding N1 ′, the secondary winding N2 and the secondary winding are supplied. Due to the imbalance of the coupling coefficient with N2 ′, the occurrence of bias is suppressed. Further, primary side parallel resonant capacitors C1 are connected to both ends of the primary winding N1 and the primary winding N1 '.

  For the secondary side, the high speed switching diode Do1 is connected to the secondary winding N2, the high speed switching diode Do2 is connected to the secondary winding N2 ′, and the choke coil Lo20 and the secondary are connected to the output side of the rectifier circuit. A DC output voltage Eo is obtained by arranging a side smoothing capacitor Co to form a double-wave rectifier circuit having a choke input smoothing circuit. The conduction angle of the high-speed switching diode Do1 and the high-speed switching diode Do2 is expanded by having a smoothing circuit for choke input.

  In addition, as described above, the gates of the switching element Q11 and the switching element Q12 are controlled by a control circuit (not shown), and the time ratio (TON / TOFF) of each of the switching element Q11 and the switching element Q12 is set to 1. That is, the length of the period TON = the length of the period TOFF = the length of the period TS / 2. Here, the period TON is the time during which the switching element Q11 or the switching element Q12 is turned on, the period TOFF is the time during which the switching element Q11 or the switching element Q12 is disconnected, and the period TS is the time of one cycle of the switching period. is there. In this way, the waveform of the voltage V4 generated between the secondary winding N2 and the secondary winding N2 'is a sine wave (see voltage V4 in FIG. 21).

  FIG. 21 shows the waveform of each part of the circuit shown in FIG. From the top of FIG. 21, voltage V3 (see FIG. 20), voltage V1 (see FIG. 20), current IQ11 (see FIG. 20), voltage V2 (see FIG. 20), current IQ12 (see FIG. 20) And voltage V4 (see FIG. 20). As shown by current IQ11 and current IQ12, body diode DD11 and body diode DD12 do not conduct, and ZVS (Zero Voltage Switching) operation is performed as shown by voltage V1 and voltage V2.

  In this operation mode, when the frequency fs, which is the switching frequency between the switching element Q11 and the switching element Q12, is increased, the switching element Q11 and the switching element Q12 become conductive during the period TOFF, and the switching loss increases. When the switching loss exceeds a predetermined value, the so-called thermal runaway state may occur and the switching element Q11 and the switching element Q12 may be destroyed.

  Therefore, when controlling the DC output voltage Eo, as shown in FIG. 22, the operational amplification OP, the saturable magnetic core SR-1 and the saturable magnetic core SR-2 on the secondary side, the diode Dr3 functioning as a reset diode, and the diode A magnetic amplifier composed of Dr4 and diode Dr5 is added, and the conduction angle of the high-speed switching diode Do1 and the high-speed switching diode Do2 is controlled by the action of this magnetic amplifier, and the DC output voltage Eo is controlled with the frequency fs fixed. A magamp control system has been proposed.

  In this case, the voltage appearing at both ends of the saturable magnetic core SR-1 and the saturable magnetic core SR-2 becomes a waveform including harmonics and the switching noise also increases. In addition, saturable magnetic core SR-1 and saturable magnetic core SR-2 having a square hysteresis characteristic, diode Dr3, diode Dr4, and diode Dr5 are required as additional components.

  Therefore, in order to eliminate the need for the saturable magnetic core SR-2, the diode Dr3, the diode Dr4, and the diode Dr5, the primary winding N1 and the primary winding N1 ′ and the secondary winding N2 are not known in the art. It can be considered that the coupling coefficient with the secondary winding N2 ′ is loosely coupled, and the values of the inductor L1, the inductor L1 ′, and the inductor L2 are increased, and the voltage is controlled using this value.

  For example, FIG. 23 shows an example of the push-pull converter shown in FIG. 21 in which the values of the inductor L1, the inductor L1 ′, and the inductor L2 are increased to decrease the value of the primary side parallel resonant capacitor C1, and the time ratio (TON FIG. 6 is a waveform diagram of voltage V1, current IQ1, and voltage V2 when the value of / TOFF is 2. When the value of the time ratio is 2, the value of the DC output voltage Eo that is the output voltage when the frequency fs that is the switching frequency (the reciprocal of the period TS that is the switching period) is changed is shown in FIG. As shown in The graph indicated by Po = 0 is a graph when the power supplied to a load (not shown) is 0 W (watts), and the graph indicated by Pomax is a graph when the power supplied to the load is maximum.

  Further, as can be seen from FIG. 24, the value of the DC output voltage Eo is lowest at the frequency fo1, the case where the value from the frequency fo1 to the frequency fs is high, and the case where the value from the frequency fo1 to the frequency fs is low. The value of the DC output voltage Eo increases. Here, the frequency fo1 is a resonance frequency of the primary side parallel resonance circuit. That is, this graph shows a primary side parallel resonance curve formed by the choke coil Lo1, the inductor L1, and the primary side parallel resonance capacitor C1, and the choke coil Lo1, the inductor L1 ′, and the primary side parallel resonance capacitor C1. It is. The inventor described in the application of the present application (hereinafter abbreviated as the inventor of the present application) has found that the DC output voltage Eo can be controlled by controlling the frequency fs from FIG.

  However, the variable frequency range for setting the value of the DC output voltage Eo to a constant value from when the power supplied to the load is 0 W to when the power supplied to the load is maximum is given by the variable frequency range Δfs. Thus, when the setting shown in FIG. 24 is adopted, the variable range of the switching frequency is, for example, 100 kHz or more and is large. For this reason, the maximum load power within the range in which the ZVS operation is possible is limited to 100 W or less.

  Therefore, the present inventor further studied and provided a voltage resonance circuit on the secondary side to form a so-called multiple resonance converter, as shown in FIG. 25, the secondary side of the primary side parallel resonance frequency fo1. By setting the frequency relationship so that the side parallel resonance frequency fo2 becomes a low frequency, control is performed according to the resonance curve having the secondary side parallel resonance frequency fo2 at the time of heavy load, and secondary at the time of intermediate load. It has been found that the size of the variable frequency range Δfs, which is the variable range of the frequency, can be reduced by changing the control according to the resonance curve having the side parallel resonance frequency fo2. It was also found that the variable frequency range Δfs can be increased in this way, and the range of the ZVS region can be expanded.

  On the other hand, as an example of a circuit of a single-ended converter that employs a single switching element, for example, a complex resonant converter having a voltage resonant converter on the primary side and a parallel resonant circuit on the secondary side is known. (See Patent Document 1). The single-ended converter shown in FIG. 26 is an example of such a complex resonant converter. A single switching element Q1 is used, and a primary side parallel resonant circuit including an inductor L1 generated in the primary winding N1 and a primary side parallel resonant capacitor C1 is provided. In this case, the secondary side is a high-speed switching diode. A half-wave rectifier circuit formed by connecting a secondary-side smoothing capacitor Co to Do1, and a secondary-side parallel resonance circuit including an inductor L2 generated in the secondary winding N2 and a secondary-side parallel resonance capacitor C3. 3 is a circuit example of a voltage resonance mode single-ended multiple resonance type converter. Here, the switching element Q1 is an NPN transistor, and the body diode DD1 is connected between the base and the emitter. Here, the withstand voltage of the switching element Q1 is set to 900V, and the range of the AC input voltage VAC is compatible with a wide range corresponding to both a so-called 100V system and a so-called 200V system.

  In order to support a wide range, a series circuit of a clamping capacitor C4 and an auxiliary switching element Q2 is connected in parallel to the primary winding N1, that is, the clamping capacitor C4 and the auxiliary switching element Q2 An active clamp circuit is formed. The auxiliary switching element Q2 is a MOS-FET and has a body diode DD2. The gate of the auxiliary switching element Q2 is driven by a voltage obtained by differentiating the voltage from the clamp winding Ng wound around the converter transformer PIT by a differentiation circuit including a capacitor Cg and a resistor Rg. By this differentiated voltage, the auxiliary switching element Q2 is turned on during the period when the switching element Q1 is cut off, and the voltage of the switching element Q1 is clamped.

  A voltage feedback power factor correction circuit is configured by the tertiary winding N3 wound around the converter transformer PIT, the filter capacitor CN, the power factor improving high speed diode Dr, the diode D1, and the power factor improving inductor Lo. .

  In general, the converter transformer PIT is a combination of two E-shaped ferrite cores having a square magnetic path cross section or an EER type ferrite core having a circular magnetic path cross section. The secondary winding and the tertiary winding N3 are wound, and a gap of about 1 mm (millimeter) is provided in the central magnetic leg of the ferrite core. By providing the gap in this way, the primary winding N1 and the secondary winding N2 are magnetically loosely coupled, and the coupling coefficient is about 0.8. As a result, the primary winding N1 functions as the inductor L1, and the secondary winding N2 functions as the inductor L2. The secondary side parallel resonant capacitor C3 and the inductor L2 generated in the secondary winding N2 form a secondary side parallel resonant circuit. On the other hand, the primary side parallel resonance capacitor C1 and the inductor L1 generated in the primary winding N1 form a primary side parallel resonance circuit.

  Specifically, the constants of the respective parts of the single-ended converter shown in FIG. 26 are set as follows. The converter transformer PIT uses EER-35 (core material name), the gap G is 1 mm (millimeter), the coupling coefficient k is 0.81, the number of turns of the primary winding N1 is 48T (turn), 3 The number of turns of the secondary winding N3 is 18T, the number of turns of the clamping coil Ng is 1T, and the number of turns of the secondary winding N2 is 50T. The capacitance value of the primary side parallel resonant capacitor C1 is 3300 pF (pico farad), the capacitance value of the secondary side parallel resonant capacitor C3 is 0.01 μF (micro farad), and the capacitance value of the clamping capacitor C4. Is 0.047 μF. The value of the resistor Rg is 470Ω (ohms), and the value of the capacitor Cg is 0.33 μF. The inductance value of the power factor improving inductor Lo is 68 μH, and the value of the filter capacitor CN is 1 μF. The value of the DC output voltage Eo (see FIG. 26) is 135V (volts), the range of power supplied to the load is from 200W (watts) to 0W of load power, and the AC input voltage VAC is from 90V to 288V. It is.

  The control circuit / oscillation / drive circuit PFM connected to the switching element Q1 is a circuit for controlling the switching element Q1, and modulates the switching element Q1 by pulse frequency. The input side of the control circuit / oscillation / drive circuit PFM is connected to the secondary side smoothing capacitor Co to obtain a constant voltage characteristic in which the value of the DC output voltage Eo is 135V.

  FIG. 27 shows the characteristics of this switching power supply circuit, and the vertical axis shows each of power conversion efficiency ηAC → DC, power factor PF, and voltage Ei of the primary side smoothing capacitor (see FIG. 26). Yes, the horizontal axis is the power supplied to the load. A solid line indicates a case where the effective voltage value of the AC input voltage VAC is 100 V, and a broken line indicates a case where the effective voltage value of the AC input voltage VAC is 230 V. At this time, the frequency range in the pulse frequency modulation ranges from 96 kHz (kilo-hertz) to 205 kHz when the load power is 200 W to no load, and the control circuit / oscillation / drive circuit is used by using a general-purpose control IC. When configuring the PFM, the load power range is up to 200W.

  Further, in the voltage feedback type power factor improvement circuit using the tertiary winding N3, the cathode voltage of the power factor improving high-speed diode Dr is increased by the resonance pulse voltage induced in the tertiary winding N3. When the anode voltage is lower than the anode voltage of the diode Dr, the current I2 flows in the order of the filter capacitor CN, the power factor improving high speed diode Dr, and the tertiary winding N3, and the current I1 flows through the diode D1 only at the peak of the input AC voltage. When the value of the AC input voltage VAC is 100 V, the region where the value of the power factor PF is 0.75 or more is the range of the load power from 200 W to 20 W, and when the value of the AC input voltage VAC is 230 V, The region where the value of the power factor PF is 0.75 or more is the range where the load power is 200 W to 100 W.

  FIG. 28 shows the characteristics of this switching power supply circuit seen from another point of view. The vertical axis represents power conversion efficiency ηAC → DC, power factor PF, and voltage Ei, and the horizontal axis represents It is an effective voltage value of the AC input voltage VAC. The power supplied to the load at this time was constant at 200W. The diagonal lines parallel to the vertical axis shown in FIG. 28 indicate the input AC voltage 100V and the input AC voltage 230V, respectively.

  FIG. 29 is a drawing-substituting photograph showing a photograph of the waveform of the main part of the switching power supply circuit measured with an oscilloscope. 29, the AC input voltage VAC (see FIG. 26), AC input current IAC (see FIG. 26), voltage V1 (see FIG. 26), current I1 (see FIG. 26), voltage, V2 (refer to FIG. 26), current I2 (refer to FIG. 26), ripple voltage ΔEi corresponding to variation of voltage Ei (refer to FIG. 26), ripple corresponding to variation of DC output voltage Eo (refer to FIG. 26) Each of the voltages ΔEo is indicated by a cycle of the AC input voltage VAC. The load power at this time is a waveform when it is relatively large.

  FIG. 30 is also a drawing-substituting photograph, and shows a photograph of the waveform of the main part of the switching power supply circuit measured with an oscilloscope. From the top to the bottom of FIG. 30, the AC input voltage VAC, the AC input current IAC, the voltage V1, the current I1, the voltage V2, the current I2, the ripple voltage ΔEi that is the fluctuation of the voltage Ei, and the fluctuation of the DC output voltage Eo. Each ripple voltage ΔEo is indicated by the period of the AC input voltage VAC. The load power at this time is a waveform when the power is relatively small.

JP 2000-324831 A Ninomiya, et al. “Operational Characteristics of Voltage Resonant Push-Pull Converters Using Magnetic Control System” Transactions of the Institute of Electronics, Information and Communication Engineers B Vol. J70-B No. 11 pp 1307-1315, November 1987

  In the push-pull converter shown in FIG. 20 described above, the configuration of means for performing constant voltage control with respect to a wide range of load power fluctuations is complicated, and the number of components greatly increases. In order to reduce noise, it is desirable to use a secondary voltage waveform as a sine wave for use in an audio device. However, in order to use a secondary voltage waveform as a sine wave, the switching on the primary side is preferable. It is necessary to set the time ratio (TON / TOFF) of the element to 1. When a magnetic amplifier is provided in order to keep the voltage supplied from the secondary side to the load constant over a wide range of load fluctuations while keeping the duty ratio (TON / TOFF) of the switching element at 1. There is a problem that the waveform of the high frequency voltage on the secondary side is not a sine wave and noise increases.

  Further, without setting the time ratio (TON / TOFF) of the switching element on the primary side to 1, for example, the switching frequency is controlled by setting the time ratio to 2, and the voltage supplied from the secondary side to the load is kept constant. In this case, the length of the period TOFF increases during light load, making it difficult to perform the ZVS operation. In such a case, the maximum load value is limited to, for example, 100 W or less, and the variable range of the switching frequency is, for example, 100 kHz or more as described above.

  In addition, the push-pull converter shown in the background art described above has a problem that a choke coil is required on the secondary side because the choke input method is adopted, which increases the size of the power supply device and increases the cost of the power supply device. is doing.

  In the single-ended converter shown in FIG. 26 described above, the value of the control range is wide (for example, 96 kHz to 205 kHz), and a commercially available general-purpose control circuit / oscillation / drive circuit IC is used as the control circuit. It was difficult. For this reason, a dedicated circuit must be newly designed, and there is a problem that the price of the switching power supply circuit becomes expensive.

  As for the power factor improvement effect, when the value of the AC input voltage VAC is 100V, the region where the power factor PF is 0.75 or more is in the range of load power from 200W to 20W. When the value of the voltage VAC is 230V system, the region where the value of the power factor PF is 0.75 or more has a problem that the load power is narrow in the range of 200W to 100W.

  In addition, the number of necessary components for configuring the power factor correction circuit is as follows: the tertiary winding N3 wound around the converter transformer PIT, the filter capacitor CN, the power factor improving high speed diode Dr, the diode D1, the power factor There are five parts for the improvement inductor Lo, and the number of parts is large.

  In view of the above-described problems, the present invention provides a switching power supply circuit that reduces the occurrence of noise, narrows the frequency change range in control, and reduces the size of the apparatus by using one switching element. Objective.

  A switching power supply circuit according to the present invention is a switching power supply circuit that obtains a DC output voltage of a predetermined voltage from an AC power source, and primary side rectification for converting AC power from the AC power source into primary DC power. A primary side rectifier circuit including an element and a primary side smoothing capacitor, and the primary side DC power is supplied to one winding end of a primary winding via a choke coil, and the primary winding and the A converter transformer having a secondary winding that is magnetically loosely coupled to the primary winding; a switching element connected to the other end of the primary winding; An oscillation / drive circuit to be disconnected, a primary side parallel resonant capacitor that forms a primary side parallel resonant circuit together with the choke coil and an inductor generated in the primary winding, and an input generated in the primary winding A primary side series resonant capacitor that forms a primary side series resonant circuit together with a transformer, a secondary side rectifier circuit that obtains a DC output voltage by converting AC power generated in the secondary winding into DC power, and the DC A secondary side parallel resonance circuit that forms a secondary side parallel resonance circuit together with a control circuit that supplies a control signal for setting the output voltage value to a predetermined value to the oscillation / drive circuit and an inductor generated in the secondary winding A resonance frequency of the primary side parallel resonance circuit is set to be substantially equal to a resonance frequency of the secondary side parallel resonance circuit, and the resonance frequency of the primary side series resonance circuit is the primary frequency The resonance frequency of the side parallel resonance circuit or the resonance frequency of the secondary side parallel resonance circuit is set to approximately ½.

  In this switching power supply circuit, AC power is once converted into primary DC power by the primary side rectifying element and the primary side smoothing capacitor. This primary side DC power is supplied to one end of the primary winding of the converter transformer via a choke coil. A switching element is connected to the other end of the primary winding of the converter transformer. This switching element is turned on or off by an oscillation / drive circuit to supply AC power to the primary winding of the converter transformer. To do. Since this converter transformer has a primary winding and a secondary winding, and the primary winding and the secondary winding are magnetically loosely coupled, the primary winding and the secondary winding Each produces an inductor. A primary side parallel resonance circuit is formed by the choke coil and the inductor generated in the primary winding and the primary side parallel resonance capacitor, and the primary side series resonance is formed by the inductor generated in the primary winding and the primary side series resonance capacitor. A circuit is formed, and a secondary parallel resonant circuit is formed by the inductor generated in the secondary winding and the secondary parallel resonant capacitor. The secondary side rectifier circuit converts AC power generated in the secondary winding to DC power to obtain a DC output voltage. The control circuit supplies a control signal to the oscillation / drive circuit to set the value of the DC output voltage to a predetermined value. The resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set to be approximately equal, and the resonance frequency of the primary side series resonance circuit is the resonance frequency of the primary side parallel resonance circuit or By setting the resonance frequency to approximately ½ of the resonance frequency of the secondary parallel resonance circuit, the ZVS operation can be performed over a wide load power range.

  According to the present invention, it is possible to provide a switching power supply circuit in which the generation of noise is reduced, the value of the frequency change range in control is narrowed, and the size of the apparatus is reduced by using one switching element.

  The best mode for carrying out the present invention (hereinafter referred to as an embodiment) will be described. The switching power supply circuit according to the embodiment is a switching power supply circuit that obtains a predetermined DC output voltage from an AC power source. This switching power supply circuit has the following configuration as a primary side. A primary-side rectifier circuit including a primary-side rectifying element and a primary-side smoothing capacitor for converting AC power from an AC power source into primary-side DC power is provided. In addition, the primary winding and the primary winding are magnetically sparse so that the primary DC power is supplied to one end of the primary winding of the inverter transformer via the choke coil. A converter transformer having a secondary winding to be coupled is provided. Moreover, the switching element connected to the other winding end of a primary winding is provided. In addition, an oscillation / drive circuit that turns on or off the switching element is provided. In addition, a primary side parallel resonant capacitor that forms a primary side parallel resonant circuit together with the inductor of the choke coil and the inductor generated in the primary winding is provided. In addition, a primary side series resonance capacitor that forms a primary side series resonance circuit together with an inductor generated in the primary winding is provided.

  Moreover, the following structure is provided as a secondary side. A secondary-side rectifier circuit is provided that converts AC power generated in the secondary winding into DC power to obtain a DC output voltage. In addition, a control circuit is provided for supplying a control signal for setting the value of the DC output voltage to a predetermined value to the oscillation / drive circuit. In addition, a secondary parallel resonant capacitor that forms a secondary parallel resonant circuit together with an inductor generated in the secondary winding is provided.

  The mutual frequency relationship among the resonance frequency of the primary side parallel resonance circuit, the resonance frequency of the primary side series resonance circuit, and the resonance frequency of the secondary side parallel resonance circuit is set as follows. That is, the resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set to be approximately equal, and the resonance frequency of the primary side series resonance circuit is equal to the resonance frequency of the primary side parallel resonance circuit or It is set to approximately ½ of the resonance frequency of the secondary side parallel resonance circuit. Here, “substantially equal” means that the resonance frequency of the other side of the resonance frequency of the primary side parallel resonance circuit or the resonance frequency of the secondary side parallel resonance circuit is in the range of about 20%. Is. Further, “substantially 1/2” means that the resonance frequency is within about 20% of the resonance frequency of the primary side parallel resonance circuit or 1/2 of the resonance frequency of the secondary side parallel resonance circuit.

(Switching power supply circuit of the first embodiment)
The switching power supply circuit according to the first embodiment is configured as a voltage resonance single-ended multiple resonance converter using a class E switching amplifier including one switching element. The multiple resonance converter includes a primary side resonance circuit (primary side resonance circuit) and a secondary side resonance circuit (secondary side resonance circuit). The primary side resonance circuit has a resonance frequency by a primary side parallel resonance circuit whose resonance frequency is governed by a primary side parallel resonance capacitor and an inductor of a choke coil, a primary side series resonance capacitor, and an inductor generated in the primary winding. And a primary-side series resonant circuit that is controlled by. The secondary side resonance circuit includes a secondary side parallel resonance circuit whose resonance frequency is governed by a secondary side parallel resonance capacitor and an inductor generated in the secondary winding. Here, the coupling coefficient between the primary winding and the secondary winding of the converter transformer is a loose coupling of 0.8 or less, so that an inductor serves as a leakage inductor in each of the primary winding and the secondary winding. Will occur. The resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set to be approximately equal, and the resonance frequency of the primary side series resonance circuit is the resonance frequency of the primary side parallel resonance circuit or It is set to approximately ½ of the resonance frequency of the secondary side parallel resonance circuit.

  The main part of the switching power supply circuit according to the first embodiment will be described with reference to FIG.

  In the switching power supply circuit of the first embodiment shown in FIG. 1, the primary side and the secondary side are separated by a converter transformer PIT (see FIG. 2). The primary side has the following main components. A primary side rectifying element Di to which a diode Di1 to a diode Di4 to which commercial AC power is supplied is bridge-connected, and a pulsating voltage obtained by rectification by the primary side rectifying element Di are smoothed to obtain a direct current on the primary side. A primary side smoothing capacitor Ci for obtaining electric power. The primary side DC power is provided with a choke coil PCC supplied to one winding end of the winding Np. The choke coil PCC is formed by winding a winding Np around a core and functions as an inductor Lp. Further, a converter transformer PIT connected to the other winding end of the winding Np of the choke coil PCC is provided. The converter transformer PIT has a primary winding N1 and a secondary winding N2 that is magnetically loosely coupled to the primary winding N1, for example, a coupling coefficient of 0.8 or less, as described above. The other winding end of the choke coil PCC is connected to one winding end of the primary winding N1 of the converter transformer PIT, and the switching element Q1 is connected to the other winding end of the converter transformer PIT. The switching element Q1 uses a MOS-FET having a body diode DD1 therein, and the drain of the MOS-FET is connected to the other winding end of the primary winding N1 of the converter transformer PIT. Moreover, the primary side parallel resonant capacitor C1 connected in parallel with the switching element Q1 is provided. Further, the source of the switching element Q1 and the other winding end of the primary winding N1 are connected via a primary side series resonance capacitor C2.

  With such a connection mode, the primary side parallel resonance circuit whose resonance frequency is dominated by the primary side parallel resonance capacitor C1 and the inductor Lp of the choke coil PCC, the primary side series resonance capacitor C2, and the primary winding N1. And a primary side series resonance circuit whose resonance frequency is governed by the inductor L1 generated in the circuit. The fact that the resonance frequency of the primary side parallel resonance circuit is governed means that the resonance frequency fp1 of the primary side parallel resonance circuit is almost determined by the primary side parallel resonance capacitor C1 and the inductor Lp. For example, the primary side smoothing capacitor Ci also affects the resonance frequency of the primary side parallel resonance circuit, but compared to the capacitance value of the primary side parallel resonance capacitor C1, the capacitance value of the primary side smoothing capacitor Ci is Since it is 1000 times larger, the primary side smoothing capacitor Ci does not greatly affect the resonance frequency of the primary side parallel resonance circuit.

  Further, the fact that the resonance frequency of the primary side series resonance circuit is governed means that the resonance frequency fos1 of the primary side series resonance circuit is almost determined by the primary side series resonance capacitor C2 and the inductor L1. For example, the inductor Lp also affects the resonance frequency of the primary side series resonance circuit, but since the inductance value of the inductor Lp is larger than the inductance value of the inductor L1, the inductor Lp is the primary side series resonance circuit. It does not greatly affect the resonance frequency.

  The switching element Q1 is controlled by the oscillation / drive circuit 2 to be turned on or off. More specifically, the gate of the switching element Q 1 is controlled by the oscillation / drive circuit 2.

  Further, the secondary side of the converter transformer PIT has the following main components. A secondary side parallel resonant capacitor C3 is connected to the secondary winding N2, and a secondary side rectifying element Do formed with a high speed switching diode Do1 to a high speed switching diode Do4 is connected to the secondary winding N2. By such a connection mode, it has a function of rectification / smoothing that converts AC power into DC power, and also functions as a secondary parallel resonance circuit.

  The function of rectification / smoothing is obtained by the high-speed switching diode Do1, the high-speed switching diode Do2, and the secondary side smoothing capacitor Co. The secondary side parallel resonance circuit has a resonance frequency fp2 that is a resonance frequency of the secondary side parallel resonance circuit by an inductor L2 that is a leakage inductor generated in the secondary winding N2 of the converter transformer PIT and a secondary side parallel resonance capacitor C3. It is formed so as to be dominant.

  Here, the dominant determination of the resonance frequency fp2 of the secondary side parallel resonance circuit means that the resonance frequency of the secondary side parallel resonance circuit is almost determined by the inductor L2 and the secondary side parallel resonance capacitor C3. is there. For example, the secondary side smoothing capacitor Co also affects the resonance frequency of the secondary side parallel resonance circuit, but the capacitance value of the secondary side smoothing capacitor Co is smaller than the capacitance value of the secondary side parallel resonance capacitor C3. Since it is 1000 times larger, the secondary side smoothing capacitor Co does not greatly affect the resonance frequency of the secondary side parallel resonance circuit.

  The secondary-side rectifier circuit is a so-called full-wave rectifier circuit, and is rectified by a secondary-side rectifier element Do formed by bridge-connecting the high-speed switching diode Do1 to the high-speed switching diode Do2, and a secondary-side smoothing capacitor. Smoothed by Co. The DC output voltage Eo is obtained from both ends of the secondary side smoothing capacitor Co. The DC output voltage Eo is detected by the control circuit 1 to obtain an error voltage which is a difference voltage between a reference potential (not shown) arranged in the control circuit 1 and the DC output voltage Eo, and this error voltage is set to zero. Thus, the control circuit 1 controls the oscillation / drive circuit 2.

  Here, the relationship between the resonance frequency fp1 of the primary side parallel resonance circuit and the resonance frequency fp2 of the secondary side parallel resonance circuit is set so that the resonance frequency hop1 and the resonance frequency hop2 are substantially equal. The resonance frequency fos1 of the primary side series resonance circuit is set to approximately ½ of the resonance frequency fopl of the primary side parallel resonance circuit or the resonance frequency fop2 of the secondary side parallel resonance circuit.

  The configuration from the AC power supply AC side to the secondary side of the switching power supply circuit shown in FIG. 1 will be described in further detail and how the switching power supply circuit operates will be described in order.

  The switching power supply circuit has a common mode filter having an across line capacitor CL1, a common mode choke coil CMC, and an across line capacitor CL2. A plug (not shown) is connected to the across line capacitor CL1 on the input side of the common mode filter via a power switch (not shown) and a fuse (not shown), and this plug is inserted into an outlet connected to the AC power supply AC. As a result, AC power is supplied to the switching power supply circuit. The common mode filter prevents the common mode component of the switching noise generated in response to the switching of the switching element Q1 from flowing out to the AC power supply AC side.

  Each end of the output side of the common mode filter (across line capacitor CL2 side) is connected to the connection point between the anode of the diode Di1 and the cathode of the diode Di3 which are the input side of the primary side rectifying element Di, and the anode of the diode Di2. The diode Di4 is connected to each of the connection points with the cathode. That is, the primary side rectifying element Di is configured by connecting the diodes Di1 to Di4 in a bridge connection. As a result, AC power supplied from the AC power supply AC is converted to DC power, and a positive voltage is obtained from the connection point between the cathode of the diode Di1 and the cathode of the diode Di2, and the anode of the diode Di3 and the anode of the diode Di4 are obtained. Negative voltage is obtained from the connection point.

  The positive side terminal of the primary side smoothing capacitor Ci is connected to the connection point between the cathode of the diode Di1 and the cathode of the diode Di2 which is the output side of the primary side rectifying element Di, and the anode of the diode Di3 and the anode of the diode Di4 Is connected to the negative terminal of the primary smoothing capacitor Ci. A choke coil PCC is connected to the positive side of the primary smoothing capacitor Ci. In this way, a primary side smoothing circuit is formed by the primary side rectifying element Di and the primary side smoothing capacitor Ci, and a voltage Ei which is a DC voltage is obtained from both ends of the primary side smoothing capacitor Ci. .

  The single-ended converter includes a switching element Q1, a primary side parallel resonant capacitor C1, a control circuit 1, an oscillation / drive circuit 2, a choke coil PCC, a converter transformer PIT, a secondary side parallel resonant capacitor C3, and a secondary side rectifier element Do. A secondary side smoothing capacitor Co is provided. This single-ended converter generates, for example, an alternating current having a frequency 1000 times higher than the commercial frequency, and converts the voltage between the primary side and the secondary side via a converter transformer PIT that is a high-frequency transformer. In addition, the primary side and the secondary side are insulated. In this way, by using a very high frequency compared with the commercial frequency, it is possible to reduce the size of the components constituting the single-ended converter including the converter transformer PIT.

  The switching element Q1 is a MOS-FET, and the body diode DD1 is a diode formed in the process of manufacturing the MOS-FET. The body diode DD1 forms a current path that flows from the source to the drain of the MOS-FET.

  The converter transformer PIT insulates the primary side from the secondary side and performs voltage conversion to form inductors L1 and L2. Here, each of inductor L1 and inductor L2 is a leakage inductor formed by loose coupling of primary winding N1 and secondary winding N2 of converter transformer PIT. Here, 0.7 is adopted as the value of the coupling coefficient k between the primary winding N1 and the secondary winding N2. How to form such a leakage inductor will be described with reference to a cross-sectional view of the converter transformer PIT shown in FIG.

  As shown in FIG. 2, the converter transformer PIT used in the switching power supply circuit of the present embodiment includes an EER type core in which an EER type core CR1 and an EER type core CR2 made of a ferrite material are combined so that their magnetic legs face each other. . The primary side and secondary side winding portions are divided so as to be independent from each other, and provided with a bobbin B formed of, for example, resin. Then, the bobbin B around which the primary winding N1 is wound as the primary winding portion and the secondary winding N2 is wound as the secondary winding portion is attached to the EER type core. The secondary winding N1 is wound in one region, the secondary winding N2 and the secondary winding N2 ′ are separated into a winding region different from this one region, and are wound around the central magnetic leg of the EER type core. Will be in a state. Thus, the converter transformer PIT is configured.

  Here, as shown in FIG. 2, since the gap G exists, a part of the magnetic flux leaks from the gap G, and the primary winding N1 and the secondary winding N2 are not linked to both. Magnetic flux is generated, and the primary winding N1 and the secondary winding N2 are loosely coupled to each other.

  In the converter transformer PIT shown in FIG. 2 used in the switching power supply circuit shown in FIG. 1, a gap G of 1.4 mm is formed with respect to the central magnetic leg of the EER type core. As a result, 0.7 was obtained as the value of the coupling coefficient k between the primary side and the secondary side. The gap G is formed by making the central magnetic legs of the EER type core CR1 and the EER type core CR2 shorter than the two outer magnetic legs. The core material was EER-40 (core material name).

  The number of turns of the primary winding N1 was 60T (turns), and the number of turns of the secondary winding N2 was 45T. At this time, the value of the DC output voltage Eo is 175 V (volts).

  Specific constants of each part on the primary side will be described. The capacitance value of the primary side parallel resonance capacitor C1 is 7500 pF (pico farad), the capacitance value of the primary side series resonance capacitor C2 is 0.047 μF (micro farad), and the capacitance value of the secondary side parallel resonance capacitor C3. The value is 0.018 μF. The choke coil uses EER-28 as a core material, and the inductance value of the inductor Lp is 865 μH (micro Henry). With this setting, the value of the resonance frequency fp1 of the primary side parallel resonance circuit is 62.5 kHz (kilo-Hells), and the value of the resonance frequency fop2 of the secondary side parallel resonance circuit is 65.6 kHz, the primary side As the value of the resonance frequency fos1 of the series resonance circuit, 31.5 kHz is obtained.

  The output of the oscillation / drive circuit 2 is connected to the gate of the switching element Q1. The oscillation / drive circuit 2 drives the gate of the switching element Q 1 in accordance with the output from the control circuit 1. By such a connection mode, it is possible to control so that the voltage value of the reference power source provided in the control circuit 1 is equal to the voltage value of the voltage obtained by dividing the DC output voltage Eo. Here, the secondary-side rectifier circuit for obtaining the DC output voltage Eo is a full-wave rectifier circuit, and this full-wave rectifier circuit includes a high-speed switching diode Do1 to a high-speed switching diode Do2 that form a bridge rectifier circuit. It consists of a secondary smoothing capacitor Co.

  FIG. 3 shows operation waveforms of each part of the switching power supply circuit shown in FIG. 1 in the switching period. The load power at this time is 300 W, and the input AC voltage is a value at 100 V. In order from the top of FIG. 3, voltage V1 (see FIG. 1), current IQ1 (see FIG. 1), current I1 (see FIG. 1), current I2 (see FIG. 1), voltage V2 (see FIG. 1) ), Voltage V3 (see FIG. 1), current I3 (see FIG. 1), and current I4 (see FIG. 1).

  FIG. 4 shows operation waveforms of the respective parts of the switching power supply circuit shown in FIG. 1 in the switching period. The load power at this time is 0 W, and the input AC voltage is 100 V. In order from the top in FIG. 3, each of voltage V1, current IQ1, current I1, current I2, voltage V2, voltage V3, current I3, and current I4 is shown.

  FIG. 5 shows power conversion efficiency ηAC → DC with respect to load fluctuation when the load power Po is in the range of 0 W (no load) to 300 W, the frequency fs of the signal for controlling the gate of the switching element Q1, and the switching element Q1 conducting. Each of the values of the period TON, which is a period during which the switching element Q1 is disconnected, and the period TOFF, which is the period during which the switching element Q1 is disconnected, are shown.

  From the above experimental data, the following becomes clear about the switching power supply circuit shown in FIG. The power conversion efficiency ηAC → DC at an AC input voltage VAC of 100 V and a load power of 300 W is 91.5%, and the power conversion efficiency ηAC → DC at a load power of 300 W to 100 W is 90% or more. The value of the period TOFF is constant when the load power is in the range of 300 W to 0 W, and the frequency fs changes in the range of 71.4 kHz to 125 kHz as the period TON changes.

  When the single-ended converter of the first embodiment shown in FIG. 1 is compared with the single-ended converter shown in the background art, the single-ended converter of the first embodiment has the following characteristics.

  In the multi-resonance converter of the background art, it is difficult to perform the ZVS operation in the entire region of 200 W or more, and the load power cannot be set to 200 W or more. However, as in the first embodiment, the resonance frequency fp1 The resonance frequency fop2 is set to be substantially equal to the resonance frequency fos1 of the primary side series resonance circuit, and the resonance frequency fop1 of the primary side parallel resonance circuit is approximately 1 / of the resonance frequency fop2 of the secondary side parallel resonance circuit. By setting to 2, load power of 200 W or more can be handled.

  In the multi-resonance converter of the background art, when the load power is 200 W or more, it is difficult to use a general-purpose control IC because the control range of the switching frequency becomes too wide, but the diagram of the first embodiment In the switching power supply circuit shown in FIG. 1, the load power ranges from 300 W to 0 W, the control range is 54 kHz, and a general-purpose control IC can be used.

  Further, if the withstand voltage of the switching element Q1 is 1800V, the ZVS region can be expanded to all ranges up to 300W or more of load power in both the 100V system and 230V system of the input AC voltage.

  Further, by configuring the switching power supply circuit as a class E switching amplifier, the current flowing from the primary side smoothing capacitor Ci becomes close to direct current, the ripple voltage is reduced, and the generation of noise is reduced.

  Further, the current I2 that is the resonance current flows through the primary winding N1, thereby reducing the unbalance of the current I4 that is the secondary side rectified current.

(Modification of the first embodiment)
A modification of the first embodiment will be described with reference to FIGS. The switching power supply circuit shown in FIG. 6 uses a double-wave rectifier circuit as the secondary rectifier circuit. The secondary winding is composed of a secondary winding N2 and a secondary winding N2 ′ formed with a center tap, and rectification is performed by the high-speed switching diode Do1 and the high-speed switching diode Do2. The secondary parallel resonant circuit is configured by connecting a secondary parallel resonant capacitor C3 to both ends of the secondary winding.

  The switching power supply circuit shown in FIG. 7 uses a half-wave rectifier circuit as a secondary side rectifier circuit. Rectification is performed by the high-speed switching diode Do1.

(Switching power supply circuit of the second embodiment)
The switching power supply circuit according to the second embodiment includes an active clamp circuit in addition to the switching power supply circuit according to the first embodiment described above. The basic configuration as a voltage resonant single-ended multiple resonant converter using a class E switching amplifier has the following configuration similar to that of the first embodiment. The multiple resonance converter includes a primary side resonance circuit (primary side resonance circuit) and a secondary side resonance circuit (secondary side resonance circuit). The primary side resonance circuit has a resonance frequency by a primary side parallel resonance circuit whose resonance frequency is governed by a primary side parallel resonance capacitor and an inductor of a choke coil, a primary side series resonance capacitor, and an inductor generated in the primary winding. And a primary-side series resonant circuit that is controlled by. The secondary side resonance circuit includes a secondary side parallel resonance circuit whose resonance frequency is governed by a secondary side parallel resonance capacitor and an inductor generated in the secondary winding. Here, the coupling coefficient between the primary winding and the secondary winding of the converter transformer is a loose coupling of 0.8 or less, so that an inductor serves as a leakage inductor in each of the primary winding and the secondary winding. Will occur. The resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set to be approximately equal, and the resonance frequency of the primary side series resonance circuit is the resonance frequency of the primary side parallel resonance circuit or It is set to approximately ½ of the resonance frequency of the secondary side parallel resonance circuit.

  The active clamp circuit is configured as a series connection circuit of a capacitor for clamping and an auxiliary switching element, and the active clamp circuit is connected to both ends of the choke coil. The auxiliary switching element is disconnected when the switching element is turned on, and is turned on when the switching element is cut off (this operation is referred to as complementary hereinafter and used). . By such an operation, the auxiliary switching element does not become a load when the switching element is conducted, and when the switching element is disconnected, the voltage generated at both ends of the switching element is clamped to reduce the withstand voltage of the switching element. And can. Control of conduction and disconnection of the auxiliary switching element is performed by a voltage generated in a clamp winding applied to the choke coil.

  A switching power supply circuit according to the second embodiment will be described with reference to FIG. In addition, the same code | symbol is attached | subjected to the component which has the structure similar to 1st Embodiment, and description about the part is abbreviate | omitted.

  The choke coil PCC used in the switching power supply circuit of the embodiment shown in FIG. 8 includes a winding Np and a clamping winding Ng. Choke coil PCC has substantially the same configuration as converter transformer PIT shown in FIG. That is, the choke coil PCC includes an EER type core in which an EER type core CR1 and an EER type core CR2 are combined so that their magnetic legs face each other, and includes a bobbin B. The bobbin B includes a winding Np and a clamp. A winding Ng is wound, and a gap G is formed with respect to the central magnetic leg of the EER type core. This prevents magnetic saturation and functions as a choke coil.

  The winding Np and the clamping winding Ng of the choke coil PCC are formed as windings wound in the same direction, and the connection point between the winding Np and the clamping winding Ng is the primary of the converter transformer PIT. It is connected to one end of the winding N1. Such a connection mode depends on the polarity of the control voltage of the auxiliary switching element Q2. In the embodiment shown in FIG. 8, an N-channel MOS-FET is used. It is what. The polarities of the winding Np and the clamping winding Ng are selected so that the switching element Q1 and the auxiliary switching element Q2 operate in a complementary manner. The winding Np functions as an inductor Lp as in the first embodiment.

  Hereinafter, the active clamp circuit which is different from the first embodiment of the second embodiment will be described in more detail.

  The choke coil PCC uses EER-28 (core material name) as the core material, and the value of the inductance of the inductor Lp generated in the winding Np is 865 μH. The clamp winding Ng is 2T.

  The auxiliary switching element Q2 is a MOS-FET and has a body diode DD2. The voltage generated in the clamp winding Ng is divided by the resistors Rg1 and Rg2, and applied to the gate of the auxiliary switching element Q2. Here, the voltage dividing ratio between the resistor Rg1 and the resistor Rg2 can be adjusted, and the period during which the auxiliary switching element Q2 is conducted can be adjusted within the range of the period during which the switching element Q1 is conducted. During the period in which the auxiliary switching element Q2 is conductive, both ends of the choke coil PCC are short-circuited via the clamping capacitor C4, whereby the voltage when the switching element Q1 is disconnected can be clamped. The value of the capacitance of the clamping capacitor C4 was 0.1 μF.

  In the active clamp circuit configured as described above, the switching loss of the auxiliary switching element Q2 increases as the AC input voltage VAC increases. As shown in the embodiment of FIG. When configured as a half-wave rectifier circuit, the conduction period of the auxiliary switching element Q2 changes, so the period during which the switching element Q1 is disconnected (period TOFF) is reduced as the load power decreases. For this reason, the decrease in the power conversion efficiency ηAC → DC due to the increase in the AC input voltage VAC is reduced, and the power conversion efficiency ηAC → DC is maintained at a substantially constant value even when the AC input voltage VAC is changed to a wide range. be able to. Further, by adding an active clamp circuit, a better ZVS operation can be obtained even when the AC input voltage VAC is changed to a wide range.

  The constants and specifications of each part in the second embodiment will be described below. The core material of the converter transformer PIT was EER-40, the gap was 1.4 mm, the number of turns of the primary winding N1 was 60T, the number of turns of the secondary winding N2 was 5T, and the value of the coupling coefficient k was 0.7. The value of the DC output voltage Eo at this time is 175V.

  The capacitance value of the primary side parallel resonance capacitor C1 was 2200 pF, the capacitance value of the primary side series resonance capacitor C2 was 0.047 μF, and the capacitance value of the secondary side parallel resonance capacitor C3 was 0.018 μF.

  A 10A / 900V MOS-FET was used as the switching element Q1, and a 5A / 900V MOS-FET was used as the auxiliary switching element Q2. Further, the range of the AC input voltage VAC that can be supported by this switching power supply circuit is a range of 90V to 288V (effective value), and the range of the load power is a range of 300W to 0W.

  FIG. 9 shows the operation waveform of each part in the switching period. The load power at this time is 300 W, and the input AC voltage is a value at 100 V. In order from the top of FIG. 9, voltage V1 (see FIG. 8), current IQ1 (see FIG. 8), current IQ2 (see FIG. 8), current I1 (see FIG. 8), current I2 (see FIG. 8) ), Voltage V2 (see FIG. 8), voltage V3 (see FIG. 8), current I3 (see FIG. 8), and current I4 (see FIG. 8).

  FIG. 10 shows an operation waveform of each part in the switching period. The load power at this time is 300 W, and the input AC voltage is a value at 230 V. In the order from the top of FIG. 10, each of voltage V1, current IQ1, current IQ2, current I1, current I2, voltage V2, voltage V3, current I3, and current I4 is shown.

  FIG. 11 shows the power conversion efficiency ηAC → DC with respect to the load fluctuation when the value of the load power Po is in the range of 0 W (no load) to 300 W, the frequency fs of the signal for controlling the gate of the switching element Q1, and the switching element Q1 conducting. Each of the period TON, which is a period during which the switching element Q1 is disconnected, and the period TOFF, which is a period during which the switching element Q1 is disconnected, are shown. Each data when the AC input voltage VAC is 100V is indicated by a solid line, and each data when the AC input voltage VAC is 230V is indicated by a broken line.

  From the above experimental data, the following becomes clear about the switching power supply circuit shown in FIG. When the AC input voltage VAC is 100 V and the load power is 300 W, the power conversion efficiency ηAC → DC is 91.2%. Further, the power conversion efficiency ηAC → DC when the AC input voltage VAC is 230 V and the load power is 300 W is 91.8%.

  Regarding the frequency fs of the signal for controlling the switching element Q1, the range of the frequency fs is 65.8 kHz to 125 kHz when the AC input voltage VAC is 100 V in the range from 300 W to 0 W as the load power, and the AC input voltage VAC Is 230V, the range of the frequency fs is 103 kHz to 169.5 kHz. Within such a frequency fs range, even if the AC input voltage VAC is in the range of 90 V to 230 V, it falls within the controllable frequency fs range of a general-purpose IC that is normally used for configuring an oscillation / drive circuit. Therefore, a general-purpose IC can be used as the oscillation / drive circuit 2 in the second embodiment.

  When the single-ended converter of the second embodiment shown in FIG. 8 is compared with the single-ended converter shown in the background art, the single-ended converter of the second embodiment has the following characteristics.

  In the multi-resonance converter of the background art, it is difficult to perform the ZVS operation in the entire region of 200 W or more, and the load power cannot be set to 200 W or more. However, as in the first embodiment, the resonance frequency fp1 The resonance frequency fop2 is set to be substantially equal to the resonance frequency fos1 of the primary side series resonance circuit, and the resonance frequency fop1 of the primary side parallel resonance circuit is approximately 1 / of the resonance frequency fop2 of the secondary side parallel resonance circuit. By setting to 2, load power of 200 W or more can be handled.

  In the multi-resonance converter of the background art, it is difficult to make the so-called wide range in which the input AC voltage corresponds to both the 100 V system and the 230 V system. However, as in the second embodiment, a voltage clamp circuit is provided. Therefore, it is possible to easily cope with a wide range. In this case, even when the rectifier circuit on the secondary side is a half-wave rectifier circuit and the conduction period of the auxiliary switching element Q2 changes and the AC input voltage VAC is changed to a wide range, the power conversion efficiency ηAC → DC is It can be kept at a substantially constant value. Thus, by providing a voltage clamp circuit, it can be set as what has a more favorable ZVS characteristic.

  Further, by configuring the switching power supply circuit as a class E switching amplifier, the current flowing from the primary side smoothing capacitor Ci becomes close to direct current, the ripple voltage is reduced, and the generation of noise is reduced.

(Modification of the second embodiment)
A modification of the second embodiment will be described with reference to FIGS. The switching power supply circuit shown in FIG. 12 uses a double-wave rectifier circuit as a secondary side rectifier circuit. The secondary winding is composed of a secondary winding N2 and a secondary winding N2 ′ formed with a center tap, and rectification is performed by the high-speed switching diode Do1 and the high-speed switching diode Do2. The secondary parallel resonant circuit is configured by connecting a secondary parallel resonant capacitor C3 to both ends of the secondary winding.

  The switching power supply circuit shown in FIG. 13 uses a full-wave rectifier circuit as a secondary side rectifier circuit. The secondary-side AC power is rectified by the secondary-side rectifying element Do configured by bridge-connecting the high-speed switching diode Do1 to the high-speed switching diode Do4, and smoothed by the secondary-side smoothing capacitor Co. The secondary side parallel resonant circuit is configured by connecting a secondary side parallel resonant capacitor C3 to both ends of the secondary winding N2.

(Switching power supply circuit of 3rd Embodiment thru | or 5th Embodiment)
The switching power supply circuit according to the third to fifth embodiments further includes a power factor correction circuit in addition to the switching power supply circuit according to the second embodiment described above. The voltage resonant single-ended multiple resonant converter using the class E switching amplifier has the following configuration similar to that of the first embodiment. That is, the multiple resonance converter includes a primary side resonance circuit (primary side resonance circuit) and a secondary side resonance circuit (secondary side resonance circuit). The primary side resonance circuit has a resonance frequency by a primary side parallel resonance circuit whose resonance frequency is governed by a primary side parallel resonance capacitor and an inductor of a choke coil, a primary side series resonance capacitor, and an inductor generated in the primary winding. And a primary-side series resonant circuit that is controlled by. The secondary side resonance circuit includes a secondary side parallel resonance circuit whose resonance frequency is governed by a secondary side parallel resonance capacitor and an inductor generated in the secondary winding. Here, the coupling coefficient between the primary winding and the secondary winding of the converter transformer is a loose coupling of 0.8 or less, so that an inductor serves as a leakage inductor in each of the primary winding and the secondary winding. Will occur. The resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set to be approximately equal, and the resonance frequency of the primary side series resonance circuit is the resonance frequency of the primary side parallel resonance circuit or It is set to approximately ½ of the resonance frequency of the secondary side parallel resonance circuit.

  The active clamp circuit is configured as a series connection circuit of a capacitor for clamping and an auxiliary switching element, and the active clamp circuit is connected to both ends of the choke coil. The auxiliary switching element operates complementarily to the switching element. By such an operation, the auxiliary switching element does not become a load when the switching element is conducted, and when the switching element is disconnected, the voltage generated at both ends of the switching element is clamped to reduce the withstand voltage of the switching element. And can. Control of conduction and disconnection of the auxiliary switching element is performed by a voltage generated in a clamp winding applied to the choke coil.

  Here, in the third embodiment, the power factor correction circuit flows between the primary side rectifier circuit and the voltage resonance single-ended multiple resonance converter using the class E switching amplifier and flows to the primary side series resonance circuit. A power regeneration type power factor correction circuit that feeds back a resonance current to a primary smoothing capacitor is used.

  In the fourth embodiment, the power factor correction circuit is a voltage regeneration type power factor correction circuit that feeds back a resonance voltage generated in a winding wound around a choke coil to a primary side smoothing capacitor.

  In the fifth embodiment, the power factor correction circuit causes the resonance current flowing in the primary side series resonance circuit to flow in the primary winding of the voltage feedback transformer, and generates the resonance voltage generated in the secondary winding of the voltage feedback transformer. It is a voltage regeneration type power factor correction circuit that feeds back to the primary side smoothing capacitor.

  (Switching power supply circuit of the third embodiment)

  A switching power supply circuit according to a third embodiment will be described with reference to FIG. The switching power supply circuit of the third embodiment has the same configuration and operation as the class E switching amplifier in the switching power supply circuit of the first embodiment described above, and the active clamp circuit in the switching power supply circuit of the second embodiment described above. Since the components have the same configuration and action, the same components as those in the first embodiment or the second embodiment are denoted by the same reference numerals, and the description thereof is omitted.

  Since the power factor correction circuit is provided in the third embodiment, this switching power supply circuit can have a good power factor. This power factor correction circuit is configured to feed back the resonance current flowing through the primary side series resonance capacitor C2 to the primary side smoothing capacitor Ci. The power factor correction circuit of the third embodiment is composed of three components, that is, a power factor improving inductor Lo, a power factor improving fast diode Dr, and a filter capacitor CN.

  One winding end of the power factor improving inductor Lo and one polar end (cathode) of the power factor improving fast diode Dr are connected, and this connection point is connected to the primary side series resonance capacitor C2. Here, in the first embodiment and the second embodiment, the primary side series resonant capacitor C2 is connected to the source of the switching element Q1, but in the third embodiment, the primary side series resonant capacitor C2 is As described above, the power factor improving inductor Lo and the cathode of the power factor improving fast diode Dr are connected to the connection point.

  The other end of the power factor improving inductor Lo is connected to a connection point between the winding Np of the choke coil PCC and the primary side smoothing capacitor Ci, and the connection point and the anode of the power factor improving fast diode Dr. Between these, a filter capacitor CN is connected. The output side of the primary side rectifying element Di is connected to the connection point between the anode of the power factor improving fast diode Dr and the filter capacitor CN.

  With the connection mode described above, the current flowing through the primary side series resonance circuit is fed back to the primary side smoothing capacitor Ci. That is, one current flowing through the primary side series resonant circuit flows from the anode to the cathode of the power factor improving fast diode Dr. Here, the high frequency component of the one current flows through the primary side smoothing capacitor Ci and the filter capacitor CN, and the low frequency component of the one current flows through the primary side rectifying element Di to increase the flow angle and increase the power factor. Improve. The other current flowing through the primary side series resonance circuit flows through the power factor improving inductor Lo and the primary side smoothing capacitor Ci, and power regeneration is performed. That is, the resonance current flowing in the primary side series resonance circuit flows from the primary winding N1 through the path of the primary side series resonance capacitor C2 and the power factor improving inductor Lo, and the cathode potential of the power factor improving fast diode Dr is The power factor improving high-speed diode Dr is oscillated at the switching cycle to intermittently flow the current I1 into the primary side smoothing capacitor Ci, and the conduction angle of the AC input current IAC is expanded to improve the power factor. .

  Here, the capacitance value of the series connection circuit of the primary side smoothing capacitor Ci and the filter capacitor CN through which one current flowing through the primary side series resonance circuit passes is equal to the capacitance value of the primary side series resonance capacitor C2. The inductance value of the power factor improving inductor Lo through which the other current flowing through the primary side series resonance circuit passes is much larger than the inductance value of the inductor L1 generated in the primary winding N1. Since it is very small, the change in the resonance frequency of the primary side series resonance circuit by providing the power factor correction circuit is greatly different from that in the first embodiment and the second embodiment in which the power factor improvement circuit is not provided. Does not occur.

  Further, in the first embodiment and the second embodiment, the primary side rectifier element Di is also used in the third embodiment in the same manner as the current from the primary side rectifier element Di flows into the primary side smoothing capacitor Ci. Current flows from the anode of the power factor improving fast diode Dr to the primary side smoothing capacitor Ci via the cathode and the power factor improving inductor Lo, and the choke coil PCC connected to the primary side smoothing capacitor Ci. Electric power is supplied to the winding Np.

  In the power factor correction circuit, the optimum value of the power factor improvement effect can be adjusted by adjusting the inductance value of the power factor improvement inductor Lo. The optimum point of the power factor improvement effect in this case is to select the power factor so that the value of the power factor increases at the intermediate load than that at the maximum load.

  Hereinafter, constants of each part of the switching power supply circuit according to the third embodiment will be described. The switching power supply circuit according to the third embodiment can handle an AC input voltage VAC from 90 V to 288 V, and a load power range from 300 W to 0 W. EER-40 is used as the core material of the converter transformer PIT, and the converter transformer PIT forms a gap G of 1.4 mm with respect to the central magnetic leg of the EER core, and the primary winding N1 and the secondary winding N2 As a value of the coupling coefficient k, 0.7 is adopted. The capacitance value of the primary side parallel resonant capacitor C1 was 2200 pF, and the capacitance value of the primary side series resonant capacitor C2 was 0.047 μF. The capacitance value of the secondary parallel resonant capacitor C3 was 0.018 μF, and the capacitance value of the clamping capacitor C4 was 0.1 μF. The value of the filter capacitor CN was 1 μF, the capacitance value of the primary smoothing capacitor was 1000 μF, and the capacitance value of the secondary smoothing capacitor was 1000 μF. The inductance value of the inductor Lp was 865 μH, and the inductance value of the power factor improving inductor Lo was 54 μH.

  FIG. 15 shows operation waveforms of respective parts in the switching cycle of the switching power supply circuit according to the third embodiment. This is an operation waveform when the load power is 300 W, the input AC voltage is 100 V, and the DC output voltage Eo is 175 V. In order from the top of FIG. 15, an AC input voltage VAC (see FIG. 14), an AC input current IAC (see FIG. 14), a voltage V1 (see FIG. 14), a current I1 (see FIG. 14), and a voltage V2 ( 14) and current I2 (see FIG. 14).

  FIG. 16 shows operation waveforms of respective parts in the switching cycle of the switching power supply circuit according to the third embodiment when the input AC voltage is 230V. This is an operation waveform when the load power is 300 W and the DC output voltage Eo is 175V. Each of the AC input voltage VAC, AC input current IAC, voltage V1, current I1, voltage V2, and current I2 is shown in order from the top of FIG.

  15 and 16, the inductance value of the power factor improving inductor Lo and the number of turns of the primary winding N1 are adjusted so that the waveform of the AC input current IAC shown in FIG. As a result, the waveform of the AC input current IAC is sinusoidal and the value of the power factor PF is improved in a region where the load is light to an intermediate load power.

  FIG. 17 shows each of the power conversion efficiency ηAC → DC, the power factor PF value, and the voltage Ei value with respect to the load fluctuation when the value of the load power Po is in the range of 0 W (no load) to 300 W. Each data when the AC input voltage VAC is 100V is indicated by a solid line, and each data when the AC input voltage VAC is 230V is indicated by a broken line.

  From the above experimental data, the following becomes clear about the switching power supply circuit shown in FIG. When the AC input voltage VAC is 100 V and the load power is 300 W, the power factor PF is 0.87, the power conversion efficiency ηAC → DC is 91.0%, and the region where the value of the power factor PF is 0.75 or more is the load power. It is in the range of 300W to 25W. In addition, the power factor PF at AC input voltage VAC 230V, load power 300W is 0.80, power conversion efficiency ηAC → DC is 91.8%, and the region where the value of power factor PF is 0.75 or more is a load The power is in the range of 300W to 75W. The value of the power factor PF satisfies the Japanese power supply harmonic distortion regulation value class A when the AC input voltage VAC is 100V, and when the AC input voltage VAC is 230V, It satisfies European power supply harmonic distortion regulation value class A.

  When the single-ended converter of the third embodiment shown in FIG. 14 is compared with the single-ended converter shown in the background art, the single-ended converter of the third embodiment has the following characteristics.

  In the multi-resonance converter of the background art, it has been difficult to make the so-called wide range in which the input AC voltage corresponds to both the 100V system and the 230V system. However, as in the third embodiment, the voltage clamp circuit and the power regeneration By providing a power factor correction circuit of the system, it was possible to secure a good power factor even in a wide range and to clear the harmonic distortion regulations in Japan and Europe. Here, the region where the harmonic distortion regulation value is cleared includes a range of 300 W to 0 W as the load power when the value of the AC input voltage VAC is 100 V, and a range of 300 W to 75 W as the load power when the value of the AC input voltage VAC is 230 V. Range and wide range.

  In the power factor improvement circuit of the background art, the number of parts constituting the power factor improvement circuit is five, but in the power factor improvement circuit of the third embodiment, the number of parts constituting the power factor improvement circuit is the power factor improvement. It can be reduced to three points: the inductor Lo, the power factor improving high speed diode Dr, and the filter capacitor CN.

  (Switching power supply circuit of the fourth embodiment)

  A switching power supply circuit according to the fourth embodiment will be described with reference to FIG. The switching power supply circuit of the fourth embodiment has the same configuration and operation as the class E switching amplifier in the switching power supply circuit of the first embodiment described above, and the active clamp circuit in the switching power supply circuit of the second embodiment described above. Since the components have the same configuration and action, the same components as those in the first embodiment or the second embodiment are denoted by the same reference numerals, and the description thereof is omitted. The switching power supply circuit of the fourth embodiment includes a power factor correction circuit as in the switching power supply circuit of the third embodiment, but the power factor correction circuit of the switching power supply circuit of the fourth embodiment is: This has a configuration different from that of the power factor correction circuit in the switching power supply circuit of the third embodiment.

  That is, the power factor correction circuit according to the fourth embodiment includes a power factor improving inductor Lo, a power factor improving high speed diode Dr, a filter capacitor CN, and a winding Ns wound around the choke coil PCC. The power factor correction circuit is configured to feed back a voltage corresponding to the resonance current flowing through the primary side series resonance capacitor C2 to the primary side smoothing capacitor Ci.

  A specific connection mode of the power factor correction circuit according to the fourth embodiment will be described below. One winding end of the power factor improving inductor Lo and one polar end (cathode) of the power factor improving fast diode Dr are connected, and the other end of the power factor improving inductor Lo is the winding of the choke coil PCC. It is connected to one end of Ns. The other polar end (anode) of the power factor improving fast diode Dr is connected to the filter capacitor CN, and the other end of the filter capacitor CN is connected to the other winding end of the winding Ns of the choke coil PCC. It is connected to the connection point with the primary side smoothing capacitor Ci. The output side of the primary side rectifying element Di is connected to the connection point between the anode of the power factor improving fast diode Dr and the filter capacitor CN.

  Here, the winding Ns of the choke coil PCC is formed as a wound winding connected to the winding Np of the choke coil PCC, and a voltage corresponding to the current flowing through the primary side series resonance circuit is applied to the winding Ns. appear. A current flows from the anode to the cathode of the power factor improving fast diode Dr by the voltage of one polarity generated in the winding Ns. Here, the high frequency component of the one current flows through the primary side smoothing capacitor Ci and the filter capacitor CN, and the low frequency component of the one current flows through the primary side rectifying element Di to increase the flow angle and increase the power factor. Improve. That is, the voltage of one polarity generated in the winding Ns is voltage-feedbacked to the primary side smoothing capacitor Ci. Further, no current flows depending on the voltage of the other polarity generated in the winding Ns.

  (Switching power supply circuit of 5th Embodiment)

  A switching power supply circuit according to a fifth embodiment will be described with reference to FIG. The switching power supply circuit of the fifth embodiment has the same configuration and operation as the class E switching amplifier in the switching power supply circuit of the first embodiment described above, and the active clamp circuit in the switching power supply circuit of the second embodiment described above. Since the components have the same configuration and action, the same components as those in the first embodiment or the second embodiment are denoted by the same reference numerals, and the description thereof is omitted. The switching power supply circuit according to the fifth embodiment includes a power factor correction circuit as in the switching power supply circuit according to the fourth embodiment and the switching power supply circuit according to the fourth embodiment, but the switching power supply according to the fifth embodiment. The circuit power factor correction circuit has a different configuration from the power factor correction circuit in the switching power supply circuit of the third embodiment and the switching power supply circuit of the fourth embodiment.

  The power factor correction circuit of the fifth embodiment is configured to feed back a voltage corresponding to the resonance current flowing through the primary side series resonance capacitor C2 to the primary side smoothing capacitor Ci. The power factor correction circuit according to the fifth embodiment includes a voltage feedback transformer VFT, a power factor improving fast diode Dr, and a filter capacitor CN.

  A series circuit of a power factor improving fast diode Dr and a filter capacitor CN is connected to the primary winding N12 of the voltage feedback transformer VFT, and is connected to a connection point between the anode of the power factor improving fast diode Dr and the filter capacitor CN. Is connected to the output side of the primary side rectifying element Di, and the primary side smoothing capacitor Ci is connected to the other end of the filter capacitor CN. The primary side series resonance capacitor C2 is connected to one winding end of the primary winding N11 of the voltage feedback transformer VFT, and the other winding end of the primary winding N11 of the voltage feedback transformer VFT is the primary side smoothing. The capacitor Ci is connected.

  Here, in the first embodiment and the second embodiment, the primary side series resonant capacitor C2 is connected to the source of the switching element Q1, but in the fifth embodiment, the primary side series resonant capacitor C2 is As described above, it is connected to one end of the primary winding N11 of the voltage feedback transformer VFT, and the primary side series resonant capacitor C2 is not connected to the source of the switching element Q1.

  With such a connection mode, a resonance current flowing through the primary side series resonant capacitor C2, that is, a voltage corresponding to a current flowing through the primary winding N11 of the voltage feedback transformer VFT is applied to the primary winding N12 of the voltage feedback transformer VFT. appear. The current flows from the anode to the cathode of the power factor improving fast diode Dr by the voltage of one polarity generated in the primary winding N12 of the voltage feedback transformer VFT. Here, the high frequency component of the one current flows through the primary side smoothing capacitor Ci and the filter capacitor CN, and the low frequency component of the one current flows through the primary side rectifying element Di to increase the flow angle and increase the power factor. Improve. Here, in the fifth embodiment, the turn ratio of the primary winding N11 of the voltage feedback transformer VFT and the secondary winding N12 of the voltage feedback transformer VFT is added as a new degree of design freedom. By setting the ratio appropriately, a good power factor improvement characteristic can be obtained.

  In addition, this invention is not limited to embodiment mentioned above, Embodiment can be changed as needed.

It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a figure which shows the structure of the converter transformer of embodiment. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment by the period of switching when it is the maximum load electric power. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment by the period of switching when there is no load. It is a figure which shows each of the power conversion efficiency with respect to the load fluctuation | variation of the switching power supply circuit of embodiment, the frequency of the signal which controls a switching element, the conduction | electrical_connection period of a switching element, and the cutting | disconnection period. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment by the period of switching when the maximum load electric power and the input AC voltage are 100V. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment by the period of switching at the time of maximum load electric power and the input alternating voltage 230V. It is a figure which shows each of the power conversion efficiency with respect to the load fluctuation | variation of the switching power supply circuit of embodiment, the frequency of the signal which controls a switching element, the conduction | electrical_connection period of a switching element, and the cutting | disconnection period. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment by the period of an input AC voltage when the maximum load electric power and the input AC voltage are 100V. It is a figure which shows the waveform of the principal part of the switching power supply circuit of embodiment by the period of an input AC voltage at the time of maximum load electric power and the input AC voltage 230V. It is a figure which shows each of the power conversion efficiency with respect to the load fluctuation | variation of the switching power supply circuit of embodiment, the frequency of the signal which controls a switching element, the conduction | electrical_connection period of a switching element, and the cutting | disconnection period. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit of background art. It is a figure which shows the waveform of the principal part of the switching power supply circuit of background art by the period of switching. It is a circuit diagram which shows the structural example of the principal part of the switching power supply circuit of background art. It is a figure which shows the waveform of the principal part of the switching power supply circuit of background art by the period of switching. It is a figure which shows the DC output voltage with respect to the switching frequency of the switching power supply circuit shown in background art. It is a figure which shows the DC output voltage with respect to the switching frequency of the switching power supply circuit shown in background art. It is a circuit diagram which shows the structural example of the switching power supply circuit of background art. It is a figure which shows each of the power conversion efficiency with respect to the load fluctuation | variation of the switching power supply circuit of a background art, the voltage of a primary side smoothing capacitor, and a power factor. It is a figure which shows each of the power conversion efficiency with respect to the input alternating voltage of the switching power supply circuit of background art, the voltage of a primary side smoothing capacitor, and a power factor. It is a drawing substitute photograph which shows the waveform of the principal part in case the load electric power of the switching power supply circuit shown to background art is comparatively large with the period of an input AC voltage. It is a drawing substitute photograph which shows the waveform of the principal part in case the load electric power of the switching power supply circuit shown to background art is comparatively small with the period of an input AC voltage.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, AC AC power supply, B bobbin, C1 primary side parallel resonance capacitor, C2 primary side series resonance capacitor, C3 secondary side parallel resonance capacitor, C4 capacitor for clamping, Ci primary side Smoothing capacitor, CL1, CL2 Across line capacitor, CMC common mode choke coil, CN filter capacitor, Co primary side smoothing capacitor, CR1, CR2 EER core, DD1, DD2 body diode, Di primary side rectifier, Di1, Di2 , Di3, Di4 diode, Do secondary side rectifier, Do1, Do2, Do3, Do4 fast switching diode, Dr power factor improving fast diode, G gap, IAC AC input current, L1, L2, Lp inductor, Lo power factor Inda for improvement N1, N11 primary winding, N12, N2, N12 secondary winding, N3 tertiary winding, Ng clamping winding, Np, Ns winding, PCC choke coil, PIT converter transformer, Q1 switching element, Q2 Auxiliary switching element, Rg1, Rg2 resistance

Claims (3)

A switching power supply circuit for obtaining a DC output voltage of a predetermined voltage from an AC power source,
A primary-side rectifier circuit comprising a primary-side rectifying element and a primary-side smoothing capacitor for converting AC power from the AC power source into primary-side DC power;
The primary side DC power is supplied to one winding end of the primary winding through the choke coil, and the secondary winding is magnetically loosely coupled to the primary winding and the primary winding. A converter transformer having windings;
A switching element connected to the other winding end of the primary winding;
An oscillation / drive circuit that turns on or off the switching element;
A primary-side parallel resonant capacitor that forms a primary-side parallel resonant circuit together with the choke coil and an inductor generated in the primary winding;
A primary-side series resonant capacitor that forms a primary-side series resonant circuit with an inductor generated in the primary winding;
A secondary-side rectifier circuit that obtains a DC output voltage by converting AC power generated in the secondary winding into DC power;
A control circuit for supplying a control signal such that the value of the DC output voltage is a predetermined value to the oscillation / drive circuit;
A secondary side parallel resonant capacitor that forms a secondary side parallel resonant circuit together with an inductor generated in the secondary winding;
The resonance frequency of the primary side parallel resonance circuit and the resonance frequency of the secondary side parallel resonance circuit are set to be substantially equal, and the resonance frequency of the primary side series resonance circuit is the resonance frequency of the primary side parallel resonance circuit. A switching power supply circuit, characterized in that the frequency or the resonant frequency of the secondary side parallel resonant circuit is set to approximately ½.   The switching power supply circuit according to claim 1, further comprising a series connection circuit of a clamping capacitor and an auxiliary switching element for clamping a voltage applied to both ends of the switching element when the switching element is disconnected.   2. The switching according to claim 1, further comprising a power factor correction circuit that improves a power factor by feeding back a resonance current flowing through the primary side series resonance circuit or a voltage corresponding to the resonance current to the primary side smoothing capacitor. Power supply circuit.

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