JP2007143275A - Rotor angle estimating method and controller of dc brushless motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a rotor angle estimating method of a DC brushless motor for suppressing a fluctuation in a level of a phase difference estimate value corresponding to a phase difference between an estimate value and an actual value of a rotor angle and accurately estimating the rotor angle, and a controller of the DC brushless motor. <P>SOLUTION: Periodic signals Vh<SB>d</SB>, Vh<SB>q</SB>wherein the sum of output voltages becomes zero in three control cycles are superimposed on drive voltages Vfb<SB>d</SB>, Vfb<SB>q</SB>due to a current feedback control. The constant drive voltage Vfb is maintained in three control cycles. A sine reference value Vs<SB>dq</SB>, a cosine reference value Vc<SB>dq</SB>and an inductance reference value L0 are calculated by using first differential currents (dI(0), dI(1), dI(2)) in each control cycle (STEP34). A normalization processing is implemented so as to divide the sine reference value Vs<SB>dq</SB>by the inductance reference value L0, and a phase difference reference value θ<SB>E1</SB>corresponding to the phase difference θ<SB>e</SB>between the estimate value and the actual value of the rotor angle is calculated (STEP35). The estimate value θ^ of the rotor angle is updated by using the phase difference θ<SB>E1</SB>(STEP40). <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a method and apparatus for estimating a rotor angle of a DC brushless motor by superimposing a high-frequency voltage on a driving voltage of the motor.

  In order to obtain a desired output torque from the DC brushless motor, it is necessary to apply a drive voltage to the armature at an appropriate phase corresponding to the electrical angle of the rotor having the magnetic pole (hereinafter referred to as the rotor angle). A method for detecting a rotor angle from a change in phase current when a high-frequency voltage is superimposed on a driving voltage of the motor so as to reduce the cost of the DC brushless motor and the driving device by omitting a position detection sensor for detecting the rotor angle. Has been proposed (see, for example, Patent Document 1 and Patent Document 2).

  In the previous application (Japanese Patent Application No. 2004-304492), the inventors of the present application also perform a predetermined control cycle including two or more control cycles when performing feedback control of the conduction current of the DC brushless motor by so-called dq vector conversion. A first-order difference of the motor d-axis current and q-axis current between adjacent control cycles is calculated by superimposing a periodic signal with the sum of output voltages in the cycle being zero on the motor drive voltage, and the first-order difference is calculated. Has proposed a method for calculating an estimated value of the rotor angle.

In this method, a sine reference value and a cosine reference value corresponding to a double angle of the phase difference between the actual value of the rotor angle and the estimated value are calculated using the calculated value of the first-order difference between the d-axis current and the q-axis current. . Then, a phase difference estimated value corresponding to a phase difference between the estimated value of the rotor angle and the actual value is calculated using the sine reference value, and the actual value and the estimated value of the rotor angle are calculated using the phase difference estimated value. The estimated value of the rotor angle is updated so as to eliminate the phase difference.
JP 2000-125589 A JP 2002-262592 A

  As described above, when the phase difference estimated value is calculated by using the sine reference value, the inventors of the present application have the phase difference estimated value extremely fluctuated due to the phase difference between the actual value and the estimated value of the rotor angle. I found out that If the phase difference estimated level fluctuates in this way, if the rotor angle is estimated based on the phase difference estimated value, an error in the estimated value with respect to the actual value of the rotor angle increases, and the estimated value is used. When the drive voltage is determined, there is a risk that the motor will step out.

  Accordingly, the present invention provides a rotor angle of a DC brushless motor that can accurately estimate the rotor angle by suppressing fluctuations in the level of the phase difference estimated value according to the phase difference between the estimated value of the rotor angle and the actual value. An object is to provide an estimation method and a control device for a DC brushless motor.

First, before explaining the present invention, a basic concept which is a premise of the present invention will be described with reference to FIG. FIG. 1 schematically shows the configuration of a DC brushless motor 1. When a salient pole type rotor 2 is used, three-phase (U, V, W) armatures 3, 4, 5 of the motor 1 are used. The relationship between the voltages V _u , V _v , and V _w applied to and the flowing currents I _u , I _v , and I _w is expressed by the following equation (13).

Where V _u , V _v , V _w : phase voltages, I _u , I _v , I _w : phase currents, r: phase resistance, Ke: induced voltage constant, l: armature self-inductance, m: electric machine Mutual inductance of the child, θ: rotor angle, ω _re : angular velocity.

  When the number of rotations of the motor 1 is almost zero, the influence of changes in the induced voltage and the rotor angle is small, and the voltage due to the phase resistance is negligibly small, the above equation (13) is approximated by the following equation (14).

  Moreover, the following formulas (15) and (16) are obtained by transforming the above formula (14) into formulas based on interphase current and interphase voltage.

Next, a so-called dq conversion model for converting the motor 1 into an equivalent circuit having a q-axis armature on the q-axis that is the magnetic flux direction of the field and a d-axis armature on the d-axis orthogonal to the q-axis is used. The relationship between the d-axis voltage V _d ^ and the q-axis voltage V _q ^ converted using the estimated rotor angle θ ^ and the d-axis current I _d ^ and the q-axis current I _q ^ Is approximately zero and the voltage effect due to the resistance of the d-axis armature and the q-axis armature is negligible, it is expressed by the following equation (17).

Where I _d ^: d-axis current, I _q ^: q-axis current, V _d ^: d-axis voltage, V _q ^: q-axis voltage, L _d : d-axis armature inductance, L _q : q-axis electric machine Child inductance, θ _e : Phase difference between the actual value θ of the rotor angle and the estimated value θ ^.

  The present invention will be described below based on the above description. According to a first aspect of the DC brushless motor rotor angle estimation method of the present invention, a DC brushless motor is divided into a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor, and a d-axis orthogonal to the q-axis. An equivalent circuit having a d-axis armature on the shaft is converted and handled, and current detection means for detecting a phase current flowing in the armature of the motor, a phase current detected by the current detection means, and a current of the motor Dq current calculation means for calculating a d-axis current flowing in the d-axis armature and a q-axis current flowing in the q-axis armature based on the estimated value of the rotor angle, and flows in the d-axis armature In order to reduce the deviation between the d-axis command current and the d-axis current, which are current command values, and the deviation between the q-axis command current and the q-axis current, which is a command value of the current flowing through the q-axis armature, D applied to the d-axis armature in each control cycle Determines the q-axis voltage applied to the voltage and the q-axis armature, in the motor control device for performing energization control of the motor, to an improved method of estimating the rotor angle of the motor.

  Then, a first step of superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage, and a calculation based on changes in the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed. , Using at least one of a sine reference value and a cosine reference value corresponding to a double angle of the phase difference between the estimated value and the actual value of the rotor angle of the motor, and the estimated value and the estimated value of the rotor angle of the motor When the phase difference reference value corresponding to the phase difference from the value is calculated and the high frequency voltage is superimposed based on the change of the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed A second step of calculating an inductance reference value according to an average inductance of the motor, and a normalization process for correcting the phase difference reference value according to a level of the inductance reference value. A third step of calculating a phase difference estimated value that changes within a predetermined range in accordance with a phase difference between the estimated value of the data angle and the actual value, and based on the phase difference estimated value, the rotor angle of the motor And a fourth step of updating the estimated value.

In such an invention, the inductances of the d-axis armature and the q-axis armature when the energization control is executed will vary according to the phase difference between the actual value and the estimated value of the rotor angle, as will be described in detail later. As shown in the above equation (17), the d-axis current I _d ^ and the q-axis current I _q ^ depend on the inductance L _{d of the d-} axis armature and the inductance L _q of the q-axis armature. Therefore, in the second step, the sine reference value and the cosine reference value calculated based on changes in the d-axis current and the q-axis current of the motor, and the sine reference value and the cosine reference value. The phase difference reference value calculated using at least one of them changes according to the inductances of the d-axis armature and the q-axis armature.

  Therefore, the inductance reference value corresponding to the average inductance of the motor when the high-frequency voltage is superimposed in the second step is calculated, and in the third step, the phase difference reference value is calculated as the inductance reference value. By performing the normalization process that corrects according to the level of the motor, the influence of changes in the inductance of the motor is suppressed, and within a predetermined range according to the phase difference between the actual value and the estimated value of the rotor angle of the motor It is possible to calculate the estimated phase difference value that changes at In the fourth step, the rotor angle of the motor can be accurately estimated by updating the estimated value of the rotor angle of the motor based on the estimated phase difference value.

  Further, according to a first aspect of the control device for a DC brushless motor of the present invention, a DC brushless motor includes a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor, and d orthogonal to the q-axis. An equivalent circuit having a d-axis armature on the shaft is converted into an equivalent circuit, current detection means for detecting a phase current flowing in the armature of the motor, and rotor angle estimation for calculating an estimated value of the rotor angle of the motor And a d-axis current flowing in the d-axis armature and a q-axis current flowing in the q-axis armature based on the phase current detected by the current detection means and the estimated value of the rotor angle of the motor. Dq current calculation means for calculating, a deviation between a d-axis command current and a d-axis current which is a command value of a current flowing through the d-axis armature, and a q-axis command current which is a command value of a current flowing through the q-axis armature To reduce the deviation between the q-axis current and the q-axis current And a DC power supply control means for determining a d-axis voltage applied to the d-axis armature and a q-axis voltage applied to the q-axis armature every predetermined control cycle, and for controlling the power supply of the motor. The present invention relates to an improvement of a brushless motor control device.

  And a high-frequency superimposing means for superposing a high-frequency voltage on the d-axis voltage and the q-axis voltage, and a rotor angle of the motor based on a change in the d-axis current and the q-axis current of the motor when the high frequency is superimposed. The phase difference between the actual value and the estimated value of the rotor angle of the motor is obtained using at least one of the sine reference value and the cosine reference value corresponding to the double angle of the phase difference between the estimated value and the actual value of And calculating a corresponding phase difference reference value and calculating an average inductance of the motor when the high-frequency voltage is superimposed based on changes in the d-axis current and the q-axis current of the motor when the high-frequency is superimposed. A reference value calculation means for calculating a corresponding inductance reference value, and a normalization process for correcting the phase difference reference value according to a level of the inductance reference value, thereby obtaining a rotor angle of the motor. Phase difference estimated value calculating means for calculating a phase difference estimated value that changes within a predetermined range in accordance with the phase difference between the marginal value and the estimated value, and the rotor angle estimating means is based on the phase difference estimated value. The estimated value of the rotor angle of the motor is updated.

In the present invention, the inductances of the d-axis armature and the q-axis armature when the energization control is executed, as will be described in detail later, vary depending on the phase difference between the actual value and the estimated value of the rotor angle. . As shown in the above equation (17), the d-axis current I _d ^{^} and the q-axis current I _q ^{^} depend on the inductance L _{d of the d-} axis armature and the inductance L _q of the q-axis armature. Therefore, the phase difference calculated by the reference value calculation means using at least one of the sine reference value and the cosine reference value based on changes in the d-axis current and the q-axis current of the motor. The reference value changes according to the inductance of the d-axis armature and the q-axis armature.

  Therefore, the reference value calculation means calculates the inductance reference value according to the average inductance of the motor when the high-frequency voltage is superimposed, and the phase difference estimation value calculation means calculates the phase difference reference value. By performing the normalization process for correcting according to the level of the inductance reference value, the influence of the change in the inductance of the motor is suppressed, and according to the phase difference between the actual value and the estimated value of the rotor angle of the motor. The phase difference estimated value that changes within a predetermined range can be calculated. Then, the rotor angle of the motor can be accurately estimated by updating the rotor angle of the motor based on the estimated phase difference value by the rotor angle estimating means.

  The high-frequency superimposing unit superimposes a periodic signal in which the sum of output voltages in a predetermined period including two or more control cycles is zero on the d-axis voltage and the q-axis voltage as the high-frequency voltage, and the reference value calculation unit The d-axis current and the q-axis voltage corresponding to the first-order difference of the motor phase current and the q-axis difference between adjacent control cycles during the predetermined period with the d-axis voltage and the q-axis voltage in the predetermined period constant. A first-order difference in current is calculated using an estimated value of the rotor angle, and the sine reference value and the cosine reference value are calculated using the first-order difference of the d-axis current and the first-order difference of the q-axis current. The inductance reference value is calculated.

  According to the present invention, the reference value calculation unit is configured to cause the high-frequency superimposing unit to superimpose a periodic signal with a total sum of output voltages of the predetermined period being zero on the d-axis voltage and the q-axis voltage. The first-order difference of d-axis current corresponding to the first-order difference of the phase current of the motor between adjacent control cycles during the predetermined period, with the d-axis voltage and the q-axis voltage being constant during the predetermined period, The first-order difference of the q-axis current is calculated. Then, the reference value calculation means uses the first-order difference of the d-axis current and the first-order difference of the q-axis current according to the double angle of the phase difference between the actual value and the estimated value of the rotor angle of the motor. By calculating the sine reference value, the cosine reference value, and the inductance reference value, the influence of a phase current detection error by the current detection unit is reduced, and the sine reference value, the cosine reference value, and the An inductance reference value can be calculated.

  Further, the reference value calculating means calculates the sine reference value, the cosine reference value, and the inductance reference value by the following formulas (18) to (24).

Where Vs _{dq is a} sine reference value, Vc _{dq is a} cosine reference value, L0 is an inductance reference value, dI _dq (i + 1) to dI _dq (i + n)... N during the predetermined period of the periodic signal. When a control cycle is included, the first-order difference between the d-axis current and the q-axis current between adjacent control cycles, i: the number of the calculation cycle of the sine reference value, cosine reference value, and inductance reference value.

Where L _{d is} the average inductance of the d-axis armature, and L _{q is} the average inductance of the q-axis armature.

However, j = 1,2, ..., n , dI d (i + j): 1 -order difference of the d-axis _{current, dI q (i + j)} : 1 -order difference of the q-axis current.

Where Vh _d (i + j): the output voltage to the d-axis armature of the jth control cycle in the predetermined period of the periodic signal, Vh _q (i + j): j of the periodic signal in the predetermined period Output voltage to the q-axis armature in the second control cycle.

  Here, when the above equation (17) is expressed in a discrete time system, the following equation (23) is obtained.

Where Vfb _d : d-axis voltage in the i-th control cycle, Vfb _q : q-axis voltage in the i-th control cycle, Vh _d (i): superposition of the periodic signal in the i-th control cycle on the d-axis voltage. Voltage, Vh _q (i): Superposed voltage on the q-axis voltage of the periodic signal in the i-th control cycle, ΔT: Control cycle time.

  The periodic signal is a signal in which n control cycles satisfying the following expression (24) are taken as one period.

Here, considering the case where the d-axis voltage Vfb _d (i) and the q-axis voltage Vfb _q (i) are zero, the above equation (23) takes the form of the following equation (25).

Therefore, if c _dq (i) is set as in the following formula (26), the following formulas (27) and (28) are established from the above formula (25).

Here, assuming that the d-axis voltage and the q-axis voltage in the predetermined period are constant values Vfb _dq , the following equation (29) is obtained.

  Therefore, the first term on the right side of the equation (28) is zero, and the following equation (30) is obtained.

Therefore, the reference value calculation means can calculate the sine reference value Vs _{dq, the} cosine reference value Vc _dq and the inductance reference value L0 by the above equation (18), and the phase difference estimated value calculation means The phase difference estimated value can be calculated using at least one of the reference value Vs _dq and the cosine reference value Vc _dq and the inductance reference value L0.

  Further, the phase difference estimated value calculation means calculates the phase difference estimated value by the following equation (31).

However, θ _E1 : estimated phase difference value, Vs _dq : sine reference value (= phase difference reference value), L 0: inductance reference value.

According to the present invention, when the phase difference between the actual value θ of the rotor angle of the motor and the estimated value θ ^ is small (θ−θ ^ ≈0), the sine reference value Vs is obtained by the above equation (31). _The phase difference estimated value can be calculated by dividing _dq as the phase difference reference value by the inductance reference value L0.

  The rotor angle estimation means sequentially updates the estimated value of the rotor angle of the motor based on the estimated phase difference value so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. The estimated value of the rotor angle of the motor is updated by an observer that is calculated.

  According to the present invention, the rotor angle estimation unit sequentially updates the estimated value of the rotor angle of the motor by the observer configured using the phase difference estimated value, and the energization control unit determines the rotor angle. The energization control of the motor can be performed based on the estimated value.

  The phase difference estimated value is further corrected by adding an offset value proportional to the d-axis command current to the phase difference estimated value to correct the phase difference estimated value.

  According to the present invention, as will be described in detail later, by performing correction by adding the offset value to the phase difference estimation value, the estimation accuracy of the rotor angle can be improved, and the phase difference estimation that differs depending on whether it is positive or negative. The range in which the phase difference estimated value can be calculated can be widened by reducing the difference in absolute value level.

  The second aspect of the rotor angle estimation method for a DC brushless motor according to the present invention is such that the deviation between the phase current flowing through the armature of the DC brushless motor and the target current is reduced every predetermined control cycle. The present invention relates to an improvement in a method for estimating a rotor angle of a motor in a motor control apparatus that controls energization of the motor by determining a drive voltage to be applied to an armature of the motor using an estimated value of a rotor angle of the motor.

  And a first step of superimposing a high frequency voltage on the driving voltage, and a rotor angle of the motor calculated based on a change in phase current of the motor when the high frequency voltage is superimposed on the driving voltage. And calculating a phase difference estimated value corresponding to the phase difference between the actual value and the estimated value of the rotor angle of the motor using at least one of a sine reference value and a cosine reference value corresponding to a double angle of A second step of calculating an inductance reference value according to an average inductance of the motor when the high-frequency voltage is superimposed, based on a change in phase current of the motor when the high-frequency voltage is superimposed; A normalization process is performed to correct the phase difference reference value according to the level of the inductance reference value, and a predetermined value is determined according to the phase difference between the estimated value and the actual value of the rotor angle of the motor. A third step of calculating a phase difference estimation value that varies 囲内, based on the phase difference estimated value, characterized in that comprising a fourth step of updating the estimated value of the rotor angle of the motor.

In the present invention, the inductance of the motor when the energization control is executed changes according to the phase difference between the actual value and the estimated value of the rotor angle, as will be described in detail later. As shown in the above equation (15), the phase currents I _u and I _w depend on the motor inductances l and m. Therefore, in the second step, the phase difference reference value calculated using at least one of the sine reference value and the cosine reference value based on a change in the phase current of the motor Varies with inductance.

  Therefore, the inductance reference value corresponding to the average inductance of the motor when the high-frequency voltage is superimposed in the second step is calculated, and in the third step, the phase difference reference value is calculated as the inductance reference value. By performing the normalization process that corrects according to the level of the motor, the influence of changes in the inductance of the motor is suppressed, and within a predetermined range according to the phase difference between the actual value and the estimated value of the rotor angle of the motor It is possible to calculate the estimated phase difference value that changes at In the fourth step, the rotor angle of the motor can be accurately estimated by updating the estimated value of the rotor angle of the motor based on the estimated phase difference value.

  According to a second aspect of the control device for a DC brushless motor of the present invention, there is provided current detection means for detecting a phase current flowing in the armature of the DC brushless motor, and rotor angle estimation for calculating an estimated value of the rotor angle of the motor. And a phase current detected by the current detection means and an estimated value of the rotor angle of the motor, so that the deviation between the predetermined target current and the phase current is reduced every predetermined control cycle. The present invention relates to an improvement of a control device for a DC brushless motor, comprising energization control means for determining energization control of the motor by determining a drive voltage to be applied to an armature of the motor.

  And a high frequency superimposing means for superimposing a high frequency voltage on the drive voltage, and a sine reference value corresponding to a double angle of a rotor angle of the motor based on a change in phase current of the motor when the high frequency voltage is superimposed. And a phase difference reference value corresponding to a phase difference between an actual value and an estimated value of the rotor angle of the motor using at least one of the cosine reference value and the cosine reference value, and when the high frequency is superimposed Reference value calculation means for calculating an inductance reference value according to an average inductance of the motor when the high-frequency voltage is superimposed based on a change in the phase current of the motor, and the phase difference reference value as the inductance reference value A phase that changes within a predetermined range according to a phase difference between an actual value and an estimated value of the rotor angle of the motor by performing a normalization process that corrects according to the level of the motor A phase difference estimation value calculating means for calculating the estimated value, the rotor angle estimation means, based on said phase difference estimates, and updates the estimated value of the rotor angle of the motor.

In the present invention, the inductance of the motor when the energization control is executed changes according to the phase difference between the actual value and the estimated value of the rotor angle, as will be described in detail later. As shown in the above equation (15), the phase currents I _u and I _w depend on the motor inductances l and m. Therefore, the phase difference reference value calculated by the reference value calculation means using at least one of the sine reference value and the cosine reference value based on the change in the phase current of the motor is the inductance of the motor. It changes according to.

  Therefore, the reference value calculation means calculates the inductance reference value according to the average inductance of the motor when the high-frequency voltage is superimposed, and the phase difference estimation value calculation means calculates the phase difference reference value. By performing the normalization process for correcting according to the level of the inductance reference value, the influence of the change in the inductance of the motor is suppressed, and according to the phase difference between the actual value and the estimated value of the rotor angle of the motor. The phase difference estimated value that changes within a predetermined range can be calculated. Then, the rotor angle of the motor can be accurately estimated by updating the rotor angle of the motor based on the estimated phase difference value by the rotor angle estimating means.

  The high-frequency superimposing unit superimposes a periodic signal with a total sum of output voltages of zero in a predetermined period including two or more control cycles as the high-frequency voltage on the driving voltage, and the reference value calculating unit includes the predetermined value calculating unit. The first phase difference of the motor phase current between adjacent control cycles is calculated with the driving voltage in a period being constant, and the sine reference value, the cosine reference value, and the inductance reference value are calculated using the first order difference. Is calculated.

  According to the present invention, the reference value calculation means is configured to cause the high-frequency superimposing means to superimpose a periodic signal in which the sum of output voltages in the predetermined period is zero on the drive voltage during the predetermined period. The first-order difference of the motor phase current between adjacent control cycles during the predetermined period is calculated with the driving voltage being constant. Then, the reference value calculation means calculates the sine reference value, the cosine reference value, and the inductance reference value according to a double angle of the rotor angle of the motor, using the first-order difference of the phase current. Thus, it is possible to calculate the sine reference value, the cosine reference value, and the inductance reference value by reducing the influence of the phase current detection error by the current detection means.

  Further, the reference value calculating means calculates the sine reference value, the cosine reference value, and the inductance reference value by the following equations (32) to (36).

Where Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, l: average value of self-inductance of each phase of motor, m: average value of mutual inductance between each phase of motor, dI (i +1) to dI (i + n): the first-order difference of phase currents between adjacent control cycles when n control cycles are included in the predetermined period of the periodic signal, i: sine reference value and Calculation cycle number of cosine reference value and inductance reference value.

  Where θ is the rotor angle of the motor.

Here, j = 1, 2,..., N, dI _u (i + j): first-order difference of U-phase phase current, dI _w (i + j): first-order difference of W-phase phase current.

Where Vh _uv (i + j) is the voltage between the U phase and V phase of the jth control cycle in the predetermined period of the periodic signal, and Vh _wv (i + j) is the jth of the periodic signal in the predetermined period. Interphase voltage of control phase W phase and V phase.

  Here, when the above equation (15) is expressed in a discrete time system, the following equation (37) is obtained.

Where Vfb _u (i), Vfb _v (i), Vfb _w (i): the drive voltage in the i-th control cycle, Vh _u (i), Vh _v (i), Vh _w (i): i-th Output voltage of the periodic signal in the control cycle, ΔT: time of the control cycle.

  The periodic signal is a signal in which n control cycles satisfying the following equation (38) are taken as one period.

  Here, considering the case where Vfb is zero, the above equation (37) takes the form of the following equation (39).

  Therefore, when c (i) is set as in the following formula (40), the following formulas (41) and (42) are established from the above formula (39).

  Here, when the driving voltage in the predetermined period is a constant value Vfb, the following equation (43) is obtained.

  Therefore, the first term on the right side of the above equation (42) is 0, and the following equation (44) is obtained.

Therefore, the sine reference value Vs, the cosine reference value Vc, and the inductance reference value L _lm can be calculated by the above equation (32), and the phase difference estimated value calculating means can calculate the sine reference value Vs and the cosine reference Vc. The phase difference reference value can be calculated using at least one of them, and the phase difference estimated value can be calculated using the phase difference reference value and the inductance reference value _Llm .

  Further, the phase difference estimated value calculating means calculates the phase difference estimated value by the following equation (45).

Where θ _{E2 is the} estimated phase difference value, θ _e is the phase difference between the actual value and the estimated value of the rotor angle, Vs is the sine reference value, Vc is the cosine reference value, and Vs · cos2θ ^ −Vc · sin2θ ^ is the phase difference. Reference value, θ ^{^} : Estimated value of rotor angle, L _lm : Inductance reference value.

According to the present invention, when the phase difference between the actual value θ of the rotor angle of the motor and the estimated value θ ^ is small (θ−θ ^ ≈0), Vs · cos2θ ^ − according to the above equation (45). by dividing the inductance reference value L _lm a vc · sin2θ ^ as the phase difference reference value, it is possible to calculate the phase difference estimates.

  The rotor angle estimation means sequentially updates the estimated value of the rotor angle of the motor based on the estimated phase difference value so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. The estimated value of the rotor angle of the motor is updated by an observer that is calculated.

  According to the present invention, the rotor angle estimation unit sequentially updates the estimated value of the rotor angle of the motor by the observer configured using the phase difference estimated value, and the energization control unit determines the rotor angle. The energization control of the motor can be performed based on the estimated value.

  The phase difference estimated value may further include a phase difference estimated value correcting means for correcting the phase difference estimated value by adding an offset value proportional to the target current to the phase difference estimated value.

  According to the present invention, as will be described in detail later, by performing correction by adding the offset value to the phase difference estimation value, the estimation accuracy of the rotor angle can be improved, and the phase difference estimation that differs depending on whether it is positive or negative. The range in which the phase difference estimation value can be calculated can be expanded by reducing the difference in value level.

  Embodiments of the present invention will be described with reference to FIGS. FIG. 1 is a configuration diagram of a DC brushless motor, FIG. 2 is a configuration diagram of a motor control device in the first embodiment, FIG. 3 is an explanatory diagram of an output mode of a periodic signal in the motor control device shown in FIG. Is a flowchart of the rotor angle estimation process in the motor control device shown in FIG. 2, FIG. 5 is a timing chart corresponding to the flowchart of FIG. 4, and FIG. 6 is a comparison showing a difference in sine reference values depending on whether or not the rotor magnetic pole is saturated. FIG. 7 is a comparison graph when the sine reference value is normalized by the sine reference value and the cosine reference value. FIG. 8 is a comparison graph showing the difference in the inductance reference value depending on whether or not the rotor magnetic pole is saturated. Is an explanatory diagram of a change in the inductance of the motor due to the phase difference between the actual value and the estimated value of the rotor angle, and FIG. 10 is a case where the sine reference value is normalized by the inductance reference value FIG. 11 is a comparison graph, FIG. 11 is a graph illustrating the effect of performing correction to add an offset value to the phase difference estimated value, FIG. 12 is a configuration diagram of a motor control device according to the second embodiment, and FIG. It is a flowchart of the estimation process of the rotor angle in embodiment.

  [First Embodiment] First, the first embodiment of the present invention will be described with reference to FIGS. The DC brushless motor control device 40 (hereinafter simply referred to as the control device 40) shown in FIG. 2 is a three-phase (U, V, W) of the DC brushless motor 1 (hereinafter simply referred to as the motor 1) shown in FIG. Feedback control of the current flowing through the armatures 3, 4 and 5 of the motor 1, the motor 1 includes a q-axis armature on the q-axis which is the magnetic flux direction of the field pole of the rotor 2, and a d-axis orthogonal to the q-axis. It is converted into an equivalent circuit based on a dq coordinate system having a d-axis armature located above.

  Then, the control device 40 determines a current flowing in the d-axis armature (hereinafter referred to as a d-axis current) and a current flowing in the q-axis armature in accordance with the d-axis command current Id_c and the q-axis command current Iq_c given from the outside. (Hereinafter referred to as q-axis current) is feedback controlled.

The control device 40 determines the rotor angle based on the voltage Vfb _d applied to the d-axis armature (hereinafter referred to as d-axis voltage Vfb _d ) and the voltage Vfb _q applied to the q-axis armature (hereinafter referred to as q-axis voltage Vfb _q ). A periodic signal superimposing unit 51 (corresponding to the high frequency voltage superimposing means of the present invention) for superimposing the periodic signals Vh _d and Vh _q for estimation (corresponding to the high frequency voltage of the present invention), a d-axis voltage Vfb _d and a q-axis voltage Vfb _q (Vfb _d + Vh _d and Vfb _q + Vh _q when the periodic signals Vh _d and Vh _q are superposed) is applied to the three-phase armature of the motor 1 Vfb _{u, Vfb v, Vfb dq /} 3 -phase conversion unit 20 converts the _w, and Vfb _u, Vfb _v, the voltage Vu corresponding to Vfb _w, Vv, Vw of the motor 1 U, V, each phase of the electrical machine W A power driver consisting of an inverter circuit configured by bridge-connecting a plurality of switching elements to be applied to the child The unit 22 is provided.

The control device 40 further includes a U-phase current sensor 23 (corresponding to current detection means of the present invention) that detects a current flowing through the U-phase armature of the motor 1, and a current flowing through the W-phase armature of the motor 1. W-phase current sensor 24 for detecting (corresponding to the current detecting means of the present invention), U-phase current detected current value I _u and W detected current value of the phase current sensor 24 I _w and d-axis current I in response to the sensor 23 A three-phase / dq conversion unit 26 (corresponding to dq current calculation means of the present invention) that calculates _d and q-axis current I _q, and an angle estimation unit 50 (which calculates an estimated value θ ^{^} of the rotor angle θ of the motor 1) Including the functions of the rotor angle estimating means, the reference value calculating means, and the phase difference estimated value calculating means of the present invention), and a non-interference operation for performing processing to cancel the influence of the speed electromotive force that interferes between the d-axis and the q-axis The unit 27 is provided.

Then, the control unit 40 subtracts the d-axis command current Id_c and the d-axis current I _d at first subtractor 28 performs PI (proportional integral) processing by the first PI calculation portion 29 on the subtraction result, the by adding the non-interference component α 1 adder 30, the following equation (46) for generating a d-axis voltage Vfb _d corresponding to the deviation of the d-axis command current Id_c and the d-axis current I _d.

  However, Kp: proportional gain, Ki: integral gain, α: non-interference component.

Similarly, the control device 40 subtracts the q-axis command current Iq_c and the q-axis current _Iq by the second subtractor 31, performs a PI process on the subtraction result by the second PI calculator 32, by adding the non-interference component β 2 adder 33 generates the q-axis voltage Vfb _q corresponding to a deviation between the q-axis command current Iq_c and the q-axis current I _q by the following equation (47).

  However, Kp: proportional gain, Ki: integral gain, β: non-interference component.

Then, the control device 40 inputs the d-axis voltage Vfb _d and the q-axis voltage Vfb _q to the dq / 3-phase conversion unit 20. The dq / 3-phase converter 20 calculates the three-phase drive voltages Vfb _u , Vfb _v , and Vfb _{w according} to the following equation (48) and outputs them to the power drive unit 22.

Thus, via the power drive unit 22, the d-axis command current Id_c and the d-axis current I _d and deviation, and q-axis command current Iq_c and the q-axis current I _q and 3-phase driving voltage Vfb _u of the deviation to reduce the , Vfb _v , Vfb _w are applied to the armature of the motor 1, and the current flowing through the armature of the motor 1 is feedback controlled.

In this manner, so as to reduce the deviation between the d-axis command current Id_c and the d-axis current I _d, and the deviation between the q-axis command current Iq_c and the q-axis current I _q, the 3-phase drive voltages Vfb _u, The configuration in which Vfb _v and Vfb _w are determined to control the phase current of the motor 1 corresponds to the energization control means of the present invention.

Here, when the _d- q / 3-phase converter 20 converts the d-axis driving pressure Vfb _d and the q-axis voltage Vfb _q into three-phase driving voltages Vfb _u , Vfb _v , Vfb _w , the rotor angle θ of the motor 1 Is required. Further, even when converting the detected current value Iw of the detected current values Iu and the W-phase current sensor 24 of the U-phase current sensor 23 to d-axis current I _d and the q-axis current I _q by 3-phase / dq converter 26, The rotor angle θ of the motor 1 is required.

Then, the control device 10 does not use a position detection sensor such as a resolver, but in the third adder 52 and the fourth adder 53, the periodic signal superimposing unit 51 converts the periodic signals Vh _d to the d-axis voltages Vfb _d and Vfb _q. , Vh _q are superimposed on each other to execute the rotor angle θ estimation process.

Note that the periodic signals Vh _d and Vh _q in the present embodiment are as follows: 3 control cycles (T ₁₁ , T ₁₂ , T ₁₃ ) of the control device 10 are 1 period as shown in FIG. The output pattern is set so that the sum of the output voltages of each phase (Vh _d , Vh _q ) becomes zero in a cycle. As another output pattern, for example, as shown in FIG. 3B, four control cycles (T ₂₁ , T ₂₂ , T ₂₃ , T ₂₄ ) are defined as one period, and two control cycles ( sum of T ₂₁ and T _22, T ₂₃ and T ₂₄₎ the output voltage for each may be set to be zero. Further, the length of one cycle of the periodic signals Vh _d and Vh _q may be two or more control cycles.

Hereinafter, the process of estimating the rotor angle of the motor 1 in the control device 40 will be described with reference to the flowchart shown in FIG. Controller 40, a counter variable ptr to 0, by the periodic signal superimposing section 51 superimposes the periodic signal Vh _q to q-axis voltage Vfb _q with superimposing the periodic signal Vh _d to d-axis voltage Vfb _d, 4 This flowchart is repeatedly executed.

Angle estimator 50, in STEP 30, the phase current I _u detected by the U-phase current sensor 23 and the W-phase current sensor 24 takes the I _w, the phase current I _u in the previous control cycle in STEP 31, and I _w First-order differences dI _u and dI _w are calculated. In the next STEP 2, dI _d and dI _q obtained by three-phase / dq conversion of the first-order differences dI _u and dI _w are calculated and held.

  If the counter variable ptr is 1 in the next STEP33, the process proceeds to STEP34 and the processes of STEP34 to STEP37 are performed. If the counter variable ptr is not 1 in STEP33, the process branches to STEP38.

Here, the counter variable ptr is incremented for each control cycle in STEP43, and cleared in STEP45 (ptr = 0) when ptr = 3 in STEP44. Therefore, every time three control cycles corresponding to one period of the periodic signals Vh _d and Vh _q elapse, ptr = 1 is set at STEP33, and the processing of STEP34 to STEP37 is executed.

In STEP 34, the angle estimation unit 50 calculates the sine reference value Vs _dq and the cosine reference value Vc _dq using the first-order differences dI _d and dI _q calculated in STEP 32 of the current, previous and previous control cycles. Specifically, the angle estimator 50 according to the following equation (49) in which n is replaced by 3 in the above equation (18), which is the number of control cycles included in one cycle of the periodic signals Vh _d and Vh _q. Calculates a sine reference value Vs _{dq, a} cosine reference value Vc _dq, and an inductance reference value L0. The configuration in which the angle estimation unit 50 calculates the sine reference value Vs _{dq, the} cosine reference value Vc _dq, and the inductance reference value L0 corresponds to the reference value calculation means of the present invention.

However, in a control cycle in which Vs _{dq is a} sine reference value, Vc _{dq is a} cosine reference value, L0 is an inductance reference value, and dI _dq (i + 1) to dI _dq (i + 3) are ptr = 2, 0, 1. First-order difference between the calculated d-axis current and q-axis current.

Here, by calculating the sine reference value Vs _{dq, the} cosine reference value Vc _dq and the inductance reference value L0 using the first-order difference between the d-axis current and the q-axis current according to the above equation (49), the d-axis current and The effect of reducing the influence of detection errors of the U-phase current sensor 23 and the W-phase current sensor 24 can be expected as compared with the case where the second-order difference of the q-axis current is used.

Then, the angle estimation unit 50 determines the phase difference θ _e between the actual value θ of the rotor angle and the estimated value θ ^ of the rotor angle used for the conversion in the dq / 3-phase conversion unit 20 and the 3-phase / dq conversion unit 26. The phase difference estimated value θ _E1 according to the above is calculated by the following equation (50).

Where, θ _E1 : phase difference estimated value, Vs _dq : sine reference value (corresponding to the phase difference reference value of the present invention), L 0: inductance reference value, θ _e : rotor angle actual value θ and estimated value θ ^ Phase difference.

  Then, the angle estimation unit 50 updates and calculates the estimated value θ ^ of the rotor angle by the follow-up calculation by the observer of the following equation (51).

Where θ ^ (n + 1): updated value of the estimated value of the rotor angle, ω ^ (n + 1): updated value of the estimated value of the angular velocity of the rotor, Δt: one period of the periodic signals Vh _d and Vh _q ( 3 control cycles), θ ^ (n): previous value of estimated rotor angle, ω ^ (n): previous value of estimated rotor angular velocity, θ _E1 (n): estimated phase difference, K1 , K2, K ~: Calculation gain, offset: Offset value proportional to d-axis command current.

Therefore, in STEP 35, the angle estimation unit 50 calculates the phase difference estimation value θ _E1 necessary for the equation (51) from the equation (50). In subsequent STEP 36, I _d and I _q calculated by the three-phase / dq converter 26 are held for current feedback control. In STEP 37, the estimated value θ ^ of the rotor angle used in the three-phase / dq conversion unit 26 and the dq / 3-phase conversion unit 20 is updated.

When the counter variable ptr becomes 0 in the next STEP 38, the process proceeds to STEP 39, and the d-axis voltage Vfb _d and the q-axis voltage Vfb _q are calculated by the above equations (46) and (47). In STEP 40, the angle estimation unit 50 calculates the estimated value θ ^ of the rotor angle by the above equation (51). On the other hand, if ptr is not zero in STEP38, the process branches to STEP41, and the processing of STEP39 to STEP40 is not executed.

Therefore, the d-axis voltage Vfb _{d, the} q-axis voltage Vfb _q and the estimated value θ ^ of the rotor angle are updated every period (every three control cycles) of the periodic signals Vh _d and Vh _q , and the periodic signals Vh _d and Vh. During one cycle of _q , the d-axis voltage Vfb _{d, the} q-axis voltage Vfb _q, and the estimated value θ ^ of the rotor angle are kept constant. Further, as described above, the estimated value θ ^ of the rotor angle used in the three-phase / dq conversion unit 26 and the dq / 3-phase conversion unit is updated in STEP 37 when ptr = 1. Therefore, the estimated value θ ^ of the rotor angle updated in STEP 40 is used in the dq / 3-phase converter 20 and the 3-phase / dq converter 26 from the next control cycle.

In this way, the update of the estimated value θ ^ of the rotor angle for dq conversion is delayed because the estimated value θ ^ of the rotor angle at the time when the d-axis voltages Vfb _d and Vfb _q are calculated in STEP 39, and STEP 32 it is to match the estimated value of the rotor angle theta ^ at the time of calculating the first-order difference dI _q of the first-order difference dI _d and q-axis current of the d-axis current. By matching the estimated value θ ^ of the rotor angle in STEP39 and STEP32, it is possible to prevent an error due to a change in the estimated value θ ^ of the rotor angle when the differential currents dI _d and dI _{q are} calculated.

In subsequent STEP 41, the periodic signal superimposing unit 51 calculates the periodic signals Vh _d and Vh _q to the d-axis voltage Vfb _d and the q-axis voltage Vfb _q , and in STEP 42, the dq / three-phase conversion unit 20 calculates the d-axis voltages Vfb _d and q The shaft voltage Vfb _q is converted into a three-phase drive voltage Vfb _u , Vfb _v , Vfb _w and output to the power drive unit 22. In subsequent STEP44 to STEP46, the counter variable ptr described above is incremented or cleared, and one control cycle is completed.

In STEP 41, the process of superimposing the periodic signals Vh _d and Vh _q by the periodic signal superimposing unit 21 corresponds to the first step of the present invention. In STEP 34, the sine reference value Vs _{dq, the} cosine reference value Vc _dq and the inductance reference The process of calculating the value L0 and calculating the phase difference estimated value θ _E1 in STEP 35 corresponds to the second step of the present invention. In STEP 40, the process of correcting by adding an offset value to the phase difference estimated value θ _E1 corresponds to the third step of the present invention, and the estimated value θ of the rotor angle is obtained using the corrected phase difference estimated value θ _E1. The process of updating ^ corresponds to the fourth step of the present invention. Further, in STEP 40, configured to correct the phase difference estimates theta _E1 adds the offset value to the phase difference estimates theta _E1 corresponds to a phase difference estimation value correcting means of the present invention.

FIG. 5 is a timing chart when the estimated value θ ^ of the rotor angle is calculated by repeatedly executing the process according to the flowchart of FIG. 4 described above. Processing is started in cycle 50, and three control cycles (cycles 51, 52, 60, cycles 61, 62, 70,...) Corresponding to ptr = 1, 2, 0 are generated by periodic signals Vh _dq (Vh _d , Vh _q ) And the periodic signal is Vh _dq (0) (when ptr = 1) → Vh _dq (1) (when ptr = 2) → Vh _dq (2) (when ptr = 0) It has been switched.

Since the d-axis voltage Vfb _d and the q-axis voltage Vfb _q (hereinafter collectively referred to as the dq voltage Vfb _dq ) are calculated in the control cycle in which ptr = 0 (step 39 in FIG. 4), the period While the signal Vh _dq switches from Vh _dq (0) → Vh _dq (1) → Vh _dq (2), the dq voltage Vfb _dq is constant (Vfb _dq (0) during cycles 51, 52 and 53, cycle 61). , 62 and 63 are maintained at Vfb _dq (1)).

Then, in control cycle 61 and control cycle 71 in which ptr = 1, a phase difference estimated value θ _E1 corresponding to the phase difference θ _e between the actual value and the estimated value of the rotor angle is calculated (processing of STEP 35 in FIG. 4). At the same time, the estimated value θ ^ of the rotor angle used in the dq / 3-phase conversion unit 20 and the 3-phase / dq conversion unit 26 is updated (processing of STEP 37 in FIG. 4). In addition, in the control cycle 61 and the control cycle 71 where ptr = 0, the dq voltage Vfb _dq and the estimated value θ ^ of the rotor angle are updated (processing of STEP39 to STEP40 in FIG. 4). In this way, the updating of the dq voltage Vfb _{dq and} the updating of the estimated value θ ^ of the rotor angle are performed at the same timing, thereby eliminating the rotor angle deviation when updating the dq voltage Vfb _dq .

In the figure, “c” indicates changes in the d-axis current and the q-axis current, and “d” indicates a response of feedback control of the d-axis current and the q-axis current. The influence of feedback control of the d-axis current and the q-axis current due to the superposition of the periodic signal Vh _dq is canceled by the dq voltage Vfb _dq calculated at the timings P ₁₂ and P _{13 in} the figure.

Next, with reference to FIGS. 6-11, the above equation by (50) calculates the phase difference estimates theta _E1, the above equation (51) on the phase difference estimated value theta _E1 correction adding an offset value offset The effect of updating the rotor angle estimated value θ ^ so that the corrected phase estimated value (θ _E1 + offset) becomes zero will be described.

First, FIG. 6A shows a sine reference value Vs _dq when the actual value of the rotor angle is 30 degrees using the inductance (constant) of the motor 1 when no current flows through the armature of the motor 1. the is a graph calculated by the above equation (49) by a computer simulation, the vertical axis is set to the sine reference value Vs _dq horizontal axis is set to a phase difference theta _e estimate and the actual value of the rotor angle. In this case, the level of the sine reference value Vs _dq changes according to the phase difference theta _e, positive (the actual value is advanced than the estimated value the advance side of the rotor angle) of the phase difference theta _e and negative ( The amplitudes on the retard side where the actual value of the rotor angle is delayed from the estimated value are equal.

On the other hand, FIG. 6B shows a sine reference value Vs when the actual value of the rotor angle is 30 degrees with the drive voltages Vfb _u , Vfb _v , and Vfb _w applied to the armature of the motor 1. the _dq, a graph was calculated by the equation (49) by a computer simulation, the vertical axis is set to the sine reference value Vs _dq horizontal axis is set to a phase difference theta _e estimate and the actual value of the rotor angle .

In this case, the amplitude of the phase difference θ _e is larger than that in the case of FIG. 6A, and the waveform on the positive side (advance side) of the phase difference θ _e is greatly distorted so that the amplitude is negative ( It is smaller than the retard side. Therefore, it is difficult to calculate the estimated value θ ^ of the rotor angle using the sine reference value Vs _dq as a phase difference estimated value corresponding to the phase difference θ _e .

Next, FIG. 7A and FIG. 7B are comparative graphs obtained by calculating the phase difference estimated value θ _E0 calculated by the following equation (52) by computer simulation under the same conditions as in FIG. is there.

FIG. 7A shows the estimated phase difference θ _E1 when the actual value of the rotor angle is 30 degrees using the inductance (constant) of the motor 1 when no current flows through the armature. 52), the vertical axis is set to the phase difference estimated value θ _E0 , and the horizontal axis is set to the phase difference θ _e between the actual value of the rotor angle and the estimated value. In this case, the level of the phase difference estimates theta _E1 varies depending on the magnitude of the phase difference theta _e, the waveform of the positive side of the phase difference estimates theta _E1 (advance side) and the negative side (retarding side) Are equal.

On the other hand, FIG. 7B shows an estimated phase difference value when the actual value of the rotor angle is 30 degrees in a state where the drive voltages Vfb _u , Vfb _v , and Vfb _w are applied to the armature of the motor 1. It is the graph which computed (theta) _E0 by said Formula (52), and the vertical axis _| shaft is set to the phase difference estimated value (theta) _E0 , and the horizontal axis is set to the phase difference (theta) _e of the actual value and estimated value of a rotor angle. In this case, the waveform of the phase difference estimates theta _E0 are distorted on the positive side of the phase difference theta _e (advance side), the phase difference theta _e in the positive side and the negative side of the phase difference estimates theta _E0 The amplitude is very different. Therefore, it is difficult to calculate the estimated value θ ^ of the rotor angle using Vs _dq / √ (Vs _dq ² + Vc _dq ² ) as a phase difference estimated value corresponding to the phase difference θ _e .

Next, FIGS. 8A and 8B show changes in the inductance reference value L0 calculated by the above equation (49). FIG. 8A shows the inductance reference value L0 when the actual value of the rotor angle is 30 degrees by computer simulation using the inductance (constant value) of the motor 1 when no current flows through the armature. is a graph calculated by the equation (49), the vertical axis is set to the inductance reference value L0 horizontal axis is set to a phase difference theta _e estimate and the actual value of the rotor angle. In this case, the difference in level of the inductance reference value L0 between the positive side (advance side) and the negative side (retard side) of the phase difference estimated value θ _E1 is small.

On the other hand, FIG. 8B shows an inductance reference value L0 when the actual value of the rotor angle is 30 degrees with the drive voltages Vfb _u , Vfb _v , and Vfb _w being applied to the armature of the motor 1. the is a graph calculated by the above equation (49) by a computer simulation, the vertical axis is set to the inductance reference value L0 horizontal axis is set to a phase difference theta _e between the estimated value and the actual value of the rotor angle. In this instance, the level of the inductance reference value L0 in the negative side of the phase difference theta _e (retard side) is larger than the level of the inductance reference value L0 in the positive side of the phase difference theta _e (advance side) ing.

Referring now to FIG. 9 (a) ~ FIG 9 (b), when the level of the inductance reference value L0 situation will be described which changes according to the sign of the phase difference theta _e, shown in FIG. 9 (a) Thus, when the phase difference θ _e is zero (actual value θ = estimated value θ ^ of the rotor angle), the direction of the magnetic field by the armature is set in the direction perpendicular to the direction of the rotor field by energization control. Therefore, the inductance of the motor 1 does not change.

In contrast, as shown in FIG. 9 (b), when the phase difference theta _e is positive (the actual state value theta is ahead estimate theta ^ of the rotor angle), the rotor magnetic field formed by the armature Since the field No. 2 is weakened, the inductance L of the motor 1 increases. Since the inductance reference value is the sum of the reciprocals of the inductances L _d and L _q of the motor 1 according to the above equation (19), the inductance reference value L 0 decreases as the inductance of the motor 1 increases.

As shown in FIG. 9C, when the phase difference θ _e is negative (the rotor angle actual value θ is delayed from the estimated value θ ^), the magnetic field M generated by energizing the armature. As a result, the magnetic field of the rotor 2 is strengthened, so that the inductance L of the motor 1 decreases. The inductance reference value L0 increases as the inductance L of the motor 1 decreases.

Here, the tendency of the level change of the inductance reference value L0 in FIG. 8B approximates the tendency of the level change of the sine reference value Vs _dq shown in FIG. 6B. Therefore, the following equation (53), a normalization process of dividing the sine reference value Vs _dq inductance reference value L0, by suppressing fluctuation in level due to positive and negative phase difference theta _e, the phase difference theta _e It is possible to calculate a phase difference estimated value θ _E1 that changes within a predetermined range according to

FIG. 10A shows the phase difference estimated value θ _E1 when the actual value of the rotor angle is 30 degrees using the inductance (constant) of the motor 1 when no current flows through the armature. 53), the vertical axis is set to the phase difference estimated value θ _E1 , and the horizontal axis is set to the phase difference θ _e between the actual value of the rotor angle and the estimated value. In this case, the level of the phase difference estimation value θ _E1 changes according to the magnitude of the phase difference θ _e , and the phase difference estimation between the positive side (advance side) and the negative side (retard side) of the phase difference θ _e The values θ _{E1 have} the same amplitude.

On the other hand, FIG. 10B shows the phase difference estimated value when the actual value of the rotor angle is 30 degrees with the drive voltages Vfb _u , Vfb _v , and Vfb _w applied to the armature of the motor 1. It is the graph which computed (theta) _E1 by the said Formula (53), A vertical axis | shaft is set to the phase difference estimated value (theta) _E1 , and a horizontal axis is set to the phase difference (theta) _e of the actual value and estimated value of a rotor angle. In this case, the waveform distortion of the phase difference estimated value θ _E1 when the phase difference θ _e is positive is suppressed as compared with the graphs of FIGS. 6B and 7B described above. Further, the amplitude of the phase difference estimated value θ _E1 is almost the same level as in FIG. 10A, and good normalization is performed.

However, when compared with the graph of FIG. 10A, it can be seen that the estimated phase difference value θ _E1 in the graph of FIG. 10B is shifted in the positive direction as a whole. As the d-axis current of the motor 1 is larger, the field strengthening / field weakening effect of the rotor 2 is increased. Therefore, it is effective to perform correction by adding the offset value offset proportional to the d-axis command current to the phase difference estimated value θ _E1 by the above equation (51).

FIG. 11 shows the effect when the offset value offset is added to the graph shown in FIG. 10B. The corrected phase difference estimated value obtained by adding the offset value according to the above equation (51). By updating the estimated value of the rotor angle so that (θ _E1 + offset) converges to zero, the convergence point of the estimated value of the updated rotor angle can be shifted from P1 to P2, which is closer to zero. Therefore, the estimation accuracy of the rotor angle can be increased.

Further, by performing the correction to add the offset value, the region where the phase difference θ _e is uniquely determined with respect to the phase difference estimated value θ _E1 can be expanded from the region S1 before adding the offset value to S2. As a result, the difference between the amplitude of the phase difference estimated value θ _E1 and the range of the stable region due to the sign of the phase difference θ _e can be reduced. The estimated value θ ^ of the rotor angle can be updated by a simple observer with a constant gain for the estimated phase difference value (θ _E1 + offset).

Although this first embodiment to calculate the sine reference value Vs _dq and the cosine reference value Vc _dq inductance reference value L0 by the above formula (49), a sine reference value Vs _dq and the cosine reference value Vc _dq and the inductance The calculation formula of the reference value L0 is determined according to the mode of the periodic signal Vh _dq .

Further, in the first embodiment, the phase difference estimated value θ _E1 is calculated by the above equation (50), but the calculation formula of the phase difference estimated value θ _E1 is a calculation mode of the phase difference reference value and the inductance reference value. It is decided according to.

  Further, in the first embodiment, the estimated value θ ^ of the rotor angle is updated by the observer of the above formula (51), but a method for updating the estimated value θ ^ of the rotor angle using the phase difference estimated value The present invention can be applied to any device.

  [Second Embodiment] Next, a second embodiment of the present invention will be described with reference to FIGS. The DC brushless motor control device 10 (hereinafter simply referred to as the control device 10) shown in FIG. 12 performs feedback control of the phase current of the motor 1 in the same manner as the control device 40 shown in FIG. The same components as those in FIG.

In the control device 10, aspects of the process of estimating the rotor angle is different from the controller 40, corresponding to the period signal Vh _u and the high-frequency voltage Vh _w (present invention for rotor position estimation in the driving voltage Vfb _u and Vfb _w ) And the angle estimation unit 25 (functions of the rotor angle estimation unit, the reference value calculation unit, and the phase difference estimation value calculation unit of the present invention). Including).

Then, the control unit 10, in a fifth adder 34 sixth adder 36, the periodic signal superimposing section 21 superimposes the periodic signal Vh _u to the drive voltage Vfb _u, a periodic signal Vh _w to the drive voltage Vfb _w The estimated value θ ^ of the rotor angle is calculated by superimposing. The output signals of the periodic signals Vh _u and Vh _w are set so that the total sum of the output voltages of the respective phases is zero in one period, with three control cycles of the motor control device 10 as one period. Further, the length of one cycle of the periodic signals Vh _u and Vh _w may be two control cycles or more.

Hereinafter, the process of estimating the rotor angle of the motor 1 in the control device 10 will be described with reference to the flowchart shown in FIG. The control device 10 sets the initial value of the counter variable ptr to 0, superimposes the periodic signals Vh _u and Vh _w on the drive voltages Vfb _u and Vfb _w by the periodic signal superimposing unit 21, and repeatedly executes the flowchart of FIG.

Angle estimator 25, phase currents I _u detected by the U-phase current sensor 23 and the W-phase current sensor 24 in STEP1, captures I _w, the phase current I _u in the previous control cycle in STEP2, 1 and I _w Calculate and hold the floor differences dI _u and dI _w . If the counter variable ptr is 1 in the subsequent STEP3, the process proceeds to STEP4 and the processes of STEP4 to STEP6 are performed. If the counter variable ptr is not 1 in STEP3, the process branches to STEP7.

Here, the counter variable ptr is incremented for each control cycle in STEP12, and cleared in STEP14 (ptr = 0) when ptr = 3 in STEP13. Therefore, the control cycle is periodic signals Vh _u, Vh _v, becomes ptr = 1 in STEP3 every three control cycles corresponding to one cycle of the Vh _w has elapsed, the processing of STEP4~STEP6 is executed.

In STEP 4, the angle estimation unit 25 uses the first-order differences dI _u and dI _w calculated in STEP 2 in the current, previous and previous control cycles, and uses the sine reference value Vs, the cosine reference value Vc, and the inductance reference value. _Llm is calculated. Specifically, the angle estimator 25 is obtained by the following equation (54) in which n is replaced by 3 which is the number of control cycles included in one cycle of the periodic signals Vh _u and Vh _{w in} the above equation (32). Calculates a sine reference value Vs, a cosine reference value Vc, and an inductance reference value _Llm . In addition, the structure which calculates the sine reference value Vs, the cosine reference value Vc, and the inductance reference value _Llm by the angle estimation part 25 is equivalent to the reference value calculation means of this invention.

Where Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, l: average value of self-inductance of each phase of motor, m: average value of mutual inductance between angular phases of motor, dI (i +1) to dI (i + 3): the first-order difference of the phase currents calculated in the control cycle where ptr = 2, 0, 1.

Here, when the second-order difference of the phase current is used by calculating the sine reference value Vs, the cosine reference value Vc, and the inductance reference value _Llm using the first-order difference of the phase current according to the above equation (54). In comparison, an effect of reducing the influence of detection errors of the U-phase current sensor 23 and the W-phase current sensor 24 can be expected.

The angle estimation unit 25 then compares the phase difference θ _e between the actual value θ of the rotor angle and the estimated value θ ^ of the rotor angle used for the conversion in the dq / 3-phase conversion unit 20 and the 3-phase / dq conversion unit 26. The phase difference estimated value θ _E2 corresponding to the above is calculated by the following equation (55).

However, θ _E2 : phase difference estimated value, Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, θ: actual value of rotor angle, θ ^: estimated value of rotor angle, l: motor value Average value of self-inductance of each phase, m: average value of mutual inductance between phases of motor, θ _e : phase difference between actual value θ of rotor angle and estimated value θ ^.

  Then, the angle estimation unit 50 updates the estimated value θ ^ of the rotor angle by the follow-up calculation by the observer of the following formula (56).

Where θ ^ (n + 1): updated value of the estimated value of the rotor angle, ω ^ (n + 1): updated value of the estimated value of the angular velocity of the rotor, Δt: one period of the periodic signals Vh _u and Vh _w ( 3 control cycles), θ ^ (n): previous value of estimated rotor angle, ω ^ (n): previous value of estimated angular velocity of rotor, θ _E2 (n): estimated phase difference, K1 , K2, K ~: calculation gain, offset: offset value proportional to the d-axis command current, Vs (n): sine reference value, Vc (n): cosine reference value, L _lm (n): inductance reference value.

Therefore, in STEP 5, the angle estimation unit 25 calculates the phase difference estimation value θ _E2 required in the above equation (56) from the above equation (55). In subsequent STEP 6, I _d and I _q calculated by the three-phase / dq converter 26 are held for current feedback control. Then, when the counter variable ptr becomes 0 in STEP 7, the process proceeds to STEP 8, and the d-axis voltage Vfb _d and the q-axis voltage Vfb _q are calculated by the above equations (46) and (47). In STEP 9, the estimated angle θ ^ is updated by the above equation (56). In this way, by updating the drive voltages Vfb _d and Vfb _q and updating the estimated value θ ^ of the rotor angle at the same timing, deviation of the rotor angle when outputting the drive voltages Vfb _d and Vfb _q is eliminated. ing.

On the other hand, when ptr = 0 in STEP7, the process branches to STEP10, and the processes of STEP8 to STEP9 are not executed. Therefore, the d-axis voltage Vfb _d and the q-axis voltage Vfb _q, estimated angle theta ^ are periodic signals Vh _u, Vh _v, is updated every cycle of the Vh _w, periodic signals Vh _u, Vh _v, the Vh _w During one period, the d-axis voltage Vfb _{d, the} q-axis voltage Vfb _d, and the estimated angle θ ^ are kept constant.

In subsequent STEP 10, the dq / 3-phase converter 20 converts the d-axis voltage Vfb _d and the q-axis voltage Vfb _q into three-phase drive voltages Vfb _u , Vfb _v , Vfb _w , and in STEP 11, the periodic signal superimposing unit 21. As a result, the periodic signals Vh _u , Vh _v , Vh _w are superimposed on the three-phase drive voltages Vfb _u , Vfb _v , Vfb _w and output to the power drive unit 22. As a result, three-phase voltages Vu, Vv, and Vw are output from the power drive unit 22 to the motor 1. Then, in the subsequent STEP12 to STEP15, the counter variable ptr described above is incremented or cleared, and one control cycle is completed.

In STEP 11, the process of superimposing the periodic signals Vh _u , Vh _v , and Vh _w by the periodic signal superimposing unit 21 corresponds to the first step of the present invention. In STEP 4, the sine reference value Vs, the cosine reference value Vc, and the inductance The process of calculating the reference value L _lm and calculating the phase difference estimated value θ _E2 in STEP 5 corresponds to the process of the second process of the present invention. In STEP 9, the process of correcting by adding an offset to the phase difference estimated value θ _E2 corresponds to the third step of the present invention, and using the corrected phase difference estimated value θ _E2 , the rotor angle estimated value θ ^ Is a process corresponding to the fourth step of the present invention. Further, in STEP 9, configured to correct the phase difference estimates theta _E2 by adding the offset value to the phase difference estimates theta _E2 by the above formula (56) corresponds to a phase difference estimation value correcting means of the present invention.

The present also in the second embodiment, the similar to the first embodiment, except the phase difference reference value by the above formula (55) to (Vs · cos2θ ^ -Vc · sin2θ ^) inductance reference value L _lm By performing the normalization process, the phase difference estimated value θ _E2 can be calculated while suppressing the influence of field weakening / field strengthening of the rotor 2.

Then, according to the above equation (56), correction is performed by adding an offset value (offset) proportional to the d-axis command current of the motor 1 to the phase difference estimated value θ _E2 , and the corrected phase difference estimated value (θ _E2 + offset) By calculating the estimated value θ ^ of the rotor angle so that becomes zero, the difference between the actual value and the estimated value of the rotor angle can be further reduced to update the rotor angle θ ^.

  In the second embodiment, the sine reference value Vs and the cosine reference value Vc are calculated by the above equation (54). It is decided according to.

Further, in the first embodiment, the phase difference estimated value θ _E2 is calculated by the above equation (55), but the calculation formula of the phase difference estimated value θ _E2 is a calculation mode of the phase difference reference value and the inductance reference value. It is decided according to.

  In the first embodiment, the estimated value θ ^ of the rotor angle is updated by the observer of the above formula (56). However, the estimated value θ ^ of the rotor angle is updated using the estimated phase difference value. The present invention can be applied to any device.

  In the first embodiment and the second embodiment, the sine reference value, the cosine reference value, and the inductance reference value are calculated using the first-order difference of the detected current. However, the second-order difference of the detected current is calculated. The present invention can also be applied to the case where the sine reference value, cosine reference value, and inductance reference value are calculated using.

In the first and second embodiments, the phase difference estimation values θ _E1 and θ _E2 are corrected by adding an offset value proportional to the d-axis command current, but are proportional to the d-axis current. Correction may be made by adding the offset value. The effect of the present invention can also be obtained when correction for adding an offset value to the phase difference estimated values θ _E1 and θ _E2 is not performed.

The block diagram of DC brushless motor. The block diagram of the motor control apparatus in 1st Embodiment. Explanatory drawing of the output aspect of the periodic signal in the motor control apparatus shown in FIG. The flowchart of the estimation process of the rotor angle in 1st Embodiment. The timing chart corresponding to the flowchart of FIG. The comparison graph which showed the difference in the sine reference value by the presence or absence of saturation of a rotor magnetic pole. The comparison graph at the time of normalizing a phase difference reference value with a sine reference value and a cosine reference value. The comparison graph which showed the difference in the inductance reference value by the presence or absence of saturation of a rotor magnetic pole. Explanatory drawing of the inductance change of the motor by the phase difference of the actual value and estimated value of a rotor angle. Comparison graph when the sine reference value is normalized by the inductance reference value. The graph explaining the effect at the time of correct | amending which adds an offset to a phase difference estimated value. The block diagram of the motor control apparatus in 2nd Embodiment. The flowchart of the estimation process of the rotor angle in 2nd Embodiment.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... DC brushless motor, 2 ... Rotor, 3 ... U-phase armature, 4 ... V-phase armature, 5 ... W-phase armature, 10 ... Motor control device (of the first embodiment), 20 ... dq / 3-phase conversion unit, 21 ... periodic signal superposition unit (of the first embodiment), 22 ... power drive unit, 23 ... U-phase current sensor, 24 ... W-phase current sensor, 25 ... (first embodiment) Angle estimation unit, 26 ... three-phase / dq conversion unit, 27 ... non-interference calculation unit, 40 ... (second embodiment) motor control device, 50 ... (second embodiment) Angle estimator 51, periodic signal superimposing unit (of the second embodiment)

Claims (14)

A DC brushless motor is handled by converting it into an equivalent circuit having a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor and a d-axis armature on the d-axis orthogonal to the q-axis. Current detecting means for detecting a phase current flowing in the armature of the motor, and d flowing in the d-axis armature based on the phase current detected by the current detecting means and the estimated value of the rotor angle of the motor. Dq current calculation means for calculating a shaft current and a q-axis current flowing in the q-axis armature, and a deviation between a d-axis command current and a d-axis current, which is a command value of a current flowing in the d-axis armature, and The d-axis voltage applied to the d-axis armature at every predetermined control cycle and the q-axis so as to reduce the deviation between the q-axis command current and the q-axis current, which is the command value of the current flowing through the q-axis armature. The q-axis voltage applied to the shaft armature is determined, and the motor The motor control device for performing energization control, a method of estimating a rotor angle of the motor,
A first step of superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage;
A sine corresponding to a double angle of the phase difference between the estimated value and the actual value of the rotor angle of the motor, which is calculated based on changes in the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed. A phase difference reference value corresponding to a phase difference between an actual value and an estimated value of the rotor angle of the motor is calculated using at least one of a reference value and a cosine reference value, and the high-frequency voltage is superimposed. A second step of calculating an inductance reference value according to an average inductance of the motor when the high-frequency voltage is superimposed based on changes in the d-axis current and the q-axis current of the motor when
A phase difference that changes within a predetermined range according to a phase difference between an estimated value and an actual value of the rotor angle of the motor by performing a normalization process for correcting the phase difference reference value according to the level of the inductance reference value. A third step of calculating an estimated value;
And a fourth step of updating an estimated value of the rotor angle of the motor based on the estimated value of the phase difference. A DC brushless motor is handled by converting it into an equivalent circuit having a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor and a d-axis armature on the d-axis orthogonal to the q-axis. Current detecting means for detecting a phase current flowing in the armature of the motor, rotor angle estimating means for calculating an estimated value of the rotor angle of the motor, phase current detected by the current detecting means, and rotor of the motor Dq current calculation means for calculating a d-axis current flowing in the d-axis armature and a q-axis current flowing in the q-axis armature based on the estimated value of the angle, and a command for the current flowing in the d-axis armature The deviation between the d-axis command current and the d-axis current, which is a value, and the deviation between the q-axis command current and the q-axis current, which is a command value of the current flowing through the q-axis armature, is reduced every predetermined control cycle. D-axis voltage applied to the d-axis armature and It determines the q-axis voltage applied to the q-axis armature, a control device of a DC brushless motor and a current supply control means for performing energization control of the motor,
High-frequency superimposing means for superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage;
A sine reference value and a cosine reference corresponding to a double angle of a phase difference between an estimated value and an actual value of the rotor angle of the motor based on changes in the d-axis current and q-axis current of the motor when the high frequency is superimposed. A phase difference reference value corresponding to a phase difference between an actual value and an estimated value of the rotor angle of the motor is calculated using at least one of the values, and the motor of the motor when the high frequency is superimposed is calculated. reference value calculating means for calculating an inductance reference value according to an average inductance of the motor when the high-frequency voltage is superimposed based on changes in the d-axis current and the q-axis current;
A phase difference that changes within a predetermined range according to the phase difference between the actual value and the estimated value of the rotor angle of the motor by performing a normalization process for correcting the phase difference reference value according to the level of the inductance reference value. A phase difference estimated value calculating means for calculating an estimated value;
The said rotor angle estimation means updates the estimated value of the rotor angle of the said motor based on the said phase difference estimated value, The control apparatus of the DC brushless motor characterized by the above-mentioned. The high-frequency superimposing means superimposes a periodic signal, in which a sum of output voltages in a predetermined period including two or more control cycles is zero, on the d-axis voltage and the q-axis voltage as the high-frequency voltage,
The reference value calculation means sets the d-axis voltage and the q-axis voltage in the predetermined period to be constant, and sets the d-axis current corresponding to the first-order difference of the motor phase current between adjacent control cycles in the predetermined period. The first-order difference and the first-order difference of the q-axis current are calculated using the estimated value of the rotor angle, and the sine reference value and the first-order difference of the d-axis current and the first-order difference of the q-axis current are calculated. The DC brushless motor control device according to claim 2, wherein the cosine reference value and the inductance reference value are calculated. 4. The DC brushless motor according to claim 3, wherein the reference value calculation means calculates the sine reference value, the cosine reference value, and the inductance reference value according to the following formulas (1) to (5). Control device.

Where Vs _{dq is a} sine reference value, Vc _{dq is a} cosine reference value, L0 is an inductance reference value, dI _dq (i + 1) to dI _dq (i + n)... N during the predetermined period of the periodic signal. When a control cycle is included, the first-order difference between the d-axis current and the q-axis current between adjacent control cycles, i: the number of the calculation cycle of the sine reference value, cosine reference value, and inductance reference value.

Where L _{d is} the average inductance of the d-axis armature, and L _{q is} the average inductance of the q-axis armature.

However, j = 1,2, ..., n , dI d (i + j): 1 -order difference of the d-axis _{current, dI q (i + j)} : 1 -order difference of the q-axis current.

Where Vh _d (i + j): the output voltage to the d-axis armature of the j-th control cycle in the predetermined period of the periodic signal, Vh _q (i + j): j of the periodic signal in the predetermined period Output voltage to the q-axis armature in the second control cycle. 5. The control apparatus for a DC brushless motor according to claim 4, wherein the phase difference estimated value calculation means calculates the phase difference estimated value by performing the normalization process according to the following equation (6).

Where θ _{E1 is an} estimated phase difference value, Vs _{dq is a} sine reference value, and L 0 is an inductance reference value.   The rotor angle estimation means calculates the estimated value of the rotor angle of the motor based on the estimated phase difference value so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. The DC brushless motor control device according to claim 2, wherein an estimated value of a rotor angle of the motor is updated by an observer.   7. The phase difference estimated value correcting means for correcting the phase difference estimated value by adding an offset value proportional to the d-axis command current to the phase difference estimated value. The control apparatus of the DC brushless motor of any one of them. A drive voltage applied to the motor armature of the motor using an estimated value of the rotor angle of the motor at every predetermined control cycle so as to reduce the deviation between the phase current flowing through the armature of the DC brushless motor and the target current. In the motor control device that determines the energization of the motor and determines the rotor angle of the motor,
A first step of superimposing a high frequency voltage on the driving voltage;
At least one of a sine reference value and a cosine reference value corresponding to a double angle of the rotor angle of the motor, which is calculated based on a change in phase current of the motor when the high-frequency voltage is superimposed on the driving voltage. Using either of them, a phase difference estimated value corresponding to the phase difference between the actual value and the estimated value of the rotor angle of the motor is calculated, and the change in the phase current of the motor when the high-frequency voltage is superimposed is calculated. A second step of calculating an inductance reference value according to an average inductance of the motor when the high-frequency voltage is superimposed,
A phase difference that changes within a predetermined range according to a phase difference between an estimated value and an actual value of the rotor angle of the motor by performing a normalization process for correcting the phase difference reference value according to the level of the inductance reference value. A third step of calculating an estimated value;
And a fourth step of updating the estimated value of the rotor angle of the motor based on the estimated value of the phase difference. Current detecting means for detecting a phase current flowing in the armature of the DC brushless motor, rotor angle estimating means for calculating an estimated value of the rotor angle of the motor, phase current detected by the current detecting means, and rotor of the motor Based on the estimated value of the angle, the drive voltage applied to the armature of the motor is determined every predetermined control cycle so as to reduce the deviation between the predetermined target current and the phase current. In a DC brushless motor control device comprising an energization control means for performing
High frequency superimposing means for superimposing a high frequency voltage on the drive voltage;
Using at least one of a sine reference value and a cosine reference value corresponding to a double angle of the rotor angle of the motor, based on a change in phase current of the motor when the high-frequency voltage is superimposed, A phase difference reference value corresponding to a phase difference between an actual value and an estimated value of the rotor angle of the motor is calculated, and the high frequency voltage is superimposed based on a change in the phase current of the motor when the high frequency is superimposed. Reference value calculating means for calculating an inductance reference value according to the average inductance of the motor when
A phase difference that changes within a predetermined range according to the phase difference between the actual value and the estimated value of the rotor angle of the motor by performing a normalization process for correcting the phase difference reference value according to the level of the inductance reference value. A phase difference estimated value calculating means for calculating an estimated value;
The said rotor angle estimation means updates the estimated value of the rotor angle of the said motor based on the said phase difference estimated value, The control apparatus of the DC brushless motor characterized by the above-mentioned. The high-frequency superimposing means superimposes a periodic signal on which the sum of output voltages in a predetermined period including two or more control cycles becomes zero as the high-frequency voltage on the driving voltage,
The reference value calculation means calculates the first difference of the phase current of the motor between adjacent control cycles with the drive voltage in the predetermined period constant, and uses the first difference to calculate the sine reference value and the 10. The control apparatus for a DC brushless motor according to claim 9, wherein a cosine reference value and the inductance reference value are calculated. 11. The DC brushless motor according to claim 10, wherein the reference value calculation means calculates the sine reference value, the cosine reference value, and the inductance reference value according to the following formulas (7) to (11). Control device.

Where Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, l: average value of self-inductance of each phase of motor, m: average value of mutual inductance between each phase of motor, dI (i +1) to dI (i + n): the first-order difference of phase currents between adjacent control cycles when n control cycles are included in the predetermined period of the periodic signal, i: sine reference value and Calculation cycle number of cosine reference value and inductance reference value.

Where θ is the rotor angle of the motor.

Here, j = 1, 2,..., N, dI _u (i + j): first-order difference of U-phase phase current, dI _w (i + j): first-order difference of W-phase phase current.

Where Vh _uv (i + j) is the voltage between the U phase and V phase of the jth control cycle in the predetermined period of the periodic signal, and Vh _wv (i + j) is the jth of the periodic signal in the predetermined period. Interphase voltage of control phase W phase and V phase. 12. The control apparatus for a DC brushless motor according to claim 11, wherein the phase difference estimated value calculating means calculates the phase difference estimated value by the following equation (12).

Where θ _{E2 is the} estimated phase difference value, θ _e is the phase difference between the actual value and the estimated value of the rotor angle, Vs is the sine reference value, Vc is the cosine reference value, and Vs · cos2θ ^ −Vc · sin2θ ^ is the phase difference. Reference value, θ ^{^} : Estimated value of rotor angle, L _lm : Inductance reference value.   The rotor angle estimation means calculates the estimated value of the rotor angle of the motor based on the estimated phase difference value so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. The DC brushless motor control device according to any one of claims 8 to 12, wherein an estimated value of a rotor angle of the motor is updated by an observer.   The phase difference estimated value correcting means for correcting the phase difference estimated value by adding an offset value proportional to the target current to the phase difference estimated value. A control device for a DC brushless motor according to claim 1.

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