JP5170505B2 - Motor control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To achieve high-precision motor drive control by highly precisely estimating a rotation position regardless of variations in motor drive target value. <P>SOLUTION: A motor controller 5 includes a current sensor 9 for detecting a motor current flowing in an electric motor 3, a command voltage generating part 13 for generating a command voltage applied to the electric motor 3, a current filter 32 for filtering the motor current, a voltage filter 31 for filtering the command voltage, a position estimation part 21 for calculating a rotation position of the electric motor 3 on the basis of the filtered motor current and the filtered motor voltage, and a cut-off frequency setting part 23 that variably sets a cut-off frequency of the current filter 32 and that of the voltage filter 31 corresponding to a current command value. <P>COPYRIGHT: (C)2008,JPO&amp;INPIT

Description

  The present invention relates to a motor control device for sensorless driving of a brushless motor. The brushless motor is used, for example, as a source for generating a steering assist force in an electric power steering apparatus.

  A motor control device for driving and controlling a brushless DC motor is generally configured to control the supply of motor current according to the output of a position sensor for detecting the rotational position of the rotor. However, the environmental resistance of the position sensor becomes a problem, and the expensive position sensor and the wiring associated therewith hinder the cost reduction and the size reduction. Therefore, a sensorless driving method for driving a brushless DC motor without using a position sensor has been proposed. The sensorless driving method is a method for estimating the phase of the magnetic pole (electrical angle of the rotor) by estimating the induced voltage accompanying the rotation of the rotor.

  In a general position sensorless control algorithm, design parameters are fixedly determined on the assumption that the motor is rotating at a constant speed. Specific examples of the design parameter are a cutoff frequency of a current filter that cuts off a high-frequency component of a motor current flowing through the motor and a cutoff frequency of a voltage filter that cuts off a high-frequency component of a motor voltage applied to the motor. The induced voltage is estimated by a disturbance observer that estimates the induced voltage of the motor as a disturbance based on the motor current and the motor voltage that have passed through the current filter and the voltage filter, respectively, and the inverse motor model. By using the estimated induced voltage obtained by the disturbance observer, the rotational position of the rotor can be obtained. Further, the rotor rotational speed can be estimated by determining the change in the rotor rotational position for each predetermined period.

Such a general position sensorless control algorithm estimates the rotor rotational position and rotor rotational speed with good accuracy when the motor current or motor voltage changes sinusoidally, as in the case where the motor is constantly rotating. can do.
JP 2000-6829 A

  However, for example, when a brushless motor is used as a drive source of the electric power steering apparatus, there are many cases where the torque to be generated changes suddenly. For example, at the time of sudden steering or overcoming a curbstone, it is necessary to change the assist force suddenly, so that the motor drive target value fluctuates greatly and the rotation speed of the motor also fluctuates greatly. In such a situation, it cannot be guaranteed that the motor current and the motor voltage change in a sine wave shape. Specifically, when the motor drive target value changes greatly, the amplitudes of the motor current waveform and the motor voltage waveform change suddenly accordingly, so that these waveforms contain high frequency components. Further, if the motor drive target value is increased, the motor current and motor voltage waveforms are distorted, resulting in a waveform containing high-frequency components. Furthermore, if the motor rotation speed is increased, the motor current and motor voltage waveform cycles are correspondingly shortened. Therefore, if the cutoff frequency of the current filter and voltage filter is constant, the rotor rotation position can be estimated. There is a risk that necessary signal components may be eliminated.

Therefore, the conventional position sensorless control algorithm may be difficult to estimate the position and the rotational speed that reflect the motor current and the motor voltage in applications where the motor drive target value and the motor rotational speed frequently fluctuate. .
Accordingly, one object of the present invention is to provide a motor control device that can accurately estimate the rotational position regardless of fluctuations in the motor drive target value, thereby realizing high-precision motor drive control. It is.

  Another object of the present invention is to provide a motor control device that can accurately estimate the rotational position regardless of fluctuations in the motor rotational speed, thereby realizing highly accurate motor drive control. is there.

The motor control device of the present invention includes a motor current detecting means (9) for detecting a motor current flowing through the motor (3), a motor voltage indicating means (13, 14) for indicating a motor voltage applied to the motor, A current filter (32) for filtering the output of the motor current detecting means; a voltage filter (31) for filtering the output of the motor voltage indicating means; a motor current filtered by the current filter; and a filtering by the voltage filter A rotational position calculating means (21) for calculating the rotational position of the motor based on the motor voltage, a differentiating means (24) for differentiating the drive target value of the motor to calculate a drive target differential value; Calculate the rotation speed of the motor from the rotation position of the motor calculated by the rotation position calculation means. That the rotational speed calculating means (22), the driving target value of the motor, the calculated driving target differential value by said differentiating means, and in accordance with the rotational speed computed by the rotational speed calculating means, said current filter and the And a cutoff frequency setting means (23) for variably setting the cutoff frequency of the voltage filter. The alphanumeric characters in parentheses indicate corresponding components in the embodiments described later. The same applies hereinafter.

  According to this configuration, since the cutoff frequency of the current filter and the voltage filter is variably set according to the motor drive target value, the motor current and the motor voltage can be filtered in accordance with the fluctuation of the drive target value. Therefore, the rotational position calculation means can calculate the rotational position with high accuracy based on the filtered motor current and motor voltage. As a result, the motor control accuracy can be increased.

The cut-off frequency setting means sets the cut-off frequency higher as the drive target value of the motor is larger, sets the cut-off frequency higher as the absolute value of the drive target differential value is larger, and the rotation speed of the motor is higher. It is preferable that the cutoff frequency is set higher.
In this arrangement, the cutoff frequency setting section, the larger the driving target value of the motor, set high cut-off frequency. As a result, the target drive value of the motor is increased, and accordingly, when the motor current waveform and the motor voltage waveform are distorted, the high frequency component included in these waveforms can be given to the rotational position calculation means. . Thereby, since the signal component required for the calculation of the motor rotation position can be supplied to the rotation position calculation means, the accuracy of the rotation position calculation can be improved.

Further, the cutoff frequency setting section, set high enough cutoff frequency is larger absolute value of the driving target differential value.

  With this configuration, since the cutoff frequency is set in consideration of the time change of the drive target value of the motor, the accuracy of the rotational position calculation can be improved. More specifically, when the drive target value changes, the amplitudes of the motor current waveform and the motor voltage waveform change, so that these waveforms contain high frequency components. This high frequency component depends on the change speed of the amplitude, but this corresponds to the change speed of the drive target value, that is, the drive target differential value (time differential value). Therefore, by variably setting the cutoff frequency based on the drive target differential value, a signal component necessary for calculating the rotational position can be supplied to the rotational position calculating means. As a result, the accuracy of the rotational position calculation can be increased.

The front Symbol cutoff frequency setting means, as the rotational speed of the motor is large, it sets high cut-off frequency.

  According to this configuration, the calculation accuracy of the rotational position can be further increased by setting the cutoff frequency according to the rotational speed of the motor. If the rotation speed of the motor increases, the motor current waveform and the motor voltage waveform change accordingly and the frequency increases accordingly. Therefore, by changing the cutoff frequency of the current filter and the voltage filter according to the rotational speed of the motor, a signal having a frequency component corresponding to the rotation of the motor can be supplied to the rotational position calculation means. Thereby, the accuracy of the rotational position calculation can be increased.

Preferably, the motor control device further includes position correction means (28, 29) for correcting the rotation position calculated by the rotation position calculation means in accordance with the rotation speed calculated by the rotation speed calculation means.
By processing the motor current and the motor voltage with the current filter and the voltage filter, respectively, a phase delay corresponding to the characteristics of these filters occurs. This phase delay depends on the frequency of the motor current and the motor voltage. The frequency of the motor current and the motor voltage depends on the rotational speed of the motor. Therefore, by correcting the calculation result by the rotation position calculation means according to the rotation speed of the motor, it is possible to correct the position estimation error caused by the phase delay in the current filter and the voltage filter. Thereby, a more accurate rotational position can be obtained.
The position correction means includes a position compensation unit (28) for generating a position compensation value according to a characteristic obtained by inverting the sign of the frequency-to-phase relationship in the current filter and the voltage filter, and a position compensation value generated by the position compensation unit. And an adding unit (29) for adding to the rotational position calculated by the rotational position calculating means.
The present invention also provides a torque sensor (1) for detecting a steering torque, a motor (3) for applying a steering assist force to the steering mechanism (2), and a drive target value corresponding to the steering torque detected by the torque sensor. An electric power steering apparatus including the motor control apparatus (5) as described above that generates and controls the drive of the motor based on the drive target value is provided.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.
FIG. 1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control apparatus according to an embodiment of the present invention is applied. The electric power steering apparatus includes a torque sensor 1 that detects a steering torque applied to a steering wheel of a vehicle, an electric motor 3 that applies a steering assist force to a steering mechanism 2 of the vehicle, and a motor control that drives and controls the electric motor 3. And a device 5. The motor control device 5 drives the electric motor 3 in accordance with the steering torque detected by the torque sensor 1, thereby realizing appropriate steering assistance according to the steering situation. In this embodiment, the electric motor 3 is a three-phase brushless DC motor. As shown schematically in FIG. 2, the electric motor 3 has a rotor 50 as a field, and stator windings 51 of U phase, V phase and W phase, 52, 53. The electric motor 3 may be an inner rotor type in which a stator is disposed outside the rotor, or may be an outer rotor type in which a stator is disposed inside a cylindrical rotor.

  The motor control device 5 includes a microcomputer 7, a drive circuit (inverter circuit) 8 that is controlled by the microcomputer 7 and supplies electric power to the electric motor 3, and currents that flow through the stator windings of each phase of the electric motor 3. And a current sensor 9 for detection. The microcomputer 7 includes a CPU and a memory (such as a ROM and a RAM), and functions as a plurality of function processing units by executing a predetermined program. The plurality of function processing units include a current command value generation unit 11, a PI (proportional integration) control unit 12, an instruction voltage generation unit 13, a γδ / αβ coordinate conversion unit 14, and an αβ / UVW coordinate conversion unit 15. A PWM control unit 16, a UVW / αβ coordinate conversion unit 17, an αβ / γδ coordinate conversion unit 18, a deviation calculation unit 19, a signal processing unit 20, a position estimation unit 21, and a rotation speed estimation unit 22. , A cutoff frequency setting unit 23 and a current command value differentiating unit 24 are provided.

The current command value generation unit 11 generates a command value I _d ^* for the d-axis current component along the rotor magnetic pole direction of the electric motor 3 and a command value I _q ^* for the q-axis current component orthogonal to the d axis. Hereinafter, these are collectively referred to as “current command value i _dq ”. However, the dq coordinate plane is a plane along the rotation direction of the rotor 50, and the d axis and the q axis define a rotation coordinate system that rotates together with the rotor 50 (see FIG. 2).

When the current command value I ^* representing the amplitude of the current (sine wave current) to be applied to the U phase, V phase and W phase of the electric motor 3 is used, the d axis current command value I _d ^* and the q axis current command value I _q ^* Is expressed by the following equations (1) and (2).

したがって、電流指令値生成部１１は、ｄ軸電流指令値Ｉ_d ^*＝０を生成する一方で、トルクセンサ１によって検出される操舵トルクに応じたｑ軸電流指令値Ｉ_q ^*を生成する。より具体的には、操舵トルクに対応したｑ軸電流指令値Ｉ_q ^*を記憶したマップ（テーブル）を用いてｑ軸電流指令値Ｉ_q ^*が生成されるようになっていてもよい。電動モータ３が発生するトルクは、モータ電流に対応するから、電流指令値ｉ_dqは、電動モータ３から発生させるべきトルクを指令するための「トルク指令値」と言い換えることもできる。

Therefore, the current command value generation unit 11 generates the d-axis current command value I _d ^* = 0, while generating the q-axis current command value I _q ^* corresponding to the steering torque detected by the torque sensor 1. More specifically, the q-axis current command value I _q ^* may be generated using a map (table) that stores the q-axis current command value I _q ^* corresponding to the steering torque. Since the torque generated by the electric motor 3 corresponds to the motor current, the current command value i _dq can be rephrased as a “torque command value” for commanding the torque to be generated from the electric motor 3.

The current sensor 9 detects the U-phase current I _U , the V-phase current I _V and the W-phase current I _w of the electric motor 3 (hereinafter, collectively referred to as “three-phase detection current I _UVW ”). The detected value is given to the UVW / αβ coordinate converter 17.
The UVW / αβ coordinate conversion unit 17 converts the three-phase detection current I _UVW into the currents I _α and I _β on the two-phase fixed coordinate system (hereinafter, collectively referred to as “two-phase detection current I _αβ ”). Convert coordinates to. The two-phase fixed coordinate system is a fixed coordinate system in which the α axis and the β axis orthogonal to the rotation center of the rotor 50 are defined (see FIG. 2). The coordinate-converted two-phase detection current I _αβ is given to the αβ / γδ coordinate conversion unit 18.

The αβ / γδ coordinate conversion unit 18 converts the two-phase detection current I _αβ into a rotational coordinate system (γ−) according to the rotor rotational position θ ^ estimated by the position estimation unit 21 (hereinafter referred to as “estimated rotational position θ ^”). The coordinates are converted into currents I _γ and I _δ on δ) (hereinafter referred to as “detection current I _γδ ” collectively). This rotational coordinate system (γ−δ) is a rotational coordinate system defined by the γ axis along the rotor magnetic pole direction and the δ axis orthogonal to the γ axis when the rotor 50 is at the estimated rotational position θ ^. is there. When the estimated rotational position θ ^ has no error and coincides with the actual rotor rotational position, the dq rotational coordinate system and the γδ rotational coordinate system coincide.

The detected current I _γδ is given to the deviation calculating unit 19. The deviation calculator 19 calculates a deviation of the γ-axis current I _{γ with} respect to the d-axis current command value I _d ^* and a deviation of the δ-axis current I _{δ with} respect to the q-axis current command value I _q ^* . These deviations are given to the PI control unit 12 and are each subjected to PI calculation processing. Then, in accordance with these calculation results, the command voltage generator 13 generates a γ-axis command voltage V _γ ^* and a δ-axis command voltage V _δ ^* (hereinafter referred to as “command voltage V _γδ ” when these are collectively referred to). It is generated and given to the γδ / αβ coordinate converter 14.

The γδ / αβ coordinate conversion unit 14 converts the γ-axis instruction voltage V _γ ^* and the δ-axis instruction voltage V _δ ^* into an α-axis instruction voltage V _α ^* and a β-axis instruction voltage V _β that are instruction voltages in the two-phase fixed coordinate system. ^* (Hereinafter, these are collectively referred to as “two-phase indicating voltage V _αβ ”). The two-phase instruction voltage V _αβ is given to the αβ / UVW coordinate conversion unit 15.
The αβ / UVW coordinate converter 15 converts the α-axis command voltage V _α ^* and the β-axis command voltage V _β ^* into the command voltages of the three-phase fixed coordinate system, that is, the command voltages V _U ^{* of the} U phase, V phase, and W phase , V _V ^* , V _W ^* (hereinafter collectively referred to as “three-phase indicating voltage V _UVW ”).

The PWM control unit 16 generates a drive signal having a duty ratio controlled according to the three-phase instruction voltages V _U ^* , V _V ^* , and V _W ^* and supplies the drive signal to the drive circuit 8. Thus, a voltage is applied to each phase of the electric motor 3 with a duty ratio corresponding to the instruction voltages V _U ^* , V _V ^* , and V _W ^* of the corresponding phase.
With such a configuration, when a steering torque is applied to a steering wheel (not shown) as an operation member coupled to the steering mechanism 2, this is detected by the torque sensor 1. Then, the current command value generation unit 11 generates a current command value I _dq corresponding to the detected steering torque. A deviation between the current command value I _dq and the detected current I _γδ is obtained by the deviation calculation unit 19, and PI calculation is performed by the PI control unit 12 so as to lead this deviation to zero. The command voltage V _γδ corresponding to the calculation result is generated by the command voltage generator 13 and converted into the three-phase command voltage V _UVW through the coordinate converters 14 and 15. Then, the drive circuit 8 operates with a duty ratio corresponding to the three-phase instruction voltage V _UVW by the action of the PWM control unit 16, whereby the electric motor 3 is driven, and an assist torque corresponding to the current command value I _dq is generated. The steering mechanism 2 will be given. Thus, steering assistance can be performed according to the steering torque. The three-phase detection current I _UVW detected by the current sensor 9 passes through the coordinate conversion units 17 and 18, and the detection current I _γδ expressed in the rotating coordinate system (γ−δ) so as to correspond to the current command value I _dq. After being converted to, it is given to the deviation calculator 19.

In order to perform coordinate conversion between the rotating coordinate system and the fixed coordinate system, a phase angle (electrical angle) θ representing the rotational position of the rotor 50 is required. An estimated rotational position θ ^ representing this phase angle is generated by the position estimation unit 21 and is given to the γδ / αβ coordinate conversion unit 14 and the αβ / γδ coordinate conversion unit 18.
The position estimation unit 21 removes a high frequency component from the disturbance observer 25 that estimates the induced voltage of the electric motor 3 as a disturbance based on the motor model that is a mathematical model of the electric motor 3, and the estimated induced voltage that is output from the disturbance observer 25. And an estimated position generation for generating an estimated rotational position θ ^ of the rotor 50 based on an estimated value filter 26 configured by a low-pass filter that performs and an estimated induced voltage (value after filtering) output by the estimated value filter 26 Part 27.

The position estimation unit 21 is supplied with the two-phase command voltage V _αβ and the two-phase detection current I _αβ after the signal processing (filtering) by the signal processing unit 20. The signal processing unit 20 includes a voltage filter 31 configured with a low-pass filter that removes a high-frequency component of the two-phase instruction voltage V _αβ and a low-pass filter that removes a high-frequency component of the two-phase detection current I _αβ. Current filter 32. Therefore, the two-phase indication voltage V _αβ filtered by the voltage filter 31 and the two-phase detection current I _αβ filtered by the current filter 32 are input to the disturbance observer 25 of the position estimation unit 21. .

The rotational speed estimator 22 generates the estimated rotational speed ω ^ (= Δθ ^) of the rotor 50 by obtaining the difference Δθ ^ of the estimated rotational position θ ^ given every predetermined control period from the position estimator 21. .
The current command value differentiating unit 24 time-differentiates the current command value i _dq (torque command value) generated by the current command value generating unit 11, and the obtained differential value (drive target differential value) is supplied to the cutoff frequency setting unit 23. give. In fact, the current command value differentiator 24 calculates the difference between the previous value and the current value of the current command value i _dq, giving the cutoff frequency setting unit 23 so as differential value of the current command value i _dq.

The cutoff frequency setting unit 23 variably sets the cutoff frequency ω _c of the estimated value filter 26, the voltage filter 31, and the current filter 32. That is, the estimated value filter 26, a voltage filter 31 and the current filter 32 are both a low-pass filter capable of variably setting the cutoff frequency omega _c, setting their cutoff frequency omega _c is the cutoff frequency setting unit 23 It has come to be. Based on the current command value i _dq (motor drive value) generated by the current command value generation unit 11 and the estimated rotational speed ω ^ calculated by the rotational speed estimation unit 22, the cutoff frequency setting unit 23 It operates to set the cut-off frequency ω _{c of} 31 and 32.

FIG. 3 is a block diagram for explaining an example of the disturbance observer 25 and a configuration related thereto. A motor model that is a mathematical model of the electric motor 3 can be expressed as, for example, (R + pL) ⁻¹ . Here, R is an armature winding resistance, L is an αβ axis inductance, and p is a differential operator. It can be considered that the two-phase command voltage V _αβ and the induced voltage E _αβ (α-axis induced voltage E _α and β-axis induced voltage E _β ) are applied to the electric motor 3.

The disturbance observer 25 receives the two-phase detection current I _αβ as an input, and an inverse motor model (inverse model of the motor model) 35 that estimates the motor voltage, and the motor voltage estimated by the inverse motor model 35 and the two-phase indication voltage V _αβ. And a voltage deviation calculation unit 36 for obtaining the deviation. The voltage deviation calculator 36 obtains a disturbance with respect to the two-phase indicating voltage V _αβ , and as is apparent from FIG. 3, this disturbance is an estimated value E ^ _αβ (α-axis induced voltage estimation corresponding to the induced voltage E _αβ. Value E ^ _α and β-axis induced voltage estimated value E ^ _β, hereinafter collectively referred to as "estimated induced voltage E ^ _αβ "). The reverse motor model 35 is represented by R + pL, for example.

As described above, in this embodiment, the disturbance observer 25 is configured to obtain the estimated induced voltage E ^ _αβ using the instruction voltage V _αβ and the detected current I _αβ of the two-phase fixed coordinate system. The induced voltage can be estimated without being affected by the speed. Thereby, it can contribute to the improvement of the rotor rotational position detection accuracy of the electric motor 3 used for the electric power steering device in which the rotational speed fluctuation is likely to occur.

The estimated value filter 26 can be configured by, for example, a low-pass filter represented by a / (s + a). a is a design parameter, and the cutoff frequency ω _c of the estimated value filter 26 is changed by variably setting the design parameter a. That is, the cut-off frequency setting unit 23 is configured to variably set the cut-off frequency ω _c by variably setting this design parameter.

The induced voltage E _αβ can be expressed by the following equation (3). However, K _E is an induced voltage constant, θ is a rotor rotational position, and ω is a rotor rotational speed.

したがって、推定誘起電圧Ｅ^_αβが求まれば、次の(4)式に従って、ロータ５０の推定回転位置θ^が求まる。この演算が、推定位置生成部２７によって行われるようになっている。

Therefore, when the estimated induced voltage E ^ _αβ is obtained, the estimated rotational position θ ^ of the rotor 50 is obtained according to the following equation (4). This calculation is performed by the estimated position generation unit 27.

図４は、遮断周波数設定部２３の働きを説明するための図であり、フィルタ２６，３１，３２の遮断周波数ω_cの設定例を示す図である。フィルタ２６，３１，３２のそれぞれに対して異なる遮断周波数を設定してもよいが、この例では、これらのフィルタ２６，３１，３２に対して共通の遮断周波数ω_cを設定するようにしている。

FIG. 4 is a diagram for explaining the function of the cutoff frequency setting unit 23 and is a diagram illustrating an example of setting the cutoff frequency ω _c of the filters 26, 31 and 32. Different cutoff frequencies may be set for the filters 26, 31, and 32, but in this example, a common cutoff frequency ω _c is set for these filters 26, 31, and 32. .

The cut-off frequency ω _c is set so as to increase monotonously with an increase in the estimated rotational speed ω ^ of the electric motor 3. More specifically, in the example of FIG. 4, the cutoff frequency ω _c is set so as to increase linearly with an increase in the estimated rotational speed ω ^. Further, the cutoff frequency omega _c is corrected so as to become larger as the current command value i _dq is large, and is corrected so as to increase the larger the absolute value of the differential value of the current command value i _dq.

More specifically, a basic cutoff frequency ω ^* _c that changes according to the estimated rotational speed ω ^ is determined in advance, and this basic cutoff frequency ω ^* _c depends on the current command value i _dq and its differential value. It is to be corrected. The relationship of the basic cutoff frequency ω ^* _{c to} the estimated rotational speed ω ^ may be stored in a memory in the form of a map (table), or the basic cutoff frequency ω ^* corresponding to the estimated rotational speed ω ^ by function calculation ^. _c may be required. The basic cut-off frequency ω ^* _c is not necessarily determined. For example, the cut-off frequency ω _c corresponding to the estimated rotation speed ω ^, the current command value i _dq and its differential value is stored in a three-dimensional map (table). It can also be set as a structure to keep.

FIG. 5 is a diagram for explaining the effect of changing the cut-off frequency ω _c according to the rotor rotational speed of the electric motor 3, and shows the time change of the motor current. In the period 61, the rotational speed of the rotor is relatively slow, and in the period 62, the rotational speed of the rotor is high. Accordingly, the period of the current waveform in the period 61 is long (the frequency of the current waveform is low), but the period of the current waveform in the period 62 is short (the frequency of the current waveform is high). If the cut-off frequency ω _c is fixed in accordance with the period of the current change in the period 61, the current signal after being filtered by the current filter 32 cannot faithfully represent the change in the motor current in the period 62. . On the other hand, if the configuration of the present embodiment is set such that the cut-off frequency ω _c is set higher as the rotor rotational speed is higher, the change in the motor current is faithfully expressed in any period, and unnecessary noise components are used. The current signal from which the current is removed can be output from the current filter 32. The same applies to the voltage filter 31 and the estimated value filter 26. Thus, by changing the cutoff frequency ω _c according to the rotor rotational speed, it becomes easier to obtain a signal necessary for the position estimation calculation in the estimated position generation unit 27, and the position estimation accuracy can be increased.

FIG. 6 is a diagram for explaining the effect of changing the cut-off frequency ω _c in accordance with the differential value of the current command value i _dq , and shows the time change of the motor current. The amplitude of the motor current varies according to the change in the current command value i _dq . That is, when the current command value i _dq is large, the amplitude of the motor current increases as indicated by reference numeral 65. When the amplitude of the motor current changes, even if the rotor rotational speed does not change, a component having a frequency higher than the rotational speed is included in the motor current waveform. Therefore, by increasing the cutoff frequency ω _c as the absolute value of the differential value of the current command value i _dq increases, a current signal that follows the amplitude of the motor current can be output from the current filter 32 and is unnecessary. Noise components can be removed. The same applies to the voltage filter 31 and the estimated value filter 26. In this way, it becomes easier to obtain a signal necessary for the position estimation calculation in the estimated position generation unit 27, so that the position estimation accuracy can be increased.

FIG. 7 is a diagram for explaining the effect of changing the cut-off frequency ω _c according to the current command value i _dq , and shows the time change of the motor current. As described above, the amplitude of the motor current increases as the current command value i _dq increases. However, as the motor current increases, the motor current waveform does not become a sine wave, and distortion is generated as indicated by reference numeral 67. It comes to occur. In this case, the motor current waveform includes a component having a frequency higher than the rotor rotational speed. Therefore, in this embodiment, the larger the current command value i _dq is, the higher the cutoff frequency ω _c is, so that a signal representing the motor current waveform more faithfully is output from the current filter 32, and unnecessary noise is generated. The component can be removed. The same applies to the voltage filter 31 and the estimated value filter 26. In this way, it becomes easier to obtain a signal necessary for the position estimation calculation in the estimated position generation unit 27, so that the position estimation accuracy can be increased.

FIG. 8 is a flowchart for explaining the flow of the rotor rotational position estimation calculation by the microcomputer 7. The control period is mainly controlled by the signal processing unit 20, the position estimation unit 21, the rotational speed estimation unit 22, and the cutoff frequency setting unit 23. The flow of processing that is repeatedly executed every time is shown.
The two-phase indication voltage V _αβ calculated by the γδ / αβ coordinate converter 14 is input to the voltage filter 31, and the two-phase detection current I _αβ calculated by the UVW / αβ coordinate converter 17 is input to the current filter 32. (Step S1).

On the other hand, the cutoff frequency setting unit 23 determines the cutoff frequency ω ^ based on the estimated rotational speed ω ^ (however, the estimated value obtained by the rotational speed estimation unit 22 in the previous control cycle), the current command value i _dq, and the differential value thereof. _c is set (step S2). That is, the design parameters of each filter are set so that the cutoff frequencies of the estimated value filter 26, the voltage filter 31, and the current filter 32 become the set value ω _c .

Thereby, in the voltage filter 31 and the current filter 32, the input voltage signal and current signal are filtered at the set cutoff frequency ω _c (step S3).
The disturbance observer 25 obtains an estimated induced voltage E ^ _αβ using the filtered voltage signal and current signal (step S4). The estimated value filter 26 performs filtering on the estimated induced voltage E ^ _αβ (step S5). This filtering is a low-pass filter process at the cutoff frequency ω _c .

Using the estimated induced voltage E ^ _αβ filtered in this manner, the estimated position generation unit 27 performs position estimation to obtain an estimated rotational position θ ^, and further, based on the estimated rotational position θ ^, rotational speed estimation is performed. A rotation speed estimation calculation is performed in the unit 22 (step S6).
As described above, according to this embodiment, the cutoff frequency ω _{c of} the voltage filter 31, the current filter 32, and the estimated value filter 26 is variable based on the estimated rotational speed ω ^, the current command value i _dq, and the differential value thereof. Is set. Thereby, current signals and voltage signals necessary and sufficient for estimating the induced voltage can be input to the disturbance observer 25, and the estimated induced voltage E ^ _αβ processed in a frequency band necessary and sufficient for the rotational position estimating calculation. Can be input to the estimated position generation unit 27. As a result, since the position estimation accuracy can be improved, the coordinate conversion calculation in the γδ / αβ coordinate conversion unit 14 and the αβ / γδ coordinate conversion unit 18 can be performed accurately, and thus the electric motor 3 is highly accurate. Sensorless control.

FIG. 9 is a block diagram for explaining a second embodiment of the present invention, and shows a configuration of a position estimation unit 211 that can be used in place of the position estimation unit 21 in the configuration of FIG. 1 described above. Yes. The position estimation unit 211 includes an estimated position compensation unit 28 that generates a position compensation value (correction value) θ _c for compensating (correcting) an error in the estimated rotational position generated by the estimated position generation unit 27, and the estimated position. In addition to the position compensation value θ _c generated by the compensation section 28 and the estimated rotation position θ ^ generated by the estimated position generation section 27, an addition section 29 that generates a corrected estimated rotation position (= θ ^ + θ _c ) I have. The corrected estimated rotational position generated by the adding unit 29 is supplied to the rotational speed estimating unit 22, the γδ / αβ coordinate converting unit 14, and the αβ / γδ coordinate converting unit 18.

FIG. 10 is a diagram for explaining an example of the position compensation value generated by the estimated position compensation unit 28. The estimated position compensator 28 is in the range of 0 ° to 90 ° according to the compensation value characteristic curve indicated by the curve (broken line) 71 according to the estimated rotational speed ω ^ (motor rotational speed) obtained by the rotational speed estimator 22. To generate a position compensation value θ _c . That is, the lower limit value of the position compensation value θ _c is 0 °, and the upper limit value thereof is 90 °. Estimated rotational speed omega ^ (motor rotational speed) is less than a predetermined first threshold value TH1, the position compensation value theta _c is the lower limit value 0 °. Further, when the estimated rotation speed omega ^ (motor rotation speed) exceeds a predetermined second threshold TH2 (> TH1), the position compensation value theta _c is an upper limit value 90 °. The position compensation value θ _c is determined so that the estimated rotational speed ω ^ (motor rotational speed) between the first and second thresholds TH1 and TH2 increases as the motor rotational speed increases. In the example of FIG. 10, the position compensation value θ _c is determined so as to change linearly with respect to the estimated rotational speed ω ^ (motor rotational speed) in the range from the first threshold TH1 to the second threshold TH2. ing.

FIG. 11 is a diagram for explaining the problem of phase delay caused by the signal processing unit 20 (the voltage filter 31 and the current filter 32). FIG. 11 shows, as an example, a Bode diagram when the signal processing unit 20 is assumed to be a simple first-order low-pass filter (low-pass filter) and its cutoff frequency is ω _c (Hz). Has been. The upper diagram shows changes in gain with respect to frequency, and the lower diagram shows changes in phase with respect to frequency.

If the cut-off frequency ω _c can be set in a low frequency region without phase delay, the rotor rotational position can be accurately estimated using the filtered signal.
However, in the case of the electric power steering device, the frequency range of the signal necessary for estimating the rotor rotational position is in the frequency range where the phase delay occurs. Therefore, the cut-off frequency ω _c is set in a frequency region where a phase delay occurs. Therefore, when the motor rotational speed is ω, an error θε is generated in the calculation of the rotational position of the rotor.

The estimated position compensation unit 28 generates a position compensation value θ _c for compensating for such an error θε.
In FIG. 10, a reference curve 72 obtained by inverting the sign of the frequency-to-phase relationship in the Bode diagram of FIG. 11 is indicated by a two-dot chain line. If the position compensation value θ _c is generated according to the reference curve 72, the error θε caused by the signal processing unit 20 can be compensated. The aforementioned curve 71 is a linear approximation of the reference curve 72. Of course, this curve 71 is only an example, and the compensation value characteristic curve may be determined by performing nonlinear approximation on the reference curve 72. Further, it is possible to modify the position compensation value θ _c obtained according to the curve 71 by adding an offset in consideration of the time delay of the signal processing unit 20 or the like.

Of course, the signal processing unit 20 does not have to be composed of a primary low-pass filter. When filter processing of another configuration is applied, the compensation value characteristic curve may be determined accordingly.
As described above, in this embodiment, an accurate estimated rotational position (θ ^ + θ _c ) obtained by compensating the phase delay generated in the signal processing unit 20 according to the estimated rotational speed ω ^ is obtained, so that the electric motor can be operated with higher accuracy. The motor 3 can be controlled. In other words, since the signal processing unit 20 does not need to consider the phase delay caused by the filter processing, the signal processing unit 20 can optimize the signal input to the position estimation unit 21. Signal processing becomes possible. Thereby, position estimation precision can be improved significantly and the performance and efficiency of the electric motor 3 at the time of sensorless control can be improved.

The estimated position compensation unit 28 can be configured by a map (table) that defines the relationship of the position compensation value θ _{c with} respect to the estimated rotational speed ω ^, as well as the model (mathematical model) of the signal processing unit 20 and the estimated rotational speed ω. Based on ^, the position compensation value corresponding to the phase delay in the signal processing unit 20 may be calculated.
While the two embodiments of the present invention have been described above, the present invention can also be implemented in other forms. For example, in the above-described embodiment, the example in which the present invention is applied to the electric motor 3 as a drive source of the electric power steering apparatus has been described. However, the present invention is applicable to the control of the electric motor for uses other than the electric power steering apparatus. It can also be applied to.

  In addition, various design changes can be made within the scope of matters described in the claims.

1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control device according to an embodiment of the present invention is applied. FIG. It is an illustration figure for demonstrating the structure of an electric motor. It is a block diagram for demonstrating an example of a disturbance observer and the structure relevant to this. It is a figure for demonstrating the function of a cutoff frequency setting part, and the example of a setting of a cutoff frequency is shown. It is a figure for demonstrating the effect by changing a cutoff frequency according to a rotor rotational speed. It is a figure for demonstrating the effect by changing a cutoff frequency according to the differential value of an electric current command value. It is a figure for demonstrating the effect by changing a cutoff frequency according to an electric current command value. It is a flowchart for demonstrating the flow of a rotor rotational position estimation calculation. It is a block diagram for demonstrating other embodiment of this invention, and shows the other structural example of the position estimation part which can be used in the structure of FIG. It is a figure for demonstrating an example of a position compensation value. It is a figure for demonstrating the problem of the phase delay resulting from the process in a signal processing part.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 5 ... Motor control apparatus, 7 ... Microcomputer, 9 ... Current sensor, 20 ... Signal processing part, 21 ... Position estimation part, 211 ... Position estimation part

Claims (5)

Motor current detecting means for detecting a motor current flowing through the motor;
Motor voltage indicating means for indicating a motor voltage applied to the motor;
A current filter for filtering the output of the motor current detection means;
A voltage filter for filtering the output of the motor voltage indicating means;
Based on the motor current filtered by the current filter and the motor voltage filtered by the voltage filter, rotational position computing means for computing the rotational position of the motor;
Differentiating means for differentiating the drive target value of the motor and calculating the drive target differential value;
A rotation speed calculation means for calculating the rotation speed of the motor from the rotation position of the motor calculated by the rotation position calculation means;
A cutoff that variably sets the cutoff frequency of the current filter and the voltage filter according to the drive target value of the motor, the drive target differential value calculated by the differentiating means, and the rotational speed calculated by the rotational speed calculating means. A motor control device including frequency setting means;   The cut-off frequency setting means sets the cut-off frequency higher as the drive target value of the motor is larger, sets the cut-off frequency higher as the absolute value of the drive target differential value is larger, and the rotation speed of the motor is higher. The motor control device according to claim 1, wherein the cutoff frequency is set higher. 3. The motor control device according to claim 1, further comprising a position correction unit that corrects a rotation position calculated by the rotation position calculation unit according to a rotation speed calculated by the rotation speed calculation unit.   The position correction unit includes a position compensation unit that generates a position compensation value according to a characteristic obtained by inverting the sign of the frequency-to-phase relationship in the current filter and the voltage filter, and the position compensation value generated by the position compensation unit is rotated. The motor control device according to claim 3, further comprising: an adding unit that adds to the rotational position calculated by the position calculating means.   A torque sensor for detecting steering torque;
  A motor for providing steering assist force to the steering mechanism;
  The motor control device according to any one of claims 1 to 4, wherein a drive target value corresponding to a steering torque detected by the torque sensor is generated, and the motor is driven and controlled based on the drive target value. Electric power steering device.

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